Order this document by MC33215/D 52 1 FB SUFFIX PLASTIC PACKAGE CASE 848B (TQFP–52) The MC33215 is developed for use in fully electronic telephone sets with speakerphone functions. The circuit performs the ac and dc line termination, 2–4 wire conversion, line length AGC and DTMF transmission. The speakerphone part includes a half duplex controller with signal and noise monitoring, base microphone and loudspeaker amplifiers and an efficient supply. The circuit is designed to operate at low line currents down to 4.0 mA enabling parallel operation with a classical telephone set. • • • • • • • • Highly Integrated Cost Effective Solution Straightforward AC and DC Parameter Adjustments 42 Efficient Supply for Loudspeaker Amplifier and Peripherals Stabilized Supply Point for Handset Microphone 1 Stabilized Supply Point for Base Microphone B SUFFIX PLASTIC PACKAGE CASE 858 (SDIP–42) Loudspeaker Amplifier can be Powered and Used Separately Smooth Switch–Over from Handset to Speakerphone Operation Adjustable Switching Depth for Handsfree Operation ORDERING INFORMATION Device Operating Temperature Range TQFP–52 MC33215FB MC33215B Package TA = –20° to +70°C SDIP–42 Simplified Application AC Impedance DC Offset Line Current Telephone Line DTMF MF Handset Microphone HM Base Microphone Current Splitter 1:10 VCC Supply Attenuator BM DC Slope Duplex Controller VCC or External Supply Base Loudspeaker Line Driver Attenuator LS Receive Signal Rx Handset Earpiece Auxiliary Input This device contains 2782 active transistors. This document contains information on a new product. Specifications and information herein are subject to change without notice. MOTOROLA ANALOG IC DEVICE DATA Motorola, Inc. 1997 Rev 0 1 MC33215 FEATURES • Separate Input for DTMF and Auxiliary Signals • Parallel Operation Down to 4.0 mA of Line Current Line Driver and Supply • AC and DC Termination of Telephone Line • Adjustable Set Impedance for Real and Complex Termination • Efficient Supply Point for Loudspeaker Amplifier and Peripherals • Two Stabilized Supply Points for Handset and Base Microphones • Separate Supply Arrangement for Handset and Speakerphone Operation Speakerphone Operation • Handsfree Operation via Loudspeaker and Base Microphone • Integrated Microphone and Loudspeaker Amplifiers • Differential Microphone Inputs • Loudspeaker Amplifier can be Powered and Used Separately from the Rest of the Circuit • Integrated Switches for Smooth Switch–Over from Handset to Speakerphone Operation • Signal and Background Noise Monitoring in Both Channels • Adjustable Switching Depth for Handsfree Operation • Switch–Over • Dial Tone Detector in the Receive Channel Handset Operation • Transmit and Receive Amplifiers • Differential Microphone Inputs • Sidetone Cancellation Network • Line Length AGC • Microphone and Earpiece Mute 43 42 41 40 LSB N/C N/C 44 VLS 45 LSO 46 PGD 47 N/C 48 VCC 49 VLN N/C 50 VHF 51 VMC 52 N/C Figure 1. Pin Connections 1 VCC PGD 42 2 VLN LSO 41 3 VHF VLS 40 4 VMC LSB 39 1 SLB LSF 39 5 SLB LSF 38 2 REG BVO 38 6 REG BVO 37 3 SLP PPL 37 7 SLP PPL 36 4 MFI LSI 36 8 MFI LSI 35 5 HM1 VOL 35 9 HM1 VOL 34 6 HM2 SWD 34 10 HM2 SWD 33 7 BM2 8 BM1 AGC 32 9 VDD Gnd 31 10 TSA RLS 30 11 TSE RSA 29 12 TBN RSE 28 13 MUT RBN 27 REF 33 N/C N/C SPS PRS SWT LSM N/C RXS RXO GRX RXI N/C N/C TQFP–52 14 15 16 17 18 19 20 21 22 23 24 25 26 (Top View) 11 BM2 SDIP–42 REF 32 12 BM1 AGC 31 13 VDD Gnd 30 14 TSA RLS 29 15 TSE RSA 28 16 TBN RSE 27 17 MUT RBN 26 18 SPS RXI 25 19 PRS GRX 24 20 SWT RXO 23 21 LSM RXS 22 (Top View) 2 MOTOROLA ANALOG IC DEVICE DATA MC33215 MAXIMUM RATINGS Rating Min Max Unit –0.5 12 V – 160 mA Voltage at VLS (if Powered Separately) –0.5 12 V Voltage at VHF (if Externally Applied) –0.5 5.5 V Voltage at SPS, MUT, PRS, LSM –0.5 7.5 V – 150 °C –65 150 °C Peak Voltage at VLN Maximum Loop Current Maximum Junction Temperature Storage Temperature Range NOTE: ESD data available upon request. RECOMMENDED OPERATING CONDITIONS Characteristic Min Max Unit Biasing Voltage at VLN 2.4 10 V Loop Current 4.0 130 mA Voltage at VLS 2.4 8.0 V Voltage at VHF (if Externally Applied) 2.4 5.0 V 0 5.0 V –20 70 °C Voltage at SPS, MUT, PRS, LSM Operating Ambient Temperature Range ELECTRICAL CHARACTERISTICS (All parameters are specified at T = 25°C, Iline = 18 mA, VLS = 2.9 V, f = 1000 Hz, PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.) ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ Characteristic Min Typ Max Unit – 3.9 4.8 2.4 4.2 5.2 – 4.5 5.6 – 1.2 1.5 mA DC Voltage at VMC (= VMC0) 1.6 1.75 1.9 V Current Available from VMC VMC = VMC0 – 200 mV 1.0 – – mA 2.6 2.8 3.0 V – 1.4 2.0 mA 2.0 – – mA Current Available from VCC VCC = 2.4 V, Iline = 20 mA 13 15 – mA DC Voltage Drop Between VLN and VCC Iline = 20 mA – 1.0 1.5 V – 1.0 1.5 mA DC LINE VOLTAGE Line Voltage Vline Parallel Operation, Iline = 4.0 mA Iline = 20 mA Iline = 70 mA V SUPPLY POINT VDD Internal Current Consumption from VDD VDD = 2.5 V SUPPLY POINT VMC SUPPLY POINT VHF DC Voltage at VHF (= VHF0) Internal Current Consumption from VHF VHF = VHF0 + 100 mV Current Available from VHF VHF = VHF0 – 300 mV SUPPLY POINT VCC SUPPLY INPUT VLS Internal Current Consumption from VLS MOTOROLA ANALOG IC DEVICE DATA 3 MC33215 ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, Iline = 18 mA, VLS = 2.9 V, f = 1000 Hz, PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.) ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ Characteristic Min Typ Max Unit Logic Low Level Pins PRS, MUT, SPS, LSM – – 0.4 V Logic High Level Pins PRS, MUT, SPS, LSM 2.0 – 5.0 V Internal Pull Up Pins PRS, MUT, LSM – 100 – kΩ Internal Pull Down Pin SPS – 100 – kΩ Voltage Gain from VHM to Vline VHM = 1.5 mVrms 46 47 48 dB Gain Reduction in Mute Condition MUT = Low or PRS = Low or SPS = High 60 – – dB Input Impedance at HM1 and HM2 14 18 22 kΩ Common Mode Rejection Ratio – 50 – dB Total Harmonic Distortion at VLN VHM = 4.5 mVrms – – 2.0 % Psophometrically Weighted Noise Level at Vline HM1 and HM2 Shorted with 200 Ω – –72 – dBmp Voltage Gain from VBM to Vline VBM = 0.5 mVrms 53 55.5 58 dB Input Impedance at BM1 and BM2 14 18 22 kΩ Common Mode Rejection Ratio – 50 – dB Total Harmonic Distortion at VLN VBM = 1.5 mV – – 2.0 % Psophometrically Weighted Noise Level at Vline BM1 and BM2 Shorted with 200 Ω – –62 – dBmp Gain Reduction in Mute Condition MUT = Low or PRS = Low or SPS = Low 60 – – dB Voltage Gain from VMF to Vline VMF = 7.5 mVrms 34 35 36 dB Input Impedance at MFI 14 18 22 kΩ Gain Reduction in Mute Condition MUT = High or PRS = Low 60 – – dB Voltage Gain from VRXI to VEAR (Note 1) Vline = 20 mVrms 23 24 25 dB Gain Reduction in Mute Condition MUT = Low or SPS = Low 60 – – dB Input Impedance at RXI 24 30 36 kΩ Psophometrically Weighted Noise Level at VEAR RXI Shorted to Gnd via 10 µF – 130 – µVrms Confidence Level During DTMF Dialing VMF = 7.5 mVrms, MUT = Low 10 15 20 mVrms Output Swing Capability into 150 Ω THD ≤2% 680 – – mVpp Output Swing Capability into 450 Ω THD ≤2%, RRXO = 360 kΩ 1800 – – mVpp LOGIC INPUTS Tx CHANNEL, HANDSET MICROPHONE AMPLIFIER Tx CHANNEL, BASE MICROPHONE AMPLIFIER (SPS = HIGH, Tx MODE FORCED) Tx CHANNEL, DTMF AMPLIFIER (MUT = LOW OR PRS = LOW) Rx CHANNEL, EARPIECE AMPLIFIER NOTE: 4 1. Corresponding to –0.6 dB gain from the line to output RXO in the typical application. MOTOROLA ANALOG IC DEVICE DATA MC33215 ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, Iline = 18 mA, VLS = 2.9 V, f = 1000 Hz, PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.) ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ Characteristic Min Typ Max Unit Voltage Gain from VRXI to VRLS (Note 2) Vline = 20 mVrms 21 24 27 dB Gain Reduction in Mute Condition SPS = Low or MUT = Low 60 – – dB Voltage Gain from VLSI to VLSP VLSI = 10 mVrms 25 26 27 dB Attenuation at Delta RVOL = 47 kΩ – 32 – dB Psophometrically Weighted Noise Level at VLSP RXI Shorted to Gnd via 10 µF – 1.2 – mVrms Confidence Level During DTMF Dialing VMF = 7.5 mVrms MUT = Low 150 200 250 mVrms Available Peak Current from LSO 110 – – mApeak Output Capability into 25 Ω THD ≤2%, VLSI = 55 mVrms 1.8 – – Vpp Output Capability into 25 Ω THD ≤2%, VLS = 5.0 V, VLSI = 90 mVrms 2.7 – – Vpp Gain Reduction in Mute Condition LSM = Low 60 – – dB Peak–to–Peak Limiter Attack Time VLSI Jumps from 40 mVrms to 120 mVrms – – 5.0 ms Peak–to–Peak Limiter Release Time VLSI Jumps from 120 mVrms to 40 mVrms – 300 – ms THD at 10 dB Overdrive VLSI = 120 mVrms – – 7.0 % Peak–to–Peak Limiter Disable Threshold at PPL – – 0.1 V 4.5 6.0 7.5 dB Gain Variation in Transmit and Receive Channel with Respect to Iline =18 mA with AGC Disabled (AGC to VDD) – – 1.5 dB Highest Line Current for Maximum Gain – 20 – mA Lowest Line Current for Minimum Gain – 50 – mA 20 – – dB – – 28 dB Voltage Gain from RXI to RSA VRXI = 15 mVrms 18 20 22 dB Voltage Gain from BMI to TSA VBM = 0.5 mVrms 17.5 18.5 19.5 dB Dynamic Range of Logarithmic Compression from TSA to TSE and RSA to RSE ITSA and IRSA from 2.5 µA to 250 µA 40 – – dB Envelope Tracking Between TSE and RSE and Between TBN and RBN – ±3.0 – dB 0.3 0.4 0.5 µA Rx CHANNEL, LOUDSPEAKER PRE–AMPLIFIER (SPS = HIGH, Rx MODE FORCED) Rx CHANNEL, LOUDSPEAKER AMPLIFIER Rx CHANNEL PEAK–TO–PEAK LIMITER AUTOMATIC GAIN CONTROL Gain Reduction in Transmit and Receive Channel with Respect to Iline = 18 mA Iline = 70 mA BALANCE RETURN LOSS Balance Return Loss with Respect to 600 Ω SIDETONE Voltage Gain from VHM to VEAR S1 in Position 2 LOGARITHMIC AMPLIFIERS AND ENVELOPE DETECTORS Maximum Source Current from TSE or RSE NOTE: 2. Corresponding to –0.6 dB gain from the line to output RLS in the typical application. MOTOROLA ANALOG IC DEVICE DATA 5 MC33215 ELECTRICAL CHARACTERISTICS (continued) (All parameters are specified at T = 25°C, Iline = 18 mA, VLS = 2.9 V, f = 1000 Hz, PRS = high, MUT = high, SPS = low, LSM = high, test figure in Figure 17 with S1 in position 1, unless otherwise stated.) ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁ Characteristic Min Typ Max Unit Maximum Sink Current into TSE or RSE 100 – – µA Maximum Sink Current into TBN and RBN 0.7 1.0 1.3 µA Maximum Source Current from TBN or RBN 100 – – µA Dial Tone Detector Threshold at Vline – 20 – mVrms Speech Noise Threshold Both Channels – 4.5 – dB Switching Depth 46 50 54 dB Adjustable Range for Switching Depth 24 – 60 dB LOGARITHMIC AMPLIFIERS AND ENVELOPE DETECTORS ATTENUATOR CONTROL Gain Variation in Idle Mode for Both Channels – 25 – dB Current Sourced from SWT Tx Mode 7.0 10 13 µA Current Sunk into SWT Rx Mode 7.0 10 13 µA PIN FUNCTION DESCRIPTION Pin ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ N Name D Description i i SDIP–42 TQFP–52 1 47 VCC Supply Output for Loudspeaker Amplifier and Peripherals 2 48 VLN Line Connection Input 3 49 VHF Supply Output for Speakerphone Section and Base Microphone 4 50 VMC Supply Output for Handset Microphone – 51 N/C Not Connected – 52 N/C Not Connected 5 1 SLB SLP Buffered Output 6 2 REG Regulation of Line Voltage Adjustment 7 3 SLP DC Slope Adjustment 8 4 MFI DTMF Input 9 5 HM1 Handset Microphone Input 1 10 6 HM2 Handset Microphone Input 2 11 7 BM2 Base Microphone Input 2 12 8 BM1 Base Microphone Input 1 13 9 VDD Supply Input for Speech Part 14 10 TSA Transmit Sensitivity Adjustment 15 11 TSE Transmit Signal Envelope Timing Adjustment 16 12 TBN Transmit Background Noise Envelope Timing Adjustment 17 13 MUT Transmit and Receive Mute Input – 14 N/C Not Connected – 15 N/C Not Connected 18 16 SPS Speakerphone Select Input 19 17 PRS Privacy Switch Input 20 18 SWT Switch–Over Timing Adjustment 21 19 LSM Loudspeaker Mute Input 6 MOTOROLA ANALOG IC DEVICE DATA MC33215 PIN FUNCTION DESCRIPTION (continued) Pin ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ Name Description SDIP–42 TQFP–52 – 20 N/C Not Connected 22 21 RXS Receive Amplifier Stability 23 22 RXO Receive Amplifier Output 24 23 GRX Earpiece Amplifier Feedback Input 25 24 RXI Receive Amplifier Input – 25 N/C Not Connected – 26 N/C Not Connected 26 27 RBN Receive Background Noise Envelope Timing Adjustment 27 28 RSE Receive Signal Envelope Timing Adjustment 28 29 RSA Receive Sensitivity Adjustment 29 30 RLS Receive Output for Loudspeaker Amplifier 30 31 Gnd Small Signal Ground 31 32 AGC Line Length AGC Adjustment 32 33 REF Reference Current Set 33 34 SWD Switching Depth Adjustment for Handsfree 34 35 VOL Volume Control Adjustment 35 36 LSI Loudspeaker Amplifier Input 36 37 PPL Peak–to–Peak Limiter Timing Adjustment 37 38 BVO Bias Voltage for Loudspeaker Amplifier Output 38 39 LSF Loudspeaker Amplifier Feedback Input – 40 N/C Not Connected – 41 N/C Not Connected 39 42 LSB Loudspeaker Amplifier Bootstrap Output 40 43 VLS Supply Input for Loudspeaker Amplifier 41 44 LSO Loudspeaker Amplifier Output 42 45 PGD Power Ground – 46 N/C Not Connected MOTOROLA ANALOG IC DEVICE DATA 7 MC33215 DESCRIPTION OF THE CIRCUIT Based on the typical application circuit as given in Figure 18, the MC33215 will be described in three parts: line driver and supplies, handset operation, and handsfree operation. The data used refer to typical data of the characteristics. LINE DRIVER AND SUPPLIES The line driver and supply part performs the ac and dc telephone line termination and provides the necessary supply points. AC Set Impedance The ac set impedance of the telephone as created by the line driver and its external components can be approximated with the equivalent circuit shown in Figure 2. Inductor CVLN 10 n Zbal CVDD 100 µ RSLB 2.2 k Inductor Slope RREG1 360 k RREG ∞ CREG 220 n With: V zener R x Slope R SLP + slope 11 VLN ) RRREG1 REG2 With the component values of the typical application, the inductor calculates as 1.6 H. Therefore, in the audio range of 300 Hz to 3400 Hz, the set impedance is mainly determined by ZVDD. As a demonstration, the impedance matching or Balance Return Loss BRL is shown in Figure 3. Figure 3. Balance Return Loss 40 1 Ǔ REG1 ) RRREG1 REG2 ǒǒ + 3.8 V ) Iline – 1.0 mA ^ 3.8 V ) Iline x 20 ǒ Ǔ Ǔ Ǔ x 20 . 12 25 BRL (dB) 10 µA x R REG2 Figure 4. Influence of RREG1 and RREG2 on the DC Mask 30 20 RREG1 = 470 k RREG2 = 220 k 10 15 RREG1 = 365 k RREG2 = 220 k 8.0 VLN (V) 10 5.0 1000 10000 f, FREQUENCY (Hz) The influence of the frequency dependent parasitic components is seen for the lower frequencies (Inductor) and the higher frequencies (CVLN) by a decreasing BRL value. DC Set Impedance The line current flowing towards the MC33215 application is partly consumed by the circuitry connected to VDD while the rest flows into Pin VLN. At Pin VLN, the current is split up 8 x Ǔ slope In the typical application this leads to a line voltage of 4.2 V at 20 mA of line current with a slope of 20 Ω. Adding a 1.5 V voltage drop for the diode bridge and the interruptor, the dc voltage at tip–ring will equal 5.7 V. If the dc mask is to be adapted to a country specific requirement, this can be done by adjusting the resistors RREG1 and RREG2, as a result, the zener voltage and the slope are varied. It is not advised to change the resistor RSLP since this changes many parameters. The influence of RREG1 and RREG2 is shown in Figure 4. 35 0 100 ILN x R If RREG2 is not mounted, the term between the brackets becomes equal to 1. With the values shown in the typical application and under the assumption that IVDD = 1.0 mA, the above formulas can be simplified to: ǒ Ǔ 1 ǒ + 0.2 x 1 ) RRREG1 ) + Iline – IVDD + RREG1 x CREG x RSLP 11 + RSLP 11 + Vzener ) ǒ Ǔǒ ǒ Ǔ VLN ILN Figure 2. Equivalent of the AC impedance ZVDD 620 into a small part for biasing the internal line drive transistor and into a large part for supplying the speakerphone. The ratio between these two currents is fixed to 1:10. The dc set impedance or dc setting of the telephone as created by the line driver and its external components can be approximated with the equivalent of a zener voltage plus a series resistor according to: 6.0 RREG1 = 365 k RREG2 = Infinite 4.0 RREG1 = 470 k RREG2 = Infinite 2.0 0 0 20 40 60 80 100 Iline (mA) As can be seen in Figure 4, for low line currents below 10 mA, the given dc mask relations are no longer valid. This is the result of an automatic decrease of the current drawn MOTOROLA ANALOG IC DEVICE DATA MC33215 from Pin REG by the internal circuit (the 10 µA term in the formulas). This built–in feature drops the line voltage and therefore enables parallel operation. The voltage over the line driver has to be limited to 12 V to protect the device. A zener of 11 V at VLN is therefore the maximum advised. VDD Supply The internal circuitry for the line driver and handset interface is powered via VDD. This pin may also be used to power peripherals like a dialer or microcontroller. The voltage at VDD is not internally regulated and is a direct result of the line voltage setting and the current consumption at VDD internally (IVDD) and externally (IPER). It follows that: ǒ + V Ǔ ) VLN – I I x R set DD VDD PER For correct operation, it must be ensured that VDD is biased at 1.8 V higher than SLP. This translates to a maximum allowable voltage drop across Z VDD of Vzener – 1.8 V. In the typical application, this results in a maximum allowable current consumption by the peripherals of 2.0 mA. VMC Supply At VMC, a stabilized voltage of 1.75 V is available for powering the handset microphone. Due to this stabilized supply, microphones with a low supply rejection can be used which reduces system costs. In order to support the parallel operation of the telephone set, the voltage at VMC will be maintained even at very low line currents down to 4.0 mA. Under normal supply conditions of line currents of 20 mA and above, the supply VMC is able to deliver a guaranteed minimum of 1.0 mA. However, for lower line currents, the supply capability of VMC will decrease. Figure 5. VMC Under Different Microphone Loads 1.8 1.7 Iline = 20 mA Iline = 4.0 mA 2.7 k VMC–VHF 1.6 VMC (V) 1.5 Iline = 4.0 mA 1.3 1.2 1.1 1.0 0.2 0.4 0.6 VHF Supply VHF is a stabilized supply which powers the internal duplex controller part of the MC33215, and which is also meant to power the base microphone or other peripherals. The base microphone however, can also be connected to VMC, which is preferred in case of microphones with a poor supply rejection. Another possibility is to create an additional filter at VHF, like is shown in the typical application. The supply capability of VHF is guaranteed as 2.0 mA for line currents of 20 mA and greater. Since in parallel operation not enough line current is available to power a loudspeaker and thus having a speakerphone working, the current internally supplied to VHF is cut around 10 mA of line current to save current for the handset operated part. A small hysteresis is built in to avoid system oscillations. When the current to VHF is cut, the voltage at VHF will drop rapidly due to the internal consumption of 1.4 mA and the consumption of the peripherals. When VHF drops below 2.0 V, the device internally switches to the handset mode, neglecting the state of the speakerphone select Pin SPS. In case an application contains a battery pack or if it is mains supplied, speakerphone operation becomes possible under all line current conditions. In order to avoid switch–over to handset operation below the 10 mA, VHF has to be supplied by this additional power source and preferably kept above 2.4 V. VCC Supply At VCC the major part of the line current is available for powering the loudspeaker amplifier and peripheral circuitry. This supply pin should be looked at as a current source since the voltage on VCC is not stabilized and depends on the instantaneous line voltage and the current to and consumed from VCC. The maximum portion of the line current which is available at VCC is given by the following relation: 10 x I I – I – I – I line VDD VHF VCC VMC 11 + 1.4 0 If, during parallel operation, a high current is required from VMC, a 2.7 k resistor between VMC and VHF can be applied. In Figure 5, the VMC voltage under different microphone currents, is shown. 0.8 1.0 IVMC (mA) MOTOROLA ANALOG IC DEVICE DATA 1.2 1.4 1.6 ǒ ǒ ǓǓ This formula is valid when the voltage drop from VLN to VCC is sufficient for the current splitter to conduct all this current to VCC. When the drop is not sufficient, the current source saturates and the surplus of current is conducted to the power ground PGD to avoid distortion in the line driver. In fact, when no current is drawn from VCC, the voltage at VCC will increase until the current splitter is in balance. In Figure 6 this behavior is depicted. 9 MC33215 Figure 6. Available Current at VCC 100 3.5 90 3.0 IVCC at 98% of IVCC(max) 80 2.5 IVCC/lline (%) 60 VLN–V CC (V) mA AND % 70 50 40 30 0 20 40 60 80 100 VCC Open 0 20 0 40 60 80 Iline (mA) Iline (mA) A. Maximum Available Current at VCC B. Voltage Drop to VCC For instance, at a line current of 20 mA a maximum of 15 mA of current is available at VCC. If all this current is taken, VCC will be 1.7 V below VLN. When not all this current is drawn from VCC, but for instance only 1.0 mA for biasing of the loudspeaker amplifier, the voltage at VCC will be 1.2 V below VLN. Although the measurements for Figure 6 are done with RREG1 = 365 k, the results are also globally valid for other dc settings. As can be seen from Figure 6, the voltage at VCC is limited by the voltage at VLN minus 1.0 V. This means that the voltage at VCC is limited by the external zener at VLN. If it is necessary to limit the voltage at VCC in order to protect peripheral circuits, a zener from VCC to Gnd can be added. If the supply of the loudspeaker VLS is also connected to VCC, it is advisable that VCC does not exceed 8.0 V. The high efficiency of the VCC power supply contributes to a high loudspeaker output power at moderate line currents. More details on this can be found in the handsfree operation paragraph. HANDSET OPERATION During handset operation, the MC33215 performs the basic telephone functions for the handset microphone and earpiece. It also enables DTMF transmission. Handset Microphone Amplifier The handset microphone is to be capacitively connected to the circuit via the differential input HM1 and HM2. The microphone signal is amplified by the HMIC amplifier and modulates the line current by the injection of the signal into the line driver. This transfer from the microphone inputs to the line current is given as 15/(RSLP/11), which makes a total transmit voltage gain AHM from the handset microphone inputs to the line of: V Z x Z set line 15 A x line HM V Z Z set R 11 HM line SLP With the typical application and Zline = 600 Ω the transmit gain calculates as 47 dB. In case an electret microphone is used, it can be supplied from the stabilized microphone supply point VMC of 1.75 V properly biased with resistors RHM1 and RHM2. This allows the setmaker to use an electret microphone with poor supply rejection to reduce total system costs. Since the transmit gain AHM is fixed by the advised RSLP = 220 Ω and the constraints of set impedance and line impedance, the transmit gain is set + 10 VCC to VLS 1.5 0.5 10 0 2.0 1.0 IVCC(max) (mA) 20 IVCC at 50% of IVCC(max) + ń ) 100 by adjusting the sensitivity of the handset microphone by adjusting the resistors RHM1 and RHM2. It is not advised to adjust the gain by including series resistors towards the Pins HM1 and HM2. A high pass filter is introduced by the coupling capacitors CHM1 and CHM2 in combination with the input impedance. A low pass filter can be created by putting capacitors in parallel with the resistors RHM1 and RHM2. The transmit noise is measured as –72 dBmp with the handset microphone inputs loaded with a capacitively coupled 200 Ω. In a real life application, the inputs will be loaded with a microphone powered by VMC. Although VMC is a stablized supply voltage, it will contain some noise which can be coupled to the handset microphone inputs, especially when a microphone with a poor supply rejection is used. An additional RC filter on VMC can improve the noise figure, see also the base microphone section. Handset Earpiece Amplifier The handset earpiece is to be capacitively connected to the RXO output. Here, the receive signal is available which is amplified from the line via the sidetone network and the Rx and EAR amplifiers. The sidetone network attenuates the receive signal from the line via the resistor divider composed of RSLB and Zbal, see also the sidetone section. The attenuation in the typical application by this network equals 24.6 dB. Then the signal from the sidetone network is pre–amplified by the amplifier Rx with a typical gain of 6.0 dB. This amplifier also performs the AGC and MUTE functions, see the related paragraphs. Finally, the signal is amplified by the noninverting voltage amplifier EAR. The overall receive gain ARX from the line to the earpiece output then follows as: R V RXO RXO A x A x 1 A RXI RX ST R V line GRX + + ǒ Ǔ ) With: AST = Attenuation of the Sidetone Network ARXI = Gain of the Pre–Amplifier Rx For the typical application an overall gain from the line to the earpiece is close to 0 dB. The receive gain can be adjusted by adjusting the resistor ratio RRXO over RGRX. However, RRXO also sets the confidence tone level during dialing which leaves RGRX to be chosen freely. A high pass filter is introduced by the coupling capacitor CRXI together with the input impedance of the input MOTOROLA ANALOG IC DEVICE DATA MC33215 Sidetone Cancellation The line driver and the receiver amplifier of the MC33215 are tied up in a bridge configuration as depicted in Figure 7. This bridge configuration performs the so–called hybrid function which, in the ideal case, prevents transmitted signals from entering the receive channel. Figure 7. Sidetone Bridge Automatic Gain Control To obtain more or less constant signal levels for transmit and receive regardless of the telephone line length, both the transmit and receive gain can be varied as a function of line current when the AGC feature is used. The gain reduction as a function of line current, and thus line length, is depicted in Figure 8. Figure 8. Automatic Gain Control 0 –1.0 RAGC = 20 k –2.0 AGC (dB) RXI. A second high pass filtering is introduced by the combination of CGRX and RGRX. A low pass filter is created by CRXO and RRXO. The coupling capacitor at the output RXO is not used for setting a high pass filter but merely for dc decoupling. In combination with dynamic ear capsules, the EAR amplifier can become unstable due to the highly inductive characteristic of some of the capsules. To regain stability, a 100 nF capacitor can be connected from RXS to Gnd in those cases. An additional 10 nF at the RXI input, as shown in the typical application, improves the noise figure of the receiver stage. RAGC = 30 k –3.0 –4.0 –5.0 –6.0 0 10 20 30 VLN Zline//Zset V Zbal x 15 ń HM R 11 SLP Transmit Gnd RXI Receive RSLP/11 RSLB Gnd SLP As can be seen from Figure 7 by inspection, the receiver will not pick up any transmit signal when the bridge is in balance, that is to say when: Z Z Z set bal line R R 11 SLB SLP The sidetone suppression is normally measured in an acoustic way. The signal at the earpiece when applying a signal on the microphone is compared with the signal at the earpiece when applying a signal on the line. The suppression takes into account the transmit and receive gains set. In fact the sidetone suppression can be given as a purely electrical parameter given by the properties of the sidetone bridge itself. For the MC33215, this so–called electrical sidetone suppression ASTE can be given as: R 11 Z SLP A 1 – bal x STE R Z Z set SLB line Values of –12 dB or better, thus ASTE < 0.25, can easily be reached in this way. + ńń ń ń ńń + 40 50 60 70 Iline (mA) For small line currents, and thus long lines, no gain reduction is applied and thus the transmit and receive gains are at their maximum. For line currents higher than Istart, the gain is gradually reduced until a line current Istop is reached. This should be the equivalent of a very short line, and the gain reduction equals 6.0 dB. For higher line currents the gain is not reduced further. For the start and stop currents the following relations are valid: 1 I stop R 11 SLP + + R 1 ń11 SLP ń 20 µ x R ń AGC 11 SLP For the typical application, where RAGC = 30 kΩ, the gain will start to be reduced at Istart = 20 mA while reaching 6.0 dB of gain reduction at Istop = 50 mA. When AGC is connected to VDD, the AGC function is disabled leading to no gain reduction for any line current. This is also sometimes called PABX mode. The automatic gain control takes effect in the HMIC and Rx amplifiers as well as in the BMIC amplifier. In this way the AGC is also active in speakerphone mode, see the handsfree operation paragraph. I start – R Privacy and DTMF Mode During handset operation a privacy and a DTMF mode can be entered according the logic Table 1. Table 1. Logic Table for Handset Mode ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ Logic Inputs Amplifiers SPS MUT PRS HMIC BMIC DTMF Rx RXatt EAR 0 1 1 Handset Normal On Off Off On Off On 0 1 0 0 0 Handset Privacy Off Off On On Off On X Handset DTMF Off Off On Off Off On M d Mode MOTOROLA ANALOG IC DEVICE DATA 11 MC33215 Table 2. Logic Table for Handsfree Mode Logic Inputs Amplifiers ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁ SPS MUT PRS HMIC BMIC DTMF Rx RXatt EAR 1 1 1 Handsfree Normal Off On Off On On Off 1 1 0 Handsfree Privacy Off Off On On On Off 1 0 X Handsfree DTMF Off Off On Off On Off M d Mode By applying a logic 0 to Pin MUT, the DTMF mode is entered where the DTMF amplifier is enabled and where the Rx amplifier is muted. A DTMF signal can be sent to the line via the MFI input for which the gain ADTMF is given as: V Z x Z set line 3.75 A x line DTMF V Z Z set R 11 MFI line SLP In the typical application, the gain equals 35 dB. The DTMF gain can be controlled by a resistor divider at the input MFI as shown in the typical application. The signal has to be capacitively coupled to the input via CMFI which creates a high pass filter with the input impedance. The line length AGC has no effect on the DTMF gains. The signal applied to the MFI input is made audible at the earpiece output for confidence tone. The signal is internally applied to the GRX pin where it is amplified via the EAR amplifier which is used as a current to voltage amplifier. The gain is therefore proportional to the feedback resistor RRXO. For RRXO = 180 kΩ the gain equals 6.0 dB. The confidence tone is also audible at the loudspeaker output when the loudspeaker amplifier is activated, see speakerphone operation. By applying a logic 0 to Pin PRS, the MC33215 enters privacy mode. In this mode, both handset and handsfree microphone amplifiers are muted while the DTMF amplifier is enabled. Through the MFI input, a signal, for example music on hold, can be sent to the line. In the same way, the MFI input can also be used to couple in signals from, for instance, an answering machine. + + ń ) HANDSFREE OPERATION Handsfree operation, including DTMF and Privacy modes, can be performed by making Pin SPS high according Table 2. The handset amplifiers will be switched off while the base amplifiers will be activated. The MC33215 performs all the necessary functions, such as signal monitoring and switch–over, under supervision of the duplex controller. With the MC33215 also a group listening–in application can be built. For more information on this subject please refer to application note AN1574. Base Microphone Amplifier The base microphone can be capacitively connected to the circuit via the differential input BM1 and BM2. The setup is identical to the one for the handset microphone amplifier. The total transmit voltage gain ABM from the base microphone inputs to the line is: V Z x Z set line 37.5 A x line BM V Z Z set R 11 BM line SLP + 12 + ń With the typical application and Zline = 600 Ω the transmit gain calculates as 55 dB. The electret base microphone can be supplied directly from VHF but it is advised to use an additional RC filter to obtain a stable supply point, as shown in the typical application. The microphone can also be supplied by VMC. The transmit gain is set by adjusting the sensitivity of the base microphone by adjusting the resistors RBM1 and RBM2. It is not advised to adjust the gain by including series resistors towards the Pins BM1 and BM2. A high pass filter is introduced by the coupling capacitors CBM1 and CBM2 in combination with the input impedance. A low pass filter can be created by putting capacitors in parallel with the resistors RBM1 and RBM2. Loudspeaker Amplifier The loudspeaker amplifier of the MC33215 has three major benefits over most of the existing speakerphone loudspeaker amplifiers: it can be supplied and used in a telephone line powered application but also stand alone, it has an all NPN bootstrap output stage which provides maximum output swing under any supply condition, and it includes a peak–to–peak limiter to limit the distortion at the output. The loudspeaker amplifier is powered at Pin VLS. In telephone line powered applications, this pin should be connected to VCC where most of the line current is available, see the VCC supply paragraph. In an application where an external power supply is used, VLS and thus the loudspeaker amplifier can be powered separately from the rest of the circuit. The amplifier is grounded to PGD, which is the circuits power ground shared by both the loudspeaker amplifier and the current splitter of the VCC supply. Half the supply voltage of VLS is at BVO, filtered with a capacitor to VLS. This voltage is used as the reference for the output amplifier. The receive signal present at RLS can be capacitively coupled to LSI via the resistor RLSI. The overall gain from RLS to LSO follows as: V R LSO A – LSF x 4.0 LS V R RLS LSI In the typical application this leads to a loudspeaker gain ALS of 26 dB. The above formula follows from the fact that the overall amplifier architecture from RLS to LSO can be looked at as an inverting voltage amplifier with an internal current gain from LSI to LSF of 4. The input LSI is a signal current summing node which allows other signals to be applied here. + + ) MOTOROLA ANALOG IC DEVICE DATA MC33215 Figure 9. Loudspeaker Output Stage 0.5 VLS 1.5 VLS 0 VLS VLS –0.5 VLS 0.5 VLS Loudspeaker LSB VLS CLSO T2 LSO T1 VLS 0.5 VLS PGD 0 Figure 10. Loudspeaker Amplifier Output Power with External Supply 140 300 120 200 P LSP (mW) P LSP (mW) 100 80 60 150 RLSP = 50 Ω 100 40 RLSP = 50 Ω 50 20 0 2.0 RLSP = 25 Ω 250 RLSP = 25 Ω 3.0 4.0 5.0 6.0 7.0 8.0 VLS (V) A. Peak–to–Peak Limiter Active The total gain from the telephone line to the loudspeaker output includes, besides the loudspeaker amplifier gain, also the attenuation of the sidetone network and the internal gain from RXI to RLS. When in receive mode, see under duplex controller, the gain from RXI to RLS is maximum and equals 24 dB at maximum volume setting. The attenuation of the sidetone network in the typical application equals 24.6 dB which makes an overall gain from line to loudspeaker of 25.4 dB. Due to the influence of the line length AGC on the Rx amplifier, the gain will be reduced for higher line currents. The output stage of the MC33215 is a modified all NPN bootstrap stage which ensures maximum output swing under all supply conditions. The major advantage of this type of output stage over a standard rail–to–rail output is the higher stability. The principle of the bootstrap output stage is explained with the aid of Figure 9. The output LSO is biased at half the supply VLS while the filtering of the loudspeaker with the big capacitor CLSO requires that LSB is biased at VLS. In fact, because of the filtering, LSB is kept at VLS/2 above the LSO output even if LSO contains an ac signal. This allows the output transistor MOTOROLA ANALOG IC DEVICE DATA 0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 VLS (V) B. Peak–to–Peak Limiter Disabled T2 to be supplied for output signals with positive excursions up to VLS without distorting the output signal. The resulting ac signal over the loudspeaker will equal the signal at LSO. As an indication of the high performance of this type of amplifier, in Figure 10, the output power of the loudspeaker amplifier as a function of supply voltage is depicted for 25 Ω and 50 Ω loads with both the peak–to–peak limiter active and disabled. As can be seen, in case the peak–to–peak limiter is disabled, the output power is roughly increased with 6.0 dB, this at the cost of increased distortion levels up to 30%. In a telephone line powered application, the loudspeaker amplifier output power is limited not only by the supply voltage but also by the telephone line current. This means that in telephones the use of 25 Ω or 50 Ω speakers is preferred over the use of the cheaper 8.0 Ω types. Figure 11 gives the output power into the loudspeaker for a line powered application and two different dc settings with the peak–to–peak limiter active. In case the peak–to–peak limiter is disabled the output power will be increased for the higher line currents up to 6.0 dB. 13 MC33215 Figure 11. Loudspeaker Amplifier Output Power when Line Powered 100 RREG1 = 365 k RREG2 = 220 k RLSP = 25 Ω 90 80 70 RREG1 = 365 k RREG2 = Infinite 50 RLSP = 25 Ω R 40 REG1 = 365 k RREG2 = Infinite 30 R = 50 Ω LSP 20 PLSP (mW) 60 loudspeaker amplifier is muted which is needed for correct handset operation. The volume of the loudspeaker signal can be varied via a potentiometer at VOL. A fixed current of 10 µA is running through the potentiometer and the resulting voltage at VOL is a measure for the gain reduction. The relation between the voltage at VOL and the obtained gain reduction is given in Figure 13. RREG1 = 365 k RREG2 = 220 k RLSP = 50 Ω Figure 13. Volume Reduction 0 –5.0 10 –10 0 20 40 60 80 100 Iline (mA) The quality of the audio output of the loudspeaker amplifier is mainly determined by the distortion level. To keep high quality under difficult supply conditions, the MC33215 incorporates a peak–to–peak limiter. The peak–to–peak limiter will detect when the output stage gets close to its maximum output swing and will then reduce the gain from LSI to LSF. The attack and release of the limiter is regulated by the CPPL capacitor. Figure 12 depicts the limiter’s attack behavior with CPPL = 100 nF. The release time is given as 3 x CPPL x RPPL. In the typical application this leads to a release time of 300 ms. Figure 12. Peak–to–Peak Limiter Dynamic Behavior 0.5 V/DIV VLSO VPPL Vin 0 1.0 2.0 3.0 4.0 5.0 6.0 t, TIME (ms) Figure 12 clearly shows that due to the action of the peak–to–peak limiter, the output swing and thus the output power is reduced with respect to the maximum possible as already indicated in Figure 10. The peak–to–peak limiter can be disabled by connecting the PPL pin to ground. On top of the peak–to–peak limiter, the MC33215 incorporates a supply limiter, which reduces the gain rapidly when the supply voltage VLS drops too much. This will avoid malfunctioning of the amplifier and unwanted oscillations. The voltage drop is detected via the BVO input and for that reason the CBVO has to be connected to VLS and not to Gnd. The amplifier can be activated by making Pin LSM high. In the typical application this pin is connected to SPS, which activates the loudspeaker amplifier automatically when the speakerphone mode is entered. When LSM is made low, the 14 dA LSP (dB) 0 –15 –20 –25 –30 –35 –40 0 100 200 300 400 500 VVOL (mV), dALSP (dB) It can be seen from Figure 13 that a linear variation of RVOL will give a logarithmic gain reduction which adapts better to the human ear than a linear gain reduction. During DTMF dialing, see Table 2, a confidence tone is audible at the loudspeaker of which the level is proportional to the feedback resistor RLSF only. At RLSF = 180 kΩ the gain from MFI to LSO equals 28.5 dB. Half Duplex Controller To avoid howling during speakerphone operation, a half duplex controller is incorporated. By monitoring the signals in both the transmit and receive channel the duplex controller will reduce the gain in the channel containing the smallest signal. A typical gain reduction will be between 40 dB and 52 dB depending on the setting, see below. In case of equal signal levels or by detection of noise only, the circuit goes into idle mode. In this mode the gain reduction in both channels is halfway, leading to 20 dB to 26 dB of reduction. In a speakerphone built around the MC33215, following the signal path from base microphone to the line and via sidetone, loudspeaker and acoustic coupling back to the microphone, the loop gain can be expressed as a sum of the gains of the different stages. However, since the transmit and receive gains are dependent on AGC and the sidetone suppression is dependent on matching with the different lines we are mostly interested by the maximum possible loop gain ALOOP(max). It follows: ALOOP(max) = ABMRX(max) + ARXBM(max) – ASWD (dB) With: ABMRX(max) = Maximum gain from BM1 and BM2 to RXI as a function of line length AGC and line impedance matching ARXBM(max) = Maximum gain from RXI to BM1 and BM2 as a function of line length AGC and acoustic coupling MOTOROLA ANALOG IC DEVICE DATA MC33215 A SWD = Switching depth as performed in the attenuators To avoid howling, the maximum possible loop gain should be below 0 dB and preferably below –10 dB for comfort. In a practical telephone design, both the ABMRX(max) and the ARXBM(max) will be less than 20 dB thus a switching depth of 50 dB will give a loop gain of less than –10 dB. An optimized sidetone network is of high importance for handsfree operation. The better the network matches with the telephone line the less local feedback and the smaller the switching range can be. The amount of gain reduction ASWD obtained by the duplex controller is set via resistor RSWD according: 2 3.6 x R SWD A 20 log (dB) SWD R REF + ǒ Ǔ In the typical application the gain reduction will be 50 dB. To compare the transmit and receive signals with each other, they have to be monitored. This is done by making a signal envelope and a background noise envelope via the CTSE, CTBN capacitors for the transmit channel and via the CRSE, CRBN capacitors for the receive channel. In Figure 14, a schematic behavior of the envelopes is depicted which is equal for both transmit and receive. The voltage signal at the input is first transferred to a current via the sensitivity adjust network. Then this current is led through a diode which gives a logarithmic compression in voltage. It is this voltage from which the signal envelope is created by means of asymmetric charge and discharge of the signal envelope capacitor. The noise envelope voltage then follows in a similar way. Based on the envelope levels, the MC33215 will switch to transmit, receive or idle mode following Table 3. The fact that in receive mode the signal on the base microphone is greater than it is in case of transmit mode, due to the coupling of the high loudspeaker signal, is automatically taken into account. In the table, two particulars can be found. At first, the set will go to idle mode if the signals are not at least 4.5 dB greater then the noise floor, which calculates as a 13 mV voltage difference in envelopes. This avoids continuous switching over between the modes under slight variations of the background noise due to, for instance, typing on a keyboard. Second, a dial tone detector threshold is implemented to avoid that the set goes to idle mode in presence of a continuous strong receive signal like a dial tone. The dial tone detector threshold is proportional to the RRSA resistor. In the typical application with RRSA = 3.3 kΩ, the threshold is at 1.26 mVrms at the input RXI or 20 mVrms at the line. Line length AGC is of influence on the dial tone detector threshold, increasing the level depending on the line current with a maximum of 6.0 dB. In order to perform a correct comparison between the signal strengths, the sensitivity of the envelope detectors can be adjusted via the resistors connected to TSA and RSA. Based on the above, and on the fact that there is an effective gain of 20 dB in the transmit monitor, it can be derived that for stable operation the following two relations are valid: ǒ Ǔt ǒ Ǔ ǒ Ǔu ǒ Ǔ 20 log R TSA 20 log R 20 log R TSA RSA 20 log R – A ) – A RSA BMRX(max) – A ) 20 (dB) RXBM(max) 20 (dB) SW By measuring the gains and choosing the RRSA, the limits for RTSA follow. The choice for the sensitivity resistors is not completely free. The logarithmic compressors and the amplifier stages have a certain range of operation and, on the receive side, the choice for RRSA is given by the desired dial tone detector threshold. Figure 15 indicates the available dynamic range for the selected value of the sensitivity resistors. Figure 14. Signal and Noise Envelopes 1.8 V Internal VHF CTSE TSE VHF CTBN TBN Microphone Input Signal TSA RTSA CTSA MOTOROLA ANALOG IC DEVICE DATA 15 MC33215 Table 3. Logic Table for Switch–Over ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁ TSE > RSE TSE > TBN + 13 mV RSE > VDDT RSE > RBN + 13 mV 1 1 X X Transmit Mode 1 0 X X Idle 0 X 1 X Receive 0 X 0 1 Receive 0 X 0 0 Idle The electrical characteristics and the behavior of the MC33215 are not the only factor in designing a handsfree speakerphone. During the design the acoustics have to be taken into account from the beginning. The choice of the transducers and the design of the cabinet are of great influence on the speakerphone performance. Also, to achieve a proper handsfree operation, the fine tuning of the components around the duplex controller have to be done with the final choice of the cabinet and the transducers. The resistors for the sensitivity setting have to be coupled capacitively to the pins for dc decoupling, and also to create a high pass filter to suppress low frequent background noises like footsteps and 50 Hz. The switch–over timing is performed by charging and discharging the CSWT capacitor. The switch–over from transmit to receive or vice versa is fast, on the order of milliseconds, and is proportional to the value of CSWT. The switch–over to idle mode is slow, in the order of a few seconds, and is proportional to the product of the values of RSWT and CSWT. Figure 16 depicts a typical switch–over behavior when applying transmit and receive stimuli. Figure 15. Compression Range of the Signal Monitors 100.0E–3 100.0E–3 Upper Limit of Compression Dial Tone Threshold 1.0E–3 10.0E–3 Lower Limit of Compression VBM1 (Vrms) VRXI (Vrms) 10.0E–3 Upper Limit of Compression Lower Limit of Compression 1.0E–3 100.0E–6 10.0E–6 100 1000 10000 100000 100.0E–6 100 1000 10000 RRSA (Ω) RTSA (Ω) A. Receive Monitor B. Transmit Monitor 100000 Figure 16. Switch–Over Behavior Receive Transmit VMC + 0.5 SWT VMC – 0.5 16 MOTOROLA ANALOG IC DEVICE DATA MC33215 Figure 17. Test Circuit ZVDD 620 CVDD 100 µ VDD Gnd RREG 360 k VDD Zbal 33 k CREG 220 n VMC Supply MC33215 CHM1 33 n HM1 VHM Driver 1x HMIC 0.2 V CHM2 33 n CBM1 33 n BM1 MBM CTSE 330 n TBN VOL SLB VMF MUT VMUT MBM MDF MRX PRS Logic Control Block VPRS LSM MRA Attenuator Control VLSM MEAR SWT CRBN RBN 4.7 µ AGC Rx Log–Amp and Envelope Detectors Analog Control Block VSWT CRSE RSE 330 n VHF CRSA 470 n RSA RRSA 3.3 k MRX Rx Attenuator Rx CRXI 47 n AGC MRA VLS 2 RXI BVO CBVO 220 n CMF1 47 n VSPS RVOL 47 k V VRLS RSLP 220 SPS Tx Log–Amp and Envelope Detectors RLS Iline CVCC 470 µ DTMF AGC CTBN 4.7 µ RAGC 30 k AGC RREF 20 k REF RSWD 100 k SWD CVHF 47 µ MFI MHM VHF VCC MDF Tx Attenuator BMIC RTSA 2.2 k TSA TSE CVMC 10 µ SLP AGC CBM2 33 n BM2 CTSA 470 n VHF MHM HM2 VBM Supply 1:10 600 Vac RSLB 2.2 k VLN REG Vline V S1 1 VRXI MEAR VLS V VLSP 25 LSB CLSO 47 µ LSO RLSO PGD LSP RXO EAR Peak Limiter GRX CEAR 10 µ RRXO 180 k V VEAR 180 k LSF PPL RPPL 1.0 M CPPL 100 n LSI RXS RLSI 36 k CLSI 47 n CRXS 100 n RGRX 24 k CGRX 47 n VLSI MOTOROLA ANALOG IC DEVICE DATA 17 MC33215 Figure 18. Typical Application T1 ZVDD 620 CVDD 100 µ VDD Gnd RREG2 RREG1 365 k REG VDD VHF Supply 1:10 MC33215 VMC VCC CHM1 33 n HM1 RHM1 1.0 k Driver 0.2 V 10 µF RBM2 1.0 k MFI Ring Dialer or Microcontroller SPS AGC MUT MHM VHF TSE CTSE 330 n MBM Tx Log–Amp and Envelope Detectors MDF MRX TBN Logic Control Block MEAR AGC RREF 20 k REF RSWD 100 k SWD Attenuator Control AGC Analog Control Block PRS Rx Log–Amp and Envelope Detectors VOL VMC RBN CRBN 4.7 µ RSWT 2.2 M RSE CRSE 330 n Rx Attenuator Rx 6 7 8 9 * 0 # VHF CRSA 470 n RXI CRXI 33 n BVO CBVO 220 n 3 5 RRSA 3.3 k MRX RLS 2 4 Speakerphone Button CSWT SWT 100 n RSA RVOL 50 k VCC Privacy Button 1 LSM MRA CTBN 4.7 µ RAGC 30 k CMF1 47 n DTMF RTSA 470 TSA CTSA 1.0 µF RSLP 220 MDF Tx Attenuator BMIC VDD CVHF 47 µ Tip MBM CBM2 33 n BM2 T2 CVMC 10 µ SLB SLP AGC CBM1 33 n BM1 RBM1 1.0 k 1.0 k 1x HMIC RHM2 1.0 k VHF Hook Switch CVCC 470 µ MHM CHM2 33 n HM2 0.01 RSLB 2.2 k VLN VMC Supply Z1 10 V Zbal 33 k CREG 220 n 10 n AGC MRA VLS MEAR CEAR 10 µ LSB 25 Ω RXO CLSO 47 µ LSO RLSO 180 k PGD LSP EAR RRXO 180 k Peak Limiter LSF PPL RPPL 1.0 M CRLS 33 n 18 150 Ω GRX LSI CPPL 100 n RXS CRXS 100 n RGRX 24 k CGRX 47 n RLSI 36 k MOTOROLA ANALOG IC DEVICE DATA MC33215 OUTLINE DIMENSIONS FB SUFFIX PLASTIC PACKAGE CASE 848B–04 (TQFP–52) ISSUE C B L B 39 27 S D C A–B V F M 0.20 (0.008) B 0.20 (0.008) M L 0.05 (0.002) A–B –B– –A– –A–, –B–, –D– DETAIL A S S H A–B S DETAIL A D 26 40 J N 14 52 1 13 BASE METAL D –D– B 0.20 (0.008) M H A–B 0.02 (0.008) S D S V M C A–B S D S DETAIL C M_ C E –H– DATUM PLANE 0.10 (0.004) H –C– M_ G U_ R Q_ K T W X DETAIL C MOTOROLA ANALOG IC DEVICE DATA C A–B S D S SECTION B–B 0.05 (0.002) A–B 0.20 (0.008) M SEATING PLANE NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DATUM PLANE –H– IS LOCATED AT BOTTOM OF LEAD AND IS COINCIDENT WITH THE LEAD WHERE THE LEAD EXITS THE PLASTIC BODY AT THE BOTTOM OF THE PARTING LINE. 4. DATUMS –A–, –B– AND –D– TO BE DETERMINED AT DATUM PLANE –H–. 5. DIMENSIONS S AND V TO BE DETERMINED AT SEATING PLANE –C–. 6. DIMENSIONS A AND B DO NOT INCLUDE MOLD PROTRUSION. ALLOWABLE PROTRUSION IS 0.25 (0.010) PER SIDE. DIMENSIONS A AND B DO INCLUDE MOLD MISMATCH AND ARE DETERMINED AT DATUM PLANE –H–. 7. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.08 (0.003) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. DAMBAR CANNOT BE LOCATED ON THE LOWER RADIUS OR THE FOOT. DIM A B C D E F G H J K L M N Q R S T U V W X MILLIMETERS MIN MAX 9.90 10.10 9.90 10.10 2.10 2.45 0.22 0.38 2.00 2.10 0.22 0.33 0.65 BSC ––– 0.25 0.13 0.23 0.65 0.95 7.80 REF 5_ 10_ 0.13 0.17 0_ 7_ 0.13 0.30 12.95 13.45 0.13 ––– 0_ ––– 12.95 13.45 0.35 0.45 1.6 REF INCHES MIN MAX 0.390 0.398 0.390 0.398 0.083 0.096 0.009 0.015 0.079 0.083 0.009 0.013 0.026 BSC ––– 0.010 0.005 0.009 0.026 0.037 0.307 REF 5_ 10_ 0.005 0.007 0_ 7_ 0.005 0.012 0.510 0.530 0.005 ––– 0_ ––– 0.510 0.530 0.014 0.018 0.063 REF 19 MC33215 OUTLINE DIMENSIONS B SUFFIX PLASTIC PACKAGE CASE 858–01 (SDIP–42) ISSUE O NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: INCH. 3. DIMENSION L TO CENTER OF LEAD WHEN FORMED PARALLEL. 4. DIMENSIONS A AND B DO NOT INCLUDE MOLD FLASH. MAXIMUM MOLD FLASH 0.25 (0.010). –A– 42 22 –B– 1 21 DIM A B C D F G H J K L M N L H C –T– SEATING PLANE 0.25 (0.010) N G F D 42 PL K M T A M J 42 PL S 0.25 (0.010) M T B INCHES MIN MAX 1.435 1.465 0.540 0.560 0.155 0.200 0.014 0.022 0.032 0.046 0.070 BSC 0.300 BSC 0.008 0.015 0.115 0.135 0.600 BSC 0_ 15 _ 0.020 0.040 MILLIMETERS MIN MAX 36.45 37.21 13.72 14.22 3.94 5.08 0.36 0.56 0.81 1.17 1.778 BSC 7.62 BSC 0.20 0.38 2.92 3.43 15.24 BSC 0_ 15_ 0.51 1.02 S Motorola reserves the right to make changes without further notice to any products herein. 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