Continuous Rate 12.3Mb/s to 2.7Gb/s Clock and Data Recovery IC w/Loop Timed SERDES ADN2865 Preliminary Technical Data FEATURES PRODUCT DESCRIPTION Serial data input: 12.3 Mb/s to 2.7 Gb/s Exceeds ITU-T Jitter Specifications Integrated Limiting Amp: 6mV sensitivity Adjustable slice level: ±100 mV Patented dual-loop clock recovery architecture Programmable LOS detect and Slice Level Integrated PRBS Generator and Detector No reference clock required Loss of lock indicator Rate Selectivity without the use of a reference clock I2C™ interface to access optional features Single-supply operation: 3.3 V Low power: 1.0W 8 mm × 8 mm 56-lead LFCSP The ADN2865 provides the receiver functions of quantization, signal level detect, and clock and data recovery for continuous data rates from 12.3 Mb/s to 2.7 Gb/s. An integrated deserialiser supports 8 bit parallel transfer to an FPGA or digital ASIC. The recovered clock can simultaneously serialize data supplied in an 8 bit parallel format. APPLICATIONS Passive Optical Network s SONET OC-1/3/12/48 and all associated FEC rates Fibre Channel, 2× Fibre Channel , GbE, HDTV, etc. WDM transponders Test equipment The ADN2865 automatically locks to all data rates without the need for an external reference clock or programming. All SONET jitter requirements are exceeded, including jitter transfer, jitter generation, and jitter tolerance. All specifications are quoted for −40°C to +85°C ambient temperature, unless otherwise noted. This device, together with a PIN diode and a TIA preamplifier, can implement a highly integrated, low cost, low power fiber optic receiver. The ADN2865 have many optional features available via an I2C interface, e.g. the user can read back the data rate that the ADN2865 is locked on to, or the user can set the device to only lock to one particular data rate if provisioning of data rates is required. FUNCTIONAL BLOCK DIAGRAM Figure 1 ADN2865 Functional Block Diagram Rev.PrA Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.326.8703 © 2006 Analog Devices, Inc. All rights reserved. ADN2865 Preliminary Technical Data TABLE OF CONTENTS Specifications..................................................................................... 3 Functional Description.................................................................. 20 Jitter Specifications....................................................................... 4 Frequency Acquisition............................................................... 20 Output and Timing Specifications ............................................. 5 Limiting Amplifier ..................................................................... 21 Absolute Maximum Ratings............................................................ 6 Slice Adjust.................................................................................. 22 Thermal Characteristics .............................................................. 7 Loss of Signal (LOS) Detector .................................................. 22 ESD Caution.................................................................................. 7 Lock Detector Operation .......................................................... 20 Timing Characteristics..................................................................... 9 Harmonic Detector .................................................................... 21 Pin Configuration and Function Descriptions........................... 10 Squelch Mode .......................... Error! Bookmark not defined. Typical Performance Characteristics ............................................. 8 I2C Interface ............................................................................... 22 I2C Interface Timing and Internal Register Description........... 12 Reference Clock (Optional) ...................................................... 23 Terminology .................................................................................... 16 Applications Information .............................................................. 26 Jitter Specifications ......................................................................... 17 PCB Design Guidelines ............................................................. 26 Jitter Generation ......................................................................... 17 DC-Coupled Application .......................................................... 28 Jitter Transfer............................................................................... 17 Coarse Data Rate Readback Look-Up Table............................... 30 Jitter Tolerance ............................................................................ 17 Outline Dimensions ....................................................................... 32 Theory of Operation ...................................................................... 18 Ordering Guide .......................................................................... 32 REVISION HISTORY Revision 0: Initial Version Revision A: Remove Minimum Supply Current Spec Revision B: Update spec table Rev. PrA | Page 2 of 33 Preliminary Technical Data ADN2865 SPECIFICATIONS TA = TMIN to TMAX, VCC = VMIN to VMAX, VEE = 0 V, CF = 0.47 μF, SLICEP = SLICEN = VEE, Input Data Pattern: PRBS 223 − 1, unless otherwise noted. Table 1. Parameter QUANTIZER—DC CHARACTERISTICS Input Voltage Range Peak-to-Peak Differential Input Input Common Mode Level Differential Input Sensitivity Input Overdrive Input Offset Input RMS Noise QUANTIZER—AC CHARACTERISTICS Data Rate S11 Input Resistance Input Capacitance QUANTIZER—SLICE ADJUSTMENT Gain Differential Control Voltage Input Control Voltage Range Slice Threshold Offset LOSS OF SIGNAL DETECT (LOS) Loss of Signal Detect Range (see Figure 2) Hysteresis (Electrical) LOS Assert Time LOS De-Assert Time LOSS OF LOCK DETECT (LOL) VCO Frequency Error for LOL Assert VCO Frequency Error for LOL De-Assert LOL Response Time ACQUISITION TIME Lock to Data Mode Conditions Min @ PIN or NIN, dc-coupled PIN – NIN DC-coupled (see Figure , Figure , and Figure ) 223 − 1 PRBS, ac-coupled,1 BER = 1 x 10–10 (see Figure ) 1.8 2 2.3 2.5 TBD TBD TBD TBD TBD TBD BER = 1 x 10–10 12.3 Max Unit 2.8 2.0 2.8 V V V mV p-p mV p-p μV μV rms 2700 Mb/s dB Ω pF TBD TBD 0.95 V/V V V mV TBD TBD TBD TBD mV mV TBD TBD TBD TBD dB dB TBD TBD TBD TBD TBD TBD dB dB ns ns With respect to nominal With respect to nominal 12.3 Mb/s OC-12 OC-48 1000 250 4 1.0 1.0 ppm ppm ms μs μs OC-48 OC-12 OC-3 OC-1 12.3 Mb/s 1.3 2.0 3.4 9.8 40.0 10.0 ms ms ms ms ms ms @ 2.5 GHz Differential SLICEP – SLICEN = ±0.5 V SLICEP – SLICEN DC level @ SLICEP or SLICEN −15 100 0.65 TBD 0.1 VEE 1 RThresh = 0 Ω RThresh = 100 kΩ OC-48 RThresh = 0 Ω RThresh = 100 kΩ OC-1 RThresh = 0 Ω RThresh = 10 kΩ DC-coupled2 DC-coupled2 Optional Lock to REFCLK Mode 1 Typ PIN and NIN should be differentially driven and ac-coupled for optimum sensitivity. When ac-coupled, the LOS assert and de-assert time is dominated by the RC time constant of the ac coupling capacitor and the 50 Ω input termination of the ADN2865 input stage. Rev. PrA | Page 3 of 33 ADN2865 Parameter DATA RATE READBACK ACCURACY Coarse Readback Fine Readback Preliminary Technical Data Conditions Min (See Table ) In addition to REFCLK accuracy Data rate < 20 Mb/s Data rate > 20 Mb/s POWER SUPPLY VOLTAGE POWER SUPPLY CURRENT OPERATING TEMPERATURE RANGE Typ Max 10 3.0 3.3 300 –40 Unit % 200 100 3.6 350 +85 ppm ppm V mA °C JITTER SPECIFICATIONS TA = TMIN to TMAX, VCC = VMIN to VMAX, VEE = 0 V, CF = 0.47 uF, SLICEP = SLICEN = VEE, Input Data Pattern: PRBS 223 − 1, unless otherwise noted. Table 2. Parameter PHASE-LOCKED LOOP CHARACTERISTICS Jitter Transfer BW Jitter Peaking Jitter Generation Conditions Min OC-48 OC-12 OC-3 OC-48 OC-12 OC-3 OC-48, 12 kHz to 20 MHz OC-12, 12 kHz to 5 MHz OC-3, 12 kHz to 1.3 MHz Jitter Tolerance Power Supply Rejection OC-48, 223 − 1 PRBS 100 kHz 1 MHz 20 MHz OC-12, 223 − 1 PRBS 25 kHz 250 kHzError! Bookmark not defined. OC-3, 223 − 1 PRBS 6500 Hz 65 kHz See Figure XX. Rev. PrA | Page 4 of 33 Typ Max Unit 0 0 0 TBD TBD TBD TBD TBD TBD 2000 500 130 0.1 0.1 0.1 TBD 0.33 TBD 0.2 TBD 0.2 kHz kHz kHz dB dB dB UI rms UI p-p UI rms UI p-p UI rms UI p-p 0.75 0.075 0.075 UI p-p UI p-p UI p-p 0.75 0.075 UI p-p UI p-p 0.75 0.075 UI p-p UI p-p dB TBD Preliminary Technical Data ADN2865 OUTPUT AND TIMING SPECIFICATIONS Table 3. Parameter CML OUPUT CHARACTERISTICS Single-Ended Output Swing Differential Output Swing Output High Voltage Output Low Voltage CML Ouput Timing Rise Time Fall Time LVDS OUPUT CHARACTERISTICS (RXCLKP/N, RXDATP/N) Differential Output Swing Output High Voltage Output Low Voltage Output Offset Voltage Output Impedance LVDS Ouputs Timing Rise Time Fall Time Setup Time Hold Time I2C INTERFACE DC CHARACTERISTICS Input High Voltage Input Low Voltage Input Current Output Low Voltage I2C INTERFACE TIMING SCK Clock Frequency SCK Pulse Width High SCK Pulse Width Low Start Condition Hold Time Start Condition Setup Time Data Setup Time Data Hold Time SCK/SDA Rise/Fall Time Stop Condition Setup Time Bus Free Time between a Stop and a Start REFCLK CHARACTERISTICS Input Voltage Range Minimum Differential Input Drive Reference Frequency Required Accuracy LVTTL DC INPUT CHARACTERISTICS Input High Voltage Input Low Voltage Input High Current 1 Conditions Min Typ Max Unit VSE (see Figure 75) VDIFF (see Figure 75) VOH VOL 300 600 350 700 VCC − 0.6 VCC − 0.35 600 1200 VCC VCC − 0.3 mV mV V V TBD TBD ps ps 320 400 1475 1200 100 1275 mV mV V V Ω 115 115 220 220 20% to 80% 80% to 20% VDIFF (see Figure 4) VOH VOL VOS Differential 20% to 80% 80% to 20% TS (see Figure 4), OC-48 TH (see Figure 4), OC-48 LVCMOS VIH VIL VIN = 0.1 VCC or VIN = 0.9 VCC VOL, IOL = 3.0 mA (See Figure ) 250 925 1125 2.61 -1.70 0.7 VCC 0.3 VCC +10.0 0.4 −10.0 400 tHIGH tLOW tHD;STA tSU;STA tSU;DAT tHD;DAT TR/TF tSU;STO tBUF Optional lock to REFCLK mode @ REFCLKP or REFCLKN VIL VIH 600 1300 600 600 100 300 20 + 0.1 Cb1 600 1300 300 0 VCC 100 12.3 200 100 VIH VIL IIH, VIN = 2.4 V 2.0 Cb = total capacitance of one bus line in pF. If mixed with Hs-mode devices, faster fall-times are allowed (see Table 6). Rev. PrA | Page 5 of 33 0.8 5 ps ps ns ns V V μA V kHz ns ns ns ns ns ns ns ns ns V V mV p-p MHz ppm V V μA ADN2865 Parameter Input Low Current LVTTL INPUT TIMING Setup Time (Sync Mode) Hold Time (Sync Mode) Setup Time (Align Mode) Hold Time (Align Mode) LVTTL DC OUTPUT CHARACTERISTICS Output High Voltage Output Low Voltage Preliminary Technical Data Conditions IIL, VIN = 0.4 V Min −5 TSSU (see Figure 3), 1.25Gb/s TSH (see Figure 3), 1.25Gb/s TASU (see Figure 4), 1.25Gb/s TAH (see Figure 4), 1.25Gb/s 3.60 0.70 TBD TBD VOH, IOH = −2.0 mA VOL, IOL = 2.0 mA 2.4 Rev. PrA | Page 6 of 33 Typ Max Unit μA ns ns ns ns 0.4 V V Preliminary Technical Data ADN2865 ABSOLUTE MAXIMUM RATINGS TA = TMIN to TMAX, VCC = VMIN to VMAX, VEE = 0 V, CF = 0.47 μF, SLICEP = SLICEN = VEE, unless otherwise noted. Table 4. Parameter Supply Voltage (VCC) Minimum Input Voltage (All Inputs) Maximum Input Voltage (All Inputs) Maximum Junction Temperature Storage Temperature Lead Temperature (Soldering 10 s) Rating 4.2 V VEE − 0.4 V VCC + 0.4 V 125°C −65°C to +150°C 300°C Stress above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. THERMAL CHARACTERISTICS Thermal Resistance 56-LFCSP, 4-layer board with exposed paddle soldered to VEE θJA = 28°C/W ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Rev. PrA | Page 7 of 33 ADN2865 Preliminary Technical Data TYPICAL PERFORMANCE CHARACTERISTICS 1000 16 ADN2812 TOLERANCE SONET REQUIREMENT MASK SONET OBJECTIVE MASK EQUIPMENT LIMIT 10 8 6 4 2 1 10 100 1k 10k 100 10 1 04228-0-030 JITTER AMPLITUDE (UI) 12 04228-0-003 TRIP POINT (mV p-p) 14 0.1 1 100k RTH (Ω) Figure 2. LOS Comparator Trip Point Programming 10 100 1k 10k 100k 1M JITTER FREQUENCY (Hz) 10M Figure 3. Typical Measured Jitter Tolerance OC-48 Rev. PrA | Page 8 of 33 100M Preliminary Technical Data ADN2865 TIMING CHARACTERISTICS Figure 4. Rx Output Timing RXCLKP/ N Tssu Tsh TXDATAP / N[ 7:0] 04228-0-002 Figure 5. Tx Input Timing (Sync Mode) Figure 6. Tx Input Timing Align Mode) OUTP VCML VSE OUTN OUTP–OUTN VDIFF 04228-0-004 VSE 0V Figure 75. Single-Ended vs. Differential Output Specifications Rev. PrA | Page 9 of 33 TXDAT4 TXCLK TXDAT3 TXDAT2 TXDAT1 TXDAT0 VREG CF1 VEE VCC VEE VCC REFN REFP 15 16 17 18 19 20 21 22 23 24 25 26 27 28 56 55 54 53 52 51 50 49 48 47 46 45 44 43 RXDATP2 RXDATN2 RXDATP3 RXDATN3 RXDATP4 RXDATN4 VCC RXDATP5 RXDATN5 RXDATP6 RXDATN6 VCC RXDATP7 RXDATN7 ADN2865 Preliminary Technical Data PIN CONFIGURATION AND FUNCTION DESCRIPTIONS Figure 86. Pin Configuration Rev. PrA | Page 10 of 33 Preliminary Technical Data ADN2865 Table 5. Pin Function Descriptions Pin # Mnemonic Type Description 1 SDOUT DO Active high, Loss of signal indicator. (LVTTL) 2 RXDATN1 DO Differential receive data output. (LVDS) 3 RXDATP1 DO 4 RXDATN0 DO Differential receive data output. (LVDS) Differential receive data output. Last bit received. (LVDS) 5 RXDATP0 DO Differential receive data output. Last bit received. (LVDS) 6 VCC3 PWR Power for CDR & Serialiser 7 VCC7 PWR Power for CML drivers 8 SERDATN DO Differential serialized data output to LDD. (CML) 9 SERDATP DO Differential serialized data output to LDD. (CML) 10 SERCLKN DO Differential clock for serialized Tx data. (CML) 11 SERCLKP DO 12 TXDAT7 DI Differential clock for serialized Tx data. (CML) Transmit data input. First bit sent. (LVTTL) 13 TXDAT6 DI Transmit data input. (LVTTL) 14 TXDAT5 DI Transmit data input. (LVTTL) 15 TXDAT4 DI Transmit data input. (LVTTL) 16 TXCLK DI Qualifying clock for transmit data input. (LVTTL) 17 TXDAT3 DI Transmit data input. (LVTTL) 18 TXDAT2 DI Transmit data input. (LVTTL) 19 TXDAT1 DI 20 TXDAT0 DI Transmit data input. (LVTTL) Transmit data input. Last bit sent. (LVTTL) 21 VREG AO Decoupling node for VCO power. 22 CF1 AO PLL loop filter capacitor. 23 VEE2 PWR Ground for VCO / PLL / Gm 24 VCC2 PWR Power for VCO / PLL / Gm 25 VEE4 PWR Ground for FLL 26 VCC4 PWR Power for FLL 27 REFN DI Reference clock input. (LVDS/LVTTL) 28 REFP DI Reference clock input. (LVDS/LVTTL) 29 THRADJ AO LOS Threshold Setting Resistor. 30 LOL DO Active high, Loss-of-Lock Indicator. (LVTTL) 31 VEE1 PWR Ground for Limamp / LOS 32 SLICEN AI Differential Slice Level Adjust Input. 33 SLICEP AI Differential Slice Level Adjust Input. 34 NIN AI Differential serial input to Limiting Amp. (CML) 35 PIN AI Differential serial input to Limiting Amp. (CML) 36 VREF AO Decoupling node for internal voltage reference. 37 VCC1 PWR Power for Limamp / LOS 38 SCK DI I2C Serial Clock Input. 39 VCC6 PWR Power for Deserialiser, LVDS pre-drivers 40 SDA DI I2C Serial Data Input. 41 RXCLKN DO Qualifying clock for Rx Data Outputs. (LVDS) 42 RXCLKP DO 43 RXDATN7 DO Qualifying clock for Rx Data Outputs. (LVDS) Differential receive data output. Last bit received. (LVDS) 44 RXDATP7 DO Differential receive data output. Last bit received. (LVDS) 45 VCC5 PWR Power for LVDS drivers 46 RXDATN6 DO Differential receive data output. (LVDS) 47 RXDATP6 DO Differential receive data output. (LVDS) 48 RXDATN5 DO Differential receive data output. (LVDS) 49 RXDATP5 DO Differential receive data output. (LVDS) 50 VCC5 PWR Power for LVDS Drivers 51 RXDATN4 DO Differential receive data output. (LVDS) 52 RXDATP4 DO Differential receive data output. (LVDS) 53 RXDATN3 DO Differential receive data output. (LVDS) 54 RXDATP3 DO Differential receive data output. (LVDS) 55 RXDATN2 DO Differential receive data output. (LVDS) 56 RXDATP2 DO Differential receive data output. (LVDS) 1 Type: P = power, AI = analog input, AO = analog output, DI = digital input, DO = digital output. Rev. PrA | Page 11 of 33 ADN2865 Preliminary Technical Data I2C INTERFACE TIMING AND INTERNAL REGISTER DESCRIPTION R/W CTRL. SLAVE ADDRESS [6...0] 1 1 0 0 0 0 0 X 0 = WR 1 = RD MSB = 1 S SLAVE ADDR, LSB = 0 (WR) A(S) SUB ADDR A(S) DATA A(S) DATA A(S) P 04228-0-006 Figure 9. Slave Address Configuration S SLAVE ADDR, LSB = 0 (WR) A(S) SUB ADDR A(S) S SLAVE ADDR, LSB = 1 (RD) A(S) DATA A(M) S = START BIT A(S) = ACKNOWLEDGE BY SLAVE DATA A(M) P P = STOP BIT A(M) = LACK OF ACKNOWLEDGE BY MASTER A(M) = ACKNOWLEDGE BY MASTER 04228-0-007 Figure 10. I2C Write Data Transfer Figure 11. I2C Read Data Transfer SDA SLAVE ADDRESS A6 SUB ADDRESS A5 STOP BIT DATA A7 A0 D7 D0 SCK S WR ACK ACK SLADDR[4...0] ACK SUB ADDR[6...1] DATA[6...1] Figure 12. I2C Data Transfer Timing tF tSU;DAT tHD;STA tBUF SDA tR tR tSU;STO tF tLOW tHIGH tHD;STA S tSU;STA S tHD;DAT Figure 13. I2C Port Timing Diagram Rev. PrA | Page 12 of 33 P S 04228-0-009 SCK P 04228-0-008 START BIT Preliminary Technical Data ADN2865 Table 6. Internal Register Map1 Reg Name R/W ADDRES S D7 FREQ0 R 0x0 MSB LSB FREQ1 R 0x1 MSB LSB FREQ2 R 0x2 0 RATE R 0x3 MISC R 0x4 CTRLA W 0x8 CTRLA_R D R 0x5 CTRLB W 0x9 CTRLB_R D R 0x6 CTRLC W 0x11 0 0 Set Signal Degrade Threshold Enable Signal Degrade LOS forces acquisition CTRLD W 0x22 CDR Bypass Power Down LVDS Drivers Power Down CML Drivers Squelch Output Buffers Initiate PBS Sequence CTRLE W 0x27 Align TX Mode FDDI_MO DE W 0x0D FDDI Mode Enable SEL_MOD E W 0x34 0 HI_CODE W 0x35 HI_CODE[8] HI_CODE[1] LO_CODE W 0x36 LO_CODE[8] LO_CODE[1] CODE_LSB W 0x39 1 D6 D5 D4 D3 D2 D1 D0 MSB LSB COARSE_RD[8] Coarse Data Readback X X Los Status Fref Range Static LOL LOL Status COARSE_RD[1] Datarate meas complete X COARSE_RD[ 0] LSB Measure Data Rate Lock to Reference 0 0 0 Config LOS Squelch Mode Boost Output Data Rate/DIV FREF Ratio Readback CTRLA Config LOL Reset MISC[4] System Reset 0 Reset MISC[2] Readback CTRLB RXCLK Alignment 0 PRBS Mode[2:0] Reverse RX Bus Reverse TX Bus 0 0 0 CLK Holdover Mode 2B 0 Subharmonic Ratio 0 0 Acq Mode Cont Rate / Single Rate 0 0 Datarate Range 0 CLK Holdover Mode 2A 0 HI_CODE[0] LO_CODE[0] All writeable registers default to 0x00. Table 7. Miscellaneous Register, MISC D7 x D6 x LOS Status D5 0 = No loss of signal 1 = Loss of signal Static LOL D4 0 = Waiting for next LOL 1 = Static LOL until reset LOL Status D3 0 = Locked 1 = Acquiring Rev. PrA | Page 13 of 33 Datarate Measurement Complete D2 0 = Measuring datarate 1 = Measurement complete D1 x Coarse Rate Readback LSB D0 COARSE_RD[0] ADN2865 Preliminary Technical Data Table 8. Control Register, CTRLA1 FREF Range D7 D6 0 0 0 1 1 0 1 1 1 12.3 MHz to 25 MHz 25 MHz to 50 MHz 50 MHz to 100 MHz 100 MHz to 200 MHz Datarate/Div_FREF Ratio D5 D4 D3 D2 0 0 0 0 0 0 0 1 0 0 1 0 n 1 0 0 0 Measure Datarate D1 Set to 1 to measure datarate 1 2 4 2n 256 Lock to Reference D0 0 = Lock to input data 1 = Lock to reference clock Where DIV_FREF is the divided down reference referred to the 12.3 MHz to 25 MHz band (see the Reference Clock (Optional) section). Table 9. Control Register, CTRLB Config LOL D7 0 = LOL pin normal operation 1 = LOL pin is static LOL Reset MISC[4] D6 Write a 1 followed by 0 to reset MISC[4] System Reset D5 Write a 1 followed by 0 to reset ADN2865 D4 Set to 0 Reset MISC[2] D3 Write a 1 followed by 0 to reset MISC[2] D2 Set to 0 D1 Set to 0 D0 Set to 0 Table 10. Control Register, CTRLC D7 Set to 0 D6 Set to 0 Signal Degrade Threshold D5 0=Set SD Threshold to 9mV 1=Set SD Threshold to 1.9x LOS Threshold Signal Degrade Mode D4 0= Disable Signal Degrade Mode 1= Enable Signal Degrade Mode D3 Set to 0 Config LOS D2 0 = Active high LOS 1 = Active low LOS SERCLK D1 0 = Power Down SERCLK buffer 1 = Enable SERCLK buffer D0 Set to 0 Table 11. Control Register, CTRLD CDR Bypass D7 0=CDR Enabled 1=CDR Disabled Buffer Control D6 0=Normal operation 1=Power Down LVDS drivers D5 0=Normal operation 1=Power Down CML drivers D4 0 Initiate PRBS D3 Write a 1 followed by 0 to initiate PRBS Generate Sequence D2 0 0 0 D1 0 0 1 PRBS Mode D0 0 Power Down Generate Mode 1 Detect Mode, 0 compares errors Table 12. Control Register, CTRLE RXCLK Alignment to RXDATA D7 D6 D5 0 0 0 0 +1 UI 1 0 0 0 0 1 0 -3 UI 1 1 0 +4 UI 0 0 1 +2 UI 1 0 1 +4 UI 0 1 1 -2 UI 1 1 1 Tx Mode D4 0=Sync Mode 1=Align D3 0=Enable align 1=Disable align Rev. PrA | Page 14 of 33 Bus Reversal D2 RXDATA[7:0] 0=Bit 0 is last received 1=Bit 7 is last received D1 TXDATA[7:0] 0=Bit 0 is last sent 1=Bit 7 is last sent D0 Set To 0 Preliminary Technical Data ADN2865 Table 13. FDDI_MODE FDDI Enable D7 0= FDDI Mode Disabled 1= FDDI Mode Enabled Subharmonic Ratio [6..2] D6 D5 D4 D3 D2 0 0 0 0 1 0 0 0 1 0 0 0 0 1 1 --------------------1 1 1 1 1 = 1 = 2 = 3 D1 Set to 0 D0 Set to 0 = 31 Table 14. SEL_MODE D7 Set to 0 D6 Set to 0 Mode Control 2 D5 0=LTD/LTR Mode 1=LTR Mode Only Mode Control 1 D4 0= Continuous Rate 1= Single Rate Mode Control 0 D3 0= Full Range (12.3M2.7G) 1= Limited Range Clock Holdover Mode 2A D2 Set to 1 for Clock Holdover Mode 2A Clock Holdover Mode 2B D1 Set to 1 for Clock Holdover Mode 2B D0 Set to 0 Table 15. CTLF D7 Set to 0 D6 Set to 0 D5 Set to 0 D4 Set to 0 D3 Set to 0 Rev. PrA | Page 15 of 33 CDR Mode D2 D1 D0 0 0 0 = NDC -> OB 1 0 1 = NDC -> PRBS ADN2865 Preliminary Technical Data TERMINOLOGY 10mVp-p Input Sensitivity and Input Overdrive VREF Sensitivity and overdrive specifications for the quantizer involve offset voltage, gain, and noise. The relationship between the logic output of the quantizer and the analog voltage input is shown in Figure . For sufficiently large positive input voltage, the output is always Logic 1 and, similarly for negative inputs, the output is always Logic 0. However, the transitions between output Logic Levels 1 and 0 are not at precisely defined input voltage levels, but occur over a range of input voltages. Within this range of input voltages, the output might be either 1 or 0, or it might even fail to attain a valid logic state. The width of this zone is determined by the input voltage noise of the quantizer. The center of the zone is the quantizer input offset voltage. Input overdrive is the magnitude of signal required to guarantee the correct logic level with 1 × 10−10 confidence level. OUTPUT 0 OFFSET PIN + Quantizer 50 50 VREF 2.5V 3k Figure 15. Single-Ended Sensitivity Measurement Driving the ADN2865 differentially (see Figure ), sensitivity seems to improve from observing the quantizer input with an oscilloscope probe. This is an illusion caused by the use of a single-ended probe. A 5 mV p-p signal appears to drive the ADN2865 quantizer. However, the single-ended probe measures only half the signal. The true quantizer input signal is twice this value, because the other quantizer input is a complementary signal to the signal being observed. NOISE 1 scope probe INPUT (V p-p) 5mV p-p SCOPE PROBE OVERDRIVE 04228-0-010 SENSITIVITY (2× OVERDRIVE) VREF PIN Figure 14. Input Sensitivity and Input Overdrive + QUANTIZER NIN AC coupling is typically used to drive the inputs to the quantizer. The inputs are internally dc biased to a commonmode potential of ~2.5 V. Driving the ADN2865 single-ended and observing the quantizer input with an oscilloscope probe at the point indicated in Figure shows a binary signal with an average value equal to the common-mode potential and instantaneous values both above and below the average value. It is convenient to measure the peak-to-peak amplitude of this signal and call the minimum required value the quantizer sensitivity. Referring to Figure , because both positive and negative offsets need to be accommodated, the sensitivity is twice the overdrive. The ADN2865 quantizer typically has 6 mV p-p sensitivity. – 50Ω VREF 50Ω VREF 5mV p-p 2.5V 3kΩ 04228-0-012 Single-Ended vs. Differential Figure 16. Differential Sensitivity Measurement LOS Response Time LOS response time is the delay between removal of the input signal and indication of loss of signal (LOS) at the LOS output, Pin 22. When the inputs are dc-coupled, the LOS assert time of the AD2817 is 500 ns typically and the de-assert time is 400 ns typically,. In practice, the time constant produced by the ac coupling at the quantizer input and the 50 Ω on-chip input termination determines the LOS response time. Rev. PrA | Page 16 of 33 Preliminary Technical Data ADN2865 JITTER SPECIFICATIONS JITTER GENERATION The jitter generation specification limits the amount of jitter that can be generated by the device with no jitter and wander applied at the input. For OC-48 devices, the band-pass filter has a 12 kHz high-pass cutoff frequency with a roll-off of 20 dB/decade, and a low-pass cutoff frequency of at least 20 MHz. The jitter generated must be less than 0.01 UI rms, and must be less than 0.1 UI p-p. JITTER TRANSFER The jitter transfer function is the ratio of the jitter on the output signal to the jitter applied on the input signal versus the frequency. This parameter measures the limited amount of the jitter on an input signal that can be transferred to the output signal (see Figure ). SLOPE = –20dB/DECADE 04228-0-013 ACCEPTABLE RANGE fC JITTER FREQUENCY (kHz) Figure 17. Jitter Transfer Curve JITTER TOLERANCE The jitter tolerance is defined as the peak-to-peak amplitude of the sinusoidal jitter applied on the input signal, which causes a 1 dB power penalty. This is a stress test intended to ensure that no additional penalty is incurred under the operating conditions (see Figure ). 15.00 SLOPE = –20dB/DECADE 1.50 0.15 f0 f1 f2 f3 JITTER FREQUENCY (kHz) Figure 18. SONET Jitter Tolerance Mask Rev. PrA | Page 17 of 33 f4 04228-0-014 The following sections briefly summarize the specifications of jitter generation, transfer, and tolerance in accordance with the Telcordia document (GR-253-CORE, Issue 3, September 2000) for the optical interface at the equipment level and the ADN2865 performance with respect to those specifications. JITTER GAIN (dB) Jitter is the dynamic displacement of digital signal edges from their long-term average positions, measured in unit intervals (UI), where 1 UI = 1 bit period. Jitter on the input data can cause dynamic phase errors on the recovered clock sampling edge. Jitter on the recovered clock causes jitter on the retimed data. 0.1 INPUT JITTER AMPLITUDE (UI p-p) The ADN2865 CDR is designed to achieve the best bit-errorrate (BER) performance and exceeds the jitter transfer, generation, and tolerance specifications proposed for SONET/SDH equipment defined in the Telcordia Technologies specification. ADN2865 Preliminary Technical Data THEORY OF OPERATION Another view of the circuit is that the phase shifter implements the zero required for frequency compensation of a second-order phase-locked loop, and this zero is placed in the feedback path and, thus, does not appear in the closed-loop transfer function. Jitter peaking in a conventional second-order phase-locked loop is caused by the presence of this zero in the closed-loop transfer function. Because this circuit has no zero in the closed-loop transfer, jitter peaking is minimized. The delay- and phase-loops together simultaneously provide wide-band jitter accommodation and narrow-band jitter filtering. The linearized block diagram in Figure shows that the jitter transfer function, Z(s)/X(s), is a second-order low-pass providing excellent filtering. Note that the jitter transfer has no zero, unlike an ordinary second-order phase-locked loop. This means that the main PLL loop has virtually zero jitter peaking (see Figure ). This makes this circuit ideal for signal regenerator applications, where jitter peaking in a cascade of regenerators can contribute to hazardous jitter accumulation. The error transfer, e(s)/X(s), has the same high-pass form as an ordinary phase-locked loop. This transfer function is free to be optimized to give excellent wide-band jitter accommodation, because the jitter transfer function, Z(s)/X(s), provides the narrow-band jitter filtering. INPUT DATA X(s) e(s) o/s d/sc 1/n Z(s) RECOVERED CLOCK d = PHASE DETECTOR GAIN o = VCO GAIN c = LOOP INTEGRATOR psh = PHASE SHIFTER GAIN n = DIVIDE RATIO JITTER TRANSFER FUNCTION Z(s) 1 = X(s) n psh cn +1 +s s2 o do 04228-0-015 TRACKING ERROR TRANSFER FUNCTION e(s) s2 = d psh do X(s) + s2 + s c cn Figure 19. ADN2865 PLL/DLL Architecture JITTER PEAKING IN ORDINARY PLL ADN2812 Z(s) X(s) o n psh d psh c 04228-0-016 The delay- and phase-loops together track the phase of the input data signal. For example, when the clock lags input data, the phase detector drives the VCO to higher frequency, and also increases the delay through the phase shifter; both these actions serve to reduce the phase error between the clock and data. The faster clock picks up phase, while the delayed data loses phase. Because the loop filter is an integrator, the static phase error is driven to zero. psh JITTER GAIN (dB) The ADN2865 is a delay- and phase-locked loop circuit for clock recovery and data retiming from an NRZ encoded data stream. The phase of the input data signal is tracked by two separate feedback loops, which share a common control voltage. A high speed delay-locked loop path uses a voltage controlled phase shifter to track the high frequency components of input jitter. A separate phase control loop, comprised of the VCO, tracks the low frequency components of input jitter. The initial frequency of the VCO is set by yet a third loop, which compares the VCO frequency with the input data frequency and sets the coarse tuning voltage. The jitter tracking phase-locked loop controls the VCO by the fine-tuning control. FREQUENCY (kHz) Figure 20. ADN2865 Jitter Response vs. Conventional PLL The delay- and phase-loops contribute to overall jitter accommodation. At low frequencies of input jitter on the data signal, the integrator in the loop filter provides high gain to track large jitter amplitudes with small phase error. In this case, the VCO is frequency modulated and jitter is tracked as in an ordinary phase-locked loop. The amount of low frequency jitter that can be tracked is a function of the VCO tuning range. A wider tuning range gives larger accommodation of low frequency jitter. The internal loop control voltage remains small for small phase errors, so the phase shifter remains close to the center of its range and thus contributes little to the low frequency jitter accommodation. Rev. PrA | Page 18 of 33 Preliminary Technical Data ADN2865 At medium jitter frequencies, the gain and tuning range of the VCO are not large enough to track input jitter. In this case, the VCO control voltage becomes large and saturates, and the VCO frequency dwells at one extreme of its tuning range or the other. The size of the VCO tuning range, therefore, has only a small effect on the jitter accommodation. The delay-locked loop control voltage is now larger, and so the phase shifter takes on the burden of tracking the input jitter. The phase shifter range, in UI, can be seen as a broad plateau on the jitter tolerance curve. The phase shifter has a minimum range of 2 UI at all data rates. cannot be tolerated. In this region, the gain of the integrator determines the jitter accommodation. Because the gain of the loop integrator declines linearly with frequency, jitter accommodation is lower with higher jitter frequency. At the highest frequencies, the loop gain is very small, and little tuning of the phase shifter can be expected. In this case, jitter accommodation is determined by the eye opening of the input data, the static phase error, and the residual loop jitter generation. The jitter accommodation is roughly 0.5 UI in this region. The corner frequency between the declining slope and the flat region is the closed loop bandwidth of the delay-locked loop, which is roughly 3 MHz at OC-48. The gain of the loop integrator is small for high jitter frequencies, so that larger phase differences are needed to make the loop control voltage big enough to tune the range of the phase shifter. Large phase errors at high jitter frequencies Rev. PrA | Page 19 of 33 ADN2865 Preliminary Technical Data FUNCTIONAL DESCRIPTION SERDES The ADN2865 has an integrated serializer / deserializer and clock divider which allows the continuous rate CDR to interface directly to an FPGA or digital ASIC, such a s a Media Access Controller (MAC), resulting in power and space savings. TXCLK TXDATA[7:0] EDGE DETECTOR REGISTER ALIGN MODE TXCLK The recovered clock is divided by 16 and is used to transfer 8 bits of receive data to the MAC on both the rising and falling edge. Both RXCLKP/N and RXDATAP/N[7:0] use LVDS signaling for noise reasons and have a relative phase which is adjustable via the I2C interface, per table 12 on page 14. RECOVERED CLOCK Dn+1[7:0] RESET DIVIDE REGISTER DIV BY 8 BY 8 Dn[7:0] COUNTER SERIALISER Figure 21. Align Mode Operation Half rate (1.25Gb/s) transmit data can also be serialised by the ADN2865 at the CML output using the recovered clock from the receive channel. An optional CML clock output is available. The parallel interface consists of 8 LVCMOS / LVTTL inputs with an optional TXCLK at the divde by 8 rate. Two timing modes are available, sync mode and align mode. Sync Mode This is the default mode of operation. and does not require a TXCLK signal. Instead, TXDATA[7:0] is timed from RXCLK and the round trip delay between these signals must meet the setup and hold time requirement specified in table 3 to avoid corrupting the serial bit stream. Sync mode is useful in applications which require a stable timing relationship between the input and output serial bit streams. Align Mode This mode is controlled using the I2C interface and requires the use of TXCLK which is used to latch TXDATA internally. When enabled, align mode centers this latched data with respect to the internal divide by 8 sampling clock, which can render the interface less sensitive to variation in the timing of TXDATA[7:0] relative to RXCLK. In a typical application, the propagation delay between RXCLK and TXDATA[7:0] will vary with process, temperature and supply voltage through the external MAC device. This variation can be calibrated out by enabling and then disabling align mode. The downside to using align mode is that the calibration process leads to additional uncertainty in the serial bit timing relative to the input bit stream by +/- 1 UI. In align mode, it is necessary to meet the setup and hold time for TXDATA[7:0] relative to TXCLK. Bit order reversal is supported for both the receive and transmit parallel buses using the I2C interface. FREQUENCY ACQUISITION The ADN2865 acquires frequency from the data over a range of data frequencies from 12.3 Mb/s to 2.7 Gb/s. The lock detector circuit compares the frequency of the VCO and the frequency of the incoming data. When these frequencies differ by more than 1000 ppm, LOL is asserted. This initiates a frequency acquisition cycle. The VCO frequency is reset to the bottom of its range, which is 12.3 MHz. The frequency detector then compares this VCO frequency and the incoming data frequency and increments the VCO frequency, if necessary. Initially, the VCO frequency is incremented in large steps to aid fast acquisition. As the VCO frequency approaches the data frequency, the step size is reduced until the VCO frequency is within 250 ppm of the data frequency, at which point LOL is de-asserted. Once LOL is de-asserted, the frequency-locked loop is turned off. The PLL/DLL pulls in the VCO frequency the rest of the way until the VCO frequency equals the data frequency. The frequency loop requires a single external capacitor between CF1 and CF2, Pins 14 and 15. A 0.47 μF ± 20%, X7R ceramic chip capacitor with < 10 nA leakage current is recommended. Leakage current of the capacitor can be calculated by dividing the maximum voltage across the 0.47 μF capacitor, ~3 V, by the insulation resistance of the capacitor. The insulation resistance of the 0.47 uF capacitor should be greater than 300 MΩ LOCK DETECTOR OPERATION The lock detector on the ADN2865 has three modes of operation: normal mode, REFCLK mode, and static LOL mode. Normal Mode In normal mode, the ADN2865 is a continuous rate CDR that locks onto any data rate from 12.3 Mb/s to 2.7 Gb/s without the use of a reference clock as an acquisition aid. In this mode, the lock detector monitors the frequency difference between the VCO and the input data frequency, and de-asserts the loss of lock signal, which appears on LOL Pin 30, when the VCO is Rev. PrA | Page 20 of 33 Preliminary Technical Data ADN2865 within 250 ppm of the data frequency. This enables the D/PLL, which pulls the VCO frequency in the remaining amount and also acquires phase lock. Once locked, if the input frequency error exceeds 1000 ppm (0.1%), the loss of lock signal is reasserted and control returns to the frequency loop, which begins a new frequency acquisition starting at the lowest point in the VCO operating range, 12.3 MHz. The LOL pin remains asserted until the VCO locks onto a valid input data stream to within 250 ppm frequency error. This hysteresis is shown in Figure 22. LOL –1000 –250 0 250 1000 fVCO ERROR (ppm) 04228-0-018 1 Figure 22. Transfer Function of LOL LOL Detector Operation Using a Reference Clock In this mode, a reference clock is used as an acquisition aid to lock the ADN2865 VCO. Lock to reference mode is enabled by setting CTRLA[0] to 1. The user also needs to write to the CTRLA[7:6] and CTRLA[5:2] bits in order to set the reference frequency range and the divide ratio of the data rate with respect to the reference frequency. For more details, see the Reference Clock (Optional) section. In this mode, the lock detector monitors the difference in frequency between the divided down VCO and the divided down reference clock. The loss of lock signal, which appears on the LOL Pin 30, is deasserted when the VCO is within 250 ppm of the desired frequency. This enables the D/PLL, which pulls the VCO frequency in the remaining amount with respect to the input data and also acquires phase lock. Once locked, if the input frequency error exceeds 1000 ppm (0.1%), the loss of lock signal is re-asserted and control returns to the frequency loop, which re-acquires with respect to the reference clock. The LOL pin remains asserted until the VCO frequency is within 250 ppm of the desired frequency. This hysteresis is shown in Figure 22. Static LOL Mode The ADN2865 implements a static LOL feature, which indicates if a loss of lock condition has ever occurred and remains asserted, even if the ADN2865 regains lock, until the static LOL bit is manually reset. The I2C register bit, MISC[4], is the static LOL bit. If there is ever an occurrence of a loss of lock condition, this bit is internally asserted to logic high. The MISC[4] bit remains high even after the ADN2865 has reacquired lock to a new data rate. This bit can be reset by writing a 1 followed by 0 to I2C Register Bit CTRLB[6]. Once reset, the MISC[4] bit remains de-asserted until another loss of lock condition occurs. Writing a 1 to I2C Register Bit CTRLB[7] causes the LOL pin, Pin 16, to become a static LOL indicator. In this mode, the LOL pin mirrors the contents of the MISC[4] bit and has the functionality described in the previous paragraph. The CTRLB[7] bit defaults to 0. In this mode, the LOL pin operates in the normal operating mode, that is, it is asserted only when the ADN2865 is in acquisition mode and de-asserts when the ADN2865 has re-acquired lock. HARMONIC DETECTOR The ADN2865 provides a harmonic detector, which detects whether or not the input data has changed to a lower harmonic of the data rate that the VCO is currently locked onto. For example, if the input data instantaneously changes from OC-48, 2.488 Gb/s, to an OC-12, 622.080 Mb/s bit stream, this could be perceived as a valid OC-48 bit stream, because the OC-12 data pattern is exactly 4× slower than the OC-48 pattern. So, if the change in data rate is instantaneous, a 101 pattern at OC-12 would be perceived by the ADN2865 as a 111100001111 pattern at OC-48. If the change to a lower harmonic is instantaneous, a typical CDR could remain locked at the higher data rate. The ADN2865 implements a harmonic detector that automatically identifies whether or not the input data has switched to a lower harmonic of the data rate that the VCO is currently locked onto. When a harmonic is identified, the LOL pin is asserted and a new frequency acquisition is initiated. The ADN2865 automatically locks onto the new data rate, and the LOL pin is de-asserted. However, the harmonic detector does not detect higher harmonics of the data rate. If the input data rate switches to a higher harmonic of the data rate the VCO is currently locked onto, the VCO loses lock, the LOL pin is asserted, and a new frequency acquisition is initiated. The ADN2865 automatically locks onto the new data rate. The time to detect lock to harmonic is 16,384 × (Td/ρ) where: 1/Td is the new data rate. For example, if the data rate is switched from OC-48 to OC-12, then Td = 1/622 MHz. ρ is the data transition density. Most coding schemes seek to ensure that ρ = 0.5, for example, PRBS, 8B/10B. When the ADN2865 is placed in lock to reference mode, the harmonic detector is disabled. LIMITING AMPLIFIER The limiting amplifier on the ADN2865 has differential inputs (PIN/NIN), which are internally terminated with 50 Ω to an on-chip voltage reference (VREF = 2.5 V typically). The inputs are typically ac-coupled externally, although dc coupling is possible as long as the input common mode voltage remains above 2.5 V (see Figure , Figure , and Figure in the Applications Rev. PrA | Page 21 of 33 ADN2865 Preliminary Technical Data The quantizer slicing level can be offset by ±100 mV to mitigate the effect of amplified spontaneous emission (ASE) noise or duty cycle distortion by applying a differential voltage input of up to ±0.95 V to SLICEP/N inputs. If no adjustment of the slice level is needed, SLICEP/N should be tied to VEE. The gain of the slice adjustment is ~0.1 V/V. If the user is not using the BER monitoring function, sample phase adjustment can be utilized to optimize the horizontal sampling point of the incoming data eye. The ADN2865 automatically centers the sampling point to the best of its ability. However, sample phase adjustment could be used to compensate for any static phase offset of the CDR and duty cycle distortion of the incoming eye. Sample phase adjustment is applied to the incoming eye via the PHASE register. It is important to note that sample phase adjustment can not be used if the user is utilising the BER monitoring capability. This is because the BER monitoring circuit requires control of the sample phase adjustment circuitry. Also, using the sample phase adjustment capability uses an additional 180mW of power. LOSS OF SIGNAL (LOS) DETECTOR The receiver front end LOS detector circuit detects when the input signal level has fallen below a user-adjustable threshold. The threshold is set with a single external resistor from Pin 9, THRADJ, to VEE. The LOS comparator trip point-versusresistor value is illustrated in Figure 2. If the input level to the ADN2865 drops below the programmed LOS threshold, the output of the LOS detector, LOS Pin 1, is asserted to a Logic 1. The LOS detector’s response time is ~500 ns by design, but is dominated by the RC time constant in ac-coupled applications. The LOS pin defaults to active high. However, by setting Bit CTRLC[2] to 1, the LOS pin is configured as active low. There is typically 6 dB of electrical hysteresis designed into the LOS detector to prevent chatter on the LOS pin. This means that, if the input level drops below the programmed LOS threshold causing the LOS pin to assert, the LOS pin is not deasserted until the input level has increased to 6 dB (2×) above the LOS threshold (see Figure ). INPUT LEVEL HYSTERESIS LOS THRESHOLD 04228-0-017 SLICE AND SAMPLE PHASE ADJUST (ADN2817 ONLY) LOS OUTPUT INPUT VOLTAGE (VDIFF) Information section). Input offset is factory trimmed to achieve better than 6 mV typical sensitivity with minimal drift. The limiting amplifier can be driven differentially or single-ended. t Figure 23. ADN2817 LOS Detector Hysteresis The LOS detector and the SLICE level adjust can be used simultaneously on the ADN2865. This means that any offset added to the input signal by the SLICE adjust pins does not affect the LOS detector’s measurement of the absolute input level. I2C INTERFACE The ADN2865 supports a 2-wire, I2C compatible, serial bus driving multiple peripherals. Two inputs, serial data (SDA) and serial clock (SCK), carry information between any devices connected to the bus. Each slave device is recognized by a unique address. The 7-bit slave address is factory programmed to binary ‘1100000’. The LSB of the word sets either a read or write operation (see Figure ). Logic 1 corresponds to a read operation, while Logic 0 corresponds to a write operation. To control the device on the bus, the following protocol must be followed. First, the master initiates a data transfer by establishing a start condition, defined by a high to low transition on SDA while SCK remains high. This indicates that an address/data stream follows. All peripherals respond to the start condition and shift the next eight bits (the 7-bit address and the R/W bit). The bits are transferred from MSB to LSB. The peripheral that recognizes the transmitted address responds by pulling the data line low during the ninth clock pulse. This is known as an acknowledge bit. All other devices withdraw from the bus at this point and maintain an idle condition. The idle condition is where the device monitors the SDA and SCK lines waiting for the start condition and correct transmitted address. The R/W bit determines the direction of the data. Logic 0 on the LSB of the first byte means that the master writes information to the peripheral. Logic 1 on the LSB of the first byte means that the master reads information from the peripheral. The ADN2865 acts as a standard slave device on the bus. The data on the SDA pin is 8 bits long supporting the 7-bit addresses plus the R/W bit. The ADN2865 has 8 subaddresses to enable the user-accessible internal registers (see Table 6 through Table 15). It, therefore, interprets the first byte as the device address and the second byte as the starting subaddress. Autoincrement mode is supported, allowing data to be read from or written to Rev. PrA | Page 22 of 33 Preliminary Technical Data ADN2865 VCC the starting subaddress and each subsequent address without manually addressing the subsequent subaddress. A data transfer is always terminated by a stop condition. The user can also access any unique subaddress register on a one-by-one basis without updating all registers. CLK OSC A reference clock is not required to perform clock and data recovery with the ADN2865. However, support for an optional reference clock is provided. The reference clock can be driven differentially or single-ended. If the reference clock is not being used, then REFCLKP should be tied to VCC, and REFCLKN can be left floating or tied to VEE (the inputs are internally terminated to VCC/2). See Figure through Figure for sample configurations. The REFCLK input buffer accepts any differential signal with a peak-to-peak differential amplitude of greater than 100 mV (for example, LVPECL or LVDS) or a standard single-ended low voltage TTL input, providing maximum system flexibility. Phase noise and duty cycle of the reference clock are not critical and 100 ppm accuracy is sufficient. ADN2817/18 REFCLKP OUT 10 Buffer REFCLKN x 11 Stop and start conditions can be detected at any stage of the data transfer. If these conditions are asserted out of sequence with normal read and write operations, then they cause an immediate jump to the idle condition. During a given SCK high period, the user should issue one start condition, one stop condition, or a single stop condition followed by a single start condition. If an invalid subaddress is issued by the user, the ADN2865 does not issue an acknowledge and returns to the idle condition. If the user exceeds the highest subaddress while reading back in autoincrement mode, then the highest subaddress register contents continue to be output until the master device issues a no-acknowledge. This indicates the end of a read. In a no-acknowledge condition, the SDATA line is not pulled low on the ninth pulse. See Figure and Figure for sample read and write data transfers and Figure 12 for a more detailed timing diagram. REFERENCE CLOCK (OPTIONAL) ADN2817/18 REFCLKP 100k 100k VCC/2 Figure 25. Single-Ended REFCLK Configuration VCC ADN2817/18 10 REFCLKP Buffer NC 11 REFCLKN 100k 100k VCC/2 Figure 26. No REFCLK Configuration The two uses of the reference clock are mutually exclusive. The reference clock can be used either as an acquisition aid for the ADN2865 to lock onto data, or to measure the frequency of the incoming data to within 0.01%. (There is the capability to measure the data rate to approximately ±10% without the use of a reference clock.) The modes are mutually exclusive, because, in the first use, the user knows exactly what the data rate is and wants to force the part to lock onto only that data rate; in the second use, the user does not know what the data rate is and wants to measure it. Lock to reference mode is enabled by writing a 1 to I2C Register Bit CTRLA[0]. Fine data rate readback mode is enabled by writing a 1 to I2C Register Bit CTRLA[1]. Writing a 1 to both of these bits at the same time causes an indeterminate state and is not supported. Using the Reference Clock to Lock onto Data In this mode, the ADN2865 locks onto a frequency derived from the reference clock according to the following equation: 10 Buffer 11 Data Rate/2CTRLA[5:2] = REFCLK/2CTRLA[7:6] REFCLKN 100k 100k VCC/2 Figure 24. Differential REFCLK Configuration The user must know exactly what the data rate is, and provide a reference clock that is a function of this rate. The ADN2865 can still be used as a continuous rate device in this configuration, provided that the user has the ability to provide a reference clock that has a variable frequency (see Application Note AN-632). The reference clock can be anywhere between 12.3 MHz and 200 MHz. By default, the ADN2865 expects a reference clock of between 12.3 MHz and 25 MHz. If it is between 25 MHz and Rev. PrA | Page 23 of 33 ADN2865 Preliminary Technical Data 50 MHz, 50 MHz and 100 MHz, or 100 MHz and 200 MHz, the user needs to configure the ADN2865 to use the correct reference frequency range by setting two bits of the CTRLA register, CTRLA[7:6]. Prior to reading back the data rate using the reference clock, the CTRLA[7:6] bits must be set to the appropriate frequency range with respect to the reference clock being used. A fine data rate readback is then executed as follows: Table 16. CTRLA Settings Step 1: Write a 1 to CTRLA[1]. This enables the fine data rate measurement capability of the ADN2865. This bit is level sensitive and does not need to be reset to perform subsequent frequency measurements. CTRLA[7:6] 00 01 10 11 Range (MHz) 12.3 to 25 25 to 50 50 to 100 100 to 200 CTRLA[5:2] 0000 0001 n 1000 Ratio 1 2 2n 256 The user can specify a fixed integer multiple of the reference clock to lock onto using CTRLA[5:2], where CTRLA should be set to the data rate/DIV_FREF, where DIV_FREF represents the divided-down reference referred to the 12.3 MHz to 25 MHz band. For example, if the reference clock frequency was 38.88 MHz and the input data rate was 622.08 Mb/s, then CTRLA[7:6] would be set to [01] to give a divided-down reference clock of 19.44 MHz. CTRLA[5:2] would be set to [0101], that is, 5, because Step 2: Reset MISC[2] by writing a 1 followed by a 0 to CTRLB[3]. This initiates a new data rate measurement. Step 3: Read back MISC[2]. If it is 0, then the measurement is not complete. If it is 1, then the measurement is complete and the data rate can be read back on FREQ[22:0]. The time for a data rate measurement is typically 80 ms. Step 4: Read back the data rate from registers FREQ2[6:0], FREQ1[7:0], and FREQ0[7:0]. Use the following equation to determine the data rate: 5 f DATARATE = (FREQ [22..0]× f REFCLK )/ 2 (14 + SEL _ RATE ) 622.08 Mb/s/19.44 MHz = 2 In this mode, if the ADN2865 loses lock for any reason, it relocks onto the reference clock and continues to output a stable clock. While the ADN2865 is operating in lock to reference mode, if the user ever changes the reference frequency, the FREF range (CTRLA[7:6]), or the FREF ratio (CTRLA[5:2]), this must be followed by writing a 0 to 1 transition into the CTRLA[0] bit to initiate a new lock to reference command. where: FREQ[22:0] is the reading from FREQ2[6:0] (MSByte), FREQ1[7:0], and FREQ0[7:0] (LSByte). Table 17. D22 D21...D17 FREQ2[6:0] D16 D15 D14...D9 D8 FREQ1[7:0] D7 D6...D1 D0 FREQ0[7:0] fDATARATE is the data rate (Mb/s). Using the Reference Clock to Measure Data Frequency The user can also provide a reference clock to measure the recovered data frequency. In this case, the user provides a reference clock, and the ADN2865 compares the frequency of the incoming data to the incoming reference clock and returns a ratio of the two frequencies to 0.01% (100 ppm). The accuracy error of the reference clock is added to the accuracy of the ADN2865 data rate measurement. For example, if a 100-ppm accuracy reference clock is used, the total accuracy of the measurement is within 200 ppm. The reference clock can range from 12.3 MHz and 200 MHz. The ADN2865 expects a reference clock between 12.3 MHz and 25 MHz by default. If it is between 25 MHz and 50 MHz, 50 MHz and 100 MHz, or 100 MHz and 200 MHz, the user needs to configure the ADN2865 to use the correct reference frequency range by setting two bits of the CTRLA register, CTRLA[7:6]. Using the reference clock to determine the frequency of the incoming data does not affect the manner in which the part locks onto data. In this mode, the reference clock is used only to determine the frequency of the data. For this reason, the user does not need to know the data rate to use the reference clock in this manner. fREFCLK is the REFCLK frequency (MHz). SEL_RATE is the setting from CTRLA[7:6]. For example, if the reference clock frequency is 32 MHz, SEL_RATE = 1, since the CTRLA[7:6] setting would be [01], because the reference frequency would fall into the 25 MHz to 50 MHz range. Assume for this example that the input data rate is 2.488 Gb/s (OC-48). After following Steps 1 through 4, the value that is read back on FREQ[22:0] = 0x26E010, which is equal to 2.5477 × 106. Plugging this value into the equation yields (2.5477e6 × 32e6)/(2 (14 +1) ) = 2.488 Gb/s If subsequent frequency measurements are required, CTRLA[1] should remain set to 1. It does not need to be reset. The measurement process is reset by writing a 1 followed by a 0 to CTRLB[3]. This initiates a new data rate measurement. Follow Steps 2 through 4 to read back the new data rate. Note: A data rate readback is valid only if LOL is low. If LOL is high, the data rate readback is invalid. Rev. PrA | Page 24 of 33 Preliminary Technical Data ADN2865 2. Additional Features Available via the I2C Interface Coarse Data Rate Readback The data rate can be read back over the I2C interface to approximately +10% without the need of an external reference clock. A 9-bit register, COARSE_RD[8:0], can be read back when LOL is de-asserted. The 8 MSBs of this register are the contents of the RATE[7:0] register. The LSB of the COARSE_RD register is Bit MISC[0]. Table provides coarse data rate readback to within ±10%. The subharmonic must be continued to be applied until LOL goes LOW, i.e. until acquisition is completed. It doesn't matter how long the subharmonic remains after LOL goes LOW. In FDDI Mode, the output of the ADN2865 is squelched until the device has acquired lock of the subharmonic input. This causes all zeros to be transmitted out of the 2865 until lock has been achieved. Once locked, the outputs are enabled and begin transmitting data. For FDDI protocol, this would be when the 'H' symbols are being transmitted during link synchronization. Fi LOS Configuration CLK HOLDOVER MODE The LOS detector output, LOS Pin 22, can be configured to be either active high or active low. If CTRLC[2] is set to Logic 0 (default), the LOS pin is active high when a loss of signal condition is detected. Writing a 1 to CTRLC[2] configures the LOS pin to be active low when a loss of signal condition is detected. System Reset A frequency acquisition can be initiated by writing a 1 followed by a 0 to the I2C Register Bit CTRLB[5]. This initiates a new frequency acquisition while keeping the ADN2865 in the operating mode that it was previously programmed to in registers CTRL[A], CTRL[B], and CTRL[C]. FDDI Mode A scheme has been implemented on the ADN2865 that enables the device to lock to input data streams that appear as subharmonics of the desired datarate, e.g. FDDI during link synchronization. This works for any code where a subharmonic down to the 31st is transmitted. FDDI uses the 5th subharmonic. The implementation requires certain programming by the user and more importantly certain assumptions about the incoming data. The user is required to program the part into FDDI mode by setting bit FDDI_MODE[7]=1. The user then needs to program the target datarate, (for FDDI this is 125MHz). This is done by programming an upper and lower 9-bit code into I2C registers HI_CODE[8..0], LO_CODE[8..0], and CODE_LSB[1..0]. See Table XX for a look-up table showing the correct register settings for each datarate. The user must also program the subharmonic ratio into I2C register FDDI_MODE[6..2] that the ADN2865 needs to lock on to, e.g. FDDI_MODE[6..2] = 00101 for FDDI (5th subharmonic). The user has to de-program FDDI mode before the next datarate is applied. Here is what is required of the incoming data: 1. The subharmonic must be a clock-type waveform i.e. transition density equal to 1 at the subharmonic frequency. CLK Holdover Mode 2A: This mode of operation will be available in all LTD modes: The output clock frequency will remain within +/-5% if the input data is removed or changed. To operate in this mode, the user would write to the I2C to put the part into CLK Holdover Mode 2A mode by setting SEL_MODE[2]=1. The user must then initiate an acquisition via a software reset. The device will then lock onto the input datarate. At this point the output frequency remains within +/- 5% of the intial acquired value regardless of whether or not the input data is taken away or the datarate changes. Only a sw reset can initiate a new acquistion in this mode. CLK Holdover Mode 2B: This mode is selected by setting SEL_MODE[1]=1. In this mode, the output clock stays within +/-5% of the initial acquired frequency, even if the input data is taken away. Unlike CLK Holdover Mode 2A, in this mode the ADN2865 will initiate a new frequency acquisition automatically if the input datarate changes. This mode requires the inputs to be DC coupled because if the inputs are AC coupled and the input is taken away, any noise present on the inputs may be large enough to trigger a new frequency acquisition which would cause the clock output frequency to change. CDR BYPASS MODE The CDR on the ADN2865 can be bypassed by setting bit CTRLD[7]=1. In this mode the ADN2865 will feed the input directly through the input amplifiers to the output buffer, completely bypassing the CDR. DISABLE OUTPUT BUFFERS The ADN2865 provides the option of disabling the output buffers for power savings. The LVDS output buffers can be disabled by setting CTRLD[6]=1. For additional power savings, e.g. in a low power standby mode, the CML output buffers can also be disabled by setting CTRLD[5]=1. Rev. PrA | Page 25 of 33 ADN2865 Preliminary Technical Data APPLICATIONS INFORMATION PCB DESIGN GUIDELINES Proper RF PCB design techniques must be used for optimal performance. Power Supply Connections and Ground Planes Use of one low impedance ground plane is recommended. The VEE pins should be soldered directly to the ground plane to reduce series inductance. If the ground plane is an internal plane and connections to the ground plane are made through vias, multiple vias can be used in parallel to reduce the series inductance. The exposed pad should be connected to the GND plane using plugged vias so that solder does not leak through the vias during reflow. By using adjacent power supply and GND planes, excellent high frequency decoupling can be realized by using close spacing between the planes. This capacitance is given by Cplane = 0.88ε r A/d (pF ) where: εr is the dielectric constant of the PCB material. A is the area of the overlap of power and GND planes (cm2). d is the separation between planes (mm). For FR-4, εr = 4.4 mm and 0.25 mm spacing, C ~15 pF/ 15 16 17 18 19 20 21 22 23 24 25 26 27 28 NC TXDAT4 TXCLK TXDAT3 TXDAT2 TXDAT1 TXDAT0 VREG CF1 VEE VCC VEE VCC REFN REFP ........ 56 55 54 53 52 51 50 49 48 47 46 45 44 43 RXDATP2 RXDATN2 RXDATP3 RXDATN3 RXDATP4 RXDATN4 VCC RXDATP5 RXDATN5 RXDATP6 RXDATN6 VCC RXDATP7 RXDATN7 .. Use of a 10 μF electrolytic capacitor between VCC and VEE is recommended at the location where the 3.3 V supply enters the PCB. When using 0.1 μF and 1 nF ceramic chip capacitors, they should be placed between the IC power supply VCC and VEE, as close as possible to the ADN2865 VCC pins. If connections to the supply and ground are made through vias, the use of multiple vias in parallel helps to reduce series inductance, especially on Pins 7,45 & 50, which supplies power to the high speed LVDS & CML output buffers. Refer to the schematic in Figure for recommended connections. Figure 27. Typical ADN2865 Applications Circuit Rev. PrA | Page 26 of 33 Preliminary Technical Data ADN2865 Transmission Lines Use of 50 Ω transmission lines is required for all LVDS and CML input and output signals to minimize reflections: PIN, NIN, RXDATAP/N[7:0], RXCLKP/N, SERDATP, SERDATN, SERCLKP, SERCLKN (also REFCLKP, REFCLKN, if a high frequency reference clock is used, such as 155 MHz). It is also necessary for the PIN/NIN input traces to be matched in length, and the parallel bus / CML output traces to be matched in length to avoid skew between the differential traces. All high speed CML outputs, SERDATP,SERDATN,SERCLKP,SERCLKN also require 100 Ω back termination chip resistors connected between the output pin and VCC. These resistors should be placed as close as possible to the output pins. These 100 Ω resistors are in parallel with on-chip 100 Ω termination resistors to create a 50 Ω back termination (see Figure ). The high speed inputs, PIN and NIN, are internally terminated with 50 Ω to an internal reference voltage (see Figure ). A 0.1 μF is recommended between VREF, Pin 36, and GND to provide an ac ground for the inputs. As with any high speed mixed-signal design, take care to keep all high speed digital traces away from sensitive analog nodes. VCC VCC 100 100 100 Choosing AC Coupling Capacitors AC coupling capacitors at the input (PIN, NIN) and output (SERDATP,SERDATN) of the ADN2865 must be chosen such that the device works properly over the full range of data rates used in the application. When choosing the capacitors, the time constant formed with the two 50 Ω resistors in the signal path must be considered. When a large number of consecutive identical digits (CIDs) are applied, the capacitor voltage can droop due to baseline wander (see Figure ), causing patterndependent jitter (PDJ). The user must determine how much droop is tolerable and choose an ac coupling capacitor based on that amount of droop. The amount of PDJ can then be approximated based on the capacitor selection. The actual capacitor value selection may require some trade-offs between droop and PDJ. Example: Assuming that 2% droop can be tolerated, then the maximum differential droop is 4%. Normalizing to Vpp: Droop = Δ V = 0.04 V = 0.5 Vpp (1 − e–t/τ) ; therefore, τ = 12t VTERM 100 package has a central exposed pad. The pad on the printed circuit board should be at least as large as this exposed pad. The user must connect the exposed pad to VEE using plugged vias so that solder does not leak through the vias during reflow. This ensures a solid connection from the exposed pad to VEE. 0.1μ where: 50 τ is the RC time constant (C is the ac coupling capacitor, R = 100 Ω seen by C). 0. 1μ 50Ω t is the total discharge time, which is equal to nΤ. 50 n is the number of CIDs. VTERM ADN2865 T is the bit period. The capacitor value can then be calculated by combining the equations for τ and t: Figure 28. Typical ADN2865 Applications Circuit VC C C = 12nT/R A D N 2 86 5 TIA 50 Ω CIN PIN C IN N IN Once the capacitor value is selected, the PDJ can be approximated as 50 0. 1u F ( VREF ) PDJ pspp = 0.5t r 1 − e ( −nT/RC ) / 0.6 50 2.5V 3k where: PDJpspp is the amount of pattern-dependent jitter allowed; < 0.01 UI p-p typical. Figure 29. ADN2865 AC-Coupled Input Configuration Soldering Guidelines for Chip Scale Package The lands on the 56 LFCSP are rectangular. The printed circuit board pad for these should be 0.1 mm longer than the package land length and 0.05 mm wider than the package land width. The land should be centered on the pad. This ensures that the solder joint size is maximized. The bottom of the chip scale tr is the rise time, which is equal to 0.22/BW, where BW ~ 0.7 (bit rate). Note that this expression for tr is accurate only for the inputs. The output rise time for the ADN2865 is ~100 ps regardless of data rate. Rev. PrA | Page 27 of 33 ADN2865 Preliminary Technical Data VCC V1 V2 DATAOU T P 50 CI N T IA AD N 2817 PIN VREF C OUT CD R L imamp 50 V1b 1 V2b DATAOU T N N IN 2 4 3 V1 V1b V2 V2b Vref Vdiff VT H Vdiff = V2-V2b VT H = AD N 2817 Quantizer T hreshold N OT ES: 1. D uring data patterns with high transition density, differential D C voltage at V1 and V2 is zero. 2. When the output of the T I A goes to CI D, V1 and V1b are driven to different D C levels. V2 and V2b discharge to the Vref level which effectively introduces a differential D C offset across the AC coupling capacitors. 3. When the burst of data starts again, the differential D C offset across the AC coupling capacitors is applied to the input levels causing a D C shift in the differential input. T his shift is large enough such that one of the states, either H I or L O depending on the levels of V1 and V1b when the T IA went to CID, is cancelled out. T he quantizer will not recognize this as a valid state. 4. T he D C offset slowly discharges until the differential input voltage exceeds the sensitivity of the AD N 2817. T he quantizer will be able to recognize both H I and L O states at this point. Figure 30. Example of Baseline Wander DC-COUPLED APPLICATION VCC The inputs to the ADN2865 can also be dc-coupled. This might be necessary in burst mode applications, where there are long periods of CIDs, and baseline wander cannot be tolerated. If the inputs to the ADN2865 are dc-coupled, care must be taken not to violate the input range and common-mode level requirements of the ADN2865 (see Figure through Figure ). If dc coupling is required, and the output levels of the TIA do not adhere to the levels shown in Figure , then level shifting and/or an attenuator must be between the TIA outputs and the ADN2865 inputs. ADN2865 PI N TIA 50Ω NIN 50 0.1uF VREF Figure 31. DC-Coupled Application Rev. PrA | Page 28 of 33 50 2.5V 3k Preliminary Technical Data ADN2865 VPP = PIN – NIN = 2 × VSE = 2.0V MAX PIN INPUT (V) VPP = PIN – NIN = 2 × VSE = 10mV AT SENSITIVITY VSE = 5mV MIN VCM = 2.3V MIN (DC-COUPLED) NIN Figure 32. Minimum Allowed DC-Coupled Input Levels 04228-0-028 NIN VCM = 2.3V (DC-COUPLED) 04228-0-027 INPUT (V) VSE = 1.0V MAX PIN Figure 33. Maximum Allowed DC-Coupled Input Levels Rev. PrA | Page 29 of 33 ADN2865 Preliminary Technical Data COARSE DATA RATE READBACK LOOK-UP TABLE Code is the 9-bit value read back from COARSE_RD[8:0]. Table 18. Look-Up Table Code 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40 41 42 43 44 45 46 47 FMID 5.1934e+06 5.1930e+06 5.2930e+06 5.3989e+06 5.5124e+06 5.6325e+06 5.7612e+06 5.8995e+06 6.0473e+06 6.2097e+06 6.3819e+06 6.5675e+06 6.7688e+06 6.9874e+06 7.2262e+06 7.4863e+06 7.4139e+06 7.4135e+06 7.5606e+06 7.7173e+06 7.8852e+06 8.0633e+06 8.2548e+06 8.4586e+06 8.6784e+06 8.9180e+06 9.1736e+06 9.4481e+06 9.7464e+06 1.0068e+07 1.0417e+07 1.0791e+07 1.0387e+07 1.0386e+07 1.0586e+07 1.0798e+07 1.1025e+07 1.1265e+07 1.1522e+07 1.1799e+07 1.2095e+07 1.2419e+07 1.2764e+07 1.3135e+07 1.3538e+07 1.3975e+07 1.4452e+07 1.4973e+07 Code 48 49 50 51 52 53 54 55 56 57 58 59 60 61 62 63 64 65 66 67 68 69 70 71 72 73 74 75 76 77 78 79 80 81 82 83 84 85 86 87 88 89 90 91 92 93 94 95 FMID 1.4828e+07 1.4827e+07 1.5121e+07 1.5435e+07 1.5770e+07 1.6127e+07 1.6510e+07 1.6917e+07 1.7357e+07 1.7836e+07 1.8347e+07 1.8896e+07 1.9493e+07 2.0136e+07 2.0833e+07 2.1582e+07 2.0774e+07 2.0772e+07 2.1172e+07 2.1596e+07 2.2049e+07 2.2530e+07 2.3045e+07 2.3598e+07 2.4189e+07 2.4839e+07 2.5527e+07 2.6270e+07 2.7075e+07 2.7950e+07 2.8905e+07 2.9945e+07 2.9655e+07 2.9654e+07 3.0242e+07 3.0869e+07 3.1541e+07 3.2253e+07 3.3019e+07 3.3834e+07 3.4714e+07 3.5672e+07 3.6694e+07 3.7792e+07 3.8985e+07 4.0273e+07 4.1666e+07 4.3164e+07 Code 96 97 98 99 100 101 102 103 104 105 106 107 108 109 110 111 112 113 114 115 116 117 118 119 120 121 122 123 124 125 126 127 128 129 130 131 132 133 134 135 136 137 138 139 140 141 142 143 Rev. PrA | Page 30 of 33 FMID 4.1547e+07 4.1544e+07 4.2344e+07 4.3191e+07 4.4099e+07 4.5060e+07 4.6090e+07 4.7196e+07 4.8378e+07 4.9678e+07 5.1055e+07 5.2540e+07 5.4150e+07 5.5899e+07 5.7810e+07 5.9890e+07 5.9311e+07 5.9308e+07 6.0485e+07 6.1739e+07 6.3081e+07 6.4506e+07 6.6038e+07 6.7669e+07 6.9427e+07 7.1344e+07 7.3388e+07 7.5585e+07 7.7971e+07 8.0546e+07 8.3333e+07 8.6328e+07 8.3095e+07 8.3087e+07 8.4689e+07 8.6383e+07 8.8198e+07 9.0120e+07 9.2179e+07 9.4392e+07 9.6757e+07 9.9356e+07 1.0211e+08 1.0508e+08 1.0830e+08 1.1180e+08 1.1562e+08 1.1978e+08 Code 144 145 146 147 148 149 150 151 152 153 154 155 156 157 158 159 160 161 162 163 164 165 166 167 168 169 170 171 172 173 174 175 176 177 178 179 180 181 182 183 184 185 186 187 188 189 190 191 FMID 1.1862e+08 1.1862e+08 1.2097e+08 1.2348e+08 1.2616e+08 1.2901e+08 1.3208e+08 1.3534e+08 1.3885e+08 1.4269e+08 1.4678e+08 1.5117e+08 1.5594e+08 1.6109e+08 1.6667e+08 1.7266e+08 1.6619e+08 1.6617e+08 1.6938e+08 1.7277e+08 1.7640e+08 1.8024e+08 1.8436e+08 1.8878e+08 1.9351e+08 1.9871e+08 2.0422e+08 2.1016e+08 2.1660e+08 2.2360e+08 2.3124e+08 2.3956e+08 2.3724e+08 2.3723e+08 2.4194e+08 2.4695e+08 2.5233e+08 2.5802e+08 2.6415e+08 2.7067e+08 2.7771e+08 2.8538e+08 2.9355e+08 3.0234e+08 3.1188e+08 3.2218e+08 3.3333e+08 3.4531e+08 Preliminary Technical Data Code 192 193 194 195 196 197 198 199 200 201 202 203 204 205 206 207 208 209 210 211 212 213 214 215 FMID 3.3238e+08 3.3235e+08 3.3876e+08 3.4553e+08 3.5279e+08 3.6048e+08 3.6872e+08 3.7757e+08 3.8703e+08 3.9742e+08 4.0844e+08 4.2032e+08 4.3320e+08 4.4719e+08 4.6248e+08 4.7912e+08 4.7449e+08 4.7447e+08 4.8388e+08 4.9391e+08 5.0465e+08 5.1605e+08 5.2831e+08 5.4135e+08 Code 216 217 218 219 220 221 222 223 224 225 226 227 228 229 230 231 232 233 234 235 236 237 238 239 ADN2865 FMID 5.5542e+08 5.7075e+08 5.8711e+08 6.0468e+08 6.2377e+08 6.4437e+08 6.6666e+08 6.9062e+08 6.6476e+08 6.6470e+08 6.7751e+08 6.9106e+08 7.0558e+08 7.2096e+08 7.3743e+08 7.5514e+08 7.7405e+08 7.9485e+08 8.1688e+08 8.4064e+08 8.6640e+08 8.9438e+08 9.2496e+08 9.5825e+08 Code 240 241 242 243 244 245 246 247 248 249 250 251 252 253 254 255 256 257 258 259 260 261 262 263 Rev. PrA | Page 31 of 33 FMID 9.4898e+08 9.4893e+08 9.6776e+08 9.8782e+08 1.0093e+09 1.0321e+09 1.0566e+09 1.0827e+09 1.1108e+09 1.1415e+09 1.1742e+09 1.2094e+09 1.2475e+09 1.2887e+09 1.3333e+09 1.3812e+09 1.3295e+09 1.3294e+09 1.3550e+09 1.3821e+09 1.4112e+09 1.4419e+09 1.4749e+09 1.5103e+09 Code 264 265 266 267 268 269 270 271 272 273 274 275 276 277 278 279 280 281 282 283 284 285 286 287 FMID 1.5481e+09 1.5897e+09 1.6338e+09 1.6813e+09 1.7328e+09 1.7888e+09 1.8499e+09 1.9165e+09 1.8980e+09 1.8979e+09 1.9355e+09 1.9756e+09 2.0186e+09 2.0642e+09 2.1132e+09 2.1654e+09 2.2217e+09 2.2830e+09 2.3484e+09 2.4187e+09 2.4951e+09 2.5775e+09 2.6666e+09 2.7625e+09 ADN2865 Preliminary Technical Data OUTLINE DIMENSIONS Figure 34. 56-Lead Frame Chip Scale Package [LFCSP] (CP-56) Dimensions shown in millimeters ORDERING GUIDE Model ADN2865ACP ADN2865ACP-RL ADN2865ACP-RL7 Temperature Range −40°C to 85°C −40°C to 85°C −40°C to 85°C Package Description 56-LFCSP 56-LFCSP, tape-reel, 2500 pcs 56-LFCSP, tape-reel, 1500 pcs Rev. PrA | Page 32 of 33 Package Option CP-56 CP-56 CP-56 Preliminary Technical Data ADN2865 PR06000-0-3/06(PrA) NOTES Rev. PrA | Page 33 of 33