LTC1435A High Efficiency Low Noise Synchronous Step-Down Switching Regulator U DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ Dual N-Channel MOSFET Synchronous Drive Programmable Fixed Frequency Wide VIN Range: 3.5V to 36V Operation Low Minimum On-Time (≤ 300ns) for High Frequency, Low Duty Cycle Applications Very Low Dropout Operation: 99% Duty Cycle Low Standby Current Secondary Feedback Control Programmable Soft Start Remote Output Voltage Sense Logic Controlled Micropower Shutdown: IQ < 25µA Foldback Current Limiting (Optional) Current Mode Operation for Excellent Line and Load Transient Response Output Voltages from 1.19V to 9V Available in 16-Lead Narrow SO and SSOP Packages U APPLICATIONS ■ ■ ■ ■ The operating frequency is set by an external capacitor allowing maximum flexibility in optimizing efficiency. A secondary winding feedback control pin, SFB, guarantees regulation regardless of load on the main output by forcing continuous operation. Burst Mode operation is inhibited when the SFB pin is pulled low, which reduces noise and RF interference. Soft start is provided by an external capacitor that can be used to properly sequence supplies. The operating current level is user-programmable via an external current sense resistor. Wide input supply range allows operation from 3.5V to 30V (36V maximum). Notebook and Palmtop Computers, PDAs Cellular Telephones and Wireless Modems Portable Instruments Battery-Operated Devices DC Power Distribution Systems , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U ■ The LTC®1435A is a synchronous step-down switching regulator controller that drives external N-channel power MOSFETs using a fixed frequency architecture. A wide duty cycle range of 5% to 99% allows high VIN to low VOUT DC/DC conversion, as well as low dropout operation that extends operating time in battery-operated systems. Burst ModeTM operation provides high efficiency at low load currents. TYPICAL APPLICATION VIN 4.5V TO 22V COSC 43pF CSS 0.1µF CC 330pF COSC VIN RUN/SS TG ITH SW M1 Si4412DY SGND BOOST + 4.7µF VOSENSE SENSE – BG RSENSE 0.033Ω VOUT 1.6V/3A R1 35.7k CB 0.1µF INTVCC 100pF CIN 22µF 35V ×2 L1 4.7µH DB CMDSH-3 LTC1435A RC 10k + M2 Si4412DY R2 102k D1 MBRS140T3 OUT + C100µF 6.3V ×2 PGND SENSE + 1000pF 1435A F01 Figure 1. High Efficiency Step-Down Converter 1 LTC1435A U W W W SYMBOL PARAMETER Main Control Loop IIN VOSENSE Feedback Current VOSENSE Feedback Voltage ∆VLINEREG Reference Voltage Line Regulation ∆VLOADREG Output Voltage Load Regulation VSFB ISFB VOVL IQ Secondary Feedback Threshold Secondary Feedback Current Output Overvoltage Lockout Input DC Supply Current Normal Mode Shutdown VRUN/SS Run Pin Threshold IRUN/SS Soft Start Current Source ∆VSENSE(MAX) Maximum Current Sense Threshold tON(MIN) Minimum On-Time TG Transition Time Rise Time Fall Time BG Transition Time BG tr Rise Time Fall Time BG t f Internal VCC Regulator VINTVCC Internal VCC Voltage VLDO INT INTVCC Load Regulation VLDO EXT EXTVCC Voltage Drop VEXTVCC EXTVCC Switchover Voltage TG t r TG t f 2 U ELECTRICAL CHARACTERISTICS W Input Supply Voltage (VIN)......................... 36V to – 0.3V Topside Driver Supply Voltage (BOOST)....42V to – 0.3V Switch Voltage (SW)............................. VIN + 5V to – 5V EXTVCC Voltage ........................................ 10V to – 0.3V SENSE +, SENSE – Voltages ...... INTVCC + 0.3V to – 0.3V ITH, VOSENSE Voltages .............................. 2.7V to – 0.3V SFB, Run/SS Voltages .............................. 10V to – 0.3V Peak Driver Output Current < 10µs (TG, BG) ............. 2A INTVCC Output Current ........................................ 50mA Operating Ambient Temperature Range LTC1435AC ............................................ 0°C to 70°C LTC1435AI ......................................... – 40°C to 85°C Junction Temperature (Note 1)............................. 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C U ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION ORDER PART NUMBER TOP VIEW COSC 1 RUN/SS 2 16 TG 15 BOOST ITH 3 14 SW SFB 4 13 VIN SGND 5 12 INTVCC VOSENSE 6 11 BG SENSE– 10 PGND 7 SENSE+ 8 LTC1435ACG LTC1435ACS LTC1435AIG LTC1435AIS 9 EXTVCC G PACKAGE S PACKAGE 16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO TJMAX = 125°C, θJA = 130°C/ W (G) TJMAX = 125°C, θJA = 110°C/ W (S) Consult factory for Military grade parts. TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted. CONDITIONS TYP MAX UNITS 10 1.19 0.002 0.5 – 0.5 1.19 –1 1.28 50 1.202 0.01 0.8 – 0.8 1.22 –2 1.32 nA V %/V % % V µA V 280 16 1.3 3 150 250 25 2 4.5 180 300 µA µA V µA mV ns CLOAD = 3000pF CLOAD = 3000pF 50 50 150 150 ns ns CLOAD = 3000pF CLOAD = 3000pF 50 40 150 150 ns ns 5.0 – 0.2 130 4.7 5.2 –1 230 V % mV V (Note 2) (Note 2) VIN = 3.6V to 20V (Note 2) ITH Sinking 5µA (Note 2) ITH Sourcing 5µA VSFB Ramping Negative VSFB = 1.5V MIN ● 1.178 ● ● ● 1.16 1.24 EXTVCC = 5V (Note 3) 3.6V < VIN < 30V VRUN/SS = 0V, 3.6V < VIN < 15V ● VRUN/SS = 0V VOSENSE = 0V, 5V Tested with Square Wave, SENSE – = 1.6V, ∆VSENSE = 20mV (Note 5 ) 6V < VIN < 30V, VEXTVCC = 4V IINTVCC = 15mA, VEXTVCC = 4V IINTVCC = 15mA, VEXTVCC = 5V IINTVCC = 15mA, VEXTVCC Ramping Positive 0.8 1.5 130 ● 4.8 ● 4.5 LTC1435A ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 15V, VRUN/SS = 5V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Oscillator fOSC Oscillator Frequency COSC = 100pF (Note 4) 112 125 138 kHz The ● denotes specifications which apply over the full operating temperature range. LTC1435ACG/LTC1435ACS: 0°C ≤ TA ≤ 70°C LTC1435AIG/LTC1435AIS: – 40°C ≤ TA ≤ 85°C Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC1435ACG/LTC1435AIG: TJ = TA + (PD)(130°C/W) LTC1435ACS/LTC1435AIS: TJ = TA + (PD)(110°C/W) Note 2: The LTC1435A is tested in a feedback loop which servos VOSENSE to the balance point for the error amplifier (VITH = 1.19V). Note 3: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 4: Oscillator frequency is tested by measuring the COSC charge and discharge currents and applying the formula: ( )( ) 8.4(108) 1 + 1 –1 fOSC (kHz) = C (pF) + 11 I OSC CHG IDIS Note 5: The minimum on-time test condition corresponds to an inductor peak-to-peak ripple current ≥ 40% of IMAX (see Minimum On-Time Considerations in the Applications Information section). U W TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Input Voltage VOUT = 3.3V Efficiency vs Input Voltage VOUT = 5V VOUT = 3.3V VOUT = 5V 90 ILOAD = 1A ILOAD = 1A 85 ILOAD = 100mA 80 90 EFFICIENCY (%) EFFICIENCY (%) 90 ILOAD = 100mA 85 80 85 80 75 Burst Mode OPERATION 70 CONTINUOUS MODE 65 60 75 75 VIN = 10V VOUT = 5V RSENSE = 0.033Ω 95 95 95 EFFICIENCY (%) Efficiency vs Load Current 100 100 100 55 70 70 0 5 10 15 20 INPUT VOLTAGE (V) 25 0 30 5 10 15 20 INPUT VOLTAGE (V) 50 0.001 30 VIN – VOUT Dropout Voltage vs Load Current Load Regulation VITH Pin Voltage vs Output Current 3.0 RSENSE = 0.033Ω ∆VOUT (%) 0.4 0.3 0.2 0.1 – 0.25 2.5 – 0.50 2.0 VITH (V) RSENSE = 0.033Ω VOUT DROP OF 5% – 0.75 0.5 1.0 1.5 2.0 LOAD CURRENT (A) 2.5 3.0 1435A G04 1.5 Burst Mode OPERATION –1.00 1.0 –1.25 0.5 –1.50 0 0 10 1435A G03 0 0.5 1 0.01 0.1 LOAD CURRENT (A) 1435A G02 1435A G01 VIN – VOUT (V) 25 CONTINUOUS MODE 0 0 0.5 1.0 1.5 2.0 LOAD CURRENT (A) 2.5 3.0 1435A G05 0 10 20 30 40 50 60 70 80 90 100 OUTPUT CURRENT (%) 1435A G06 3 LTC1435A U W TYPICAL PERFORMANCE CHARACTERISTICS Input Supply and Shutdown Current vs Input Voltage 100 VOUT = 5V EXTVCC = VOUT 60 VOUT = 3.3V EXTVCC = OPEN 1.0 40 0.5 20 200 VEXTVCC = 0V 180 70°C 0 25°C – 0.3 10 15 20 INPUT VOLTAGE (V) 25°C 120 100 – 55°C 80 60 20 0 5 140 40 SHUTDOWN 0 0 70°C 160 0.3 EXTVCC – INTVCC (mV) SUPPLY CURRENT (mA) 80 SHUTDOWN CURRENT (µA) 2.0 0.5 ∆INTVCC (%) 2.5 1.5 EXTVCC Switch Drop vs INTVCC Load Current INTVCC Regulation vs INTVCC Load Current 25 – 0.5 30 0 0 10 15 5 INTVCC LOAD CURRENT (mA) 1435A G07 20 0 2 4 6 8 10 12 14 16 18 20 INTVCC LOAD CURRENT (mA) 1435A G09 1435A G08 RUN/SS Pin Current vs Temperature Normalized Oscillator Frequency vs Temperature 10 4 5 3 SFB Pin Current vs Temperature 0 fO –5 SFB CURRENT (µA) RUN/SS CURRENT (µA) FREQUENCY (%) – 0.25 2 – 0.50 – 0.75 –1.00 1 –1.25 –10 – 40 –15 60 35 85 10 TEMPERATURE (°C) 110 135 0 – 40 –15 85 10 35 60 TEMPERATURE (°C) 110 135 –1.50 – 40 –15 60 35 85 10 TEMPERATURE (°C) 1435A G11 1435A G10 Maximum Current Sense Threshold Voltage vs Temperature 110 135 1435A G12 Transient Response Transient Response CURRENT SENSE THRESHOLD (mV) 154 152 VOUT 50mV/DIV VOUT 50mV/DIV 150 148 ILOAD = 50mA to 1A 146 – 40 –15 85 10 35 60 TEMPERATURE (°C) 110 135 1435A G13 4 1435A G14 ILOAD = 1A to 3A 1435A G15 LTC1435A U W TYPICAL PERFORMANCE CHARACTERISTICS Soft Start: Load Current vs Time Burst Mode Operation VOUT 20mV/DIV RUN/SS 5V/DIV INDUCTOR CURRENT 1A/DIV VITH 200mV/DIV ILOAD = 50mA 1435A G16 1435A G17 U U U PIN FUNCTIONS COSC (Pin 1): External capacitor COSC from this pin to ground sets the operating frequency. RUN/SS (Pin 2): Combination of Soft Start and Run Control Inputs. A capacitor to ground at this pin sets the ramp time to full current output. The time is approximately 0.5s/µF. Forcing this pin below 1.3V causes the device to be shut down. In shutdown all functions are disabled. ITH (Pin 3): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0V to 2.5V. SFB (Pin 4): Secondary Winding Feedback Input. Normally connected to a feedback resistive divider from the secondary winding. This pin should be tied to: ground to force continuous operation; INTVCC in applications that don’t use a secondary winding; and a resistive divider from the output in applications using a secondary winding. SGND (Pin 5): Small-Signal Ground. Must be routed separately from other grounds to the (–) terminal of COUT. VOSENSE (Pin 6): Receives the feedback voltage from an external resistive divider across the output. SENSE – (Pin 7): The (–) Input to the Current Comparator. SENSE + (Pin 8): The (+) Input to the Current Comparator. Built-in offsets between SENSE– and SENSE+ pins in conjunction with RSENSE set the current trip thresholds. EXTVCC (Pin 9): Input to the Internal Switch Connected to INTVCC. This switch closes and supplies VCC power when- ever EXTVCC is higher than 4.7V. See EXTVCC connection in Applications Information section. Do not exceed 10V on this pin. Connect to VOUT if VOUT ≥ 5V. PGND (Pin 10): Driver Power Ground. Connects to source of bottom N-channel MOSFET and the (–) terminal of CIN. BG (Pin 11): High Current Gate Drive for Bottom N-Channel MOSFET. Voltage swing at this pin is from ground to INTVCC. INTVCC (Pin 12): Output of the Internal 5V Regulator and EXTVCC Switch. The driver and control circuits are powered from this voltage. Must be closely decoupled to power ground with a minimum of 2.2µF tantalum or electrolytic capacitor. VIN (Pin 13): Main Supply Pin. Must be closely decoupled to the IC’s signal ground pin. SW (Pin 14): Switch Node Connection to Inductor. Voltage swing at this pin is from a Schottky diode (external) voltage drop below ground to VIN. BOOST (Pin 15): Supply to Topside Floating Driver. The bootstrap capacitor is returned to this pin. Voltage swing at this pin is from INTVCC to VIN + INTVCC. TG (Pin 16): High Current Gate Drive for Top N-Channel MOSFET. This is the output of a floating driver with a voltage swing equal to INTVCC superimposed on the switch node voltage SW. 5 LTC1435A W FUNCTIONAL DIAGRA U U VIN + CIN COSC 1 COSC 4 SFB 13 VIN SGND 5 INTVCC 1.19V REF 1µA DB BOOST 15 – 1.19V CB + SHUTDOWN OSC + TG 16 DROP OUT DET OV S Q R – 1.28V 0.6V SWITCH LOGIC + – VOSENSE 6 VFB – – I1 EA R2 + Ω 1.19V gm = 1m 180k VSEC D1 4k + SW 14 I2 – + VIN + INTVCC INTVCC CSEC + 12 + R1 5V LDO REG – SHUTDOWN 3µA RUN SOFT START 6V + 4.8V 30k BG 11 8k VOUT – RC 2 RUN/SS CSS 3 ITH CC SENSE+ 8 7 SENSE DFB* – 9 EXTVCC COUT PGND 10 + RSENSE 1435A • FD * FOLDBACK CURRENT LIMITING OPTION U OPERATION (Refer to Functional Diagram) Main Control Loop The LTC1435A uses a constant frequency, current mode step-down architecture. During normal operation, the top MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the main current comparator I1 resets the RS latch. The peak inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH pin , which is the output of error amplifier EA. The VOSENSE pin, described in the Pin Functions section, allows EA to receive an output feedback voltage VFB from an external resistive divider. When the load current increases, it causes a slight decrease in VFB relative to the 1.19V ref- 6 erence, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle. The top MOSFET driver is biased from floating bootstrap capacitor CB, which normally is recharged during each off cycle. However, when VIN decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector counts the number of oscillator cycles that the top MOSFET remains LTC1435A U OPERATION (Refer to Functional Diagram) on and periodically forces a brief off period to allow CB to recharge. either of which causes drive to be returned to the TG pin on the next cycle. The main control loop is shut down by pulling the RUN/SS pin low. Releasing RUN/SS allows an internal 3µA current source to charge soft start capacitor CSS. When CSS reaches 1.3V, the main control loop is enabled with the ITH voltage clamped at approximately 30% of its maximum value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume. Two conditions can force continuous synchronous operation, even when the load current would otherwise dictate low current operation. One is when the common mode voltage of the SENSE+ and SENSE – pins is below 1.4V and the other is when the SFB pin is below 1.19V. The latter condition is used to assist in secondary winding regulation as described in the Applications Information section. Comparator OV guards against transient overshoots > 7.5% by turning off the top MOSFET and keeping it off until the fault is removed. INTVCC/EXTVCC Power Low Current Operation The LTC1435A is capable of Burst Mode operation in which the external MOSFETs operate intermittently based on load demand. The transition to low current operation begins when comparator I2 detects current reversal and turns off the bottom MOSFET. If the voltage across RSENSE does not exceed the hysteresis of I2 (approximately 20mV) for one full cycle, then on following cycles the top and bottom drives are disabled. This continues until an inductor current peak exceeds 20mV/RSENSE or the ITH voltage exceeds 0.6V, Power for the top and bottom MOSFET drivers and most of the other LTC1435A circuitry is derived from the INTVCC pin. The bottom MOSFET driver supply pin is internally connected to INTVCC in the LTC1435A. When the EXTVCC pin is left open, an internal 5V low dropout regulator supplies INTVCC power. If EXTVCC is taken above 4.8V, the 5V regulator is turned off and an internal switch is turned on to connect EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency external source such as the output of the regulator itself or a secondary winding, as described in the Applications Information section. U W U U APPLICATIONS INFORMATION The basic LTC1435A application circuit is shown in Figure 1, High Efficiency Step-Down Converter. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, COSC and L can be chosen. Next, the power MOSFETs and D1 are selected. Finally, CIN and COUT are selected. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 28V (limited by the external MOSFETs). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. The LTC1435A current comparator has a maximum threshold of 150mV/RSENSE and an input common mode range of SGND to INTVCC. The current comparator threshold sets the peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half the peak-to-peak ripple current ∆IL. Allowing a margin for variations in the LTC1435A and external component values yields: RSENSE = 100mV IMAX The LTC1435A works well with RSENSE values ≥ 0.005Ω. COSC Selection for Operating Frequency The LTC1435A uses a constant frequency architecture with the frequency determined by an external oscillator capacitor COSC. Each time the topside MOSFET turns on, the voltage COSC is reset to ground. During the on-time, COSC is charged by a fixed current. When the voltage on the capacitor reaches 1.19V, COSC is reset to ground. The process then repeats. 7 LTC1435A U W U U APPLICATIONS INFORMATION The value of COSC is calculated from the desired operating frequency: 1.37(104 ) – 11 COSC (pF) = Frequency (kHz) A graph for selecting COSC vs frequency is given in Figure 2. As the operating frequency is increased the gate charge losses will be higher, reducing efficiency (see Efficiency Considerations). The maximum recommended switching frequency is 400kHz. 300 The inductor value also has an effect on low current operation. The transition to low current operation begins when the inductor current reaches zero while the bottom MOSFET is on. Lower inductor values (higher ∆IL) will cause this to occur at higher load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. The Figure 3 graph gives a range of recommended inductor values vs operating frequency and VOUT. 250 200 60 150 50 100 50 0 0 100 200 300 400 OPERATING FREQUENCY (kHz) 500 INDUCTOR VALUE (µH) COSC VALUE (pF) greater core losses. A reasonable starting point for setting ripple current is ∆IL = 0.4(IMAX). Remember, the maximum ∆IL occurs at the maximum input voltage. VOUT = 5.0V VOUT = 3.3V VOUT ≤ 2.5V 40 30 20 10 1435A F02 0 Figure 2. Timing Capacitor Value 0 250 100 150 200 50 OPERATING FREQUENCY (kHz) Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses. In addition to this basic tradeoff, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN or VOUT: V 1 ∆IL = VOUT 1– OUT VIN ( f)(L) Accepting larger values of ∆IL allows the use of low inductances, but results in higher output voltage ripple and 8 300 1435A F03 Figure 3. Recommended Inductor Values For low duty cycle, high frequency applications where the required minimum on-time, tON(MIN) = (V VOUT IN(MAX ) )(f) , is less than 350ns, there may be further restrictions on the inductance to ensure proper operation. See Minimum OnTime Considerations section for more details. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. Actual core loss is independent of core size for Kool Mµ is a registered trademark of Magnetics, Inc. LTC1435A U W U U APPLICATIONS INFORMATION a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, designs for surface mount are available which do not increase the height significantly. Power MOSFET and D1 Selection Two external power MOSFETs must be selected for use with the LTC1435A: an N-channel MOSFET for the top (main) switch and an N-channel MOSFET for the bottom (synchronous) switch. The peak-to-peak gate drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see EXTVCC Pin Connection). Consequently, logic level threshold MOSFETs must be used in most LTC1435A applications. The only exception is applications in which EXTVCC is powered from an external supply greater than 8V (must be less than 10V), in which standard threshold MOSFETs (VGS(TH) < 4V) may be used. Pay close attention to the BVDSS specification for the MOSFETs as well; many of the logic level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the “ON” resistance RDS(ON), reverse transfer capacitance CRSS, input voltage and maximum output current. When the LTC1435A is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: V Main Switch Duty Cycle = OUT VIN (V − V ) Synchronous Switch Duty Cycle = IN OUT VIN The MOSFET power dissipations at maximum output current are given by: V 2 PMAIN = OUT (IMAX ) (1 + δ )RDS(ON) + VIN k(VIN ) 1.85 (IMAX )(CRSS )( f) V −V 2 PSYNC = IN OUT (IMAX ) (1 + δ )RDS(ON) VIN where δ is the temperature dependency of RDS(ON) and k is a constant inversely related to the gate drive current. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CRSS actual provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage or during a short circuit when the duty cycle in this switch is nearly 100%. Refer to the Foldback Current Limiting section for further applications information. The term (1 + δ) is generally given for a MOSFET in the form of a normalized R DS(ON) vs Temperature curve, but δ = 0.005/°C can be used as an approximation for low voltage MOSFETs. CRSS is usually specified in the MOSFET characteristics. The constant k = 2.5 can be used to estimate the contributions of the two terms in the main switch dissipation equation. The Schottky diode D1 shown in Figure 1 conducts during the dead-time between the conduction of the two large power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on and storing charge during the dead-time, which could cost as much as 1% in efficiency. A 1A Schottky is generally a good size for 3A regulators. 9 LTC1435A U U W U APPLICATIONS INFORMATION CIN and COUT Selection In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/ VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: CIN required IRMS ≈ IMAX [ ( VOUT VIN − VOUT )] 1/ 2 VIN This formula has a maximum at V IN = 2V OUT , where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied the capacitance is adequate for filtering. The output ripple (∆VOUT) is approximated by: 1 ∆VOUT ≈ ∆IL ESR + 4 fC OUT where f = operating frequency, COUT = output capacitance and ∆IL= ripple current in the inductor. The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. With ∆IL = 0.4IOUT(MAX) the output ripple will be less than 100mV at max VIN assuming: COUT required ESR < 2RSENSE Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR(size) product of any aluminum electrolytic at a somewhat higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. 10 In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. INTVCC Regulator An internal P-channel low dropout regulator produces the 5V supply that powers the drivers and internal circuitry within the LTC1435A. The INTVCC pin can supply up to 15mA and must be bypassed to ground with a minimum of 2.2µF tantalum or low ESR electrolytic. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. High input voltage applications, in which large MOSFETs are being driven at high frequencies, may cause the maximum junction temperature rating for the LTC1435A to be exceeded. The IC supply current is dominated by the gate charge supply current when not using an output derived EXTVCC source. The gate charge is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 1 of the Electrical Characteristics. For example, the LTC1435A is limited to less than 17mA from a 30V supply: TJ = 70°C + (17mA)(30V)(100°C/W) = 126°C To prevent maximum junction temperature from being exceeded, the input supply current must be checked when operating in continuous mode at maximum VIN. EXTVCC Connection The LTC1435A contains an internal P-channel MOSFET switch connected between the EXTVCC and INTVCC pins. The switch closes and supplies the INTVCC power whenever the EXTVCC pin is above 4.8V, and remains closed until EXTVCC drops below 4.5V. This allows the MOSFET driver and LTC1435A U W U U APPLICATIONS INFORMATION control power to be derived from the output during normal operation (4.8V < VOUT < 9V) and from the internal regulator when the output is out of regulation (start-up, short circuit). Do not apply greater than 10V to the EXTVCC pin and ensure that EXTVCC < VIN. Significant efficiency gains can be realized by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be scaled by a factor of Duty Cycle/Efficiency. For 5V regulators this supply means connecting the EXTVCC pin directly to VOUT. However, for 3.3V and other lower voltage regulators, additional circuitry is required to derive INTVCC power from the output. + 1N4148 VIN OPTIONAL EXT VCC CONNECTION 5V ≤ VSEC ≤ 9V R6 4. EXTVCC connected to an external supply. If an external supply is available in the 5V to 10V range (EXTVCC ≤ VIN), it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements. When driving standard threshold MOSFETs, the external supply must always be present during operation to prevent MOSFET failure due to insufficient gate drive. Topside MOSFET Driver Supply (CB, DB) An external bootstrap capacitor CB connected to the Boost pin supplies the gate drive voltage for the topside MOSFET. Capacitor CB in the Functional Diagram is charged through diode DB from INTVCC when the SW pin is low. When the VSEC + L1 1:N 1µF RSENSE EXTVCC VOUT + LTC1435A COUT SW SFB BG R5 SGND N-CH PGND 1435A F04a Figure 4a. Secondary Output Loop and EXTVCC Connection + + VIN 1µF CIN 1. EXTVCC left open (or grounded). This will cause INTVCC to be powered from the internal 5V regulator resulting in an efficiency penalty of up to 10% at high input voltages. 3. EXTVCC connected to an output-derived boost network. For 3.3V and other low voltage regulators, efficiency gains can still be realized by connecting EXTVCC to an output-derived voltage which has been boosted to greater than 4.8V. This can be done with either the inductive boost winding as shown in Figure 4a or the capacitive charge pump shown in Figure 4b. The charge pump has the advantage of simple magnetics. N-CH TG The following list summarizes the four possible connections for EXTVCC: 2. EXTVCC connected directly to VOUT. This is the normal connection for a 5V regulator and provides the highest efficiency. VIN CIN BAT85 0.22µF BAT85 VIN TG BAT85 N-CH EXTVCC VN2222LL L1 RSENSE VOUT LTC1435A + SW BG COUT N-CH 1435A F04b PGND Figure 4b. Capacitive Charge Pump for EXTVCC topside MOSFET is to be turned on, the driver places the CB voltage across the gate source of the MOSFET. This enhances the MOSFET and turns on the topside switch. The switch node voltage SW rises to VIN and the Boost pin rises to VIN + INTVCC. The value of the boost capacitor CB needs to be 100 times greater than the total input capacitance of the topside MOSFET. In most applications 0.1µF is adequate. The reverse breakdown on DB must be greater than VIN(MAX). Output Voltage Programming The output voltage is set by a resistive divider according to the following formula: R2 VOUT = 1.19V 1 + , VOUT ≥ 1.19 V R1 11 LTC1435A U W U U APPLICATIONS INFORMATION The external resistive divider is connected to the output as shown in Figure 5 allowing remote voltage sensing. 1.19V ≤ VOUT ≤ 9V R2 VOSENSE 100pF LTC1435A SGND R1 1435A F05 Figure 5. Setting the LTC1435A Output Voltage Run/ Soft Start Function The RUN/SS pin is a dual purpose pin that provides the soft start function and a means to shut down the LTC1435A. Soft start reduces surge currents from VIN by gradually increasing the internal current limit. Power supply sequencing can also be accomplished using this pin. An internal 3µA current source charges up an external capacitor CSS. When the voltage on RUN/SS reaches 1.3V the LTC1435A begins operating. As the voltage on RUN/SS continues to ramp from 1.3V to 2.4V, the internal current limit is also ramped at a proportional linear rate. The current limit begins at approximately 50mV/RSENSE (at VRUN/ SS = 1.3V) and ends at 150mV/RSENSE (VRUN/SS > 2.7V). The output current thus ramps up slowly, charging the output capacitor. If RUN/SS has been pulled all the way to ground there is a delay before starting of approximately 500ms/µF, followed by an additional 500ms/µF to reach full current. tDELAY = 5(10 5)CSS Seconds Pulling the RUN/SS pin below 1.3V puts the LTC1435A into a low quiescent current shutdown (IQ < 25µA). This pin can be driven directly from logic as shown in Figure 6. Diode D1 in Figure 6 reduces the start delay but allows CSS to ramp up slowly for the soft start function; this diode and CSS can be deleted if soft start is not needed. The RUN/SS pin has an internal 6V Zener clamp (See Functional Diagram). 3.3V OR 5V RUN/SS RUN/SS D1 CSS CSS 1435 F06 Figure 6. RUN/SS Pin Interfacing 12 Foldback Current Limiting As described in Power MOSFET and D1 Selection, the worstcase dissipation for either MOSFET occurs with a shortcircuited output, when the synchronous MOSFET conducts the current limit value almost continuously. In most applications this will not cause excessive heating, even for extended fault intervals. However, when heat sinking is at a premium or higher RDS(ON) MOSFETs are being used, foldback current limiting should be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding diode DFB between the output and the ITH pin as shown in the Functional Diagram. In a hard short (VOUT = 0V) the current will be reduced to approximately 25% of the maximum output current. This technique may be used for all applications with regulated output voltages of 1.8V or greater. SFB Pin Operation When the SFB pin drops below its ground referenced 1.19V threshold, continuous mode operation is forced. In continuous mode, the large N-channel main and synchronous switches are used regardless of the load on the main output. In addition to providing a logic input to force continuous synchronous operation, the SFB pin provides a means to regulate a flyback winding output. Continuous synchronous operation allows power to be drawn from the auxiliary windings without regard to the primary output load. The SFB pin provides a way to force continuous synchronous operation as needed by the flyback winding. The secondary output voltage is set by the turns ratio of the transformer in conjunction with a pair of external resistors returned to the SFB pin as shown in Figure 4a. The secondary regulated voltage, VSEC, in Figure 4a is given by: R6 VSEC ≈ (N + 1)VOUT > 1.19 1 + R5 where N is the turns ratio of the transformer and VOUT is the main output voltage sensed by VOSENSE. Minimum On-Time Considerations Minimum on-time, tON(MIN), is the smallest amount of time that the LTC1435A is capable of turning the top MOSFET LTC1435A U W U U APPLICATIONS INFORMATION on and off again. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit. If the duty cycle falls below what can be accommodated by the minimum on-time, the LTC1435A will begin to skip cycles. The output voltage will continue to be regulated, but the ripple current and ripple voltage will increase. Therefore this limit should be avoided. The minimum on-time for the LTC1435A in a properly configured application is less than 300ns but increases at low ripple current amplitudes (see Figure 7). If an application is expected to operate close to the minimum on-time limit, an inductor value must be chosen that is low enough to provide sufficient ripple amplitude to meet the minimum on-time requirement. To determine the proper value, use the following procedure: 1. Calculate on-time at maximum supply, t ON(MIN) = (1/f)(VOUT/VIN(MAX)). 2. Use Figure 7 to obtain the peak-to-peak inductor ripple current as a percentage of IMAX necessary to achieve the calculated tON(MIN). 3. Ripple amplitude ∆IL(MIN) = (% from Figure 7)(IMAX) where IMAX = 0.1/RSENSE. VIN(MAX ) – VOUT 4. LMAX = tON(MIN) ∆IL(MIN) Choose an inductor less than or equal to the calculated LMAX to ensure proper operation. MINIMUM ON-TIME (ns) 400 350 RECOMMENDED REGION FOR MIN ON-TIME AND MAX EFFICIENCY 300 250 200 0 50 60 70 10 20 30 40 INDUCTOR RIPPLE CURRENT (% OF IMAX) 1435A F07 Figure 7. Minimum On-Time vs Inductor Ripple Current Because of the sensitivity of the LTC1435A current comparator when operating close to the minimum on-time limit, it is important to prevent stray magnetic flux generated by the inductor from inducing noise on the current sense resistor, which may occur when axial type cores are used. By orienting the sense resistor on the radial axis of the inductor (see Figure 8), this noise will be minimized. INDUCTOR L 1435A F08 Figure 8. Allowable Inductor/RSENSE Layout Orientations Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1435A circuits. LTC1435A VIN current, INTVCC current, I2R losses, and topside MOSFET transition losses. 1. The VIN current is the DC supply current given in the electrical characteristics which excludes MOSFET driver and control currents. V IN current results in a small (< 1%) loss which increases with VIN. 2. INTVCC current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the control circuit current. In continuous mode, IGATECHG = f(QT + QB), where QT and QB are the gate charges of the topside and bottom side MOSFETs. 13 LTC1435A U W U U APPLICATIONS INFORMATION By powering EXTVCC from an output-derived source, the additional VIN current resulting from the driver and control currents will be scaled by a factor of Duty Cycle/Efficiency. For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately 3mA of VIN current. This reduces the midcurrent loss from 10% or more (if the driver was powered directly from VIN) to only a few percent. 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current shunt. In continuous mode the average output current flows through L and RSENSE, but is “chopped” between the topside main MOSFET and the synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and RSENSE to obtain I2R losses. For example, if each R DS(ON) = 0.05Ω, RL = 0.15Ω, and RSENSE = 0.05Ω, then the total resistance is 0.25Ω. This results in losses ranging from 3% to 10% as the output current increases from 0.5A to 2A. I2R losses cause the efficiency to drop at high output currents. 4. Transition losses apply only to the topside MOSFET(s), and only when operating at high input voltages (typically 20V or greater). Transition losses can be estimated from: Transition Loss = 2.5 (VIN)1.85(IMAX)(CRSS)(f) Other losses, including CIN and COUT ESR dissipative losses, Schottky conduction losses during dead-time, and inductor core losses, generally account for less than 2% total additional loss. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD)(ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT which generates a feedback error signal. The regulator loop then acts to return VOUT to its steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing, which would indicate a stability problem. The ITH external components shown in 14 the Figure 1 circuit will provide adequate compensation for most applications. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25)(CLOAD). Thus a 10µF capacitor would require a 250µs rise time, limiting the charging current to about 200mA. Automotive Considerations: Plugging into the Cigarette Lighter As battery-powered devices go mobile, there is a natural interest in plugging into the cigarette lighter in order to conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main battery line in an automobile is the source of a number of nasty potential transients, including load dump, reverse battery and double battery. Load dump is the result of a loose battery cable. When the cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes several hundred milliseconds to decay. Reverse battery is just what it says, while double battery is a consequence of tow truck operators finding that a 24V jump start cranks cold engines faster than 12V. The network shown in Figure 9 is the most straightforward approach to protect a DC/DC converter from the ravages of an automotive battery line. The series diode prevents current from flowing during reverse battery, while the transient suppressor clamps the input voltage during load dump. Note that the transient suppressor should not 12V 50A IPK RATING VIN TRANSIENT VOLTAGE SUPPRESSOR GENERAL INSTRUMENT 1.5KA24A LTC1435A 1435A F09 Figure 9. Automotive Application Protection LTC1435A U U W U APPLICATIONS INFORMATION conduct during double battery operation, but must still clamp the input voltage below breakdown of the converter. Although the LTC1435A has a maximum input voltage of 36V, most applications will be limited to 30V by the MOSFET BVDSS. Design Example As a design example, assume VIN = 12V(nominal), VIN = 22V(max), VOUT = 1.6V, IMAX = 3A and f = 250kHz, RSENSE and COSC can immediately be calculated: COSC = 1.37(104)/250 – 11 = 43pF Referring to Figure 3, a 4.7µH inductor falls within the recommended range. To check the actual value of the ripple current the following equation is used: V V ∆IL = OUT 1– OUT ( f)(L) VIN The highest value of the ripple current occurs at the maximum input voltage: 1.6V 1.6V ∆IL = 1– = 1.3A 250kHz 4.7µH 22V ) The lowest duty cycle also occurs at maximum input voltage. The on-time during this condition should be checked to make sure it doesn’t violate the LTC1435A’s minimum on-time and cause cycle skipping to occur. The required ontime at VIN(MAX) is: tON(MIN) = ( VOUT = 1.6V )( ) ( )( VIN(MAX ) f 22V 250kHz 1.6V 3 22V 2 1.85 The most stringent requirement for the synchronous N-channel MOSFET occurs when VOUT = 0 (i.e. short circuit). In this case the worst-case dissipation rises to: ( PSYNC = ISC( AVG) ) (1+ δ ) RDS(ON) 2 With the 0.033Ω sense resistor ISC(AVG) = 4A will result, increasing the Si4412DY dissipation to 950mW at a die temperature of 105°C. RSENSE = 100mV/3A = 0.033Ω ( ( ) [1+ (0.005)(50°C − 25°C)](0.042Ω) + 2.5 (22V ) (3A )(100pF )(250kHz) = 88 mW PMAIN = ) = 291ns The ∆IL was previously calculated to be 1.3A, which is 43% of IMAX. From Figure 7, the LTC1435A minimum on-time at 43% ripple is about 235ns. Therefore, the minimum ontime is sufficient and no cycle skipping will occur. The power dissipation on the topside MOSFET can be easily estimated. Choosing a Siliconix Si4412DY results in: RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input voltage with T(estimated) = 50°C: CIN is chosen for an RMS current rating of at least 1.5A at temperature. COUT is chosen with an ESR of 0.03Ω for low output ripple. The output ripple in continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR(∆IL) = 0.03Ω(1.3A) = 39mVP-P PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1435A. These items are also illustrated graphically in the layout diagram of Figure 10. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1435A signal ground pin must return to the (–) plate of COUT. The power ground connects to the source of the bottom N-channel MOSFET, anode of the Schottky diode, and (–) plate of CIN, which should have as short lead lengths as possible. 2. Does the VOSENSE pin connect directly to the feedback resistors? The resistive divider R1, R2 must be connected between the (+) plate of COUT and signal ground. The 100pF capacitor should be as close as possible to the LTC1435A. 3. Are the SENSE – and SENSE + leads routed together with minimum PC trace spacing? The filter capacitor between SENSE + and SENSE – should be as close as possible to the LTC1435A. 15 LTC1435A U U W U APPLICATIONS INFORMATION 4. Does the (+) plate of CIN connect to the drain of the topside MOSFET(s) as closely as possible? This capacitor provides the AC current to the MOSFET(s). 6. Keep the switching node SW away from sensitive smallsignal nodes. Ideally the switch node should be placed at the furthest point from the LTC1435A. 5. Is the INTVCC decoupling capacitor connected closely between INTVCC and the power ground pin? This capacitor carries the MOSFET driver peak currents. 7. SGND should be exclusively used for grounding external components on COSC, ITH, VOSENSE and SFB pins. 8. If operating close to the minimum on-time limit, is the sense resistor oriented on the radial axis of the inductor? See Figure 8. + M1 1 CSS COSC 2 CIN 15 BOOST VIN CC1 RC 16 TG RUN/SS + COSC 3 CC2 ITH 4 LTC1435A SFB 5 14 SW 13 VIN INTVCC SGND DB VOSENSE BG 7 SENSE – PGND 8 SENSE + EXTVCC D1 – 100pF 6 CB 0.1µF 12 + 11 M2 4.7µF 10 1000pF 9 L1 – R1 + R2 COUT VOUT RSENSE BOLD LINES INDICATE HIGH CURRENT PATHS + 1435A F10 Figure 10. LTC1435A Layout Diagram U TYPICAL APPLICATIONS Intel Mobile CPU VID Power Converter 1 COSC 43pF 2 CSS 0.1µF 3 CC 1000pF CC2 220pF COSC VIN RUN/SS TG ITH SW INTVCC 50pF 6 SGND BOOST VOSENSE SENSE – 7 BG PGND SENSE + 1000pF 8 VIN 4.5V TO 22V 16 CF 0.1µF 14 DB CMDSH-3 LTC1435A RC 10k 5 4.7Ω 13 12 M1 Si4410 + CIN 10µF 30V ×2 RSENSE 0.015Ω VOUT 1.3V TO 2.0V 7A L1 3.3µH 3 0.22µF 15 5 + 11 10 4.7µF M2 Si4410 D1 MBRS140T3 6 VCC LTC1706-19 FB VID 0 1 2 3 GND 7 8 1 2 FROM µP 1435A TA07 16 SENSE 4 + COUT 820µF 4V ×2 LTC1435A U TYPICAL APPLICATIONS Dual Output 5V and Synchronous 12V Application VIN 5.4V TO 28V COSC 68pF 1 CSS 0.1µF RC 10k 2 CC1 470pF 3 CC2 51pF 4 COSC TG RUN/SS BOOST ITH SW SFB VIN LTC1435A 5 M1 Si4412DY 14 T1 10µH 1:1.42 13 7 VOSENSE BG SENSE – PGND 0.1µF 12 + RSENSE 0.033Ω + 100pF 6 IRLL014 4.7k 15 CMDSH-3 INTVCC SGND 16 0.01µF CIN 22µF 35V ×2 + M2 Si4412DY MBRS140T3 SENSE + EXTVCC COUT 100µF 10V ×2 + 10 R2 20k 1% 1000pF 8 VOUT 5V/3.5A R1 35.7k 1% 4.7µF 11 CSEC 3.3µF 35V 9 100Ω SGND 100Ω 11.3k 1% 100k 1% 1435A TA04 T1: DALE LPE6562-A236 VOUT2 12V 120mA 3.3V/4.5A Converter with Foldback Current Limiting VIN 4.5V TO 28V COSC 68pF CSS 0.1µF RC 10k CC2 51pF CC1 330pF INTVCC CIN 22µF 35V ×2 + 1 2 3 4 COSC TG RUN/SS BOOST ITH SW SFB LTC1435A 5 VIN 15 14 ITH PIN 3 13 7 VOSENSE SENSE – IN4148 BG PGND SENSE + EXTVCC RSENSE 0.025Ω + 11 M2 Si4410DY 10 9 MBRS140T3 100pF OPTIONAL: CONNECT TO 5V VOUT 3.3V/4.5A R1 35.7k 1% 4.7µF 1000pF 8 L1 10µH 0.1µF 12 100pF 6 M1 Si4410DY CMDSH-3 INTVCC SGND 16 R2 20k 1% + COUT 100µF 10V ×2 SGND (PIN 5) 1435A TA01 17 LTC1435A U TYPICAL APPLICATIONS Constant-Current/Constant-Voltage High Efficiency Battery Charger E1 VIN + C1* 22µF 35V E3 GND E3 SHDN + C2* 22µF 35V R7 1.5M C4 0.1µF C11 56pF C5 0.1µF LTC1435A C13 0.033µF R5 1k 1 C12 0.1µF 2 3 C14 1000pF 4 5 C9 100pF C15 0.1µF LT1620 1 2 3 4 AVG SENSE IOUT PROG GND VCC NIN 6 PIN COSC TG RUN/SS BOOST SW ITH VIN SFB SGND INTVCC VOSENSE BG 7 SENSE – PGND 8 SENSE + EXTVCC 16 Q1 Si4412DY 15 D1 14 13 + C6 0.33µF D2 11 Q2 Si4412DY 10 9 + 7 C8 100pF C7 4.7µF 16V 6 R2 1M 0.1% 5 R3 105k 0.1% JP1A C16 0.33µF R6 10k 1% C17 0.01µF R1 0.025Ω 12 C10 100pF 8 L1 27µH C18 0.1µF R4 76.8k 0.1% JP1B 1435A TA06 E5 GND RPROG E4 IPROG *CONSULT CAPACITOR MANUFACTURER FOR RECOMMENDED ESR RATING FOR CONTINUOUS 4A OPERATION Current Programming Equation )(R6) – 0.04 (I IBATT = PROG 10(R1) Efficiency 100 VIN = 24V VBATT = 16V 95 EFFICIENCY (%) VBATT = 12V 90 VBATT = 6V 85 80 75 0 1 3 4 2 BATTERY CHARGE CURRENT (A) 5 1435A TA05 18 C3 22µF 35V E6 BATT E7 GND LTC1435A U TYPICAL APPLICATIONS Dual Output 5V and 12V Application VIN 5.4V TO 28V COSC 68pF 1 CSS 0.1µF RC 10k CIN 22µF 35V ×2 + 2 CC1 510pF 3 CC2 51pF 4 COSC RUN/SS ITH VIN CMDSH-3 7 SENSE – VOUT 5V/3.5A RSENSE 0.033Ω 4.7µF 11 M2 IRF7403 R1 35.7k 1% MBRS140T3 PGND SENSE + COUT 100µF 10V ×2 + 10 R2 20k 1% 1000pF 8 CSEC 3.3µF 25V 0.1µF + BG VOSENSE + 12 100pF 6 24V T1 10µH 1:2.2 13 INTVCC SGND MBRS1100T3 14 SW SFB M1 IRF7403 15 BOOST LTC1435A 5 16 TG 9 EXTVCC 100Ω SGND 100Ω 10k 90.9k VOUT2 12V 1435A TA02 T1: DALE LPE6562-A092 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. G Package 16-Lead Plastic SSOP (0.209) (LTC DWG # 05-08-1640) 0.205 – 0.212** (5.20 – 5.38) 0.239 – 0.249* (6.07 – 6.33) 16 15 14 13 12 11 10 9 0.068 – 0.078 (1.73 – 1.99) 0° – 8° 0.005 – 0.009 (0.13 – 0.22) 0.022 – 0.037 (0.55 – 0.95) 0.0256 (0.65) BSC *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.301 – 0.311 (7.65 – 7.90) 0.010 – 0.015 (0.25 – 0.38) 0.002 – 0.008 (0.05 – 0.21) 1 2 3 4 5 6 7 8 G16 SSOP 1197 S Package 16-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0.004 – 0.010 (0.101 – 0.254) 0.386 – 0.394* (9.804 – 10.008) 16 15 14 13 12 11 10 9 0° – 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.050 (1.270) TYP 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) S16 0695 1 2 3 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 4 5 6 7 8 19 LTC1435A U TYPICAL APPLICATION Low Dropout 2.9V/3A Converter VIN 3.5V TO 20V COSC 68pF 1 CSS 0.1µF RC 10k 2 CC1 330pF CC2 51pF INTVCC 3 4 COSC TG RUN/SS BOOST ITH SW SFB VIN LTC1435A 5 SGND 16 14 13 CMDSH-3 INTVCC 7 VOSENSE BG SENSE – PGND CIN 22µF 35V ×2 + 15 L1 10µH 0.1µF 12 RSENSE 0.033Ω VOUT 2.9V/3A + 100pF 6 M1 1/2 Si9802DY 4.7µF 11 M2 1/2 Si9802DY MBRS140T3 100pF SENSE + EXTVCC + 10 R2 24.9k 1% 1000pF 8 R1 35.7k 1% 9 OPTIONAL: CONNECT TO 5V COUT 100µF 10V ×2 SGND 1435A TA03 L1: SUMIDA CDRH125-10 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1142HV/LTC1142 Dual High Efficiency Synchronous Step-Down Switching Regulators Dual Synchronous, VIN ≤ 20V LTC1148HV/LTC1148 High Efficiency Sychronous Step-Down Switching Regulator Controllers Synchronous, VIN ≤ 20V LTC1159 High Efficiency Synchronous Step-Down Switching Regulator Synchronous, VIN ≤ 40V, For Logic Threshold FETs LT 1375/LT1376 1.5A, 500kHz Step-Down Switching Regulators High Frequency, Small Inductor, High Efficiency Switchers, 1.5A Switch LTC1430 High Power Step-Down Switching Regulator Controller High Efficiency 5V to 3.3V Conversion at Up to 15A LTC1436A/LTC1436A-PLL/ LTC1437A High Efficiency Low Noise Synchronous Step-Down Switching Regulators Full-Featured Single Controller LTC1438/LTC1439 Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulators Full-Featured Dual Controllers ® LT1510 Constant-Voltage/ Constant-Current Battery Charger 1.3A, Li-Ion, NiCd, NiMH, Pb-Acid Charger LTC1538-AUX Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulator 5V Standby in Shutdown LTC1539 Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulator 5V Standby in Shutdown LTC1706-19 VID Voltage Programmer Creates a Programmable 1.3V to 2V Supply for Intel Mobile Pentium® II Processor When Used with the LTC1435A Pentium is a registered trademark of Intel Corporation. 20 Linear Technology Corporation 1435af LT/GP 0798 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1998