19-3505; Rev 0; 11/04 KIT ATION EVALU E L B A AVAIL Low-Dropout, Wide-Input-Voltage, Step-Down Controllers The MAX8597/MAX8598/MAX8599 voltage-mode PWM step-down controllers are designed to operate from a 4.5V to 28V input supply and generate output voltages down to 0.6V. A proprietary switching algorithm stretches the duty cycle to >99.5% for low-dropout design. Unlike conventional step-down regulators using a pchannel high-side MOSFET to achieve high duty cycle, the MAX8597/MAX8598/MAX8599 drive n-channel MOSFETs resulting in high efficiency and high-currentcapability designs. The MAX8597 is available in a 20-pin thin QFN package and is designed for applications that use an analog signal to control the output voltage with an adjustable offset, such as DC fan-speed control. This is achieved with an internal uncommitted operational amplifier. The MAX8597 is also targeted for tracking output-voltage applications for chipsets, ASIC and DSP cores, and I/O supplies. The MAX8598/MAX8599 are available in a 16pin thin QFN package and do not have the uncommitted operational amplifier, reference input, and reference output, but offer an open-drain, power-OK output. The MAX8597/MAX8598/MAX8599 allow startup with prebias voltage on the output for applications where a backup supply or a tracking device may charge the output capacitor before the MAX8597/MAX8598/ MAX8599 are enabled. In addition, the MAX8599 features output overvoltage protection. These controllers also feature lossless high-side peak inductor current sensing, adjustable current limit, and hiccup-mode short-circuit protection. Switching frequency is set with an external resistor from 200kHz to 1.4MHz. This wide frequency range combined with a wide-bandwidth error amplifier enables the loop compensation scheme to give the user ample flexibility to optimize for cost, size, and efficiency. Features ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ Low Dropout with >99.5% Duty Cycle Lossless High-Side Current Limit Wide 4.5V to 28V Input Range Dynamic Output Voltage Adjustment with Adjustable Offset (MAX8597) Remote Voltage Sensing for Both Positive and Negative Rails (MAX8597) Tracking Output Through REFIN (MAX8597) Adjustable Switching Frequency from 200kHz to 1.4MHz Adjustable Soft-Start Prebias Startup Enable and Power-OK (MAX8598/MAX8599) for Flexible Sequencing 25MHz Error Amplifier Adjustable Hiccup Current Limit for Output Short-Circuit Protection Output Overvoltage Protection (MAX8599) Small, Low-Profile Thin QFN Package Ordering Information PART TEMP RANGE PIN-PACKAGE MAX8597ETP+ -40°C to +85°C 20 Thin QFN 4mm x 4mm (T2044-3) MAX8598ETE+ -40°C to +85°C 16 Thin QFN 4mm x 4mm (T1644-4) MAX8599ETE+ -40°C to +85°C 16 Thin QFN 4mm x 4mm (T1644-4) +Denotes lead-free package. Pin Configurations Nonisolated Power Modules DH BST PGND DL Applications LX TOP VIEW 15 14 13 12 11 ILIM 16 10 VL Variable-Speed DC Fan Power Supplies (MAX8597) FREQ 17 9 V+ Tracking Power Supplies (MAX8597) AOUT 18 8 REFOUT AIN- 19 7 EN AIN+ 20 6 COMP 1 2 3 4 5 AVL REFIN GND SS FB Chipset Power Supplies MAX8597 THIN QFN 4mm x 4mm Pin Configurations continued at end of data sheet ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX8597/MAX8598/MAX8599 General Description MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers ABSOLUTE MAXIMUM RATINGS V+, ILIM to GND .....................................................-0.3V to +30V AVL, VL to GND........................................................-0.3V to +6V PGND to GND .......................................................-0.3V to +0.3V FB, EN, POK, AIN-, AIN+, REFIN to GND ................-0.3V to +6V AOUT, REFOUT, FREQ, SS, COMP to GND .....................................................-0.3V to (VAVL + 0.3V) BST to GND ............................................................-0.3V to +36V DH to LX ....................................................-0.3V to (VBST + 0.3V) LX to GND ........................-2V (-2.5V for less than 50ns) to +30V LX to BST..................................................................-6V to +0.3V DL to PGND.................................................-0.3V to (VVL + 0.3V) Continuous Power Dissipation 16- or 20-Pin Thin QFN Up to +70°C (derate 16.9mW/°C above +70°C)........1349mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VV+ = VVL = VAVL = VEN = VREFIN = 5V, VBST = 6V, VLX = 1V, CVL = 4.7µF, CREFOUT = 1µF, VAIN- = VAOUT, VAIN+ = 2.5V, VILIM = VLX - 0.2V, VFB = 0.65V, GND = PGND = 0V, CSS = 0.01µF, RFREQ = 20kΩ, TA = 0°C to +85°C, typical values are at TA = +25°C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS 5.5 28.0 V 4.5 5.5 V 5.0 mA GENERAL V+ Operating Range V+/VL Operating Range V+ = VL V+ Operating Supply Current VV+ = 12V, VL unloaded, no MOSFETs connected, VFB = 0V 3.4 V+ Standby Supply Current VV+ = 12V, VL unloaded, VFB = 0V 2.0 mA VL REGULATOR Output Voltage 5.5V < VV+ < 28V, 1mA < ILOAD < 35mA 4.7 5.0 5.3 V VL Undervoltage-Lockout Trip Level Rising edge, typical hysteresis = 460mV 4.05 4.2 4.35 V Thermal Shutdown Rising temperature, typical hysteresis = 10°C +160 °C REFERENCE (MAX8597 only) REFOUT Output Voltage IREFOUT = 150µA, VV+ = VVL = 4.5V or 5.5V REFOUT Load Regulation IREFOUT = 10µA to 1mA REFOUT Internal Discharge Switch On-Resistance During VL UVLO 2.49 2.50 2.51 V 10 mV Ω 15 CURRENT-LIMIT COMPARATOR (all current limits are tested at VV+ = VVL = 4.5V and 5.5V) ILIM Sink Current 1.8V < VLX < 28V, VBST = VLX + 5V 180 Comparator Input Offset Voltage Error VLX = 28V, VBST = VLX + 5V -10 200 220 µA +10 mV SOFT-START Soft-Start Source Current VSS = 100mV 3 5 7 µA Soft-Start Sink Current VSS = (0.6V or VREFIN) 3 5 7 µA FREQUENCY Frequency 2 RFREQ = 100kΩ 150 200 240 RFREQ = 20.0kΩ 800 1000 1200 RFREQ = 14.3kΩ 1100 1400 1700 _______________________________________________________________________________________ kHz Low-Dropout, Wide-Input-Voltage, Step-Down Controllers (VV+ = VVL = VAVL = VEN = VREFIN = 5V, VBST = 6V, VLX = 1V, CVL = 4.7µF, CREFOUT = 1µF, VAIN- = VAOUT, VAIN+ = 2.5V, VILIM = VLX - 0.2V, VFB = 0.65V, GND = PGND = 0V, CSS = 0.01µF, RFREQ = 20kΩ, TA = 0°C to +85°C, typical values are at TA = +25°C, unless otherwise noted.) PARAMETER CONDITIONS DH Minimum Off-Time MIN TYP MAX UNITS 180 200 220 ns 115 140 ns 100 nA 0.600 0.606 V -10 mV DH Minimum On-Time FB ERROR AMPLIFIER FB Input Bias Current FB Input Voltage Set Point Over load and line FB Offset Error VREFIN = 1.25V and 2.5V, measured with respect to REFIN Error-Amp Open-Loop Voltage Gain VCOMP = 1.2V to 2.4V Slew Rate CLOAD = 80pF 0.594 +10 72 90 dB 18 V/µs UNCOMMITTED OPERATIONAL AMPLIFIER (MAX8597 only) Open-Loop Voltage Gain (AVOL) RLOAD = 100kΩ 90 RLOAD = 10kΩ 70 Output-Voltage Swing High VAIN+ = 2.5V, VAIN- = (VAIN+ - 100mV), ISOURCE = 100µA Output-Voltage Swing Low VAIN+ = 2.5V, VAIN- = (VAIN+ + 100mV), ISINK = 100µA dB VAVL 20mV V 20 Unity-Gain BW 1.5 CLOAD = 10pF, RLOAD = 10kΩ to 100kΩ +80 CLOAD = 100pF, RLOAD = 10kΩ to 100kΩ +40 Slew Rate CLOAD = 100pF 3.5 Input Offset Voltage VCM = 1.25V and 2.5V Phase Margin Input Leakage Current Input Common-Mode Range (CMVR) mV MHz Degrees V/µs -3 +3 mV -10 +10 nA +0.50 VAVL 2.0 V Common-Mode Rejection Ratio (CMRR) 75 dB DRIVERS DH, DL Break-Before-Make Time CLOAD = 2000pF 20 DH On-Resistance in Low State VBST - VLX = 5V 1.0 2.5 ns Ω DH On-Resistance in High State VBST - VLX = 5V 1.5 3.3 Ω DL On-Resistance in Low State VVL = VV+ = 5V 0.45 1.0 Ω DL On-Resistance in High State VVL = VV+ = 5V 1.3 2.5 Ω BST Bias Current VBST = 33V, VLX = 28V, VEN = 0V 230 520 µA LX Bias Current VBST = 33V, VLX = 28V, VEN = 0V -230 -520 µA BST/LX Leakage Current VBST = VLX = 28V, VEN = 0V 50 µA 0.80 V LOGIC INPUTS (EN) Input Low Level 4.5V < VVL = VV+ = VAVL < 5.5V Input High Level 4.5V < VVL = VV+ = VAVL < 5.5V Input Bias Current VVL = VV+ = VAVL = 5.5V, VEN = 0 to 5.5V 1.14 2.40 -1 1.73 V +1 µA _______________________________________________________________________________________ 3 MAX8597/MAX8598/MAX8599 ELECTRICAL CHARACTERISTICS (continued) MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers ELECTRICAL CHARACTERISTICS (continued) (VV+ = VVL = VAVL = VEN = VREFIN = 5V, VBST = 6V, VLX = 1V, CVL = 4.7µF, CREFOUT = 1µF, VAIN- = VAOUT, VAIN+ = 2.5V, VILIM = VLX - 0.2V, VFB = 0.65V, GND = PGND = 0V, CSS = 0.01µF, RFREQ = 20kΩ, TA = 0°C to +85°C, typical values are at TA = +25°C, unless otherwise noted.) PARAMETER CONDITIONS MIN TYP MAX UNITS 0 2.75 V VAVL 1.0 VAVL 0.5 V -250 +250 nA REFIN INPUT (MAX8597 only) REFIN Input Voltage Range REFIN Dual Mode™ Threshold REFIN Input Bias Current VREFIN = 1.25V or 2.5V OV AND UV FAULT COMPARATORS Upper FB Fault Threshold (OV) Rising edge, hysteresis = 15mV (MAX8599 only) 115 117 120 % Lower FB Fault Threshold (UV) Falling edge, hysteresis = 15mV 67 70 73 % POWER-OK OUTPUT (POK) (MAX8598/MAX8599 only) Clock cycles POK Delay For both FB rising and falling edges 8 Lower FB POK Threshold FB falling, hysteresis = 20mV POK Output Low Level ISINK = 2mA 0.4 V POK Output High Leakage VPOK = 5.5V 5 µA 85 88 90 % ELECTRICAL CHARACTERISTICS (VV+ = VVL = VAVL = VEN = VREFIN = 5V, VBST = 6V, VLX = 1V, CVL = 4.7µF, CREFOUT = 1µF, VAIN- = VAOUT, VAIN+ = 2.5V, VILIM = VLX - 0.2V, VFB = 0.65V, GND = PGND = 0V, CSS = 0.01µF, RFREQ = 20kΩ, TA = -40°C to +85°C, typical values are at TA = +25°C, unless otherwise noted.) (Note 1) PARAMETER CONDITIONS MIN TYP MAX UNITS 5.5 28.0 V 4.5 5.5 V 5.0 mA GENERAL V+ Operating Range V+/VL Operating Range V+ = VL V+ Operating Supply Current VV+ = 12V, VL unloaded, no MOSFETs connected, VFB = 0V VL REGULATOR Output Voltage 5.5V < VV+ < 28V, 1mA < ILOAD < 35mA 4.7 5.3 V VL Undervoltage-Lockout Trip Level Rising edge, typical hysteresis = 460mV 4.05 4.35 V REFOUT Output Voltage IREFOUT = 150µA, VV+ = VVL = 4.5V or 5.5V 2.47 2.51 V REFOUT Load Regulation IREFOUT = 10µA to 1mA 10 mV 180 220 µA -10 +10 mV REFERENCE (MAX8597 only) CURRENT-LIMIT COMPARATOR (all current limits are tested at VV+ = VVL = 4.5V and 5.5V) ILIM Sink Current VILIM = VLX - 0.2V, 1.8V < VLX < 28V, VBST = VLX + 5V Comparator Input Offset Voltage Error SOFT-START Soft-Start Source Current VSS = 100mV 3 7 µA Soft-Start Sink Current VSS = (0.6V or VREFIN) 3 7 µA Dual Mode is a trademark of Maxim Integrated Products, Inc. 4 _______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers (VV+ = VVL = VAVL = VEN = VREFIN = 5V, VBST = 6V, VLX = 1V, CVL = 4.7µF, CREFOUT = 1µF, VAIN- = VAOUT, VAIN+ = 2.5V, VILIM = VLX - 0.2V, VFB = 0.65V, GND = PGND = 0V, CSS = 0.01µF, RFREQ = 20kΩ, TA = -40°C to +85°C, typical values are at TA = +25°C, unless otherwise noted.) (Note 1) PARAMETER CONDITIONS MIN TYP MAX UNITS FREQUENCY Frequency RFREQ = 100kΩ 140 240 RFREQ = 20.0kΩ 800 1200 RFREQ = 14.3kΩ 1100 1700 180 230 ns 140 ns 150 nA DH Minimum Off-Time DH Minimum On-Time kHz FB ERROR AMPLIFIER FB Input Bias Current FB Input Voltage Set Point Over load and line FB Offset Error VREFIN = 1.25V and 2.5V, measured with respect to REFIN Error-Amp Open-Loop Voltage Gain VCOMP = 1.2V to 2.4V 0.591 0.606 V +20 -20 mV 72 dB VAVL 20mV V UNCOMMITTED OPERATIONAL AMPLIFIER (MAX8597 only) Output Voltage Swing High VAIN+ = 2.5V, VAIN- = (VAIN+ - 100mV), ISOURCE = 100µA Output Voltage Swing Low VAIN+ = 2.5V, VAIN- = (VAIN+ + 100mV), ISINK = 100µA Input Offset Voltage VCM = 1.25V and 2.5V Input Common-Mode Range (CMVR) 20 mV -3 +3 mV +0.50 VAVL 2.0 V DRIVERS DH On-Resistance in Low State VBST - VLX = 5V 2.5 Ω DH On-Resistance in High State VBST - VLX = 5V 3.3 Ω DL On-Resistance in Low State VVL = VV+ = 5V 1.0 Ω DL On-Resistance in High State VVL = VV+ = 5V 3.5 Ω BST Bias Current VBST = 33V, VLX = 28V, VEN = 0V 520 µA LX Bias Current VBST = 33V, VLX = 28V, VEN = 0V -520 µA BST/LX Leakage Current VBST = VLX = 28V, VEN = 0V 50 µA 0.8 V LOGIC INPUTS (EN) Input Low Level 4.5V < VVL = VV+ = VAVL < 5.5V Input High Level 4.5V < VVL = VV+ = VAVL < 5.5V 2.4 Input Bias Current VVL = VV+ = VAVL = 5.5V, VEN = 0 to 5.5V -1 +1 µA REFIN Input Voltage Range 0 2.75 V REFIN Dual-Mode Threshold VAVL 1.0 VAVL 0.5 V -250 +250 nA V REFIN INPUT (MAX8597 only) REFIN Input Bias Current VREFIN = 1.25V or 2.5V _______________________________________________________________________________________ 5 MAX8597/MAX8598/MAX8599 ELECTRICAL CHARACTERISTICS (continued) ELECTRICAL CHARACTERISTICS (continued) (VV+ = VVL = VAVL = VEN = VREFIN = 5V, VBST = 6V, VLX = 1V, CVL = 4.7µF, CREFOUT = 1µF, VAIN- = VAOUT, VAIN+ = 2.5V, VILIM = VLX - 0.2V, VFB = 0.65V, GND = PGND = 0V, CSS = 0.01µF, RFREQ = 20kΩ, TA = -40°C to +85°C, typical values are at TA = +25°C, unless otherwise noted.) (Note 1) PARAMETER CONDITIONS MIN TYP MAX UNITS OV AND UV FAULT COMPARATORS Upper FB Fault Threshold (OV) Rising edge, hysteresis = 15mV (MAX8599 only) 115 120 % Lower FB Fault Threshold (UV) Falling edge, hysteresis = 15mV 67 73 % 85 90 % 0.4 5 µA POWER-OK OUTPUT (POK) (MAX8598/MAX8599 only) Lower FB POK Threshold FB falling, hysteresis = 20mV POK Output Low Level ISINK = 2mA POK Output High Leakage VPOK = 5.5V V Note 1: Limits to -40°C are guaranteed by design and characterization. Typical Operating Characteristics (Circuit of Figure 4, TA = +25°C, 500kHz switching frequency, VIN = 12V, unless otherwise noted.) EFFICIENCY vs. LOAD CURRENT CIRCUIT OF FIGURE 2 90 80 85 EFFICIENCY (%) EFFICIENCY (%) 90 VOUT = 3.3V VOUT = 6V 80 75 OUTPUT VOLTAGE (V) VOUT = 9V VOUT = 2.5V VOUT = 1.8V 70 VOUT = 1.2V 60 50 70 1.210 MAX8597 toc02 VOUT = 11.5V 95 100 MAX8597 toc01 100 OUTPUT VOLTAGE vs. LOAD CURRENT MAX8597 toc03 EFFICIENCY vs. LOAD CURRENT CIRCUIT OF FIGURE 1 1.205 1.200 1.195 40 65 30 60 0.1 1.190 1 10 1 100 10 LOAD CURRENT (A) LOAD CURRENT (A) OUTPUT VOLTAGE vs. INPUT VOLTAGE POWER-UP WAVEFORMS 0 2 4 6 8 10 12 14 16 18 20 ILOAD (A) POWER-DOWN WAVEFORMS MAX8597 toc06 MAX8597 toc05 MAX8597 toc04 1.210 OUTPUT VOLTAGE (V) MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers 5V/div VAVL 5V/div VAVL 1.205 ILOAD = 0A 1.200 ILX 10A/div ILX 10A/div 1V/div IOUT 1V/div 10V/div VIN 10V/div ILOAD = 20A VOUT 1.195 VIN 1.190 10.0 10.5 11.0 11.5 12.0 12.5 13.0 13.5 14.0 2ms/div 2ms/div VIN (V) 6 _______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers STARTUP/SHUTDOWN WITH EN (ILOAD = 20A) OUTPUT PREBIASED STARTUP MAX8597 toc07 MAX8597 toc08 5V/div VPOK 10V/div VLX 1.2V ILX 10A/div 1.0V VOUT 5V/div 1V/div VOUT VIN 5V/div VDL 5V/div VEN 1ms/div 2ms/div ENTERING DROPOUT WAVEFORMS CIRCUIT OF FIGURE 1 OUTPUT VOLTAGE vs. VADJ (VIN = 12V) MAX8597 toc10 MAX8597 toc09 13 12 OUTPUT VOLTAGE (V) 11 VLX 10V/div 11V 10 VOUT 9 8 7 VCOMP 500mV/div VIN (AC-COUPLED) 100mV/div 6 CIRCUIT OF FIGURE 1 RLOAD = 1.2Ω 5 4 0 1 2 3 4 5 6 2µs/div VADJ (V) HEAVY-DROPOUT WAVEFORMS CIRCUIT OF FIGURE 1 OUTPUT TRACKING REFIN MAX8597 toc11 VLX MAX8597 toc12 VREFIN 10V/div 11.9V VOUT VOUT 1V/div 1ms RISE TIME VREFIN 500mV/div VCOMP VOUT 5ms RISE TIME 1V/div 500mV/div VIN (AC-COUPLED) CIRCUIT OF FIGURE 3 10µs/div 1ms/div _______________________________________________________________________________________ 7 MAX8597/MAX8598/MAX8599 Typical Operating Characteristics (continued) (Circuit of Figure 4, TA = +25°C, 500kHz switching frequency, VIN = 12V, unless otherwise noted.) MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers Typical Operating Characteristics (continued) (Circuit of Figure 4, TA = +25°C, 500kHz switching frequency, VIN = 12V, unless otherwise noted.) 90% LOAD STEP AT 5A/µs 50% LOAD STEP AT 5A/µs MAX8597 toc14 MAX8597 toc13 VOUT (AC-COUPLED) 100mV/div 50mV/div VOUT (AC-COUPLED) 20A 20A 10A IOUT IOUT 2A 40µs/div 40µs/div OUTPUT OVERVOLTAGE PROTECTION SHORT-CIRCUIT RESPONSE MAX8597 toc16 MAX8597 toc15 VOUT 1V/div IIN VFB 500mV/div VDH 10V/div 2A/div ILX 5V/div 10A/div VDL 4ms/div 8 10µs/div _______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers ILIM MAX8597 MAX8598 MAX8599 200µA BST DH FREQ CONTROL LOGIC OSC LX 1VP-P PWM VL 1/20 COUNTER DL PGND AVL REFERENCE REFOUT (MAX8597) SOFT-START EN BIAS VREG COMP V+ EAMP VL VL GND 1.17 x VREG UVP SS REFIN (MAX8597) OVP (MAX8599) 0.7 x VREG FB POK (MAX8598/ MAX8599) AOUT (MAX8597) N AIN+ (MAX8597) 0.88 x VREG AIN(MAX8597) _______________________________________________________________________________________ 9 MAX8597/MAX8598/MAX8599 Block Diagram MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers Pin Description PIN MAX8597 MAX8598/ MAX8599 NAME 1 1 AVL 2 — REFIN 3 2 GND 4 3 SS Soft-Start Programming Input. Connect a capacitor from SS to GND to set the soft-start time. See the Selecting the Soft-Start Capacitor section for details. 5 4 FB Feedback Input. Connect to the center tap of an external resistor-divider to set the output voltage. Regulates to 0.6V for the MAX8598/MAX8599 and MAX8597 when REFIN is connected to AVL. Regulates to VREFIN (MAX8597) when using an external reference. 6 5 COMP 7 6 EN 8 — REFOUT 9 7 V+ Input Supply Voltage for Internal VL Regulator. Connect to an input supply in the 4.5V to 28V range. Bypass to GND with a 1µF or larger ceramic capacitor through a 3Ω resistor. 10 8 VL Internal 5V Linear-Regulator Output. VL provides power for the internal MOSFET gate drivers. Bypass to PGND with a 1µF or larger ceramic capacitor. VL is always enabled except in thermal shutdown. See the Internal 5V Linear Regulator section for details. 11 9 DL Low-Side Gate-Driver Output. Connect to the gate of the synchronous rectifier. DL swings from PGND to VL. DL is held low during shutdown. 12 10 PGND 13 11 BST Bootstrap Input Supply for the High-Side MOSFET Driver. Connect to the cathode of an external diode from VL and connect a 0.1µF or larger capacitor from BST to LX. 14 12 DH High-Side Gate-Driver Output. Connect to the gate of the high-side MOSFET. DH swings from LX to BST. DH is low (connected to LX) during shutdown. 15 13 LX External Inductor Connection. LX is the low supply for the DH gate driver as well as the sense connection for the current-limit circuitry. Connect LX to the switched side of the inductor as well as the source of the high-side MOSFET and the drain of the synchronous rectifier. 16 14 ILIM 10 FUNCTION Filtered VL Input. Connect to VL through a 10Ω resistor. Bypass to GND with a 0.22µF or larger ceramic capacitor. External Reference Input. FB tracks the voltage input to REFIN. Connect REFIN to AVL to use the internal 0.6V reference. Analog Ground. Connect to the exposed paddle and analog ground plane and then connect to PGND at the output ground. Compensation Input. Connect to the required compensation network. See the Compensation Design section for details. Enable Input. Drive EN high to enable the IC. Drive low to shut down the IC. Internal Reference Output. REFOUT regulates to 2.5V and can source up to 1mA. REFOUT discharges to GND during UVLO. Power Ground. Connect to the synchronous rectifier’s source and PGND plane. Current-Limit Sense Input. Connect a resistor from ILIM to the current-sense point to set the output current limit. See the Setting the Current Limit section for details. ______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers PIN MAX8597 MAX8598/ MAX8599 NAME FUNCTION 17 15 FREQ Frequency Adjust Input. Connect a resistor from FREQ to GND to set the switching frequency. The range of the FREQ resistor is 14.3kΩ to 100kΩ (corresponding to 1400kHz to 200kHz). 18 — AOUT Output of the Uncommitted Operational Amplifier. AOUT is high impedance during undervoltage lockout. 19 — AIN- Inverting Input of the Uncommitted Operational Amplifier 20 — AIN+ Noninverting Input of the Uncommitted Operational Amplifier — 16 POK Power-OK Output. POK is an open-drain output that goes high impedance when the regulator output is greater than 88% of the regulation threshold. POK is low during shutdown. — — EP Exposed Paddle. Connect to analog ground plane for improved thermal performance. Detailed Description The MAX8597/MAX8598/MAX8599 voltage-mode PWM step-down controllers are designed to operate from 4.5V to 28V input and generate output voltages down to 0.6V. A proprietary switching algorithm stretches the duty cycle to >99.5% for low-dropout design. Unlike conventional step-down regulators using a p-channel high-side MOSFET to achieve high duty cycle, the MAX8597/MAX8598/MAX8599 drive n-channel MOSFETs permitting high efficiency and high-current designs. The MAX8597 is available in a 20-pin thin QFN package and is designed for applications that use an analog signal to control the output voltage with adjustable offset, such as DC fan speed control. For example, a 12VDC fan can be driven from 6V to 12V with 12V input power source depending on the system’s cooling requirement to minimize fan noise and power consumption. This is achieved with an internal uncommitted operational amplifier. With the addition of an external RC filter, a PWM input can also be used to control the output voltage. The MAX8597 also generates a tracking output for chipsets, ASICs, and DSP where core and I/O supplies are split and require tracking. In applications where tighter output tolerance is required, the MAX8597 output can be set by an external precision reference source feeding to REFIN. The MAX8598/ MAX8599 are available in a 16-pin thin QFN package and do not have the uncommitted operational amplifier, reference input, and reference output, but offer a powerOK output (POK). With the enable input and POK output, the MAX8598/MAX8599 can easily be configured to have power sequencing of multiple supply rails. The MAX8597/MAX8598/MAX8599 allow startup with prebias voltage on the output for applications where a backup supply or a tracking device may charge the output capacitor before the MAX8597/MAX8598/ MAX8599 are enabled. The MAX8599 has output overvoltage protection. These controllers feature lossless high-side peak inductor current sensing, adjustable current limit, and hiccup-mode short-circuit protection. Switching frequency is set with an external resistor from 200kHz to 1.4MHz. This wide frequency range combined with a wide-bandwidth error amplifier enable the loop-compensation scheme to give the user ample flexibility to optimize for cost, size, and efficiency. DC-DC Controller The MAX8597/MAX8598/MAX8599 step-down DC-DC controllers use a PWM voltage-mode control scheme. An internal high-bandwidth (25MHz) operational amplifier is used as an error amplifier to regulate the output voltage. The output voltage is sensed and compared with an internal 0.6V reference or REFIN (MAX8597) to generate an error signal. The error signal is then compared with a fixed-frequency ramp by a PWM comparator to give the appropriate duty cycle to maintain output voltage regulation. The high-side MOSFET turns on at the rising edge of the internal clock 20ns after DL (the low-side MOSFET gate drive) goes low. The high-side MOSFET turns off once the internal ramp voltage reaches the error-amplifier output voltage. The process repeats for every clock cycle. During the high-side MOSFET on-time, current flows from the input through the inductor to the output capacitor and load. At the moment the high-side MOSFET turns off, the energy stored in the inductor during the on-time is released to support the load as the inductor ______________________________________________________________________________________ 11 MAX8597/MAX8598/MAX8599 Pin Description (continued) MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers current ramps down through the low-side MOSFET body diode; 20ns after DH goes low, the low-side MOSFET turns on, resulting in a lower voltage drop to increase efficiency. The low-side MOSFET turns off at the rising edge of the next clock pulse, and when its gate voltage discharges to zero, the high-side MOSFET turns on and another cycle starts. These controllers also sense peak inductor current and provide hiccup-overload and short-circuit protection (see the Current Limit section). The MAX8597/ MAX8598/MAX8599 operate in forced-PWM mode where the inductor current is always continuous. The controller maintains constant switching frequency under all loads, except under dropout conditions where it skips DL pulses. Current Limit The MAX8597/MAX8598/MAX8599 DC-DC step-down controllers sense the peak inductor current either with the on-resistance of the high-side MOSFET for lossless sensing, or a series resistor for more accurate sensing. When the voltage across the sensing element exceeds the current-limit threshold set with ILIM, the controller immediately turns off the high-side MOSFET. The lowside MOSFET is then turned on to let the inductor current ramp down. As the output load current increases above the ILIM threshold, the output voltage sags because the truncated duty cycle is insufficient to support the load current. When FB falls 30% below its nominal threshold, the output undervoltage protection is triggered and the controller enters hiccup mode to limit power dissipation. This current-limit method allows the circuit to withstand a continuous output short circuit. The MAX8597/MAX8598/MAX8599 current-limit threshold is set by an external resistor that works in conjunction with an internal 200µA current sink (see the Setting the Current Limit section for more details). Synchronous-Rectifier Driver (DL) Synchronous rectification reduces the conduction loss in the rectifier by replacing the normal Schottky catch diode with a low-resistance MOSFET switch. The MAX8597/MAX8598/MAX8599 also use the synchronous rectifier to ensure proper startup of the boost gate-drive circuit. 12 High-Side Gate-Drive Supply (BST) Gate-drive voltage for the high-side n-channel MOSFET is generated by an external flying capacitor and diode boost circuit (D1 and C5 in Figure 1). When the synchronous rectifier is on, C5 is charged from the VL supply through the Schottky diode. When the synchronous rectifier is turned off, the Schottky is reverse biased and the voltage on C5 is stacked above LX to provide the necessary turnon voltage for the high-side MOSFET. A low-current Schottky diode, such as Central Semiconductor’s CMDSH-3, works well for most applications. The capacitor should be large enough to prevent it from charging to excessive voltage, but small enough to adequately charge during the minimum low-side MOSFET on-time, which occurs at minimum input voltage. A capacitor in the 0.1µF to 0.47µF range works well for most applications. Internal 5V Linear Regulator The MAX8597/MAX8598/MAX8599 contain a lowdropout 5V regulator that provides up to 35mA to supply gate drive for the external MOSFETs, and supplies AVL, which powers the IC’s internal circuitry. Bypass the regulator’s output (VL) with 1µF per 10mA of VL load, or greater ceramic capacitor. The current required to drive the external MOSFET can be estimated by multiplying the total gate charge (at VGS = 5V) of the MOSFETs by the switching frequency. Undervoltage Lockout (UVLO) When V VL drops below 3.75V (typ), the MAX8597/ MAX8598/MAX8599s’ undervoltage-lockout (UVLO) circuitry inhibits switching, forces POK (MAX8598/ MAX8599) low, and forces DH and DL low. Once VVL rises above 4.2V (typ), the controller powers up the output in startup mode (see the Startup section). Startup The MAX8597/MAX8598/MAX8599 start switching once all the following conditions are met: 1) EN is high. 2) VVL > 4.2V (typ). 3) Soft-start voltage VSS exceeds VFB. 4) Thermal limit is not exceeded. The third condition ensures that the MAX8597/ MAX8598/MAX8599 do not discharge a prebiased output. Once all of these conditions are met, the IC begins switching and the soft-start cycle is initiated. ______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers The power-OK signal (POK) is an open-drain output that goes high impedance when FB is above 91% of its nominal threshold. There is an eight clock-cycle delay before POK goes high impedance. For 500kHz switching frequency, this delay is typically 16µs. To obtain a logic voltage output, connect a pullup resistor from POK to AVL. A 100kΩ resistor works well for most applications. If unused, connect POK to GND or leave it unconnected. Enable and Soft-Start The MAX8597/MAX8598/MAX8599 are enabled using the EN input. A logic high on EN enables the output of the IC. Conversely, a logic low on EN disables the output. On the rising edge of EN, the controllers enter softstart. Soft-start gradually ramps up the reference voltage seen at the error amplifier to control the output rate of rise and reduce the inrush current during startup. The soft-start period is determined by a capacitor connected from SS to GND (C6 in Figure 1). A 5µA current source charges the external capacitor to the reference voltage (0.6V or VREFIN). The capacitor value is determined as follows: C6 = 5µA × t SS VFB where tSS is the soft-start time in seconds and VFB is 0.6V or VREFIN. The output reaches regulation when soft-start is completed. Output Undervoltage Protection (UVP) Output UVP begins when the controller is at its current limit and VFB is 30% below its nominal threshold. This condition causes the controller to drive DH and DL low and discharges the soft-start capacitor with a 5µA pulldown current until VSS reaches 50mV. Then the controller begins in soft-start mode. If the overload condition still exists, the UVP process begins again. The result is “hiccup” mode, where the controller attempts to restart periodically as long as the overload condition exists. In hiccup mode, the soft-start capacitor voltage ramps up to 112% of the nominal VFB threshold and then ramps down to 50mV. For the MAX8597, VREFIN must be greater than 450mV to trigger UVP. The softstart capacitor voltage then ramps up to 112% of VREFIN and then down to 50mV. Output Overvoltage Protection (OVP, MAX8599) The output voltage is continuously monitored for overvoltage (MAX8599 only). If the output voltage is more than 117% of its nominal set value, OVP is triggered after a 12µs (typ) delay. The MAX8599 latches DH low to turn off the high-side MOSFET, and DL high to turn on the low-side MOSFET to clamp the output to PGND. The latch is reset either by toggling EN or by cycling V+ below the UVLO threshold. Note that DL latching high causes a negative spike at the output due to the energy stored in the output LC at the instant of OVP trip. If the load cannot tolerate this negative spike, add a power Schottky diode across the output to act as a reverse polarity clamp. Thermal-Overload Protection Thermal-overload protection limits the total power dissipation in the MAX8597/MAX8598/MAX8599. When the junction temperature exceeds +160°C, a thermal sensor shuts down the device, forcing DH and DL low, allowing the IC to cool. The thermal sensor turns the part on after the junction temperature cools by 10°C, resulting in a pulsed output during continuous thermaloverload conditions. During a thermal event, the switching converter is turned off, the reference is turned off, the VL regulator is turned off, POK is high impedance, and the soft-start capacitor is discharged. ______________________________________________________________________________________ 13 MAX8597/MAX8598/MAX8599 Power-OK Signal (POK, MAX8598/MAX8599 Only) MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers Design Procedure VIN (10.8V TO 13.2V) R15 3Ω R1 33.2kΩ C2A 10µF C1 1µF VL EN ON OFF D1 CMDSH-3 AVL R4 100kΩ VADJ (0V TO 5V) C15 0.01µF 19 R5 32.4kΩ 20 C6 0.033µF 4 R6 48.7kΩ 18 R9 6.04kΩ R7 48.7kΩ R10 93.1kΩ R8 24.9kΩ 5 C8 4.7pF R12 47kΩ R11 5.1kΩ 2 7 REFIN EN 17 9 FREQ V+ 16 ILIM AIN- BST SS MAX8597 AOUT LX DL FB PGND AVL 1 COMP GND 3 6 C9 820pF R13 10Ω REFO 8 C11 0.22µF Q1 IRF7821 14 C5 0.22µF R16 3Ω 15 VOUT 6V TO 12V/10A L1 1µH Q2 IRF7821 11 R1 2Ω C7A 47µF C7B 47µF C14 2200pF 12 VL 10 C13 1µF C10 100pF C3 0.01µF 13 AIN+ DH C2B 10µF R2 1.21kΩ C12 4.7µF VL Figure 1. MAX8597 (600kHz): Live Adjustable Output Voltage from 6V to 12V at 10A C2A 10µF VIN (10.8V TO 13.2V) R14 3Ω R1 40.2kΩ ON EN OFF D1 CMDSH-3 AVL R3 10kΩ 19 R4 10kΩ 20 BST 4 R9 12.1kΩ 5 C8 39pF R10 16kΩ C3 0.01µF R2 1.65kΩ 13 Q1 R16 3Ω 15 AOUT PGND FB COMP GND 3 6 C9 6800pF C10 0.22µF AVL 1 R11 10Ω REFO 8 VL 10 11 (Q3 = Q4 = IRF7832) 12 Q4 R12 3Ω C7A 470µF C13 2200pF C12 4.7µF C11 1µF VL Figure 2. 1.2V at 20A Output with Remote Sensing 14 VOUT 1.2V/20A L1 0.7µH Q3 DL Q2 (Q1 = Q2 = C5 IRF7807Z) 0.22µF MAX8597 SS C2C 10µF 14 AIN+ LX 18 C8 1800pF 16 ILIM AIN- C4 0.033µF R8 7.2kΩ 17 9 FREQ V+ DH R5 10kΩ R6 10kΩ R7 12.1kΩ 2 7 REFIN EN C2B 10µF VL C1 1µF ______________________________________________________________________________________ C7B 470µF Low-Dropout, Wide-Input-Voltage, Step-Down Controllers VIN (10.8V TO 13.2V) R1 20kΩ ON OFF R12 3Ω EN C2A 10µF C1 1µF VL REFIN R3 70kΩ C4 1000pF D2 CMPD914 19 R4 18.2kΩ R5 6.98kΩ 2 7 17 REFIN EN FREQ 9 V+ 16 ILIM AIN- 20 D1 CMDSH-3 BST AIN+ DH 18 R8 5.6kΩ C10 1000pF DL FB PGND AVL 1 COMP GND 3 6 C8 56pF R7 390Ω LX AOUT 5 R6 10kΩ MAX8597 SS C9 8200pF R9 10Ω REFO 8 C11 0.22µF C3 0.01µF 13 Q1 IRF7807Z 14 C6 2200pF 4 C2B 10µF R2 1.5kΩ C5 0.22µF R16 3Ω 15 Q2 IRF7821 11 C7A R10 2Ω 100µF C7B 100µF C14 2200pF 12 VL 10 VOUT 1.8V/10A L1 0.56µH C13 4.7µF C12 1µF VL Figure 3. MAX8597 1MHz Tracking Supply with Clamp (Output voltage tracks VREFIN from 0V up to the nominal output regulation voltage.) C2A 10µF VIN (10.8V TO 13.2V) R11 3Ω R1 40.2kΩ ON EN OFF D1 CMDSH-3 AVL 6 EN R3 100kΩ POK 16 15 FREQ 7 V+ 14 ILIM BST POK C4 0.033µF DH 3 R4 12.1kΩ 4 FB LX C8 39pF R5 12.1kΩ R6 1.2kΩ C8 1800pF 5 R7 16kΩ DL 12 13 R16 3Ω 9 COMP C9 6800pF GND 2 AVL 1 VL 8 AVL R8 10Ω C10 0.22µF R2 2.26kΩ C3 0.01µF C2C 10µF 11 SS MAX8598 MAX8599 C2B 10µF VL C1 1µF C5 0.22µF Q2 HAT2165H Q1 HAT2168H VOUT 1.2V/20A L1 0.7µH Q3 HAT2165H R9 3Ω C7A 470µF C7B 470µF C12 2200pF PGND 10 C11 4.7µF VL Figure 4. MAX8598/MAX8599 500kHz, 1.2V, 20A Output Power Supply ______________________________________________________________________________________ 15 MAX8597/MAX8598/MAX8599 Design Procedure (continued) MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers Setting the Output Voltage Fixed Output Voltage The output voltage is set by a resistor-divider network from the output to GND with FB at the center tap (R4 and R5 in Figure 4). Select R4 between 5kΩ and 15kΩ and calculate R5 by: R5 = R4 x [( VOUT / VFB) - 1] Live Adjustable Output Voltage (see Figure 1) Using the uncommitted operational amplifier, the MAX8597 can be configured such that the output voltage is adjustable using a voltage source (VADJ). The following parameters must be defined before starting the design: • The minimum desired output voltage, VOUT_MIN • The maximum desired output voltage, VOUT_MAX • The desired input that corresponds to the minimum output voltage, VADJ_MIN • The desired input that corresponds to the maximum output voltage, VADJ_MAX Select VAOUT (uncommitted operational-amplifier output) between 0.05V and 3V and V AOUT_MAX higher than VAOUT_MIN. Calculate the required AIN+ reference (VAIN+) as: VAIN+ = VAOUT _ MAX × VADJ _ MAX − VAOUT _ MIN × VADJ _ MIN (VADJ _ MAX − VADJ _ MIN ) + (VAOUT _ MAX − VAOUT _ MIN ) VAIN+ is set using a resistor-divider from REFOUT to GND (R6 and R7). Select R7 to be approximately 50kΩ as a starting point and then calculate R6 as: R6 = R7 x [(2.5V / VAIN+) - 1] Select R4 to be 100kΩ and calculate R5 as: R5 = (VAIN+ − VAOUT _ MIN ) × R4 (VADJ _ MAX − VAIN+ ) Additionally, to minimize error, R6 and R7 should be chosen such that: R6 × R7 R4 × R5 = R6 + R7 R4 + R5 Inductor Selection There are several parameters that must be examined when determining which inductor is to be used: input voltage, output voltage, load current, switching frequency, and LIR. LIR is the ratio of inductor current ripple to DC load current. A higher LIR value allows for a smaller inductor but results in higher losses and higher output ripple. A good compromise between size and efficiency is a 30% LIR. Once all the parameters are chosen, the inductor value is determined as follows: L = VOUT x ( VIN − VOUT ) VIN x fS x ILOAD(MAX) x LIR where fS is the switching frequency. Choose a standard value close to the calculated value. The exact inductor value is not critical and can be adjusted in order to make trade-offs among size, cost, and efficiency. Lower inductor values minimize size and cost, but also increase the output ripple and reduce the efficiency due to higher peak currents. On the other hand, higher inductor values increase efficiency, but eventually resistive losses due to extra turns of wire exceed the benefit gained from lower AC current levels. Find a lowloss inductor having the lowest possible DC resistance that fits the allotted dimensions. Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well up to 300kHz. The chosen inductor’s saturation current rating must exceed the peak inductor current determined as: ⎛ LIR ⎞ IPEAK = ILOAD(MAX) + ⎜ ⎟ × ILOAD(MAX) ⎝ 2 ⎠ Select R9 between 5kΩ and 15kΩ, then calculate R8 and R10 as follows: R8 = [(VOUT _ MIN − VFB ) × (VFB − VAOUT _ MIN ) + (VOUT _ MAX − VFB ) × (VAOUT _ MAX − VFB )] × R9 ((VOUT _ MAX − VFB ) − (VOUT _ MIN − VFB )) × VFB R10 = ( R8 × R9 × VOUT _ MAX − VFB ) (VFB × R8) + [(VFB − VAOUT _ MIN ) × R9] where VFB is the feedback regulation voltage (0.6V with REFIN connected to AVL). 16 Input Capacitor The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit’s switching. The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents defined by the following equation: IRMS = ILOAD × VOUT × ( VIN − VOUT ) VIN ______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers Output Capacitor The key selection parameters for the output capacitor are the actual capacitance value, the equivalent series resistance (ESR), the equivalent series inductance (ESL), and the voltage-rating requirements, which affect the overall stability, output ripple voltage, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, voltage drop across the capacitor’s ESR, and voltage drop across the capacitor’s ESL, caused by the current into and out of the capacitor. The following equations estimate the worst-case ripple: VRIPPLE = VRIPPLE(ESR) + VRIPPLE(ESL) + VRIPPLE(C) VRIPPLE(ESR) = IP−P × ESR V × ESL VRIPPLE(ESL) = IN L + ESL IP−P VRIPPLE(C) = 8 × COUT × fS ⎛V −V ⎞ ⎛V ⎞ IP−P = ⎜ IN OUT ⎟ × ⎜ OUT ⎟ ⎝ fS × L ⎠ ⎝ VIN ⎠ where IP-P is the peak-to-peak inductor current. The response to a load transient depends on the selected output capacitor. After a load transient, the output instantly changes by (ESR x ∆ILOAD) + (ESL x di/dt). Before the controller can respond, the output deviates further depending on the inductor and output capacitor values. After a short period of time (see the Typical Operating Characteristics), the controller responds by regulating the output voltage back to its nominal state. The controller response time depends on the closedloop bandwidth. With higher bandwidth, the response time is faster, preventing the output capacitor voltage from further deviation from its regulation value. Do not exceed the capacitor’s voltage or ripple current ratings. MOSFET Selection The MAX8597/MAX8598/MAX8599 controllers drive external, logic-level, n-channel MOSFETs as the circuitswitch elements. The key selection parameters are: • On-resistance (RDS(ON)): the lower, the better. • Maximum drain-to-source voltage (VDSS): should be at least 20% higher than the input supply rail at the high-side MOSFET’s drain. • Gate charges (Qg, Qgd, Qgs): the lower, the better. Choose MOSFETs with RDS(ON) rated at VGS = 4.5V. For a good compromise between efficiency and cost, choose the high-side MOSFET that has conduction loss equal to the switching loss at the nominal input voltage and maximum output current. For the low-side MOSFET, make sure it does not spuriously turn on due to dv/dt caused by the high-side MOSFET turning on, resulting in efficiency degrading shoot-through current. MOSFETs with a lower Qgd/Qgs ratio have higher immunity to dv/dt. For proper thermal-management design, the power dissipation must be calculated at the desired maximum operating junction temperature, maximum output current, and worst-case input voltage (for low-side MOSFET, worst case is at VIN(MAX); for high-side MOSFET, it could be either at VIN(MIN) or VIN(MAX)). High-side and low-side MOSFETs have different loss components due to the circuit operation. The low-side MOSFET operates as a zero-voltage switch; therefore, the major losses are the channel-conduction loss (PLSCC) and the body-diode conduction loss (PLSDC): PLSCC = [1 - (VOUT / VIN)] x (ILOAD)2 x RDS(ON) PLSDC = 2 x ILOAD x VF x tdt x fS where VF is the body-diode forward-voltage drop, tdt is the dead-time between the high-side MOSFET and the low-side MOSFET switching transitions, and fS is the switching frequency. The high-side MOSFET operates as a duty-cycle control switch and has the following major losses: the channel-conduction loss (PHSCC), the V-I overlapping switching loss (PHSSW), and the drive loss (PHSDR). The high-side MOSFET does not have body-diode conduction loss because the diode never conducts current: PHSCC = (VOUT / VIN) x ILOAD2 x RDS(ON) Use RDS(ON) at TJ(MAX): PHSSW = VIN x ILOAD x fS x [(Qgs + Qgd) / IGATE] where IGATE is the average DH-high driver output-current capability determined by: IGATE = 2.5 / (RDH + RGATE) ______________________________________________________________________________________ 17 MAX8597/MAX8598/MAX8599 I RMS has a maximum value when the input voltage equals twice the output voltage (VIN = 2 x VOUT), so IRMS(MAX) = ILOAD / 2. Ceramic capacitors are recommended due to their low ESR and ESL at high frequency, with relatively lower cost. Choose a capacitor that exhibits less than 10°C temperature rise at the maximum operating RMS current for optimum long-term reliability. MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers where RDH is the high-side MOSFET driver’s average on-resistance (1.25Ω typ) and RGATE is the internal gate resistance of the MOSFET (typically 0.5Ω to 2Ω): the LX voltage waveform can interfere with the current limit. Below is the procedure for selecting the value of the series RC snubber circuit (R14 and C14 in Figure 1): PHSDR = Qgs x VGS x fS x RGATE / (RGATE + RDH) where VGS ~ VVL = 5V. In addition to the losses above, add approximately 20% more for additional losses due to MOSFET output capacitances and low-side MOSFET body-diode reverse-recovery charge dissipated in the high-side MOSFET that exists, but is not well defined in the MOSFET data sheet. Refer to the MOSFET data sheet for thermal-resistance specification to calculate the PC board area needed to maintain the desired maximum operating junction temperature with the abovecalculated power dissipation. To reduce EMI caused by switching noise, add a 0.1µF or larger ceramic capacitor from the high-side switch drain to the lowside switch source or add resistors in series with DH and DL to slow down the switching transitions. However, adding a series resistor increases the power dissipation of the MOSFETs, so be sure this does not overheat the MOSFETs. The minimum load current must exceed the high-side MOSFET’s maximum leakage plus the maximum LX bias current over temperature. 1) Connect a scope probe to measure VLX to GND, and observe the ringing frequency, fR. 2) Find the capacitor value (connected from LX to GND) that reduces the ringing frequency by half. The circuit parasitic capacitance (CPAR) at LX is then equal to 1/3 the value of the added capacitance above. The circuit parasitic inductance (LPAR) is calculated by: Setting the Current-Limit The MAX8597/MAX8598/MAX8599 controllers sense the peak inductor current to provide constant-current and hiccup current limit. The peak current-limit threshold is set by an external resistor (R2 in Figure 1) together with the internal current sink of 200µA. The voltage drop across the resistor R2 due to the 200µA current sets the maximum peak inductor current that can flow through the high-side MOSFET or the optional currentsense resistor (between the high-side MOSFET source and LX) by the equations below: IPEAK(MAX) = 200µA x R2 / RDSON(HSFET) IPEAK(MAX) = 200µA x R2 / RSENSE The actual corresponding maximum load current is lower than the IPEAK(MAX) by half of the inductor ripple current. If the RDS(ON) of the high-side MOSFET is used for current sensing, use the maximum RDS(ON) at the highest operating junction temperature to avoid false tripping of the current limit at elevated temperature. Consideration should also be given to the tolerance of the 200µA current sink. When the RDS(ON) of the highside MOSFET is used for current sensing, ringing on 18 LPAR = 1 (2π × fR )2 × CPAR The resistor for critical dampening (R14) is equal to 2π x fR x LPAR. Adjust the resistor value up or down to tailor the desired damping and the peak voltage excursion. The capacitor (C14) should be at least 2 to 4 times the value of the CPAR in order to be effective. The power loss of the snubber circuit is dissipated in the resistor (R14) and is calculated as: PR14 = C14 x (VIN)2 x fS where VIN is the input voltage and fS is the switching frequency. Choose an R14 power rating that meets the specific application’s derating rule for the power dissipation calculated. Additionally, there is parasitic inductance of the current-sensing element, whether the high-side MOSFET (L SENSE_FET ) or the optional current-sense resistor (LRSENSE) are used, which is in series with the output filter inductor. This parasitic inductance, together with the output inductor, forms an inductive divider and causes error in the current-sensing voltage. To compensate for this error, a series RC circuit can be added in parallel with the sensing element (see Figure 5). The RC time constant should equal LRSENSE / RSENSE, or LSENSE_FET / RDS(ON). First, set the value of R equal to or less than R2 / 100. Then, the value of C is calculated as: C = LRSENSE / (RSENSE x R) or C = LSENSE_FET / (RDS(ON) x R) Any PC board trace inductance in series with the sensing element and output inductor should be added to the specified FET or resistor inductance per the respective manufacturer’s data sheet. For the case of ______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers ILIM ILIM DH R C3 DH RDS(ON) R C3 C RSENSE LX C LX DL DL Figure 5. Adding RC for More Accurate Sensing the MOSFET, it is the inductance from the drain to the source lead. Alternately, to save board space and cost, the RC networks above can be omitted; however, the value of RILIM should be raised to account for the voltage step caused by the inductive divider. An additional switching noise filter may be needed at ILIM by connecting a capacitor in parallel with R2 (in the case of RDS(ON) sensing) or from ILIM to LX (in the case of resistor sensing). For the case of RDS(ON) sensing, the value of the capacitor should be: C3 > 15 / (π x fS x R2) For the case of resistor sensing: C3 < 25 x 10-9 / R2 Selecting the Soft-Start Capacitor An external capacitor from SS to GND is charged by an internal 5µA current source, to the corresponding feedback threshold. Therefore, the soft-start time is calculated as: tSS = CSS x VFB / 5µA For example, 0.033µF from SS to GND yields approximately a 3.96ms soft-start period. In the tracking application (see Figure 3), the output voltage is required to track REFIN during REFIN rise and fall time. CSS must be chosen so that tss is less than REFIN rise and fall time. Compensation Design The MAX8597/MAX8598/MAX8599 use a voltage-mode control scheme that regulates the output voltage by comparing the error-amplifier output (COMP) with a fixed internal ramp to produce the required duty cycle. The error amplifier is an operational amplifier with 25MHz bandwidth to provide fast response. The output lowpass LC filter creates a double pole at the resonant frequency that introduces a gain drop of 40dB per decade and a phase shift of 180 degrees per decade. The error amplifier must compensate for this gain drop and phase shift to achieve a stable high-bandwidth closed-loop system. The Type III compensation scheme (Figure 6) is used to achieve this stability. The basic regulator loop can be thought of as consisting of a power modulator and an error amplifier. The power modulator has a DC gain set by VIN / VRAMP, with a double pole, fP_LC, and a single zero, fZ_ESR, set by the output inductor (L), the output capacitor (CO), and its equivalent series resistance (RESR). Below are the equations that define the power modulator: GMOD(DC) = fP _ LC = fZ _ ESR = VIN , where VRAMP = 1V (typ) VRAMP 1 2π L × C O 1 2π × RESR × CO where CO is the total output capacitance and RESR is the total ESR of the output capacitors. ______________________________________________________________________________________ 19 MAX8597/MAX8598/MAX8599 R2 R2 MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers When the output capacitor is comprised of paralleling n number of the same capacitors, then: CO = n x CEACH and RESR = RESR_EACH / n Thus, the resulting fZ_ESR is the same as that of a single capacitor. The total closed-loop gain must be equal to unity at the crossover frequency, where the crossover frequency is less than or equal to 1/5 the switching frequency (fS): fC ≤ fS / 5 So the loop-gain equation at the crossover frequency is: GEA(FC) x GMOD(FC) = 1 where GEA(FC) is the error-amplifier gain at fC, and GMOD(FC) is the power-modulator gain at fC. The loop compensation is affected by the choice of output filter capacitor due to the position of its ESR-zero frequency with respect to the desired closed-loop crossover frequency. Ceramic capacitors are used for higher switching frequencies and have low capacitance and low ESR; therefore, the ESR-zero frequency is higher than the closed-loop crossover frequency. Electrolytic capacitors (e.g., tantalum, solid polymer, and OS-CON) are needed for lower switching frequencies and have high capacitance (and some have higher ESR); therefore, the ESR-zero frequency can be lower than the closed-loop crossover frequency. Thus, the compensation design procedures are separated into two cases: Case 1: Crossover frequency is less than the output-capacitor ESR-zero (fC < fZ_ESR). The modulator gain at fC is: GMOD(FC) = GMOD(DC) x (fP_LC / fC)2 Since the crossover frequency is lower than the output capacitor ESR-zero frequency and higher than the LC double-pole frequency, the error-amplifier gain must have a +1 slope at fC so that, together with the -2 slope of the LC double pole, the loop crosses over at the desired -1 slope. The error amplifier has a dominant pole at a very low frequency (~0Hz), and two additional zeros and two additional poles as indicated by the equations below and illustrated in Figure 7: fZ1_EA = 1 / (2 π x R4 x C2) fZ2_EA = 1 / (2 π x (R1 + R3) x C1) 20 fP2_EA = 1 / (2 π x R3 x C1) fP3_EA = 1 / (2 π x R4 x (C2 x C3 / (C2 + C3))) Note that fZ2_EA and fP2_EA are chosen to have the converter closed-loop crossover frequency, fC, occur when the error-amplifier gain has +1 slope, between fZ2_EA and fP2_EA. The error-amplifier gain at fC must meet the requirement below: GEA(FC) = 1 / GMOD(FC) The gain of the error amplifier between f Z1_EA and fZ2_EA is: GEA(fZ1_EA - fZ2_EA) = GEA(FC) x fZ2_EA / fC = fZ2_EA / (fC x GMOD(FC)) This gain is set by the ratio of R4/R1 (Figure 6), where R1 is calculated as illustrated in the Setting the Output Voltage section. Thus: R4 = R1 x fZ2_EA / (fC x GMOD(FC)) where fZ2_EA = fP_LC. Due to the underdamped (Q > 1) nature of the output LC double pole, the first error-amplifier zero frequency must be set less than the LC double-pole frequency in order to provide adequate phase boost. Set the erroramplifier first zero, fZ1_EA, at 1/4 of the LC double-pole frequency. Hence: C2 = 2 / (π x R4 x fP_LC) Set the error amplifier fP2_EA at fZ_ESR and f f fp3 _ EA at s if fZ _ ESR is less than s . 2 2 fs If fZ _ ESR is greater than , then set 2 fs fp2 _ EA at and fp3 _ EA at fZ _ ESR. 2 The error-amplifier gain between fP2_EA and fP3_EA is set by the ratio of R4/RM and is equal to: GEA(fZ1_EA - fZ2_EA) x (fP2_EA / fP_LC) where RM = R1 x R3 / (R1 + R3). Then: RM = R4 x fP_LC / (GEA(fZ1_EA - fZ2_EA) x fP2_EA) = R4 x fC x GMOD(FC) / fP2_EA The value of R3 can then be calculated as: R3 = R1 x RM / (R1 – RM) Now we can calculate the value of C1 as: C1 = 1 / (2 π x R3 x fp2_EA) and C3 as: C3 = C2 / ((2 π x C2 x R4 x fP3_EA) - 1) ______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers GMOD(FC) = GMOD(DC) x (fP_LC)2 / (fZ_ESR x fC) Since the output-capacitor ESR-zero frequency is higher than the LC double-pole frequency but lower than the closed-loop crossover frequency, where the modulator already has -1 slope, the error-amplifier gain must have zero slope at fC so the loop crosses over at the desired -1 slope. The error-amplifier circuit configuration is the same as case 1 above; however, the closed-loop crossover frequency is now between fP2 and fP3 as illustrated in Figure 8. The equations that define the error amplifier’s zeros (f Z1_EA , f Z2_EA ) and poles (f P2_EA , f P3_EA ) are the same as case 1; however, fP2_EA is now lower than the closed-loop crossover frequency. Therefore, the erroramplifier gain between fZ1_EA and fZ2_EA is now calculated as: Set the error-amplifier third pole, fP3_EA, at half the switching frequency, and let RM = (R1 x R3) / (R1 + R3). The gain of the error amplifier between fP2_EA and f P3_EA is set by the ratio of R4/RM and is equal to GEA(FC) = 1 / GMOD(FC). Then: RM = R4 x GMOD(FC) Similar to case 1, R3, C1, and C3 are calculated as: R3 = R1 x RM / (R1 - RM) C1 = 1 / (2π x R3 x fZ_ESR) C3 = C2 / ((2π x C2 x R4 x fP3_EA) - 1) GAIN (dB) CLOSED-LOOP GAIN EA GAIN GEA(fZ1_EA - fZ2_EA) = GEA(FC) x fZ2_EA / fP2_EA = fZ2_EA / (fP2_EA x GMOD(FC)) This gain is set by the ratio of R4/R1, where R1 is calculated as illustrated in the Setting the Output Voltage section. Thus: R4 = R1 x fZ2_EA / (fP2_EA x GMOD(FC)) where fZ2_EA = fP_LC and fP2_EA = fZ_ESR. Similar to case 1, C2 is calculated as: C2 = 2 / (π x R4 x fP_LC) 0 fZ1 fZ2 fC fP2 fP3 FREQUENCY Figure 7. Closed-Loop and Error-Amplifier Gain Plot for Case 1 GAIN (dB) L CO MAX8597 MAX8598 MAX8599 R3 COMP C2 C1 R1 CLOSED-LOOP GAIN EA GAIN C3 R4 R2 FB REF 0 Figure 6. Type III Compensation Network fZ1 fZ2 fP2 fC fP3 FREQUENCY Figure 8. Closed-Loop and Error-Amplifier Gain Plot for Case 2 ______________________________________________________________________________________ 21 MAX8597/MAX8598/MAX8599 Case 2: Crossover frequency is greater than the output-capacitor ESR zero (fC > fZ_ESR). The modulator gain at fC is: Applications Information PC Board Layout Guide Careful PC board layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention. Follow these guidelines for good PC board layout: 1) Place the high-side MOSFET close to the low-side MOSFET and arrange them in such a way that the drain of the high-side MOSFET and the source of the low-side MOSFET can be tightly decoupled with a 10µF or larger ceramic capacitor. The MOSFETs should also be placed close to the controller IC, preferably not more than 1.5in away from the IC. 2) Place the IC’s pin decoupling capacitors as close to pins as possible. 3) A current-limit setting resistor must be connected from ILIM directly to the drain of the high-side MOSFET. 4) Try to keep the LX node connection to the IC pin separate from the connection to the flying boost capacitor. Pin Configurations (continued) 14 FREQ 15 POK 16 BST PGND DL 10 9 MAX8598 MAX5899 1 2 3 4 FB ILIM 11 SS 13 12 GND LX 5) Keep the power ground plane (connected to the source of the low-side MOSFET, PGND pin, input and output capacitors’ ground, VL decoupling ground) and the signal ground plane (connected to GND pin and the rest of the circuit ground returns) separate. Connect the two ground planes together at the ground of the output capacitor(s). 6) Place the RC snubber circuit as close to the lowside MOSFET as possible. 7) Keep the high-current paths as short as possible. 8) Connect the drains of the MOSFETs to a large copper area to help cool the devices and further improve efficiency and long-term reliability. 9) Ensure the feedback connection is short and direct. Place the feedback resistors as close to the IC as possible. 10) Route high-speed switching nodes, such as LX, DH, and DL away from sensitive analog areas (FB, COMP, ILIM, AIN+, AIN-). Refer to the MAX8597/MAX8598/MAX8599 evaluation kit for a sample board layout. Chip Information TRANSISTOR COUNT: 4493 PROCESS: BiCMOS DH TOP VIEW AVL MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers 8 VL 7 V+ 6 EN 5 COMP THIN QFN 4mm x 4mm 22 ______________________________________________________________________________________ Low-Dropout, Wide-Input-Voltage, Step-Down Controllers 24L QFN THIN.EPS PACKAGE OUTLINE 12, 16, 20, 24L THIN QFN, 4x4x0.8mm 21-0139 C 1 2 ______________________________________________________________________________________ 23 MAX8597/MAX8598/MAX8599 Package Information (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.) MAX8597/MAX8598/MAX8599 Low-Dropout, Wide-Input-Voltage, Step-Down Controllers Package Information (continued) (The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.) PACKAGE OUTLINE 12, 16, 20, 24L THIN QFN, 4x4x0.8mm 21-0139 C 2 2 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 24 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2004 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.