19-0881; Rev 2; 1/10 KIT ATION EVALU E L B AVAILA High-Power Synchronous HBLED Drivers with Rapid Current Pulsing The MAX16821A/MAX16821B/MAX16821C pulsewidth-modulation (PWM) LED driver controllers provide high output-current capability in a compact package with a minimum number of external components. The MAX16821A/MAX16821B/MAX16821C are suitable for use in synchronous and nonsynchronous step-down (buck), boost, buck-boost, SEPIC, and Cuk LED drivers. A logic input (MODE) allows the devices to switch between synchronous buck and boost modes of operation. These devices are the first high-power drivers designed specifically to accommodate common-anode HBLEDs. The ICs offer average current-mode control that enable the use of MOSFETs with optimal charge and on-resistance figure of merit, thus minimizing the need for external heatsinking even when delivering up to 30A of LED current. The differential sensing scheme provides accurate control of the LED current. The ICs operate from a 4.75V to 5.5V supply range with the internal regulator disabled (VCC connected to IN). These devices operate from a 7V to 28V input supply voltage with the internal regulator enabled. The MAX16821A/MAX16821B/MAX16821C feature a clock output with 180° phase delay to control a second out-of-phase LED driver to reduce input and output filter capacitor size and to minimize ripple currents. The wide switching frequency range (125kHz to 1.5MHz) allows the use of small inductors and capacitors. Additional features include programmable overvoltage protection and an output enable function. Features ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ Up to 30A Output Current True-Differential Remote Output Sensing Average Current-Mode Control 4.75V to 5.5V or 7V to 28V Input-Voltage Range 0.1V/0.03V LED Current-Sense Options Maximize Efficiency (MAX16821B/MAX16821C) Thermal Shutdown Nonlatching Output Overvoltage Protection Low-Side Buck Mode with or without Synchronous Rectification High-Side Buck and Low-Side Boost Mode with or without Synchronous Rectification 125kHz to 1.5MHz Programmable/Synchronizable Switching Frequency Integrated 4A Gate Drivers Clock Output for 180° Out-of-Phase Operation for Second Driver -40°C to +125°C Operating Temperature Range Ordering Information TEMP RANGE MAX16821AATI+ -40°C to +125°C 28 TQFN-EP* MAX16821BATI+ -40°C to +125°C 28 TQFN-EP* MAX16821CATI+ -40°C to +125°C 28 TQFN-EP* +Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad. Simplified Diagram 7V TO 28V Applications Front Projectors/Rear Projection TVs Portable and Pocket Projectors PINPACKAGE PART C1 IN EN DH Q1 Automotive Exterior Lighting LCD TVs and Display Backlight I.C. Automotive Emergency Lighting and Signage VLED L1 MAX16821 DL Q2 C2 Q3 OVI CSP CLP PGND R1 Typical Operating Circuit and Selector Guide appear at end of data sheet. . HIGH-FREQUENCY PULSE TRAIN ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX16821A/MAX16821B/MAX16821C General Description MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing ABSOLUTE MAXIMUM RATINGS All Other Pins to SGND...............................-0.3V to (VCC + 0.3V) Continuous Power Dissipation (TA = +70°C) 28-Pin TQFN 5mm x 5mm (derate 34.5mW/°C above +70°C) ............................................................2758mW Operating Temperature Range .........................-40°C to +125°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C IN to SGND.............................................................-0.3V to +30V BST to SGND..........................................................-0.3V to +35V BST to LX..................................................................-0.3V to +6V DH to LX ...........................................-0.3V to (VBST - VLX) + 0.3V DL to PGND................................................-0.3V to (VDD + 0.3V) VCC to SGND............................................................-0.3V to +6V VCC, VDD to PGND ...................................................-0.3V to +6V SGND to PGND .....................................................-0.3V to +0.3V VCC Current ......................................................................300mA Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VCC = 5V, VDD = VCC, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER SYMBOL Input-Voltage Range VIN Quiescent Supply Current IQ CONDITIONS Internal LDO on Internal LDO off (VCC connected to VIN) MIN TYP MAX 7 28 4.75 5.50 VEN = VCC or SGND, no switching 2.7 5.5 UNITS V mA LED CURRENT REGULATOR Differential Set Value (VSENSE+ to VSENSE-) (Note 2) Soft-Start Time VIN = VCC = 4.75V to 5.5V, fSW = 500kHz (MAX16821A) 0.594 0.600 0.606 VIN = 7V to 28V, fSW = 500kHz (MAX16821A) 0.594 0.600 0.606 VIN = VCC = 4.75V to 5.5V, fSW = 500kHz (MAX16821B) 0.098 0.100 0.102 V VIN = 7V to 28V, fSW = 500kHz (MAX16821B) 0.098 0.100 0.102 VIN = VCC = 4.75V to 5.5V, fSW = 500kHz (MAX16821C) 0.028 0.030 0.032 VIN = 7V to 28V, fSW = 500kHz (MAX16821C) 0.028 0.030 0.032 tSS Clock Cycles 1024 STARTUP/INTERNAL REGULATOR VCC Undervoltage Lockout (UVLO) UVLO VCC rising UVLO Hysteresis VCC falling VCC Output Voltage VIN = 7V to 28V, ISOURCE = 0 to 60mA 4.1 4.3 4.5 200 4.85 V mV 5.10 5.30 V 1.1 3 Ω MOSFET DRIVER Output Driver Impedance Output Driver Source/Sink Current Nonoverlap Time 2 Low or high output, ISOURCE/SINK = 20mA IDH, IDL tNO CDH/DL = 5nF 4 A 35 ns _______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing (VCC = 5V, VDD = VCC, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 1500 kHz OSCILLATOR Switching Frequency Range Switching Frequency 125 fSW Switching Frequency Accuracy CLKOUT Phase Shift with Respect to DH (Rising Edges) RT = 500kΩ 120 125 130 RT = 120kΩ 495 521 547 RT = 39.9kΩ 1515 1620 1725 120kΩ < RT ≤ 500kΩ -5 +5 40kΩ ≤ RT ≤ 120kΩ -8 +8 fSW = 125kHz, MODE connected to SGND kHz % 180 Degrees CLKOUT Phase Shift with Respect to DL (Rising Edges) fSW = 125kHz, MODE connected to VCC CLKOUT Output-Voltage Low VOL ISINK = 2mA CLKOUT Output-Voltage High VOH ISOURCE = 2mA SYNC Input High Pulse Width 180 0.4 V 4.5 V tSYNC 200 ns SYNC Input Clock High Threshold VSYNCH 2 SYNC Input Clock Low Threshold VSYNCL SYNC Pullup Current ISYNC_OUT SYNC Power-Off Level VSYNC_OFF VRT/SYNC = 0V V 250 0.4 V 500 µA 0.4 V 33.0 mV INDUCTOR CURRENT LIMIT Average Current-Limit Threshold VCL CSP to CSN Reverse Current-Limit Threshold VCLR CSP to CSN -2.0 mV CSP to CSN 60 mV VCSP to VCSN = 75mV 260 ns 4 kΩ Cycle-by-Cycle Current Limit Cycle-by-Cycle Overload 26.4 27.5 CURRENT-SENSE AMPLIFIER CSP to CSN Input Resistance Common-Mode Range Input Offset Voltage Amplifier Voltage Gain 3dB Bandwidth RCS VCMR(CS) VIN = 7V to 28V 0 5.5 V VOS(CS) 0.1 mV AV(CS) 34.5 V/V f3dB 4 MHz gm 550 µS AVL(CE) 50 dB CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER) Transconductance Open-Loop Gain _______________________________________________________________________________________ 3 MAX16821A/MAX16821B/MAX16821C ELECTRICAL CHARACTERISTICS (continued) MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing ELECTRICAL CHARACTERISTICS (continued) (VCC = 5V, VDD = VCC, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 1.0 V LED CURRENT SIGNAL DIFFERENTIAL VOLTAGE AMPLIFIER (DIFF) Common-Mode Voltage Range DIFF Output Voltage Input Offset Voltage Amplifier Voltage Gain 3dB Bandwidth SENSE+ to SENSE- Input Resistance VCMR(DIFF) VCM VOS(DIFF) AV(DIFF) f3dB RVS 0 VSENSE+ = VSENSE- = 0V 0.6 V MAX16821A -3.7 +3.7 MAX16821B/MAX16821C -1.5 +1.5 MAX16821A 0.992 1 MAX16821B 5.85 6 6.1 MAX16821C 18.5 20 21.5 MAX16821A, CDIFF = 20pF 1.7 MAX16821B, CDIFF = 20pF 1600 MAX16821C, CDIFF = 20pF 550 MAX16821A 50 100 MAX16821B 30 60 MAX16821C 10 20 mV 1.008 V/V MHz kHz kΩ OUTV AMPLIFIER Gain-Bandwidth Product VOUTV = 2V 3dB Bandwidth VOUTV = 2V 4 MHz 1 MHz Output Sink Current 30 µA Output Source Current 80 µA Maximum Load Capacitance 50 OUTV to (CSP - CSN) Transfer Function 4mV ≤ CSP - CSN ≤ 32mV 132.5 Input Offset Voltage 135 pF 137.7 V/V 1 mV 70 dB VOLTAGE-ERROR AMPLIFIER (EAOUT) Open-Loop Gain AVOLEA Unity-Gain Bandwidth fGBW EAN Input Bias Current IB(EA) Error Amplifier Output Clamping Voltage VCLAMP(EA) 3 MHz VEAN = 2V -0.2 +0.03 +0.2 µA With respect to VCM 905 930 940 mV INPUTS (MODE AND OVI) MODE Input-Voltage High 2 V MODE Input-Voltage Low MODE Pulldown Current OVI Trip Threshold OVPTH OVI Hysteresis OVIHYS OVI Input Bias Current 4 IOVI VOVI = 1V 0.8 V 4 5 6 µA 1.244 1.276 1.308 V 200 mV 0.2 µA _______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing (VCC = 5V, VDD = VCC, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 2.437 2.5 2.562 V 16.5 µA ENABLE INPUT (EN) EN Input-Voltage High EN rising EN Input Hysteresis 0.28 EN Pullup Current IEN 13.5 15 V THERMAL SHUTDOWN Thermal Shutdown 165 °C Thermal-Shutdown Hysteresis 20 °C Note 1: All devices are 100% production tested at +25°C. Limits over temperature are guaranteed by design. Note 2: Does not include an error due to finite error amplifier gain. See the Voltage-Error Amplifier section. Typical Operating Characteristics (VIN = 12V, VDD = VCC = 5V, TA = +25°C, unless otherwise noted.) SUPPLY CURRENT (IQ) vs. FREQUENCY VIN = 24V 6 5 VIN = 12V 4 VIN = 5V 3 2 55 500 700 VIN = 24V 5.1 VIN = 12V 5.0 4.9 50 900 1100 1300 1500 VIN = 7V 4.8 4.7 VIN = 12V CDH/DL = 22nF 40 300 5.3 5.2 45 -40 -15 10 35 60 4.6 4.5 85 0 15 30 45 60 75 90 105 120 135 150 FREQUENCY (kHz) TEMPERATURE (°C) VCC LOAD CURRENT (mA) DRIVER RISE TIME vs. DRIVER LOAD CAPACITANCE DRIVER FALL TIME vs. DRIVER LOAD CAPACITANCE HIGH-SIDE DRIVER (DH) SINK AND SOURCE CURRENT 100 MAX16821A toc04 200 180 160 MAX16821A toc06 MAX16821A toc05 100 5.4 60 1 0 MAX16821A toc03 65 VCC (V) 7 5.5 MAX16821A toc02 SUPPLY CURRENT (mA) 8 SUPPLY CURRENT (mA) EXTERNAL CLOCK NO DRIVER LOAD 9 VCC LOAD REGULATION vs. VIN SUPPLY CURRENT vs. TEMPERATURE 70 MAX16821A toc01 10 80 CLOAD = 22nF VIN = 12V 120 fF (ns) tR (ns) 140 100 80 60 2A/div 40 DL DH 60 DH 40 20 DL 20 0 0 0 5 10 15 LOAD CAPACITANCE (nF) 20 25 0 5 10 15 20 25 100ns/div LOAD CAPACITANCE (nF) _______________________________________________________________________________________ 5 MAX16821A/MAX16821B/MAX16821C ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (continued) (VIN = 12V, VDD = VCC = 5V, TA = +25°C, unless otherwise noted.) LOW-SIDE DRIVER (DL) SINK AND SOURCE CURRENT HIGH-SIDE DRIVER (DH) FALL TIME HIGH-SIDE DRIVER (DH) RISE TIME MAX16821A toc09 MAX16821A toc08 MAX16821A toc07 CLOAD = 22nF VIN = 12V VIN = 12V DH RISING 2V/div 2V/div 3A/div 40ns/div 40ns/div 100ns/div LOW-SIDE DRIVER (DL) FALL TIME LOW-SIDE DRIVER (DL) RISE TIME FREQUENCY vs. RT MAX16821A toc11 MAX16821A toc10 CLOAD = 22nF VIN = 12V 10,000 CLOAD = 22nF VIN = 12V fSW (kHz) VIN = 12V 2V/div 2V/div MAX16821A toc12 CLOAD = 22nF VIN = 12V 1000 100 40ns/div 40ns/div 30 70 110 150 190 230 270 310 350 390 430 470 510 550 RT (kΩ) FREQUENCY vs. TEMPERATURE SYNC, CLKOUT, AND DH WAVEFORMS SYNC, CLKOUT, AND DL WAVEFORMS MAX16821A toc14 MAX16821A toc13 260 VIN = 12V 258 256 MAX16821A toc15 RT/SYNC 5V/div 0V 254 fSW (kHz) MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing RT/SYNC 5V/div 0V MODE = SGND 252 250 248 MODE = VCC CLKOUT 5V/div 0V CLKOUT 5V/div 0V DH 5V/div 0V DL 5V/div 0V 246 244 242 240 0 5 10 15 20 25 30 35 1µs/div 1µs/div TEMPERATURE (°C) 6 _______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing PIN NAME 1 PGND FUNCTION 2, 7 N.C. 3 DL Low-Side Gate-Driver Output 4 BST Boost-Flying Capacitor Connection. Reservoir capacitor connection for the high-side MOSFET driver supply. Connect a ceramic capacitor between BST and LX. 5 LX 6 DH Power-Supply Ground No Connection. Not internally connected. High-Side MOSFET Source Connection High-Side Gate-Driver Output Signal Ground. SGND is the ground connection for the internal control circuitry. Connect SGND and PGND together at one point near the IC. 8, 22, 25 SGND 9 CLKOUT 10 MODE 11 EN 12 RT/SYNC 13 OUTV 14 I.C. Internally Connected. Connect to SGND for proper operation. 15 OVI Overvoltage Protection. When OVI exceeds the programmed output voltage by 12.7%, the low-side and the high-side drivers are turned off. When OVI falls 20% below the programmed output voltage, the drivers are turned on after power-on reset and soft-start cycles are completed. 16 CLP Current-Error-Amplifier Output. Compensate the current loop by connecting an RC network to ground. 17 EAOUT 18 EAN Voltage-Error-Amplifier Inverting Input 19 DIFF Differential Remote-Sense Amplifier Output. DIFF is the output of a precision amplifier with SENSE+ and SENSE- as inputs. 20 CSN Current-Sense Differential Amplifier Negative Input. The differential voltage between CSN and CSP is amplified internally by the current-sense amplifier (Gain = 34.5) to measure the inductor current. 21 CSP Current-Sense Differential Amplifier Positive Input. The differential voltage between CSP and CSN is amplified internally by the current-sense amplifier (Gain = 34.5) to measure the inductor current. 23 SENSE- Differential LED Current-Sensing Negative Input. Connect SENSE- to the negative side of the LED currentsense resistor or to the negative feedback point. 24 SENSE+ Differential LED Current-Sensing Positive Input. Connect SENSE+ to the positive side of the LED currentsense resistor, or to the positive feedback point. 26 IN 27 VCC Internal +5V Regulator Output. VCC is derived from VIN. Bypass VCC to SGND with 4.7µF and 0.1µF ceramic capacitors. 28 VDD Low-Side Driver Supply Voltage — EP Oscillator Output. If MODE is low, the rising edge of CLKOUT phase shifts from the rising edge of DH by 180°. If MODE is high, the rising edge of CLKOUT phase shifts from the rising edge of DL by 180°. Buck/Boost Mode Selection Input. Drive MODE low for low-side buck mode operation. Drive MODE high for boost or high-side buck mode operation. MODE has an internal 5µA pulldown current to ground. Output Enable. Drives EN high or leave unconnected for normal operation. Drive EN low to shut down the power drivers. EN has an internal 15µA pullup current. Switching Frequency Programming. Connect a resistor from RT/SYNC to SGND to set the internal oscillator frequency. Drive RT/SYNC to synchronize the switching frequency with an external clock. Inductor Current-Sense Output. OUTV is an amplifier output voltage proportional to the inductor current. The voltage at OUTV = 135 x (VCSP - VCSN). Voltage-Error-Amplifier Output. Connect EAOUT to the external gain-setting network. Supply Voltage Input. Connect IN to VCC, for a 4.75V to 5.5V input supply range. Exposed Pad. EP is internally connected to SGND. Connect EP to a large-area ground plane for effective power dissipation. Connect EP to SGND. Do not use as a ground connection. _______________________________________________________________________________________ 7 MAX16821A/MAX16821B/MAX16821C Pin Description MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Detailed Description The MAX16821A/MAX16821B/MAX16821C are high-performance average current-mode PWM controllers for high-power and high-brightness LEDs (HBLEDs). The average current-mode control technique offers inherently stable operation, reduces component derating and size by accurately controlling the inductor current. The devices achieve high efficiency at high currents (up to 30A) with a minimum number of external components. A logic input (MODE) allows the LED driver to switch between buck and boost modes of operation. The MAX16821A/MAX16821B/MAX16821C feature a CLKOUT output 180° out-of-phase with respect to either the high-side or low-side driver, depending on MODE’s logic level. CLKOUT provides the drive for a second out-of-phase LED driver for applications requiring reduced input capacitor ripple current while operating another LED driver. The MAX16821A/MAX16821B/MAX16821C consist of an inner average current regulation loop controlled by an outer loop. The combined action of the inner current loop and outer voltage loop corrects the LED current errors by adjusting the inductor current resulting in a tightly regulated LED current. The differential amplifier (SENSE+ and SENSE- inputs) senses the LED current using a resistor in series with the LEDs and produces an amplified version of the sense voltage at DIFF. The resulting amplified sensed voltage is compared against an internal 0.6V reference at the error amplifier input. Input Voltage The MAX16821A/MAX16821B/MAX16821C operate with a 4.75V to 5.5V input supply range when the internal LDO is disabled (VCC connected to IN) or a 7V to 28V input supply range when the internal LDO is enabled. For a 7V to 28V input voltage range, the internal LDO provides a regulated 5V output with 60mA of sourcing capability. Bypass VCC to SGND with 4.7µF and 0.1µF low-ESR ceramic capacitors. The MAX16821A/MAX16821B/MAX16821C’s VDD input provides supply voltage for the low-side and the highside MOSFET drivers. Connect VDD to VCC using an R-C filter to isolate the analog circuits from the MOSFET drivers. The internal LDO powers up the MAX16821A/ MAX16821B/MAX16821C. For applications utilizing a 5V input voltage, disable the internal LDO by connecting IN and VCC together. The 5V power source must be in the 4.75V to 5.5V range of for proper operation of the MAX16821A/MAX16821B/MAX16821C. 8 Undervoltage Lockout (UVLO) The MAX16821A/MAX16821B/MAX16821C include UVLO and a 2048 clock-cycle power-on-reset circuit. The UVLO rising threshold is set to 4.3V with 200mV hysteresis. Hysteresis at UVLO eliminates chattering during startup. Most of the internal circuitry, including the oscillator, turns on when the input voltage reaches 4V. The MAX16821A/MAX16821B/MAX16821C draw up to 3.5mA of quiescent current before the input voltage reaches the UVLO threshold. Soft-Start The MAX16821A/MAX16821B/MAX16821C include an internal soft-start for a glitch-free rise of the output voltage. After 2048 power-on-reset clock cycles, a 0.6V reference voltage connected to the positive input of the internal error amplifier ramps up to its final value after 1024 clock cycles. Soft-start reduces inrush current and stress on system components. During soft-start, the LED current will ramp monotonically towards its final value. Internal Oscillator The internal oscillator generates a clock with the frequency inversely proportional to the value of RT (see the Typical Operating Circuit). The oscillator frequency is adjustable from 125kHz to 1.5MHz range using a single resistor connected from RT/SYNC to SGND. The frequency accuracy avoids the overdesign, size, and cost of passive filter components like inductors and capacitors. Use the following equation to calculate the oscillator frequency: For 120kΩ ≤ RT ≤ 500kΩ: fSW = 6.25 x 1010 (Hz) RT For 40kΩ ≤ RT ≤ 120kΩ: fSW = 6.40 x 1010 (Hz) RT The oscillator also generates a 2VP-P ramp signal for the PWM comparator and a 180° out-of-phase clock signal at CLKOUT to drive a second out-of-phase LED current regulator. _______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C VCC EN 0.5 x VCC UVLO POR TEMP SEN +5V LDO IN VCC TO INTERNAL CIRCUIT I.C. CLP CSP AV = 34.5 CSN VCM AV = 4 gm VCLAMP LOW OUTV BST VCLAMP HIGH PWM COMPARATOR CLK RT/SYNC S OSCILLATOR Q DH MUX R Q LX VDD DL PGND 2 x fS RAMP GENERATOR CLKOUT VTH DIFF VCM SENSEDIFF AMP SENSE+ MODE EAOUT EAN ERROR AMP VREF = 0.6V 0.12 x VREF OVP COMPARATOR SOFTSTART ENABLE MAX16821A MAX16821B MAX16821C UVLO VCM OVI SGND Figure 1. Internal Block Diagram _______________________________________________________________________________________ 9 MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Synchronization MAX16821C outer LED-current control loop consists of a differential amplifier (DIFF), a reference voltage, and a voltage-error amplifier (VEA). The MAX16821A/MAX16821B/MAX16821C synchronize to an external clock connected to RT/SYNC. The application of an external clock at RT/SYNC disables the internal oscillator. Once the MAX16821A/MAX16821B/ MAX16821C are synchronized to an external clock, the external clock cannot be removed if reliable operation is to be maintained. Inductor Current-Sense Amplifier The differential current-sense amplifier (CSA) provides a 34.5V/V DC gain. The typical input offset voltage of the current-sense amplifier is 0.1mV with a 0 to 5.5V commonmode voltage range (VIN = 7V to 28V). The current-sense amplifier senses the voltage across RS. The maximum common-mode voltage is 3.2V when VIN = 5V. Control Loop The MAX16821A/MAX16821B/MAX16821C use an average current-mode control scheme to regulate the output current (Figure 2). The main control loop consists of an inner current regulation loop for controlling the inductor current and an outer current regulation loop for regulating the LED current. The inner current regulation loop absorbs the double pole of the inductor and output capacitor combination reducing the order of the outer current regulation loop to that of a single-pole system. The inner current regulation loop consists of a current-sense resistor (RS), a current-sense amplifier (CSA), a current-error amplifier (CEA), an oscillator providing the carrier ramp, and a PWM comparator (CPWM) (Figure 2). The MAX16821A/MAX16821B/ Inductor Peak-Current Comparator The peak-current comparator provides a path for fast cycle-by-cycle current limit during extreme fault conditions, such as an inductor malfunction (Figure 3). Note the average current-limit threshold of 27.5mV still limits the output current during short-circuit conditions. To prevent inductor saturation, select an inductor with a saturation current specification greater than the average current limit. The 60mV threshold for triggering the peak-current limit is twice the full-scale average current-limit voltage threshold. The peak-current comparator has only a 260ns delay. RCF RIN DIFF RF EAN CCZ CF EAOUT CCP CSN CSP CLP CA VIN SENSE+ CEA DIFF VEA LED STRING L CPWM DRIVER SENSECOUT VREF RS MODE = SGND Figure 2. MAX16821A/MAX16821B/MAX16821C Control Loop 10 ______________________________________________________________________________________ RLS High-Power Synchronous HBLED Drivers with Rapid Current Pulsing PWM Comparator and R-S Flip-Flop An internal PWM comparator sets the duty cycle by comparing the output of the current-error amplifier to a 60mV 2VP-P ramp signal. At the start of each clock cycle, an R-S flip-flop resets and the high-side driver (DH) turns on if MODE is connected to SGND, and DL turns on if MODE is connected to VCC. The comparator sets the flip-flop as soon as the ramp signal exceeds the CLP voltage, thus terminating the ON cycle. See Figure 3. Differential Amplifier The differential amplifier (DIFF) allows LED current sensing (Figure 2). It provides true-differential LED current sensing, and amplifies the sense voltage by a factor of 1 (MAX16821A), 6 (MAX16821B), and 20 (MAX16821C), while rejecting common-mode voltage errors. The VEA provides the difference between the differential amplifier output (DIFF) and the desired LED current-sense voltage. The differential amplifier has a bandwidth of 1.7MHz (MAX16821A), 1.6MHz (MAX16821B), and 550kHz (MAX16821C). The difference between SENSE+ and SENSE- is regulated to +0.6V (MAX16821A), +0.1V (MAX16821B), or +0.03V (MAX16821C). PEAK-CURRENT COMPARATOR CLP CSP AV = 34.5 CSN gm = 550µS PWM COMPARATOR IN MODE = GND BST S Q RAMP DH LX VDD R CLK Q DL PGND SHDN Figure 3. MAX16821A/MAX16821B/MAX16821C Phase Circuit ______________________________________________________________________________________ 11 MAX16821A/MAX16821B/MAX16821C Current-Error Amplifier The MAX16821A/MAX16821B/MAX16821C include a transconductance current-error amplifier with a typical gm of 550µS and 320µA output sink and source capability. The current-error amplifier output (CLP) is connected to the inverting input of the PWM comparator. CLP is also externally accessible to provide frequency compensation for the inner current regulation loop (Figure 2). Compensate CEA so the inductor current negative slope, which becomes the positive slope to the inverting input of the PWM comparator, is less than the slope of the internally generated voltage ramp (see the Compensation section). In applications without synchronous rectification, the LED driver can be turned off and on instantaneously by shorting or opening the CLP to ground. MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Voltage-Error Amplifier (VEA) Current Limit The VEA sets the gain of the voltage control loop, and determines the error between the differential amplifier output and the internal reference voltage. The VEA output clamps to 0.93V relative to the internal commonmode voltage, VCM (+0.6V), limiting the average maximum current. The maximum average current-limit threshold is equal to the maximum clamp voltage of the VEA divided by the gain (34.5) of the current-sense amplifier. This results in accurate settings for the average maximum current. The error amplifier (VEA) output is clamped between -0.050V and +0.93V with respect to common-mode voltage (VCM). Average current-mode control limits the average current sourced by the converter during a fault condition. When a fault condition occurs, the VEA output clamps to +0.93V with respect to the commonmode voltage (0.6V) to limit the maximum current sourced by the converter to ILIMIT = 0.0275 / RS. MOSFET Gate Drivers The high-side (DH) and low-side (DL) drivers drive the gates of external n-channel MOSFETs. The drivers’ 4A peak sink- and source-current capability provides ample drive for the fast rise and fall times of the switching MOSFETs. Faster rise and fall times result in reduced cross-conduction losses. Size the high-side and low-side MOSFETs to handle the peak and RMS currents during overload conditions. The driver block also includes a logic circuit that provides an adaptive nonoverlap time to prevent shoot-through currents during transition. The typical nonoverlap time is 35ns between the high-side and low-side MOSFETs. Overvoltage Protection The OVP comparator compares the OVI input to the overvoltage threshold. The overvoltage threshold is typically 1.127 times the internal 0.6V reference voltage plus VCM (0.6V). A detected overvoltage event trips the comparator output turning off both high-side and lowside MOSFETs. Add an RC delay to reduce the sensitivity of the overvoltage circuit and avoid unnecessary tripping of the converter (Figure 4). After the OVI voltage falls below 1.076V (typ.), high-side and low-side drivers turn on only after a 2048 clock-cycle POR and a 1024 clock-cycle soft-start have elapsed. Disable the overvoltage function by connecting OVI to SGND. BST The MAX16821A/MAX16821B/MAX16821C provide power to the low-side and high-side MOSFET drivers through VDD. A bootstrap capacitor from BST to LX provides the additional boost voltage necessary for the high-side driver. VDD supplies power internally to the low-side driver. Connect a 0.47µF low-ESR ceramic capacitor between BST and LX and a Schottky diode from BST to VDD. COVI RA OVI VOUT RB MAX16821A MAX16821B MAX16821C DIFF Protection The MAX16821A/MAX16821B/MAX16821C include output overvoltage protection (OVP). During fault conditions when the load goes to high impedance (output opens), the controller attempts to maintain LED current. The OVP disables the MAX16821A/MAX16821B/ MAX16821C whenever the output voltage exceeds the OVP threshold, protecting the external circuits from undesirable voltages. 12 RIN EAN RF EAOUT Figure 4. Overvoltage Protection Input Delay ______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing inductor discharges to the output. The output voltage cannot go below the input voltage in this configuration. Resistor R1 senses the inductor current and resistor R2 senses the LED current. The outer LED current regulation loop programs the average current in the inductor, thus achieving tight LED current regulation. Boost LED Driver Figure 5 shows the MAX16821A/MAX16821B/MAX16821C configured as a synchronous boost converter with MODE connected to VCC. During the on-time, the input voltage charges the inductor. During the off-time, the VCC R4 VLED ON/OFF C3 R9 VIN 7V TO 28V R3 C2 R10 C11 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 8 SGND CLKOUT N.C. 7 15 OVI C10 L1 VLED R8 Q2 16 CLP DH 6 17 EAOUT LX 5 R7 Q1 C9 C4 C8 MAX16821A MAX16821B MAX16821C 18 EAN R5 LED STRING R5 19 DIFF DL 3 20 CSN N.C. 2 R2 D1 R1 PGND 1 21 CSP SGND 22 C1 BST 4 SENSE23 SENSE+ 24 SGND 25 IN 26 VDD 28 VCC 27 VIN C7 C6 C5 Figure 5. Synchronous Boost LED Driver (Output Voltage Not to Exceed 28V) ______________________________________________________________________________________ 13 MAX16821A/MAX16821B/MAX16821C Applications Information MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Input-Referenced Buck-Boost LED Driver span from the output to the input. This effectively removes the boost-only restriction of the regulator in Figure 5, allowing the voltage across the LED to be greater or less than the input voltage. LED currentsensing is not ground-referenced, so a high-side current-sense amplifier is used to measure current. The circuit in Figure 6 shows a step-up/step-down regulator. It is similar to the boost converter in Figure 5 in that the inductor is connected to the input and the MOSFET is essentially connected to ground. However, rather than going from the output to ground, the LEDs VCC R4 VLED ON/OFF VIN 7V TO 28V C3 R8 R3 C2 R9 C11 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 8 SGND CLKOUT LED STRING 1 TO 6 LEDS R2 C2 L1 D1 N.C. 7 15 OVI VLED C10 Q1 R7 16 CLP DH 6 17 EAOUT LX 5 VCC R6 C1 C9 C8 MAX16821A MAX16821B MAX16821C 18 EAN R5 DL 3 20 CSN N.C. 2 R1 PGND 1 21 CSP SGND 22 RS- BST 4 19 DIFF SENSE23 SENSE+ 24 SGND 25 IN 26 VDD 28 VCC 27 VIN C7 C6 C5 Figure 6. Typical Application Circuit for an Input-Referred Buck-Boost LED Driver (7V to 28V Input) 14 RS+ ______________________________________________________________________________________ OUT High-Power Synchronous HBLED Drivers with Rapid Current Pulsing provide current to recharge C1 and supplies the load current. Since the voltage waveform across L1 and L2 are exactly the same, it is possible to wind both inductors on the same core (a coupled inductor). Although voltages on L1 and L2 are the same, RMS currents can be quite different so the windings may require a different gauge wire. Because of the dual inductors and segmented energy transfer, the efficiency of a SEPIC converter is lower than the standard buck or boost configurations. As in the boost driver, the current-sense resistor connects to ground, allowing the output voltage of the LED driver to exceed the rated maximum voltage of the MAX16821A/MAX16821B/MAX16821C. VCC R4 VLED ON/OFF VIN 7V TO 28V C2 R8 R3 R9 C10 C9 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 8 SGND CLKOUT 15 OVI N.C. 7 16 CLP DH 6 17 EAOUT LX 5 R7 L1 C3 VLED D1 Q1 R6 L2 C8 C7 MAX16821A MAX16821B MAX16821C 18 EAN R5 LED STRING BST 4 19 DIFF DL 3 20 CSN N.C. 2 R2 R1 PGND 1 21 CSP SGND 22 C1 SENSE23 SENSE+ 24 SGND 25 IN 26 VDD 28 VCC 27 VIN C6 C5 C4 Figure 7. Typical Application Circuit for a SEPIC LED Driver ______________________________________________________________________________________ 15 MAX16821A/MAX16821B/MAX16821C SEPIC LED Driver Figure 7 shows the MAX16821A/MAX16821B/ MAX16821C configured as a SEPIC LED driver. While buck topologies produce an output always lower than the input, and boost topologies produce an output always greater than the input, a SEPIC topology allows the output voltage to be greater than, equal to, or less than the input. In a SEPIC topology, the voltage across C3 is the same as the input voltage, and L1 and L2 have the same inductance. Therefore, when Q1 turns on (ontime), the currents in both inductors (L1 and L2) ramp up at the same rate. The output capacitor supports the output voltage during this time. When Q1 turns off (offtime), L1 current recharges C3 and combines with L2 to MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Low-Side Buck Driver with Synchronous Rectification In Figure 8, the input voltage goes from 7V to 28V and, because of the ground-based current-sense resistor, the output voltage can be as high as the input. The synchronous MOSFET keeps the power dissipation to a minimum, especially when the input voltage is large compared to the voltage on the LED string. For the inner average current-loop inductor, current is sensed by resistor R1. To regulate the LED current, R2 creates a voltage that the differential amplifier compares to 0.6V. Capacitor C1 is small and helps reduce the ripple current in the LEDs. Omit C1 in cases where the LEDs can tolerate a higher ripple current. The average currentmode control scheme converts the input voltage to a current source feeding the LED string. VCC R4 VLED ON/OFF C3 R9 R3 VIN 7V TO 28V R10 C11 C10 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 8 SGND CLKOUT 15 OVI N.C. 7 16 CLP DH 6 C2 R9 Q1 R7 17 EAOUT LX 5 C9 C8 VLED L1 R5 MAX16821A MAX16821B MAX16821C 18 EAN R6 C4 BST 4 19 DIFF DL 3 20 CSN N.C. 2 LED STRING Q2 C1 D2 R2 R1 PGND 1 21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 IN 26 VDD 28 VCC 27 VIN C7 C6 C5 Figure 8. Application Circuit for a Low-Side Buck LED Driver 16 ______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing amplifier, U2. The voltage appearing across resistor R11 becomes the average inductor current-sense voltage for the inner average current loop. To regulate the LED current, R2 creates a voltage that the differential amplifier compares to its internal reference. Capacitor C1 is small and is added to reduce the ripple current in the LEDs. In cases where the LEDs can tolerate a higher ripple current, capacitor C1 can be omitted. In Figure 9, the input voltage goes from 7V to 28V, the LED load is connected from the positive side to the currentsense resistor (R1) in series with the inductor, and MODE is connected to VCC. For the inner average current-loop inductor, current is sensed by resistor R1 and is then transferred to the low side by the high-side current-sense VCC R4 ON/OFF C3 R3 C11 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 CLKOUT N.C. 7 15 OVI C10 VIN 7V TO 28V 8 SGND R8 16 CLP DH 6 17 EAOUT LX 5 C1 Q1 R7 L1 C9 C8 LED STRING C2 I.C. 18 EAN R6 BST 4 DL 3 20 CSN N.C. 2 U2 Q2 RS- R2 OUT R11 PGND 1 21 CSP SGND 22 RS+ D1 19 DIFF VCC R1 C4 R5 MAX16821A MAX16821B MAX16821C SENSE23 SENSE+ 24 SGND 25 IN 26 VDD 28 VCC 27 VIN C7 C6 C5 Figure 9. Application Circuit for a High-Side Buck LED Driver ______________________________________________________________________________________ 17 MAX16821A/MAX16821B/MAX16821C High-Side Buck Driver with Synchronous Rectification MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Inductor Selection Switching MOSFETs The switching frequency, peak inductor current, and allowable ripple at the output determine the value and size of the inductor. Selecting higher switching frequencies reduces inductance requirements, but at the cost of efficiency. The charge/discharge cycle of the gate and drain capacitance in the switching MOSFETs create switching losses worsening at higher input voltages, since switching losses are proportional to the square of the input voltage. The MAX16821A/ MAX16821B/MAX16821C operate up to 1.5MHz. Choose inductors from the standard high-current, surface-mount inductor series available from various manufacturers. Particular applications may require custom-made inductors. Use high-frequency core material for custom inductors. High ∆IL causes large peak-topeak flux excursion increasing the core losses at higher frequencies. The high-frequency operation coupled with high ∆IL reduces the required minimum inductance and makes the use of planar inductors possible. The following discussion is for buck or continuous boost-mode topologies. Discontinuous boost, buckboost, and SEPIC topologies are quite different in regards to component selection. Use the following equations to determine the minimum inductance value: When choosing a MOSFET for voltage regulators, consider the total gate charge, RDS(ON), power dissipation, and package thermal impedance. The product of the MOSFET gate charge and on-resistance is a figure of merit, with a lower number signifying better performance. Choose MOSFETs optimized for high-frequency switching applications. The average current from the MAX16821A/MAX16821B/MAX16821C gate-drive output is proportional to the total capacitance it drives from DH and DL. The power dissipated in the MAX16821A/MAX16821B/MAX16821C is proportional to the input voltage and the average drive current. The gate charge and drain capacitance losses (CV2), the cross-conduction loss in the upper MOSFET due to finite rise/fall time, and the I2R loss due to RMS current in the MOSFET RDS(ON) account for the total losses in the MOSFET. Estimate the power loss (PDMOS_) in the high-side and low-side MOSFETs using the following equations: PDMOS _ HI = (QG × VDD × fSW ) + ⎡ VIN × ILED × (tR + t f ) × fSW ⎤ ⎢ ⎥ + 2 ⎢⎣ ⎥⎦ RDSON × I2RMS−HI Buck regulators: LMIN = (VINMAX − VLED ) × VLED VINMAX × fSW × ∆IL Boost regulators: LMIN = (VLED − VINMAX ) × VINMAX VLED × fSW × ∆IL where VLED is the total voltage across the LED string. The average current-mode control feature of the MAX16821A/MAX16821B/MAX16821C limits the maximum peak inductor current and prevents the inductor from saturating. Choose an inductor with a saturating current greater than the worst-case peak inductor current. Use the following equation to determine the worstcase current in the average current-mode control loop. ILPEAK = VCL ⎛ ∆I ⎞ + ⎜ CL ⎟ ⎝ 2 ⎠ RS where RS is the sense resistor and VCL = 0.030V. For the buck converter, the sense current is the inductor current and for the boost converter, the sense current is the input current. 18 where QG, RDS(ON), tR, and tF are the upper-switching MOSFET’s total gate charge, on-resistance, rise time, and fall time, respectively. IRMS−HI = D ⎛I2 VALLEY + I2 PK + I ⎞ VALLEY × IPK ⎠ × ⎝ 3 For the buck regulator, D is the duty cycle, IVALLEY = (IOUT - ∆IL / 2) and IPK = (IOUT + ∆IL / 2). PDMOS _ LO = (QG × VDD × fSW ) + RDSON × I2 RMS−LO IRMS−LO = (1− D) ⎛I2 VALLEY + I2 PK + I ⎞ VALLEY × IPK ⎠ × ⎝ 3 Input Capacitors The discontinuous input-current waveform of the buck converter causes large ripple currents in the input capacitor. The switching frequency, peak inductor current, and the allowable peak-to-peak voltage ripple reflected back to the source dictate the capacitance requirement. The input ripple is comprised of ∆V Q (caused by the capacitor discharge) and ∆V ESR (caused by the ESR of the capacitor). ______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Output Capacitors The function of the output capacitor is to reduce the output ripple to acceptable levels. The ESR, ESL, and the bulk capacitance of the output capacitor contribute to the output ripple. In most of the applications, the output ESR and ESL effects can be dramatically reduced by using low-ESR ceramic capacitors. To reduce the ESL effects, connect multiple ceramic capacitors in parallel to achieve the required bulk capacitance. In a buck configuration, the output capacitance, COUT, is calculated using the following equation: (VINMAX − VLED ) × VLED COUT ≥ ∆VR × 2 × L × VINMAX × fSW 2 where ∆VR is the maximum allowable output ripple. In a boost configuration, the output capacitance, COUT, is calculated as: (VLED − VINMIN ) × 2 × ILED COUT ≥ ∆VR × VLED × fSW where ILED is the output current. In a buck-boost configuration, the output capacitance, COUT is: 2 × VLED × ILED COUT ≥ ∆VR × (VLED + VINMIN ) × fSW where VLED is the voltage across the load and ILED is the output current. Average Current Limit The average current-mode control technique of the MAX16821A/MAX16821B/MAX16821C accurately limits the maximum output current in the case of the buck configuration. The MAX16821A/MAX16821B/MAX16821C sense the voltage across the sense resistor and limit the peak inductor current (IL-PK) accordingly. The on-cycle terminates when the current-sense voltage reaches 26.4mV (min). Use the following equation to calculate the maximum current-sense resistor value: ⎛ 0.0264 ⎞ RSENSE = ⎜ ⎟ ⎝ ILED ⎠ Select a 5% lower value of RS to compensate for any parasitics associated with the PCB. Select a non-inductive resistor with the appropriate wattage rating. In the case of the boost configuration, the MAX16821A/ MAX16821B/MAX16821C accurately limits the maximum input current. Use the following equation to calculate the current-sense resistor value: ⎛ 0.0264 ⎞ RSENSE = ⎜ ⎟ ⎝ IIN ⎠ where IIN is the input current. Compensation The main control loop consists of an inner current loop (inductor current) and an outer LED current regulation loop. The MAX16821A/MAX16821B/MAX16821C use an average current-mode control scheme to regulate the LED current (Figure 2). The VEA output provides the controlling voltage for the current source. The inner current loop absorbs the inductor pole reducing the order of the LED current loop to that of a single-pole system. The major consideration when designing the current control loop is making certain that the inductor downslope (which becomes an upslope at the output of the CEA) does not exceed the internal ramp slope. This is a necessary condition to avoid subharmonic oscillations similar to those in peak current mode with insufficient slope compensation. This requires that the gain at the output of the CEA be limited based on the following equation: Buck: RCF ≤ VRAMP × fSW × L AV × RS × VLED × gm where VRAMP = 2V, gm = 550µS, AV = 34.5V/V, and VLED is the voltage across the LED string. The crossover frequency of the inner current loop is given by: fC = RS VIN × × 34.5 × gm × RCF VRAMP 2× π ×L For adequate phase margin place the zero formed by RCF and CCZ at least 3 to 5 times below the crossover frequency. The pole formed by RCF and CCP may not be required in most applications but can be added to minimize noise at a frequency at or above the switching frequency. ______________________________________________________________________________________ 19 MAX16821A/MAX16821B/MAX16821C Use low-ESR ceramic capacitors with high ripple-current capability at the input. In the case of the boost topology where the inductor is in series with the input, the ripple current in the capacitor is the same as the inductor ripple and the input capacitance is small. MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Boost: RCF ≤ For adequate phase margin at crossover, place the zero formed by RCF and CCZ at least 3 to 5 times below the crossover frequency. The pole formed by RCF and CCP is added to eliminate noise spikes riding on the current waveform and is placed at the switching frequency. VRAMP × fSW × L AV × RS × (VLED − VIN ) × gm The crossover frequency of the inner current loop is given by: fC = RS VRAMP × PWM Dimming Even though the MAX16821A/MAX16821B/MAX16821C do not have a separate PWM input, PWM dimming can be easily achieved by means of simple external circuitry. See Figures 10 and 11. VLED × 34.5 × gm × RCF 2× π ×L VCC R4 VLED ON/OFF C3 R9 R3 VIN 7V TO 28V R10 C11 C10 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 8 SGND CLKOUT 15 OVI N.C. 7 16 CLP DH 6 C2 R9 Q1 R7 17 EAOUT LX 5 C9 C8 R5 MAX16821A MAX16821B MAX16821C 18 EAN R6 C4 BST 4 19 DIFF DL 3 20 CSN N.C. 2 Q2 Q3 D2 R2 PGND 1 21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 IN 26 VDD 28 VCC 27 VIN C7 LED STRING PWM DIM R1 C6 C5 Figure 10. Low-Side Buck LED Driver with PWM Dimming 20 VLED L1 ______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing MAX16821A/MAX16821B/MAX16821C VCC VLED R4 VCC R8 ON/OFF VIN 7V TO 28V C3 R10 R3 Q5 R9 PWM DIM C2 Q4 C11 C10 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 8 SGND CLKOUT 15 OVI N.C. 7 16 CLP DH 6 R7 L1 VLED D1 Q1 LED STRING R6 17 EAOUT C1 LX 5 C9 PWM DIM C8 MAX16821A MAX16821B MAX16821C 18 EAN Q3 R5 Q2 19 DIFF DL 3 20 CSN N.C. 2 R2 R1 PGND 1 21 CSP SGND 22 PWM DIM BST 4 SENSE23 SENSE+ 24 SGND 25 IN 26 VCC 27 VDD 28 VIN C7 C6 C5 Figure 11. Boost LED Driver with PWM Dimming Power Dissipation Calculate power dissipation in the MAX16821A/ MAX16821B/MAX16821C as a product of the input voltage and the total VCC regulator output current (ICC). ICC includes quiescent current (IQ) and gate-drive current (IDD): PD = VIN x ICC ICC = IQ + [fSW x (QG1 + QG2)] where QG1 and QG2 are the total gate charge of the low-side and high-side external MOSFETs at VGATE = 5V, IQ is the supply current, and fSW is the switching frequency of the LED driver. Use the following equation to calculate the maximum power dissipation (PDMAX) in the chip at a given ambient temperature (TA): PDMAX = 34.5 x (150 – TA) mW ______________________________________________________________________________________ 21 Selector Guide PCB Layout MAX16821A 0.60 1 MAX16821B 0.10 6 MAX16821C 0.03 20 DIFF EAOUT CLP OVI 20 19 18 17 16 15 SGND 22 *EP SENSE- 23 SENSE+ 24 MAX16821A MAX16821B MAX16821C SGND 25 IN 26 VCC 27 + *EP = EXPOSED PAD. 4 5 6 7 DH 3 N.C. 2 LX 1 DL VDD 28 8) Provide enough copper area at and around the switching MOSFETs, inductor, and sense resistors to aid in thermal dissipation. 9) Use 2oz or thicker copper to keep trace inductances and resistances to a minimum. Thicker copper conducts heat more effectively, thereby reducing thermal impedance. Thin copper PCBs compromise efficiency in applications involving high currents. 21 EAN TOP VIEW CSN Pin Configuration BST 7) Distribute the power components evenly across the board for proper heat dissipation. DIFFERENTIAL AMP GAIN (V/V) N.C. 2) Minimize the area and length of the high-current switching loops. 3) Place the necessary Schottky diodes that are connected across the switching MOSFETs very close to the respective MOSFET. 4) Use separate ground planes on different layers of the PCB for SGND and PGND. Connect both of these planes together at a single point and make this connection under the exposed pad of the MAX16821A/MAX16821B/MAX16821C. 5) Run the current-sense lines CSP and CSN very close to each other to minimize the loop area. Run the sense lines SENSE+ and SENSE- close to each other. Do not cross these critical signal lines with power circuitry. Sense the current right at the pads of the current-sense resistors. The current-sense signal has a maximum amplitude of 27.5mV. To prevent contamination of this signal from high dv/dt and high di/dt components and traces, use a ground plane layer to separate the power traces from this signal trace. 6) Place the bank of output capacitors close to the load. PART DIFFERENTIAL SET VALUE (VSENSE+ - VSENSE-) (V) CSP Use the following guidelines to layout the LED driver. 1) Place the IN, V CC , and V DD bypass capacitors close to the MAX16821A/MAX16821B/MAX16821C. PGND MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing I.C. 13 OUTV 12 RT/SYNC 11 EN 10 MODE 9 CLKOUT 8 SGND TQFN Chip Information PROCESS: BiCMOS 22 14 ______________________________________________________________________________________ High-Power Synchronous HBLED Drivers with Rapid Current Pulsing VCC R4 VLED ON/OFF R9 C3 R3 VIN 7V TO 28V R10 C11 C10 14 13 12 I.C. OUTV RT/SYNC 10 MODE 11 EN 9 8 SGND CLKOUT 15 OVI N.C. 7 16 CLP DH 6 C2 R9 Q1 R7 17 EAOUT LX 5 C9 C8 VLED L1 R6 C4 R5 MAX16821A MAX16821B MAX16821C 18 EAN BST 4 19 DIFF DL 3 20 CSN N.C. 2 LED STRING Q2 C1 D2 R2 R1 PGND 1 21 CSP SGND 22 SENSE23 SENSE+ 24 SGND 25 IN 26 VDD 28 VCC 27 VIN C7 C6 C5 Package Information For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. PACKAGE TYPE PACKAGE CODE DOCUMENT NO. 28 TQFN-EP T2855-8 21-0140 ______________________________________________________________________________________ 23 MAX16821A/MAX16821B/MAX16821C Typical Operating Circuit MAX16821A/MAX16821B/MAX16821C High-Power Synchronous HBLED Drivers with Rapid Current Pulsing Revision History REVISION NUMBER REVISION DATE 0 7/07 Initial release 1 3/09 Updated Electrical Characteristics table. DESCRIPTION PAGES CHANGED — 3, 4 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 24 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2009 Maxim Integrated Products Heaney Maxim is a registered trademark of Maxim Integrated Products, Inc.