LM363 Precision Instrumentation Amplifier General Description The LM363 is a monolithic true instrumentation amplifier. It requires no external parts for fixed gains of 10, 100 and 1000. High precision is attained by on-chip trimming of offset voltage and gain. A super-beta bipolar input stage gives very low input bias current and voltage noise, extremely low offset voltage drift, and high common-mode rejection ratio. A two-stage amplifier design yields an open loop gain of 10,000,000 and a gain bandwidth product of 30 MHz, yet remains stable for all closed loop gains. The LM363 operates with supply voltages from g 5V to g 18V with only 1.5 mA current drain. The LM363’s low voltage noise, low offset voltage and offset voltage drift make it ideal for amplifying low-level, lowimpedance transducers. At the same time, its low bias current and high input impedance (both common-mode and differential) provide excellent performance at high impedance levels. These features, along with its ultra-high common-mode rejection, allow the LM363 to be used in the most demanding instrumentation amplifier applications, replacing expensive hybrid, module or multi-chip designs. Because the LM363 is internally trimmed, precision external resistors and their associated errors are eliminated. The 16-pin dual-in-line package provides pin-strappable gains of 10, 100 or 1000. Its twin differential shield drivers eliminate bandwidth loss due to cable capacitance. Compensation pins allow overcompensation to reduce bandwidth and output noise, or to provide greater stability with capacitive loads. Separate output force, sense and reference pins permit gains between 10 and 10,000 to be programmed using external resistors. On the 8-pin metal can package, gain is internally set at 10, 100 or 500 but may be increased with external resistors. The shield driver and offset adjust pins are omitted on the 8-pin versions. The LM363 is rated for 0§ C to 70§ C. Features Y Y Y Y Y Y Y Y Offset and gain pretrimmed 12 nV/0Hz input noise (G e 500/1000) 130 dB CMRR typical (G e 500/1000) 2 nA bias current typical No external parts required Dual shield drivers Can be used as a high performance op amp Low supply current (1.5 mA typ) Typical Connections 16-Pin Package 8-Pin Package G e 10 2, 3, 4, open G e 100 3–4 shorted G e 1000 2–4 shorted TL/H/5609 – 33 TL/H/5609 – 1 Connection Diagrams Metal Can Package Order Number LM363H-10, LM363H-100 or LM363H-500 See NS Package Number H08C C1995 National Semiconductor Corporation TL/H/5609 16-Pin Dual-In-Line Package TL/H/5609 – 2 Order Number 363D See NS Package Number D16C RRD-B30M115/Printed in U. S. A. LM363 Precision Instrumentation Amplifier April 1991 Absolute Maximum Ratings (Note 5) Input Voltage If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/Distributors for availability and specifications. Supply Voltage g 18V Differential Input Voltage Input Current g 10V Equal to Supply Voltage Reference and Sense Voltage g 25V Lead Temp. (Soldering, 10 sec.) ESD rating to be determined. 300§ C g 20 mA LM363 Electrical Characteristics (Notes 1 and 2) LM363 Parameter Conditions Typ Tested Limit (Note 3) Design Limit (Note 4) 150 250 2.5 400 700 6 mV mV mV 4 8 75 mV/§ C mV/§ C mV/§ C Units FIXED GAIN (8-PIN) Input Offset Voltage G e 500 G e 100 G e 10 30 50 0.5 Input Offset Voltage Drift G e 500 G e 100 G e 10 1 2 20 Gain Error ( g 10V Swing, 2 kX Load) G e 500 G e 100 G e 10 0.1 0.07 0.05 0.8 0.7 0.6 0.9 0.8 0.7 % % % Input Offset Voltage G e 1000 G e 100 G e 10 50 100 1 250 450 3.5 500 900 8 mV mV mV Input Offset Voltage Drift G e 1000 G e 100 G e 10 1 2 10 5 10 100 mV/§ C mV/§ C mV/§ C Gain Error ( g 10V Swing, 2 kX Load) G e 1000 G e 100 G e 10 2.0 0.1 0.6 3.5 0.8 2.3 % % % Gain Temperature Coefficient G e 1000 G e 500 G e 100, 10 40 20 10 Gain Non-Linearity ( g 10V Swing, 2 kX Load) G e 10, 100 G e 500, 1000 PROGRAMMABLE GAIN (16-PIN) 3.0 0.7 2.0 FIXED GAIN AND PROGRAMMABLE 0.01 0.01 2 ppm/§ C ppm/§ C ppm/§ C 0.03 0.05 0.04 0.06 % % LM363 Electrical Characteristics (Continued) (Notes 1 and 2) LM363 Parameter Conditions Typ Tested Limit (Note 3) Design Limit (Note 4) Units Common-Mode Rejection Ratio (b10VsVCMs10V) G e 1000, 500 G e 100 G e 10 130 120 105 114 94 90 104 84 80 dB dB dB Positive Supply Rejection Ratio (5V to 15V) G e 1000, 500 G e 100 G e 10 130 120 100 110 100 85 100 95 78 dB dB dB Negative Supply Rejection Ratio (b5V to b15V) G e 1000, 500 G e 100 G e 10 120 106 86 100 85 70 90 75 60 dB dB dB Input Bias Current 2 10 20 nA Input Offset Current 1 3 5 100 8 Common-Mode Input Resistance Differential Mode Input Resistance G e 1000, 500 G e 100 G e 10 0.2 2 20 Input Offset Current Change b 11V s VCM s 13V 20 Reference and Sense Resistance GX GX GX 100 300 pa/V 30 80 27 83 kX kX kX 50 Min Max nA GX Open Loop Gain GCL e 1000, 500 10 1 Supply Current Positive Negative 1.2 1.6 2.4 2.8 V/mV 3.0 3.4 mA mA Note 1: These conditions apply unless otherwise noted; V a e 15V, Vb eb 15V, VCM e 0V, RL e 2 kX, reference pin grounded, sense pin connected to output and Tj e 25§ C. Note 2: Boldface limits are guaranteed over full temperature range. Operating ambient temperature range is 0§ C to 70§ C for the LM363. Note 3: Guaranteed and 100% production tested. Note 4: Guaranteed but not 100% tested. These limits are not used in determining outgoing quality levels. Note 5: Maximum rated junction temperature is 100§ C for the LM363. Thermal resistance, junction to ambient, is 150§ C/W for the TO-99(H) package and 100§ C/W for the ceramic DIP (D). 3 Typical Performance Characteristics TA e 25§ C Fixed Gain and Programmable Parameter 1000/500 Units 100 10 Input Voltage Noise, rms, 1 kHz 12 18 90 Input Voltage Noise (Note 6) 0.4 1.5 10 nV/ SHz mVp-p Input Current Noise, rms, 1 kHz 0.2 0.2 0.2 pA/ SHz pAp-p Input Current Noise (Note 6) 40 40 40 Bandwidth 30 100 200 kHz Slew Rate 1 0.36 0.24 V/ms Settling Time, 0.1% of 10V 70 25 20 ms Offset Voltage Warm-Up Drift (Note 7) 5 15 50 mV Offset Voltage Stability (Note 8) 5 10 100 mV 0.01 0.005 0.05 % Gain Stability (Note 8) Note 6: Measured for 100 seconds in a 0.01 Hz to 10 Hz bandwidth. Note 7: Measured for 5 minutes in still air, V a e 15V, Vb eb 15V. Warm-up drift is proportionally reduced at lower supply voltages. Common-Mode Input Voltage Limit Supply Current vs Supply Voltage Input Bias Current vs Temperature Output Swing Referred to Supplies Supply Current vs Temperature Input Offset Current vs Temperature TL/H/5609 – 3 4 Typical Performance Characteristics (Continued) Output Current Limit Input Noise Voltage Input Current Noise Input Current vs Voltage Overdrive Gain Non-Linearity Gain Error vs Frequency* *Trimmed to zero at 100 Hz Gain Error vs Frequency* Positive Power Supply Rejection Negative Power Supply Rejection Negative Power Supply Rejection Negative Power Supply Rejection *Trimmed to zero at 100 Hz Negative Power Supply Rejection TL/H/5609 – 4 5 Typical Performance Characteristics (Continued) CMRR with Balanced Source Resistance CMRR with Balanced Source Resistance CMRR with Balanced Source Resistance CMRR with Unbalanced Source Resistance CMRR with Unbalanced Source Resistance CMRR with Unbalanced Source Resistance CMRR with Balanced Source Resistance CMRR with Balanced Source Resistance CMRR with Balanced Source Resistance CMRR with Unbalanced Source Resistance CMRR with Unbalanced Source Resistance CMRR with Unbalanced Source Resistance TL/H/5609 – 5 6 Typical Performance Characteristics (Continued) Shield Driver Bias Voltage Shield Driver Loading Error Shield Driver Loading Error Shield Driver Loading Error Small Signal Transient Response Small Signal Transient Response Small Signal Transient Response Small Signal Transient Response Large Signal Transient Response Large Signal Transient Response Large Signal Transient Response Large Signal Transient Response TL/H/5609 – 6 7 Simplified Schematic (pin numbers in parentheses are for 8-pin package) TL/H/5609 – 7 Theory of Operation This voltage divided by the attenuation factor R2 R4 e R3 a R4 R1 a R2 is equal to the output-to-reference voltage. Hence, the overall gain is given by VOUT R3 a R4 RE3-4 e c . Ge VIN R4 RE1-2 Referring to the Simplified Schematic, it can be seen that the input voltage is applied across the bases of Q1 and Q2 and appears between their emitters. If RE1-2 is the resistance across these emitters, a differential current equal to VIN/RE1-2 flows from Q1’s emitter to Q2’s. The second stage amplifier shown maintains Q1 and Q2 at equal collector currents by negative feedback to Q4. The emitter currents of Q3 and Q4 must therefore be unbalanced by an amount equal to the current flow across RE1-2. Defining RE3-4 e R5 a R6, the differential voltage across the emitters of Q4 to Q3 is equal to VIN c RE3-4. RE 1-2 8 Application Hints overdrives these diodes conduct, greatly increasing input currents. This behavior is illustrated in the IIN vs VIN plot in the Typical Performance Characteristics. (The graph is not symmetrical because at large input currents a portion of the current into the device flows out the V b terminal.) The input protection resistors allow a full 10V differential input voltage without degradation even at G e 1000. At input voltages more than one diode drop below Vb or two diode drops above V a input, current increases rapidly. Diode clamps to the supplies, or external resistors to limit current to 20 mA, will prevent damage to the device. The LM363 was designed to be as simple to use as possible, but several general precautions must be taken. The differential inputs are directly coupled and need a return path to power supply common. Worst-case bias currents are only 10 nA for the LM363, so the return impedance can be as high as 100 MX. Ground drops between signal return and IC supply common should not be ignored. While the LM363 has excellent common-mode rejection, signals must remain within the proper common-mode range for this specification to apply. Operating common-mode range is guaranteed from b10V to a 10V with g 15V supplies. The high-gain (500 or 1000) versions have large gain-bandwidth products (15 MHz or 30 MHz) so board layout is fairly critical. The differential input leads should be kept away from output force and sense leads, especially at high impedances. Only 1 pF from output to positive input at 100 kX source impedance can cause oscillations. The gain adjust leads on the 16-pin package should be treated as inputs and kept away from the output wiring. REFERENCE AND SENSE INPUTS The equivalent circuit is shown in the schematic diagram. Limitations for correct operation are as follows. Maximum differential swing between reference and sense pins is typically g 15V ( g 10V guaranteed). If this limit is exceeded, the sense pin no longer controls the output, which then pegs high or low. The negative common-mode limit is 1.5V below Vb. (This is permissible because R2 and R4 are returned to a node biased higher than Vb.) If large positive voltages are applied to the reference and sense pins, the common-mode range of the signal inputs begins to suffer as the drop across R13 and R16 increases. For example, at g 15V supplies, VREF e VSENSE e 0V, signal input range is typically b 12V to a 13.5V. At VREF e VSENSE e 15V, signal input range drops to b11V to a 13.5V. The reference and sense pins can be as much as 10V above V a as long as a restricted signal common-mode range (b10V min) can be tolerated. For maximum bipolar output swing at g 15V supplies, the reference pin should be returned to a voltage close to ground. At lower supply voltages, the reference pin need not be halfway between the supplies for maximum output swing. For example, at V a e a 12V and Vb e b5V, grounding the reference pin still allows a a 11V to b4V swing. For single-supply systems, the reference pin can be tied to either supply if a single output polarity is all that is required. For a bipolar input and output, create a low impedance reference with an op amp and voltage divider or a regulator (e.g., LM336, LM385, LM317L). This forms the reference for all succeeding signal-processing stages. (Don’t connect the reference terminal directly to a voltage divider; this degrades gain error.) See Figure 1 . POWER SUPPLY The LM363 may be powered from split supplies from g 5V to g 18V (or single-ended supplies from 10V to 36V). Positive supply current is typically 1.2 mA independent of supply voltage. The negative supply current is higher than the positive by the current drawn through the voltage dividers for the reference and sense inputs (typ 600 mA total). The LM363’s excellent PSRR often makes regulated supplies unnecessary. Actually, supply voltage can be as low as 7V total but PSRR is severely degraded, so that well-regulated supplies are recommended below 10V total. Split supplies need not be balanced; output swing and input common-mode range will simply not be symmetrical with unbalanced supplies. For example, at a 12V and b5V supplies, input common-mode range is typically a 10.5V to b2V and output swing is a 11V to b4V. When using ultra-low offset versions, best results are obtained at g 15V supplies. For example, the LM363-500’s offset voltage is guaranteed within 150 mV at g 15V at 25§ C. Running at g 5V results in a worst-case negative PSRR error of 10V (b15V to b5V) multiplied by 3.2X10b6 (110 dB) or 32 mV, increasing the worst-case offset. Positive PSRR results in another 10 mV worst-case change. INPUTS The LM363 input circuitry is depicted in the Simplified Schematic. The input stage is run relatively rich (50 mA) for low voltage noise and wide bandwidth; super-beta transistors and bias-current cancellation (not shown) keep bias currents low. Due to the bias-current cancellation circuitry, bias current may be either polarity at either input. While input current noise is high relative to bias current, it is not significant until source resistance approaches 100 kX. Input common-mode range is typically from 3V above V b to 1.5V below V a , so that a large potential drop between the input signal and output reference can be accommodated. However, a return path for the input bias current must be provided; the differential input stage is not isolated from the supplies. Differential input swing in the linear region is equal to output swing divided by gain, and typically ranges from 1.3V at G e 10 to 13 mV at G e 1000. Clamp diodes are provided to prevent zener breakdown and resulting degradation of the input transistors. At large input a. Usual configuration maximizes bipolar output swing. TL/H/5609 – 8 b. Unequal supplies, output ground referred. Full output swing preserved referred to supplies. FIGURE 1. Reference Connections 9 Application Hints (Continued) TL/H/5609 – 9 c. Single Supply, Unipolar Output d. Single Supply, Bipolar Output FIGURE 1. Reference Connections (Continued) duce an offset shift. A simple low-pass RC filter will usually cure this problem (Figure 2 ). Use film type resistors for their low thermal EMF. In highly noisy environments, LC filters can be substituted for increased RF attenuation. OUTPUTS The LM363’s output can typically swing within 1V of the supplies at light loads. While specified to drive a 2 kX load to g 10V, current limit is typically 15 mA at room temperature. The output can stably drive capacitive loads up to 400 pF. For higher load capacitance, the amplifier may be overcompensated (see COMPENSATION section, following). The output may be continuously shorted to ground without damaging the device. OFFSET VOLTAGE The LM363’s offset voltage is internally trimmed to a very low value. Note that data sheet values are given at Tj e 25§ C, VCM e 0V and V a e Vb e 15V. For other conditions, warm-up drift, temperature drift, common-mode rejection and power supply rejection must be taken into account. Warm-up drift, due to chip and package thermal gradients, is an effect separate from temperature drift. Typical warm-up drift is tabulated in the Electrical Characteristics; settling time is approximately 5 minutes in still air. At load currents up to 5 mA, thermal feedback effects are negligible (DVOSs2mV at G e 1000). Care must be taken in measuring the extremely low offset voltages of the high gain amplifiers. Input leads must be held isothermal to eliminate thermocouple effects. Oscillations, due to either heavy capacitive loading or stray capacitance from input to output, can cause erroneous readings. In either case, overcompensation will help. High frequency noise fed into the inputs may be rectified internally, and pro- TL/H/5609 – 10 FIGURE 2. Low Pass Filter Prevents RF Rectification Instrumentation amplifiers have both an input offset voltage (VIOS) and an output offset voltage (VOOS). The total inputreferred offset voltage (VOSRTI) is related to the instrumentation amplifier gain (G) as follows: VOSRTI e VIOS a VOOS/ G. The offset voltage given in the LM363 specifications is the total input-referred offset. As long as only one gain is used, offset voltage can be nulled at either input or output as shown in Figures 3a and 3b . When the 16-pin device is used at multiple gain settings, both VIOS and VOOS should be nulled to get minimum offset at all gains, as shown in Figure 3c . The correct procedure is to trim VOOS for zero output at G e 10, then trim VIOS at G e 1000. TL/H/5609 – 11 FIGURE 3. Offset Voltage Trimming 10 Application Hints (Continued) Because the LM363’s offset voltage is so low to begin with, offset nulling has a negligible effect on offset temperature drift. For example, zeroing a 100 mV offset, assuming external resistor TC of 200 ppm/§ C and worst-case internal resistor TC, results in an additional drift component of 0.08 mV/§ C. For this reason, drift specifications are guaranteed, with or without external offset nulling. worst-case output offset of 50 mV, creating an input-referred error of 5 mV at G e 10 or 50 mV at G e 1000. Increasing gain this way increases output offset error. An LM363H-100 may have an output offset of 5 mV, resulting in input referred offset component of 50 mV. Raising the gain to 200 yields a 10 mV error at the output and changes input referred error by an additional 50 mV. External resistors connected to the reference and sense pins can only increase the gain. If ultra-low output impedance is not critical, the technique in Figure 5 can be used to trim the gain to nominal value. Alternatively, the VOS adjustment terminals on the 16-pin package may be used to trim the gain (Figure 10b ). GAIN ADJUSTMENT Gain may be increased by adding an external voltage divider between output force and sense and reference; the preferred connection is shown in Figure 4 . Since both the sense and reference pins look like 50 kX ( g 20 kX) to Vb, impedances presented to both pins must be equal to avoid offset error. For example, a 100X imbalance can create a R1 and R2 should be as low as possible to avoid errors due to 50 kX input impedance of reference and sense pins. Total resistance (R2 a 2R1) should be above 4 kX, however, to prevent excessive load on the LM363 output. The exact formula for calculating gain (G) is: # G e GO 1 a 2R1 R1 a R2 50k GO e preset gain J The last term may be ignored in applications where gain accuracy is not critical. The table below gives suggested values for R1 and R2 along with the calculated error due to ‘‘closest value’’ standard 1% resistors. Total gain error tolerance includes contributions from LM363 GO error and resistor tolerance ( g 1%) and works out to approximately 2.5% in every case. Pinout shown is for 16-pin package. This same technique can also be used with 8-pin versions. TL/H/5609–12 Gain Increase 1.5 2 2.5 3 4 5 6 7 8 9 10 R1 1.21k 1.21k R2 5k 2.49k 2k 2k 1.78k 2k 2.49k 2.94k 3.48k 3.92k 4.42k 2.74k 2.05k 1.21k 1k 1k 1k 1k 1k Error (typ) a 0.6% b 0.2% 0 1k b 0.3% b 0.6% a 0.8% a 0.5% b 0.9% a 0.4% b 0.9% b 0.7% FIGURE 4. Increasing Gain Pinout shown is for 8-pin versions. This same technique can also be used with 16-pin version. TL/H/5609 – 13 FIGURE 5. Adjusting Gain, Alternate Technique 11 Application Hints (Continued) Heavy Miller overcompensation on the 16-pin package can degrade AC PSRR. A large capacitor between pins 15 and 16 couples transients on the positive supply to the output buffer. Since the amplifier bandwidth is severely rolled off it cannot keep the output at the correct state at moderate frequencies. Hence, for good PSRR, either keep the Miller capacitance under 1000 pF or use the pin 15-to-ground compensation shown in Table I. COMPENSATION AND OUTPUT CLAMPING The LM363 is internally compensated for unity feedback from output to sense. Increasing gain with external dividers will decrease the bandwidth and increase stability margin. Without external compensation, the amplifier can stably drive capacitive loads up to 400 pF. When used as an op amp (sense and reference pins grounded, feedback to inverting input), the LM363 is stable for gains of 100 or more. For greater stability, the device may be over-compensated as in Figure 6 . Tables I and II depict suggested compensation components along with the resulting changes in large and small signal bandwidth for the 8-pin and 16-pin packages, respectively. Note that the RC network from pin 8 of the 8-pin device to ground has a large effect on power bandwidth, especially at low gains. The Miller capacitance utilized for overcompensating the 16-pin device permits higher slew rate and larger load capacitance for the same bandwidth, and is preferred when bandwidth must be greatly reduced (e.g., to reduce output noise). TL/H/5609 – 14 FIGURE 6. Overcompensation TABLE I. Overcompensation on 8-Pin Package Compensation Network (Pin 8 to Ground) ² Gain Ð 100 pF, 15k 1000 pF, 5k 0.01 mF,500X 0.1 mF Ð 100 pF, 15k 1000 pF, 5k 0.01 mF, 500X 0.1 mF Ð 100 pF, 15k 1000 pF, 5k 0.01 mF, 500X 0.1 mF 500 100 10 Small Signal 3 dB Bandwidth (kHz) 125 95 45 10 1 240 170 80 20 2 240 170 90 20 2 Power Bandwidth ( g 10V Swing) (Hz) 100k 15k 1.8k 200 20 100k 15k 1.8k 200 20 100k 15k 1.8k 200 20 Maximum Capacitive Load (pF) 400 600 800 1000* 1000* 400 900 1200 1600* 2000* 400 900 1200 1600* 2000* *Also stable for CL t 0.05 mF ² Pin 15 to ground on 16-pin package TABLE II. Overcompensation on 16-Pin Package Gain 1000 100 10 Compensation Capacitor (Pin 15 to 16) Ð 10 pF 100 pF 1000 pF 0.01 mF Ð 10 pF 100 pF 1000 pF 0.01 mF Ð 10 pF 100 pF 1000 pF 0.01 mF Small Signal 3 dB Bandwidth (Hz) 45k 16k 2.5k 250 25 140k 50k 7.5k 750 75 180k 60k 9k 900 90 *Also stable for CL t 0.05 mF 12 Power Bandwidth ( g 10V Swing) (Hz) 45k 16k 2.5k 250 25 100k 50k 7.5k 750 75 90k 50k 9k 900 90 Maximum Capacitive Load (pF) 1000* 2000* 2500* 3000* 3000* 900 1600 2000* 2000* 2000* 600 1100 1600 2000* 2000* Application Hints (Continued) 50 pF to ground at both shield driver outputs. Do not use only one shield driver for a single-ended signal as oscillations can result; shield driver to input capacitance must be roughly balanced ( g 30%). To further reduce noise pickup, the shielded signal lines may be enclosed together in a grounded shield. If a large amount of RF noise is the problem, the only sure cure is a filter capacitor at both inputs; otherwise the RFI may be internally rectified, producing an offset. DC loading on the shield drivers should be minimized. The drivers can only source approximately 40 mA; above this value the input stage bias voltages change, degrading VOS and CMRR. While the shield drivers can sink several mA, VOS may degrade severely at loads above 100 mA (see Shield Driver Loading Error curve in Typical Performance Characteristics). Because the shield drivers are one diode drop above the input levels, unbalanced leakage paths from shield to input can produce an input offset at high source impedances. Buffering with emitter-followers (Figure 8b ) reduces this leakage current by reducing the voltage differential and eliminates any loading on the amplifier. Because the LM363’s output voltage is approximately one diode drop below the voltage at pin 15 (pin 8 for the 8-pin device), this point may be used to limit output swing as seen in Figure 7a . Current available from this pin is only 50 mA, so that zeners must have a sharp breakdown to clamp accurately. Alternatively, a diode tied to a voltage source could be used as in Figure 7b . TL/H/5609 – 15 FIGURE 7. Output Clamp SHIELD DRIVERS When differential signals are sent through long cables, three problems occur. First, noise, both common-mode and differential, is picked up. Second, signal bandwidth is reduced by the RC low-pass filter formed by the source impedance and the cable capacitance. Finally, when these RC time constants are not identical (unbalanced source impedance and/or unbalanced capacitance), AC common-mode rejection is degraded, amplifying both induced noise and ‘‘ground’’ noise. Either filtering at the amplifier inputs or slowing down the amplifier by overcompensating will indeed reduce the noise, but the price is slower response. The LM363D’s dual shield drivers can actually increase bandwidth while reducing noise. The way this is done is by bootstrapping out shield capacitance. The shield drivers follow the input signal. Since both sides of the shield capacitance swing the same amount, it is effectively out of the circuit at frequencies of interest. Hence, the input signal is not rolled off and AC CMRR is not degraded (Figure 8 ). The LM363D’s shield drivers can handle capacitances (shield to center conductor) as high as 1000 pF with source resistances up to 100 kX. For best results, identical shielded cables should be used for both signal inputs, although small mismatches in shield driver to ground capacitance (s500 pF) do not cause problems. At certain low values of cable capacitance (50 pF – 200 pF), high frequency oscillations can occur at high source resistance (t 10 kX). This is alleviated by adding TL/H/5609 – 16 FIGURE 8. Driving Shielded Cables MISCELLANEOUS TRIMMING The VOS adjust and shield driver pins available on the 16pin package may be used to trim the other parameters besides offset voltage, as illustrated in Figure 10 . The bias-current trim relies on the fact that the voltage on the shield driver and gain setting pins is one diode drop respectively above and below the input voltage. Input bias current can be held to within 100 pA over the entire common-mode range, and input offset current always stays under 30 pA. The CMRR trims use the shield driver pins to drive the VOS adjust pins, thus maintaining the LM363’s ultra-high input impedance. 13 Application Hints (Continued) VOS. Both the gain and DC CMRR trims can degrade positive PSRR; the positive PSRR can then be nulled out if desired. The correct order of trimming from first to last is bias current, gain, CMRR, negative PSRR, positive PSRR and VOS. If power supply rejection is critical, frequently only the negative PSRR need be adjusted, since the positive PSRR is more tightly specified. Any or all of the trim schemes of Figure 10 can be combined as desired. As long as the center tap of the 100k trimpot is returned to a voltage 200 mV below V a , the trim schemes shown will not greatly affect Top Trace: Cable Shield Grounded Bottom Trace: Cable Shield Bootstrapped TL/H/5609–17 TL/H/5609 – 18 FIGURE 9. Improved Response using Shield Drivers TL/H/5609 – 19 FIGURE 10. Other Trims for 16-Pin Package 14 Typical Applications 4 mA-20 mA Two Wire Current Transmitter TL/H/5609 – 20 The LM329 reference provides excellent line regulation and gain stability. When bridge is balanced (IOUT e 4 mA), there’s no drop across R3 and R4, so that gain and offset adjustments are non-interactive. The LM334 configured as a zero-TC current source supplies quiescent current to circuit. R11 provides current limiting. Design Equations IOS e (IR6 a IR7) Gain e #1 a J R2 e 4 mA R1 DIOUT AV R2 a R3 a R4 10 mA j X j DVIN R1 R3 a R4 mV when AV e LM363 voltage gain Pick I334 e 0.68V 68 mV j 3.8 mA a R9 R10 VZ b 2.4V e 26 mA IMAX e I334 a R11 IBRIDGE(MAX) j I334-I363-IZ j 1.5mA Precision Current Source (Low Output Current) R1 e R2 IOUT e VIN , l VIN l s 10V GR1 TL/H/5609 – 21 Precision Voltage to Current Converter (Low Input Voltage) R1 e R2 Req e R1 ll 50 kX IOUT e G VIN G VIN e Req 1 kX TL/H/5609 – 22 15 Typical Applications (Continued) Curvature Corrected Platinum RTD Thermometer *70k and 2k should track to 5 ppm/§ C **Less than 5 ppm/§ C drift ² Less than 100 ppm/§ C drift ² ² These resistors should track to 20 ppm/§ C ³ Equivalent circuit, showing lead resistance This thermometer is capable of 0.01§ C accuracy over b 50§ C to a 150§ C. A unique trim arrangement eliminates cumbersome trim interactions so that zero, gain, and nonlinearity correction can be trimmed in one oven trip. Extra op amps provide full Kelvin sensing on the sensor without adding drift and offset terms found in other designs. A2 is configured as a Howland current pump, biasing the sensor with a fixed current. TL/H/5609 – 23 Resistors R2, R3, R4 and R5 from a bridge driven into balance by A1. In balance, both inputs of A1 are at the same voltage. Since R6 e R7, A1 draws equal currents from both legs of the bridge. Any loading of the R4/R5 leg by the sensor would unbalance the bridge; therefore, both bridge taps are given to the sensor open circuit voltage and no current is drawn. Precision Temperature Controller TL/H/5609 – 24 *Ultronix 105A wirewound Thermistor e Yellow Springs Ý44032 Setpoint stability e 2.5X10b4§ C/Hr 16 Typical Applications (Continued) Low Frequency Rolloff (AC Coupling) f1 e 1 e 1 Hz 2qC1(50 kX) f2 e 100 f1 e 100Hz Reduced DC voltage gain attenuates offset error and 1/f noise by a factor of 100. TL/H/5609 – 25 Precision Comparator with Balanced Inputs and Variable Offset Boosted Current Source with Limiting R1 e R2 IO e G VIN R2 IMAX e VBE R2 j 60 mA tpd j 15 mS at 1 mV overdrive DVOUT e V2 a 0.6V Hysteresis e DVOUT e 2 mV G(R1 a R2) Offset e VSENSE/G g 1.3V range TL/H/5609 – 26 Thermocouple Amplifier with Cold Junction Compensation Input protection circuitry allows thermocouple to short to 120 VAC without damaging amplifier. Calibration: 1) Apply 50 mV signal in place of thermocouple. Trim R3 for VOUT e 12.25V. 2) Reconnect thermocouple. Trim R9 for correct output. TL/H/5609 – 27 17 Typical Applications (Continued) Synchronous Demodulator TL/H/5609 – 28 *Use square wave drive produced by optical chopper to run LF13333 switch inputs. Pulsed Bridge Driver/Amplifier TL/H/5609 – 29 18 Typical Applications (Continued) Precision Barometer **Parallel trim for 28.00× Hg e 0V ² Parallel trim for 32.00× Hg e 4V out *B.L.H. Electronics ÝDHF-444114 Pressure Transducer, 350X input impedance. Output e 1 mV/volt excitation/psi TL/H/5609 – 30 Removing Large DC Offsets *Optional bandlimiting to reduce noise. Pick R1C1 e R2C2 e R3C3/10 e 1 2 q fl TL/H/5609 – 31 fl e 0.1 Hz for values shown. Integrator nulls out offset error to LM363 bias currents flowing into R1 and R2. Removing Small DC Offsets *Optional bandlimiting to reduce noise. Low frequency break frequency fl e 1 e 0.01 Hz 2qR1C1 Accommodates out referred offset of several volts. Limit is set by max differential between reference and sense terminals. TL/H/5609 – 32 19 20 Physical Dimensions inches (millimeters) Metal Can Package (H) Order Number LM363H-10, LM363H-100 or LM363H-500 NS Package Number H08C 21 LM363 Precision Instrumentation Amplifier Physical Dimensions inches (millimeters) (Continued) Hermetic Dual-In-Line Package (D) Order Number LM363D NS Package Number D16C LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. 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Tel: 81-043-299-2309 Fax: 81-043-299-2408 National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.