FAIRCHILD AN4105

www.fairchildsemi.com
Application Note AN4105
Design Considerations for Switched Mode Power
Supplies Using A Fairchild Power Switch (FPS) in a
Flyback Converter
Introduction
power MOSFET (SenseFET) and pulse width modulation
(PWM) based control IC in one package. Moreover, they
provide enhanced IC functionality, thereby minimizing the
number of additional components needed in an SMPS.
Fairchild Power Switch (SPS) ICs are widely used in the
power circuits of a variety of equipment, such as color TVs,
printers, PCs, monitors, battery chargers and ac adapters.
They typically incorporate a variety of enhanced protection
functions and they permit a much reduced power
consumption in standby modes.
Flyback switched mode power supplies (SMPS) are among
the most frequently used power circuits in household and
consumer electronics. The basic function of an SMPS is to
supply regulated power to the load on the secondary, or
output side. An SMPS typically incorporates a power
transformer, secondary-side rectifier diodes, a switching
semiconductor device with control IC, and peripheral
circuitry. If the level of integration of the switching and
control circuitry is not high enough, then additional, separate
circuits will be required to accommodate all functions. Such
additional components raise the overall SMPS cost and not
uncommonly reduce reliability.
This application note considers the three major functional
blocks of an SMPS: Fairchild Power Switch (SPS) , flyback
converter, and transformer. It discusses a variety of issues
important to their design and use in the overall SMPS.
Fairchild Power Switches are highly-integrated ICs for
power supply applications. They combine a high-voltage
#3 Vcc
6.3V
2uA
32V
Sync.
Voltage
Ref.
UVLO
Vck
1mA
#4
Feedback
OSC.
#1
Drain
2.5R
#5
Soft Start
& Sync.
S
R
R
Q
LEB
Voffset
#2
Source
GND
Sense
5V
Rsense
Reset
Sense
FET
S
7.5V
Reset
Q
Thermal
Protection
R
OVP
Control IC
Figure1. Internal block diagram of a Fairchild Power Switch(FPS).
Rev. 1.0.2
©2002 Fairchild Semiconductor Corporation
AN4105
APPLICATION NOTE
1. Block Diagram and Basic Operation
of a Fairchild Power Switch(FPS)
1.1 Block Diagram
Figure 1 presents a block diagram of a Fairchild Power
Switch (FPS). It can be divided into several large, functional
sections: under voltage lockout circuitry (UVLO); reference
voltage; oscillator (OSC); pulse width modulation (PWM)
block; protection circuits; and gate driving circuits.
IC current is less than 300µA. IC operation starts when CCC
(Figure 2) charges to 15V. Because only a small current (<1
mA) is allowed to flow in through the resistor during normal
operation this technique reduces the current dissipation in
the SMPS start up resistor.
DC
L INK
Rg
Ccc
V cc
3
1.2 Under Voltage Lockout (UVLO)
In te rn al Bia s
A Fairchild Power Switch (FPS) under voltage lockout
(UVLO) circuitry (Figure 2) guarantees stable operation of
the IC’s control circuit by stopping and starting it as a
function of the value of VCC (Figure 3). The tur off and turn
on voltage thresholds are fixed internally at 10V and 15V,
respectively. Therefore the UVLO circuitry turns off the
control circuit when VCC is lower than 10V and starts it
when VCC is higher than 15V. Once the control circuit starts
operating, VCC must drop below the 10V level for the
UVLO to stop the circuit again. Before switching starts, the
P o we r O n
Re se t
L a tch
Co mp a ra to r
Fairchild Power
Switch(FPS)
SPS
+
5V
V re f
1 5V /1 0 V
-
UV L
Vz
O
6V
Good Logic
Figure 2. Detail of the undervoltage lockout (UVLO)
circuitry in a Fairchild Power Switch. The gate
operating circuit holds in a low state during UVLO,
thereby maintaining the SenseFET at turnoff.
Icc
[mA]
20
Power On
Reset Range
0.3
6V
10V
15V
Vz
Vc
[V]
[V]
c
Figure 3. Fairchild Power Switch(FPS) control circuit status vs. Vcc.
1.3 Feedback Control Circuit
The Fairchild Power Switch(FPS) control IC uses a current
mode PWM and operates such that MOSFET current is
proportional to the feedback voltage Vfb. This limits the
MOSFET current at every cycle. It also offers other
advantages, such as a well regulated SMPS output voltage
with input voltage changes. This method of control also
works successfully in SMPSs used for monitors, which may
have a broad range of synchronizing frequencies to deal
with. As shown in Figure 4, the Fairchild Power Switch
(FPS) oscillator turns on the MOSFET. The feedback
comparator operates to turn it off again, when the MOSFET
2
current reaches a set value proportional to Vfb. The
MOSFET turn off operation is as follows:(1) the internal
(R+2.5R) voltage divider sets the voltage fed back to one
input of the feedback comparator at Vfb /3.5; (2) a current
proportional to the drain current flows to the MOSFET sense
terminal making Vsense proportional to the drain current;
and, (3) when Vsense becomes greater than Vfb, the output of
the feedback comparator goes high, turning off the
MOSFET. Figure 4 also shows that the circuit is designed to
use an opto isolator in the feedback loop. This is appropriate
for an off line design where input to output
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
isolation is required. Cfb improves the noise characteristics.
If the control IC incorporates an error amp however, as in
Fairchild’s KA3842B/3B/4B/5B current mode PWM
controller ICs for SMPSs then a resistor and capacitor are
required to provide the feedback to the error amp. This
would provided the same functions as those provided by the
circuit of Figure 4, e.g., fine control of the output voltage
through Vfb. Similarly, other appropriate devices here are
Fairchild’s LM431/TL431/KA431 series of three terminal
shunt regulators. These have a very sharp turn on
characteristic much like a Zener diode and are widely used in
SMPS secondary side error amplifiers.
Fairchild Power Switch(FPS)
SPS
Precision
Current Source
2uA
Vck
1mA
Idrain
OSC.
Vfb
#4
#1
5V
S
2.5R
On
Cfb
Q
Vfb*
Vss
#2
Sense
R
R
Off
Ioffset
#5
Css
Reset
Rsense
Vsense
Isense
Figure 4. Fairchild Power Switch(FPS) feedback circuit appropriate for off line SMPS use (current mode PWM).
1.4 Example Fairchild Power Switch (FPS)
Control Circuit
Figure 5 shows two approaches to control feedback with a
Fairchild Power Switch (FPS). The design in Figure 5a uses
the LM431 regulator, and that of Figure 5b uses a Zener diode.
Even though zenor diode approach is cost effective, the output
regulation is relatively poor.
In Figure 5a, C1 together with R1 produce high frequency
pole formed by the internal 3.5kΩ resistor and Cfb in the
compensation networrk. R4 limits the maximum current of the
photodiode to 2.3mA [(12V - 2.5V - 2V) /3.3KΩ, where 2.5V
is the LM431's saturation voltage and 2 V is the photodiode's
voltage drop]. Cfb should be determined by considering the
shutdown delay time (see Section 2.1). In Figure 5b, R3 sets a
fixed current to the Zener diode to stabilize its voltage.
©2002 Fairchild Semiconductor Corporation
3
AN4105
APPLICATION NOTE
2uA
Vo, 80V
12V
PC817
Vck
1mA
Vfb
Fairchild Power Switch(FPS)
SPS
OSC.
#4
C1
33n
R3
1k
R1
33k
Cfb
33n
2k
S
2R
Q
Vfb*
R4
3.3k
1k
R
R
Ioffset
10n
R2
1k
KA431
C2
Sense
Rsense
LM431
Control IC
(a) Control
ControlCircuit
circuitusing
thatKA431(LM431)
used KA431(LM431)
Fairchild Power Switch(FPS)
2uA
Vo, 80V
R1
33k
12V
PC817
Vfb
2k
S
2R
Q
Vfb*
1k
2.2k
C1
0.1u
OSC.
#4
Cfb
33n
R3
1k
SPS
Vck
1mA
R
R
Ioffset
Rsense
Sense
R2
Vz
Zener the
Diode
Control
IC
(b) Control
ControlCircuit
circuitusing
thata used
zener
diode
Figure 5. Fairchild Power Switch(FPS) feedback control circuit.
1.5 Soft Start Operation
Normally, the SMPS output voltage increases from start up
with a fixed time constant. This is due to the capacitive
component of the load. At start up, therefore, the feedback
signal applied to the PWM comparator's inverting input
reaches its maximum value (1V), This is because the
feedback loop is effectively open. Also at this time, the drain
current is at its peak value (Ipeak) and maximum allowable
power is being delivered to the secondary load. With that
said, note that when the SMPS pushes maximum power to
the secondary side for this initial fixed time, the entire circuit
is seriously stressed. Use of a soft start function avoids such
stresses. Figure 6 shows how to implement a soft start for a
Fairchild Power Switch(FPS). At turn on, the soft start
capacitor CS on pin 5 of the Fairchild Power Switch(FPS)
starts to charge through the 1mA current source. When the
4
voltage across CS reaches 3V, diode DS turns off. No more
current flows to it from the 1mA current source. Cs then
continues to charge to 5V through the 50kΩ resistor.
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
Fa ircPower
hild Switch(FPS)
Pow e r
Fairchild
1 0V
2㎂
Fairchild Power Switch(FPS)
SPS
Sw itc h (SPS)
1㎃
1㎃
PW M
Co mp a ra to r
5V
Ds
-
5 0k
+
#4
Figure 6. Soft start circuit.
Note that when the voltage across CS exceeds 3V, The voltage
at the comparator’s inverting terminal no longer follows the
voltage across CS. Instead, it follows the output voltage
feedback signal. In shutdown or protection circuit operation,
capacitor CS is discharged, in order to enable it to charge from
0V at restart.
1.6 Synchronization
In an SMPS intended for use with monitors, synchronization
is handled differently than in a general purpose SMPS. For
monitor use, it is necessary to prevent noise from appearing on
the monitor display. To accomplish this, it is necessary to
synchronize the SMPS switching frequency with the
monitor’s horizontal sync frequency. The monitor’s horizontal
scan flyback signal is commonly used as the external sync
signal for the SMPS. By synchronizing the switching with the
horizontal scan’s flyback, the switching noise is positioned at
the far left of the monitor display where it cannot be seen.
Figure 7 shows how to implement the external circuit for
synchronization. The external sync signal, applied across
resistor Rs, cannot drop below 0.6V because of diode Dsync.
After the conclusion of the initial soft start, the voltage across
Cs remains at 5V until the external sync signal is applied, at
which point it looks like VRs of Figure 8. The sync comparator
compares VCs against a 6.3V level and produces the
comparator output waveform, Vcomp of Figure 8.
A Fairchild Power Switch (FPS) has an internal timing
capacitor, Ct. Figure 8 shows that when the voltage on Ct, VCt,
reaches an upper threshold, it begins to discharge; then, when
it reaches a lower threshold, it again starts to charge. This
operation is controlled by the internal oscillator. The oscillator
output signal, VCk in Figure 8, which goes low.
©2002 Fairchild Semiconductor Corporation
V C O MP
V RS
Rs
Dsyn c
-
6 .3V
+
OSC .
S yn c
c o m p a ra t o r
S eries
B
+
#5
Cs
K A5XX Series
Exte rna l
Syn c In p ut
KA2SXX
Rs K
A 5S X X
KA3SXX
K
A 5Q X X
A
V CS
E x t e rn a l
S yn c
I n p ut
#5
Cs
PW M
c o m p a ra to r
-
5V
Figure 7. Synchronization circuit.
when Ct recharges and high when it discharges, is applied to
the Fairchild Power Switch (FPS) S/R Latch Set terminal. In
the absence of an external sync signal, the voltage across Ct
oscillates at the basic frequency of 20kHz. In the presence of
a sync signal, however, the Set signal goes high because VCt
charges to the high threshold following the external sync
signal, and, ultimately, the Set signal, which determines the
switching frequency, synchronizes to the external sync
signal. It is necessary to limit the Set signal's high duration to
5% or less of the full cycle. As the Set signal drops low the
gate turns on. If the device were not synchronized to the
horizontal scan of the monitor, noise would appear on the
screen. When the Set signal goes high, the sync is synchronized with the horizontal scan flyback. Because the high
duration is 5% (maximum) of the full cycle, the start of the
horizontal scan (as the Set signal goes low) turns on the
switch. The switch turn on noise, therefore, is hidden in the
horizontal blanking period at the far left of the monitor
display.
V RS
2V
0V
V CS
5V
0V
V C O MP
0V
V ThH
V CT
0V
V ThL
V ck
0V
Figure 8. Synchronization circuit operation.
5
AN4105
APPLICATION NOTE
2. Fairchild Power Switch(FPS) Built
In Protection Circuits
When the Fairchild Power Switch (FPS) senses an overload,
it waits for a specified time. If the overload is still present
after this, it is considered a true overload and the device
shuts down.
The Fairchild Power Switch (FPS) has a current control that
prohibits current flow above a set maximum, which means
the maximum input power is limited at any given voltage.
Therefore, if the output load tries to draw more than this
level, Vo (Figure 9) drops below the set voltage and LM431
can draw only a given minimum current. As a result, the opto
coupler secondary current drops to almost zero. Almost the
entire current flow at the node is from the Fairchild Power
Switch (FPS) 1mA current source. Hence, the internal 3kΩ
resistor (2.5R + R = 3KΩ) moves Vfb to 3V. From this point
on, however, the 5µA current source starts to charge Cfb,
and, because the opto coupler secondary current is almost
zero, Vfb continues to increase. When Vfb reaches 7.5V, the
Fairchild Power Switch (FPS) shuts down.
Since a Fairchild Power Switch (FPS)’s built in protection
circuits do not require additional, external components,
reliability is increased without increasing cost. Note that the
protective circuitry can completely stop the SMPS operation
(latch mode protection) until the power is turned off and on
again, and can make the control voltage restart above the
ULVO level should the latch be released below ULVO
(Auto Restart Mode protection).
2.1 Output Overload Protection
An overload is any load greater than the load defined as
normal for operation. This is not a short circuit. The
Fairchild Power Switch (FPS) overload protection
determines whether the overload is true or merely transient.
Only a true overload will trigger the overload protection.
2uA
Vck
1mA
Vfb
Vo
Fairchild Power Switch(FPS)
SPS
OSC.
#4
D1
Cfb
D2
S
2.5R
2R
Q
Vfb*
R
R
Ioffset
LM431
KA431
Sense
Rsense
Reset
7.5V
R
Q
Thermal
Shutdown
S
Shutdown
Figure 9. Fairchild Power Switch(FPS) overload protection circuit.
The shutdown delay interval is therefore determined by Cfb.
When Cfb is 10nF, the shutdown delay t2 (see Figure 10b) is
about 9mS. With Cfb at 0.1µF, t2 is about 90ms. Such delay
intervals do not allow the typical transients observed to shut
down the Fairchild Power Switch (FPS). Note that, if a
longer delay is needed, Cfb cannot be made arbitrarily large
because it is important in determining the dynamic response
of the SMPS.
When a large value of Cfb is necessary, a series connected
capacitor and Zener diode can be connected across Cfb as
shown in Figure 10a. The combination works in this way:
When Vfb is below 3V, the low valued Cfb allows the SMPS
to have a good dynamic response. When Vfb is above 3.9V,
the high valued Cd extends the delay time to the desired
6
shutdown point. Of course, where transients are insignificant
and good dynamic response is not required, then eliminate
Cd and the Zener, thereby eliminating their added costs.
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
Fairchild Power Switch(FPS) V
SPS
2uA 1mA
7.5V
Vfb
Vo
#
Cd
Cf
4
D1
D2
Vfb*
b
Vz=3.9V
3.9V
3V
0
t
LM431
KA431
t1
7.5V
Tim e Constant
= 3.5R*Cfb
(a)
t3
t2
2uA = Cfb*0.9V/t2
Shutdown
2uA = Cd*3.6V/t3
< SPS
Delayed
Shutdown
>
<Fairchild
Powerlong
Switch
(FPS) long
Delayed Shutdown>
(b)
Figure 10. Long delay shutdown.
2.2 Output Short Circuit Protection
When the SMPS’s output terminal is short circuited the input
current is at maximum. The output power however, is not a
maximum. This is because there can be no load voltage since
the load is a short circuit. Under such conditions the output
short circuit protection operates as follows.
When the output load short circuits a relatively large
transformer winding in the SMPS, the Fairchild Power
Switch(FPS)’s MOSFET current becomes much greater than
Ipeak because the inductor’s magnetic core is unable to reset.
This is due to low transformer coil voltage at turn off. This
occurs because the current remaining in the inductor during
the Fairchild Power Switch(FPS)’s minimum turn on time
cannot decrease by that amount during the remaining turn off
time. Even though it is a large current, it does not unduly
stress the MOSFET but does greatly stress the transformer’s
secondary coils and the secondary side rectifiers.
In most flyback or forward converters the controller gets its
power from a small secondary bias winding in the
transformer. Furthermore, this controller voltage is proportional to the output voltage (see circuit of Figure 11a). This
is fairly straightforward because, for a flyback converter, the
coil voltage when the switch turns off is proportional to the
output voltage.
For a forward converter, the transformer average voltage at
turn off is proportional to the output voltage. For a flyback
converter SMPS, the output short circuit protection circuit
can be operated in either latch mode or auto restart mode.
The change in Vcc as a function of Rsd is shown in Figure
11b. Rsd is shown in Figure 11a. When Rsd is zero, Vcc
reaches the maximum value of Vtx (see lower primary
winding, Figure 11a), i.e., n'Vsn/n, which is proportional to
the maximum transformer current. For this case, if the output
short circuits, Vcc increases and, after a specified delay, the
©2002 Fairchild Semiconductor Corporation
protection circuit operates, entering the latch mode.
However, when Rsd is made sufficiently large, Vcc can
become smaller than n'Vo. Therefore, if the output short
circuits, Vcc drops, but, if Ccc (see Figure 11a) is sufficiently
large, Vcc stays at a level higher than the UVLO's lower
threshold voltage (10V) until Vfb reaches 7.5V (Figure 11c2) and latch mode protection starts. In contrast, if Ccc is
small enough Vcc approaches the UVLO's low threshold
before Vfb reaches 7.5V (Figure 11c-3), and the UVLO
operates instead of the protection circuit, stopping the device
switching. In such a case, if Vcc exceeds the ULVO's upper
threshold voltage (15V), auto restart operates again. Rsd,
Ccc, and Cfb have affects even at start up and power down, so
their values must be decided upon carefully. Our
experiments have indicated that 10-20W is most appropriate
for Rsd.
7
AN4105
APPLICATION NOTE
Vo
n :1
Vin
V sn
V in
Fairchild Power Switch(FPS)
SPS
1
LM431
C o n t ro l
IC
5
K A 4 31
2
4
3
V fb
V cc
Rsd
C fb
n *: 1
V tx
Ccc
(a ) F ly b a ck c o n v e rt e r
V tx
No Rsd
n *V s n / n
n *V o
L ow Rsd
L a rg e R s d
0
n * V in / n
The voltage Vtx waveform of NB and VCC(the rectified VNB) depending on Rsd
(b ) F la yb a c k co n v e rt e r c o n t ro lle r v o lt a g e a n d R s d d e p e n d e n t V c c
(1 )
V cc
S h u t do w n
(2 )
(3 )
UVLO
S h u t do w n
1 0V
7 . 5V
3V
V fb
Cvcc
t
0V
O u t p ut s h o rt
UVLO
S h u t do w n
(C ) V c c a n d V f b w a v e f o rm s d e p e n d in g o n t h e re la t iv e s iz e o f C v c c a t o u t p u t sh o rt
Figure 11. Operation of the SMPS flyback converter’s output short circuit protection (latch mode ).
8
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
2.3 Fast Protection Without Delay
can be made to operate sufficiently fast. In such a case, when
the transistor is turned off, the protection shifts to auto restart
mode for normal operation. It is also possible to use this
circuit as an output enable circuit. Fast latch mode protection
can be implemented by adding a photocoupler. Thus, when
the output terminal latch mode transistor turns on, a large
current flows through the photodiode PC2. A large current
therefore flows through the primary phototransistor, which
increases Vfb rapidly. This executes fast latch mode
protection with no time delay.
It was mentioned above that the Fairchild Power
Switch(FPS) shutdown capability is associated with a delay,
to allow normal transients to occur without shutting down
the system. In effect, this restricts the protection range. If
fast protection is required, then additional circuitry is called
for. Figure 12 shows how it’s done. A transistor is used to
force the feedback photodiode current to increase causing
the primary side Vfb to be forced below 0.3V. Therefore the
Fairchild Power Switch (FPS) stops switching. Depending
on the magnitude of the photodiode current, the protection
n:1
Vin
Vo
V sn
Vin
P C1
Fairchild Power Switch(FPS)
SPS
5
1
Co n tro l
IC
2
KA4 31
LM431
B urst Mo de
S hu t Do wn
4
3
V fb
Vcc
Rsd
Ccc
PC1
Vo
PC2
n *:1
Vtx
P C2
L atch Mod e
Cfb
Sh u t Do wn
Figure 12. A fast protection circuit without a shutdown delay.
2.4 Overvoltage Protection
The Fairchild Power Switch (FPS) has a self protection
feature that operates even when faults exist in the feedback
path. These could include an open or short circuit. On the
primary side, if the feedback terminal is short circuited, then
the voltage on it is zero and hence the Fairchild Power
Switch (FPS) is unable to start switching.
If the feedback path open circuits, the protection circuit
operates as though a secondary overload is present. Further,
if the feedback terminal looks open due say, to some fault in
the primary feedback circuit, the primary side could switch
at the set maximum current level until the protection circuit
operates. This causes the secondary voltage to become much
greater than the rated voltage. Note that in such a case if
there were no protection circuit, the fuse could blow or, more
serious, a fire could start.
It is possible that, without a regulator, devices directly
connected to the secondary output could be destroyed.
Instead, however, the Fairchild Power Switch (FPS) over
©2002 Fairchild Semiconductor Corporation
voltage protection circuit operates. Since VCC is proportional to the output, in an over voltage situation it will also
increase. In the Fairchild Power Switch (FPS), the protection
circuit operates when Vcc exceeds 25V. Therefore in normal
operation Vcc must be set below 25V.
9
AN4105
APPLICATION NOTE
3. Noise Considerations at switch Turn On
3.1 SMPS Current Sensing
(1) Reverse recovery current is generated when the SMPS
operates in continuous conduction mode (CCM; see Section
4.1). If the MOSFET is turned on while the rectifier diode is
conducting, the diode, during its reverse recovery time will
act like a short circuit. Therefore a large current spike will
flow in the MOSFET. There are three ways to reduce the
magnitude of this (reverse recovery) current: use a fast
recovery diode; reduce the MOSFET’s gate operating
current at turn on; or, increase the transformer leakage
inductance.
(2) The total capacitance on the MOSFET’s drain side
includes the parasitic capacitor Cds between the drain and the
source terminals, the junction capacitance of the snubber
diode, and the capacitance of the transformer windings. At
turn on, the total equivalent capacitor discharges through the
MOSFET.
(3) As shown in the diagram, the MOSFET gate operating
current also flows through the current sensing resistor.
Whether an SMPS is current mode or voltage mode
controlled, or uses some form of non linear control, it
requires the protection afforded by current sensing
capabilities. Even though most current sensing is done using
a sensing resistor or a current transformer, there are
instances where a MOSFET is used by the SMPS to further
reduce current sensing losses. The Fairchild Power Switch
(FPS) switching element is a SenseFET. This minimises any
power losses in the sense resistor, which is integrated onto
the controller chip.
3.2 Current Sensing Waveform Noise After
Turn On
A leading edge spike is present on the current sense line
when the SMPS switching device turns on. It arises from
three causes, as shown in Figure 13: (1) reverse recovery
current; (2) charge/discharge current of the MOSFET shunt
capacitance; and (3) MOSFET gate operating current.
Vo
Vin
Vo
Vin
Rg
Rg
Control
IC
Vo
Vin
Control
IC
Rf
Rs
Rg
Cf
Cf
Control
IC
Rf
Rf
Rs
R
Cf
s
Reverse Recovery Current
Charge/Discharge Current of
the equvalent capacitor
MOSFET Gate Operating
Current
Figure 13. Current sensing waveform noise after turn on.
10
©2002 Fairchild Semiconductor Corporation
AN4105
APPLICATION NOTE
3.3 Dealing With Leading Edge Noise
3.3.1 Burst Mode Operation
Among the measures taken to reduce leading edge noise, the
most commonly used technique is the RC filter. As shown in
Figure 14a the RC filter is effective against the noise, but it
has the disadvantage that it distorts the current sensing signal
so that accurate current sensing becomes difficult.
Furthermore, a large RC value may be difficult to implement
on an IC and may even require a bigger chip. The technique
of leading edge blanking, as presented in Figure 14b and
14c, overcomes the distortion disadvantage of the RC
technique and works as follows. Since the problem noise
arises just after turn on, if a circuit is inserted that ignores the
current sensing line for a fixed time just after turn on,
operation can continue normally regardless of the noise.
Whatever the details of the location and type of circuit used,
the basic idea is to maintain a minimum turn on time i.e.,
touse the shortest turn on time that cannot be terminated
once turn on starts. Duty ratio control with a minimum turn
on time is implemented through a non linear control method
having a very wide control range relative to a linear control.
The non linear control operates such that if load conditions
require a turn on time of 400ns when the minimum turn on
time is set at 500ns, then one switching cycle will turn on at
800ns. The next cycle will be missed, ensuring that the
average turn on time is 400ns. In this case every other cycle
is missed. This is pulse skipping. In this case the switching
frequency will be half that of a linearly controlled system,
thereby improving SMPS efficiency at light loads. The input
power is therefore minimised. The Fairchild KA34063 dc/dc
converter is an example of a non linear control IC.
The aforementioned method can be viewed as an example of
burst mode operation. Burst mode operation, by reducing the
switching frequency, is one of the most useful ways to
improve SMPS efficiency at light loads and to reduce the
standby input power of household appliances, etc. Note that
burst mode operation is not a burst oscillation (as in ringing
choke conversion circuits), which can bring about reliability
problems. There are mainly two types of true burst mode
operation: one type lowers the switching frequency equally.
The other switches at normal frequency for a fixed time and
stops the control IC operation for a large number of cycles.
Even though in the first method the control IC continues to
consume power, the output voltage ripple is minimized.
The second method can be a useful way to reduce the
minimum input power at standby (since obviously the
standby power is greatly reduced when the IC is stopped).
Indeed, it is often used in cell phones to reduce the dc/dc
converter’s power consumption in standby mode. However,
it has the disadvantage of a larger output voltage ripple.
Currently, Europe restricts a household appliance’s standby
input power to less than 5W, and in time it will be required
to be less than 3W. For such needs, burst mode operation
will be a powerful method to satisfy the requirement for
reduced standby input power.
11
©2002 Fairchild Semiconductor Corporation
AN4105
APPLICATION NOTE
Rg
PW M
IC
MO S FE T
Rf
C u rre n t
S e n s in g
Cf
Rs
(a)
< N o ise elim inat ion u sing R C f ilter >
V ck
Vi
OSC
Vo
LEB
Ifb
V fb
S
Va
Q
Vq
R
C fb
Vb
Ifb
V is
F e e d ba c k
C ir cu it
(b)
< In te rn al blo ck d iag ra m f or no ise elim inat ion u sing L E B >
V ck
Va
0
V fb
V is
0
Vb
Vq
0
0
(c) Noise elimination and waveforms in LEB
< N oise e lim ina tio n a nd w an efo rm s in LE B >
Figure 14. Leading edge blanking.
12
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
4. Flyback Converter Operation
4.1 Operation In Continuous Conduction Mode
(CCM)
Figure 15 shows a typical flyback converter. When the current
through the converter’s inductor is always greater than zero
within a switching cycle, the converter is said to operating in
the continuous conduction mode (CCM). Figure 16 shows the
waveforms of CCM operation, which operates as follows.
V i TON = nV O T OFF
nV O
T ON
D
------------ = --------------- = ------------Vi
T OFF
D–1
The input and output currents become:
I i = DI Lm, AVG
I O = n ( 1 – D )I Lm, AVG
4.1.1 For t0 ~ t1 = TON
At t0 the MOSFET turns on. Immediately before t0, the
inductor current was flowing through diode D, but when the
MOSFET turns on, D turns off thereby isolating the output
terminal from the input terminal. At MOSFET turn on, its
VDS goes to zero, hence VD becomes (Vo + Vi/n). During the
interval after MOSFET turn on (i.e., from t0 on), Vi is
applied to Lm, so ILm increases linearly with a slope as
shown by the following equation:
V
Slope = ------iLm
Hence the input and output powers are equal. An ideal
waveform is shown here because the effect of leakage
inductance has been ignored. In reality leakage inductance
will cause ringing.
ID
IO
D
IL m
IC
VD
VL m
Lm
V in
n:1
ID S
When the energy flow is examined, it is seen that the input
power source supplies energy to Lm while the MOSFET is on.
However, since the energy in Lm continues to increase while
the output terminal is isolated from the input terminal, it is Co
that has to supply the output current during this interval.
RO
VO
CO
V DS
Figure 15. A typical flyback converter.
V GS
ON
4.1.2 For t1 ~ t2 = TOFF
At t1 the MOSFET turns off. At the instant of turn off, the
inductor current that had been flowing through the MOSFET
starts to flow through diode D. When D turns on, VDS
becomes (Vi +nVo). During this interval the voltage nVo is
applied to Lm so that ILm decreases in a straight line with the
following slope:
nV O
Slope = ----------Lm
When the energy flow during this interval is examined, it is
seen that the inductor energy is delivered to the output. The
energy in Lm is reduced by the amount of energy it delivers
to the output. When the MOSFET turns on again, at t2, one
switching cycle will end.
0
OFF
T ON
T O FF
VDS
V i+ n V O
0
V Lm
A
Vi
0
B
nV 0
VD
V 0+ V i/ n
0
S lo p e = V i/ L m
S lp o e = n V O /L m
IL m
0
ID S
0
4.1.3 Relationship Between Input And Output
As shown in Figure 16, the colored areas A and B of the
waveform VLm (the voltage applied to the inductor) must be
equal because the average voltage of the inductor or
transformer in steady state is always zero. Therefore:
©2002 Fairchild Semiconductor Corporation
IC
ID
0
IO
0
t0
t1
t2
t3
Figure 16. Flyback converter operating waveforms in
continuous current mode (CCM).
13
AN4105
APPLICATION NOTE
.
4.2 Discontinuous Conduction Mode (DCM)
The appearance of an interval in which the inductor current
becomes zero during a switching cycle marks flyback
converter operation as discontinuous conduction mode
(DCM). As shown in Figure 17, the voltage waveform
applied to the inductor, VLm, becomes more complex in
DCM. Hence, to avoid difficulties in computation TOFF is
not used. Instead, three input and output relationships of a
converter are derived by using TOFF * , the time when the
output rectifier diode is actually conducting.
V GS
ON
OFF
0
VT
= nV T*
i ON
O OFF
VO
T ON
D*
-------- = ------------------ = ---------------Vi
T* OFF
1 – D*
Deriving the above equation again, using Io and the fact that
the input and output powers are equal, Vo is obtained as:
2
( V i T ON )
V O = ----------------------------------------------------------------IO
2 ----- L m ( T ON + T OFF ) + V i
n
The following equation represents the input power:
2
1
P IN = --- L m I Lm,(peak) fsw
2
VDS
V i+nV O
where fsw is the switching frequency.
Vi
0
VLm
Vi
A
4.3 Flyback Converter Design
V i+nV O
0
B
nV 0
4.3.1 Turns Ratio Considerations
VD
V O +V i/n
0
T OFF *
TON
Slope
Slpoe=V i/Lm
IL m
The turns ratio of an SMPS’s flyback converter transformer
is an important variable. It affects the voltage and current
levels associated with the primary side switching device and
the secondary side rectifier, as well as the number of turns on
the transformer and the current through it. A frequently
discussed design concept suggests operating at maximum
duty ratio when the input voltage is a minimum. For
simplified calculations, here it is assumed that operating
conditions change as listed immediately below.
- Vac input: 85 ~ 265Vac
VO
T OFF
Slope=nV O /L m
0
ID S
0
- Vdc (rectified voltage): 100 ~ 400Vdc
ID
IC
0
IO
0
t0
t1
t2
t3
Figure 17. Flyback converter operating waveforms in
discontinuous current mode (DCM).
The boundary condition between DCM and CCM is:
Vi
Ii + I O = ---------- T ON
2Lm
The following input output relationship in DCM is derived
by using the fact that the colored areas A and B of VLm in
Figure 17 must always be equal because, in steady state, the
average inductor (or transformer) voltage is always zero.
14
- Output voltage: 50Vdc
- Inductor current: Continuous conduction mode (CCM)
operation assumed.
The input power taken by the dc source is the product of the
dc voltage and average input current. Using a wide duty
cycle to deliver equal average current reduces efficiency.
A narrow duty cycle increases the effective current on the
primary side, increasing the operating temperature of the
primary winding and the MOSFET. Also, it is best to decide
on a turns ratio, n, based on the device used. If the voltage on
the primary side MOSFET is relatively low (e.g., 600V),
make n small; if it is on the high side (e.g., 800V), make n
large. As the value of n increases, the primary side switching
device current and the secondary side rectifier diode voltage
decreases Hence, with high output voltage and multiple
secondary side outputs, it is advantageous to increase n.
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
4.3.2 Deciding On The Operating Current Mode
As discussed in Sections 4.1 and 4.2, there are two different
operating current modes possible in a flyback converter: the
continuous conduction mode (CCM); and the discontinuous
conduction mode (DCM). Here the advantages and
disadvantages of each are reviewed, to help the designer
make a proper choice between them.
4.3.2.1 Characteristics Of The Discontinuous
Conduction Mode
In a flyback converter design, if discontinuous conduction
occurs just at minimum input voltage and maximum output
power, then discontinuous conduction must be considered to
be the case for all input conditions. The flyback converter’s
input power in discontinuous conduction mode can be
expressed as:
2
1
P I = --- L m I P fsw
2
Regardless of any changes in input voltage, the power
equation indicates that the input current is limited by the
peak value of the current flowing through the MOSFET in
the transformer primary. A Fairchild Power Switch (FPS)
has an integrated overcurrent protection feature (see Section
2.1, above). This feature does not require external
components and operates across the range of input current.
However, the fixed operating current of DCM, tends
somewhat to offset the effect of the larger effective primary
©2002 Fairchild Semiconductor Corporation
side current. The gain is at the low frequency end where core
loss is not a problem since only a minimum number of turns
need be wound. Also, turn on loss is not a serious problem
due to the low input current. Other losses such as eddy
current, skin effect, proximity effect, etc., are not significant.
A more clearly defined advantage of DCM operation is that
it permits the use of a slow and hence, low cost secondary
rectifier diode. In contrast to the continuous conduction
mode, in the discontinuous conduction mode the effective
current is higher, requiring the use of heavier wire and hence
thicker coils. Therefore DCM does not bring an advantage
insofar as transformer construction is concerned. Moreover,
DCM causes the MOSFET operating temperature to increase
because of the large effective primary side current, as was
described above (Section 4.3.1).
4.3.2.2 Characteristics Of The
Continuous Conduction Mode
Since the coils’ effective current is decreased CCM brings
the advantage of lighter wire. The smaller effective current
also reduces MOSFET heating. This is a definite advantage
for average input current. On the other hand, CCM operation
brings with it a need to consider the rectifier diode’s reverse
recovery current. Depending on the diode’s reverse recovery
time (trr), the reverse recovery current may stress the diode
and increase the loss at its end terminals. It is therefore
necessary to use a diode with the minimum trr possible
within the allowable cost range.
15
AN4105
APPLICATION NOTE
n
V i =10 0V
V i= 400 V
V DS
M e ri t s & de m e ri ts
n V O =6 7 V
V DS
n V O =6 7 V
- V o ltag e
V i=1 0 0V
0
V i=4 0 0V
0
VD
VD
V i/n =7 5 V
1
V O= 50 V
0
V i/n =2 6 7 V
V D S=1 67V
V D =1 25 V
V O= 50 V
0
ap p li e d to sw itch d ev ic e is
lo w
-E ffect iv e cu rren t in sw itc h i n g d ev ice
an d p rim ary w in d in g is lo w
-V o ltag e ap p li ed to rect ifi er d io d e is
h ig h
-O u tp u t v o lta g e rip p l e is sm al l
-C o n tro l m u st b e d o n e th ro u g h sh o rt
tu rn - o n tim e
V D S =46 7V
V D =31 7V
V DS
n V O =1 0 0 V
n:1
V DS
n V O =1 0 0 V
VLm
V i=1 0 0V
V i=4 0 0V
0
VD
Vi
VDS
0
VD
2
VD
- Int e rm e di a te D e si gn M e t hod
V i/n =5 0 V
V i/n =2 0 0 V
V O= 50 V
0
V D S=2 00V
V O= 50 V
0
V D =1 00 V
V D S =50 0V
V D =25 0V
V DS
V DS
n V O =1 5 0 V
n V O =1 5 0 V
0
V i=1 0 0V
V i=4 0 0V
0
3
VD
VD
V i/n =3 3 V
V i/n =1 3 3 V
V O= 50 V
0
V D S=2 50V
V D = 83V
0
V O= 50 V
V D S =55 0V
- V o ltag e
ap p li e d to sw itch d ev ic e is
h ig h
-E ffect iv e cu rren t in sw itc h i n g d ev ice
an d p rim ary w in d in g is h ig h
-V o ltag e ap p li ed to rect ifi er d io d e is
lo w
-O u tp u t v o lta g e rip p l e is larg e l
-C o n tro l m u st
tu rn - o n tim e
be
done
th ro u g h
lo n g
V D =18 3V
Figure 18. The current and voltage ratings required on the primary side switching device and the secondary side
rectifier diode depend on the turns ratio (n) selected.
4.3.2.3 Designer’s Choice
As is clear from the above discussion, DCM operation can
be advantageous, in terms of cost and efficiency, if the input
is small and it is required to precisely control the input power
through the primary side switch (MOSFET) current.
On the other hand, for large inputs and where switching turn
16
on loss could be a major problem, a CCM design would be
more advantageous. In conclusion, therefore, the system
designer must decide between the two modes according to
which mode best fits the characteristics of the system being
designed.
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
5. The Transformer
5.1 Why a Transformer Is Needed
There are three reasons for needing a transformer in a power
conversion circuit. The first reason is safety. A transformer
affords electrical isolation between the primary and
secondary sides, as shown in Figure 19a. In addition, a true
ground on the output side helps prevent electric shock. The
second reason is for voltage conversion. For example, if a
dc/dc converter (such as the buck converter shown in Figure
19b) switching at 50kHz is used to obtain 5V from 100V.
The duty cycle would only be 5%. Using a 50kHz switching
frequency, the control circuit may have only about 1µs to
act, which is not an easy task. Even if this were possible, the
internal voltage and current for each element would be very
large reducing efficiency. The problem is aggravated at high
output current. In the above example, using a transformer to
AC220V
DC
& Rectifier
Output
lower the voltage to 10V would then allowing an on time of
about 10µs. This is an advantageous strategy for lowering
cost and raising efficiency. The third reason to use a
transformer has to do with high voltages and voltage fluctuations. For example, even though all control for a 1000V
supply is done at the 5V power source on the GND side, a
transformer is necessary if power is needed for current
sensing at the 1000V output terminal or for other control. If
the isolation voltage between the transformer windings is
sufficient, a 1000V potential difference can be safely
maintained between the primary and secondary windings
and power can be delivered. Further, a transformer is also
required when the power GND has a sudden potential
fluctuation as in a half bridge converter, gate drive power
source.
GND
Vin=100V
100V
Vo=5V
---->
Electric Shock
short when touched Safe to touch
Duty 5%
---->
5V
50kHz --> Ton=1uS
(a) For safty
Safety
5V
Vo=5V
Vin=100V
10:1
1000V
5V
100V
--> 10V
-->
Duty 50%
-->
5V
50kHz -> Ton=10uS
(b) For voltage change
(c) For potential fluctuation
Figure 19. Why a transformer is needed.
5.2 The Ideal Transformer
The transformer is a device that uses inductive coupling
between its windings to deliver power or signals from one
winding to the another. This is usually from the primary
winding to the secondary winding. The voltages across the
windings can be raised or lowered with respect to each other,
and, if necessary, the primary and secondary sides can be
isolated from each other. Figure 20 shows an ideal
transformer, a simplistic model useful in describing the
transformer concept. An ideal transformer which is actually
a fictional concept must satisfy the following three
©2002 Fairchild Semiconductor Corporation
conditions:
1.The coupling coefficient between the windings
is unity (i.e., the leakage flux is zero);
2.The coil loss is zero (the device has no losses).
3.The inductance of each coil is infinite.
17
AN4105
APPLICATION NOTE
IP
IS
I d e a l T ra n sfo rm e r
VP
VS
NP : N
VP : VS =N
VP
S
P
I P*
IP
: N
IS
Lm
VS
S
IP : IS = N S : N P
Figure 20. Ideal transformer.
The input to output voltage ratio of an ideal transformer is
directly proportional to the turns ratio. This is the ratio of the
number of turns on the primary winding to the number of
turns on the secondary. The polarity is represented schematically by the placement of a dot on each winding. Since n =
Vp/Vs, and an ideal transformer has no loss, the current ratio
is inversely proportional to the turns ratio. The current direction is such that it enters on one side and leaves at another.
Thus the sum of all the nI that flow into the dot is zero. The
dot, indicating the winding polarity, is placed to make the
flux direction in the transformer core uniform when current
flows into the dot. Furthermore, in the case of an ideal transformer, if the path on the secondary side windings is opened,
there is no secondary current flow, and the current on the primary side also goes to zero.
5.3 The Real Transformer
Significant differences exist between an ideal transformer
and a real one. In a real transformer:
1. The coupling coefficient between each coil is finite, and
when a gap is placed in the core as is done in many power
transformers, the coupling coefficient becomes still smaller
(i.e., there is leakage flux);
2. There are losses, such as iron (hysteresis) loss, eddy current
loss, coil resistance loss, etc.; and,
3. The inductance of each coil is finite. When a gap is
placed in the core the inductance becomes still smaller.
NP : N S
V P : V S =N P : N S
IP * : I S = N S : N P
Figure 21. A model of an actual transformer showing the
magnetizing inductance Lm, which accounts for energy
storage.
Rp
Llp
Ideal Transformer
Rm
Lm
Lls
Rs
- Rp:Primary side winding resistance
- Rs:Secondary side winding resistance
- Llp:Primary side leakage inductance
- Lls:Secondary side leakage inductance
- Lm:Magnetizing inductance
- Rm:Transform er core loss resistance
Figure 22. A more complete equivalent circuit of an
actual transformer, showing inductances and loss
resistances.
In a real transformer, should the secondary side be opened
current would continue to flow in the primary (as in 3,
above). So, while energy cannot be stored in the ideal
transformer, it is stored in a real transformer. The so-called
magnetizing inductance accounts for this energy storage
phenomenon. The circuit of Figure 21 is a simplistic view of
an actual transformer and shows the magnetizing inductance.
Figure 22 presents a more complete equivalent circuit of a
real transformer, showing inductances and loss resistances
18
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4105
6. Transformer Design
6.1 Core Selection
The maximum power that a transformer core can deliver and
the maximum energy a transformer inductor can store
depends on the shape and size of the core. In general, as the
effective cross sectional area (Ae) increases, more power can
be delivered. Also, as the window area (Aw) on which the
coils are wound increases, more and thicker windings can be
used, allowing a further increase in the power that can be
delivered. The product of Aw and Ae is called the area
product, AP, and the maximum power a transformer can
deliver is proportional to an exponential power of AP.
Indeed, recent transformer theory shows designs depending
almost entirely on AP. In the broader view, a flyback
converter transformer can be viewed as a coupled inductor,
so it's common to design a flyback transformer using
inductor design methods. The two equations below, (a) and
(b), represent two ways to calculate AP. Equation (a) below,
is a method based on whether or not the core is saturated, is
appropriate at low operating frequencies. Equation (b),
limited by core loss, is appropriate at high frequencies. For
any given design, it is necessary to calculate AP using both
equations, and the equation that gives the higher value is the
one that must be considered correct. Equation (a) assumes
that all losses are wire losses and ignores the core (iron) loss.
4 1.31
 LIP I RMS 10 
AP =  -------------------------------
 420KB MAX 
2
[ cm ]
a)
L = Inductance of Transformer
Ip = Operating peak current
Bmax = MAximum operating flux density
Irms = RMS current
L and BMAX are in Henries and Tesla units, respectively, and
K is listed in Table 1, below. From the above equation the
current density (J) per unit area of wire is obtained from the
current by the relationship below, which assumes that the
temperature of the inductor’s “hot spot” is 30°C above
ambient.
J 30 = 420 AP
– 0.24
2
[ A ⁄ cm ]
At any operating frequency high enough so that the core
losses become large, the following equation should be used.
Specifically, it assumes that the total transformer losses are
split equally (50/50) between the wire and the core.
©2002 Fairchild Semiconductor Corporation
4 1.58
 L∆I I

0.66
2
m RMS 10
(K f
+ K ( f )2 )
[ cm ]
AP =  ------------------------------------------
H SW
E sw


130K


b)
L = Inductance of Transformer
Ip = Operating peak current
Irms = RMS current
In equation (b), KH is the hysteresis coefficient (typically 4 x
10-5 for ferrite cores) and KE is the eddy current coefficient
(typically 4 x 10-10 for ferrite cores).
The current density relationship here, represented by the
equation for J30, below, assumes a hot spot temperature of
15°C above ambient, with the iron loss adding on an
additional 15°C (again, a total of 30°C above ambient).
J 30 = 297AP
– 0.24
2
[ A ⁄ cm ]
The parameter K in equation (a) is the product of the window
utilization factor KU with the primary area factor KP.
(See Table 1.) KU is the ratio of the cross sectional area of
the winding’s copper to the entire window area, and it is
significant in setting the isolation between the primary and
secondary sides. Because it is related to the transformer
shape and winding method, the designer should know the
value of KU for the transformer usually used.
The value of KU can vary greatly, depending particularly on
how closely the isolation safety standards (re isolation) are
followed; the KU of Table 1 assumes a general bobbin is
used. KP is the ratio of the area of the primary winding to
that of the total winding. In Table 1, it is unity for the
inductors because a buck boost inductor has no secondary
windings. For the flyback transformer coupled inductor, KP
is usually 0.5, as Table 1 shows; such an inductor has the
highest efficiency when the primary and secondary winding
areas are equal. When there are more secondary windings,
however, the KP for a flyback transformer coupled inductor
can be lower than 0.5. Note that the ease and speed of
obtaining an accurate AP from equations (a) and (b) depends
on the designer's experience. A reasonably good knowledge
of the probable values of the three parameters KU, KP, and J
for the flyback transformer being designed will reduce the
number of trial and error attempts necessary.
19
AN4105
APPLICATION NOTE
Table 1. KU, KP, and K
KU
KP
K = KU KP
CCM Buck, Boost Inductor
0.7
1.0
0.7
DCM Buck, Boost Inductor
0.7
1.0
0.7
CCM Flyback Transformer
0.4
0.5
0.2
DCM Flyback Transformer
0.4
0.5
0.2
6.2 Determining The Number of Turns
6.4 Determining The Gap
Although the equation for the minimum number of flyback
transformer turns can be determined using applied voltage
and maximum turn on time, the equations used here are
derived from the relationship between L and IP.
For an AP determined from equation (a), above, calculate
NMIN using the following equations:
It is not easy to precisely calculate the required gap.
However, the gap can be calculated from the following
equation. It is based on the fringing effect of the surrounding
flux. Note that the calculated value of L is usually larger than
required.
LI P
4
N MIN = ---------------------- 10
B MAX A C
L = Inductance of Transformer
Ip = Operating peak current
Bmax = Maximum operating flux density
Ae = Effective cross-sectional area of core
For an AP determined from equation (b), above, calculate
NMIN using the following equations:
L∆I m
4
N MIN = ------------------- 10
∆B m AC
L = Inductance of Transformer
Ip = Operating peak current
Bmax = Maximum operating flux density
Ae = Effective cross-sectional area of core
6.3 The Windings
The cross sectional area of the winding must be obtained
from the calculated effective current and the current density
(from the appropriate equation above), factoring in the
number of turns (as determined from a calculation of
NMIN). If eddy current loss is not a serious problem simply
divide the effective current by the current density, thereby
determining the coil cross sectional area.
Be aware, however, that as the coil becomes thicker, the
problem of eddy current loss will arise. Using twisted thin
coil strands (Litz wire), instead of a single heavy wire, can
reduce the eddy current loss, but KU will become smaller.
20
2
uO ur N AC –2
- 10 [ cm ]
I g = --------------------------L
Uo = Permeability of free space
Ur = Relative permeability
Therefore, the gap must be changed to obtain the required
value of L.
References
1. Transformer and Inductor Design Handbook. 2nd ed.
Col. Wm. T. McLyman. Marcel Dekker, Inc., 1988.
2. Switch Mode Power Supply Handbook. Keith H.
Billings. McGraw-Hill, Inc., 1989.
KA1H0280RB, KA1M0280RB, KA1H0380RB, KA1M0380RB,
KA1L0380RB, KA1L0380B, KA1H0680B, KA1M0680B,
KA1H0680RFB, KA1M0680RB, KA1M0880B, KA1L0880B,
KA1M0880BF, KA1M0880D, KA2S0680B, KA2S0880B,
KA3S0680RB, KA3S0680RFB, KA3S0880RB, KA3S0880RFB,
KA1H0165RN, KA1H0165R, KA1H0265R, KA1M0265R,
KA1H0365R, KA1M0365R, KA1L0365R, KA1H0565R,
KA1M0565R, KA1M0765RC, KA1M0765R, KA1M0965R,
KA2S0765, KA2S0965, KA2S09655, KA2S1265, KA3S0765R,
KA3S0765RF, KA3S0965R, KA3S0965RF, KA3S1265R,
KA3S1265RF, KA5H0280R, KA5M0280R, KA5H0380R,
KA5M0380R, KA5L0380R, KA5P0680C, KA5H0165R,
KA5M0165R, KA5L0165R, KA5H0165RV, KA5M0165RV,
KA5L0165RV, KA5M0165RI, KA5L0165RI, KA5H0165RN,
KA5M0165RN, KA5L0165RN, KA5H0165RVN,
KA5M0165RVN, KA5L0165RVN, KA5H02659RN,
KA5M02659RN, KA5H0265RC, KA5M0265R, KA5L0265R,
KA5M0365RN, KA5L0365RN, KA5H0365R, KA5M0365R,
KA5L0365R, KA5M0765RQC, KA5M0965Q, KA5S0765C,
KA5S09654QT, KA5S0965, KA5S12656, KA5S1265,
KA5Q0740RT, KA5Q0945RT, KA5Q0765RT, KA5Q12656RT,
KA5Q1265RF, KA5Q1565RF, FSDH0165, FSDH565,
FS6M07652RTC, FS6M12653RTC, FS6S0765RCB,
FS6S0965RT, FS6S0965RC, FS6S1265RB, FS6S1565RB,
FS7M0880, FS8S0765RC
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
©2002 Fairchild Semiconductor Corporation
AN4105
21
AN4105
APPLICATION NOTE
DISCLAIMER
FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY
PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION, OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY
LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER
DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICODUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
2. A critical component is any component of a life support
which (a) are intended for surgical implant into the body, or
device or system whose failure to perform can be
(b) support or sustain life, or (c) whose failure to per-form
reasonably expected to cause the failure of the life support
when properly used in accordance with instructions for use
device or system, or to affect its safety or effectiveness.
provided in the labeling can be reasonably expected to
result in significant injury to the user.
www.fairchildsemi.com
3/19/04 0.0m 002
 2002 Fairchild Semiconductor Corporation