19-4320; Rev 0; 10/08 KIT ATION EVALU E L B AVAILA 4A, 2MHz Step-Down Regulator with Integrated Switches Features The MAX15038 high-efficiency switching regulator delivers up to 4A load current at output voltages from 0.6V to 90% of VIN. The IC operates from 2.9V to 5.5V, making it ideal for on-board point-of-load and postregulation applications. Total output error is less than ±1% over load, line, and temperature ranges. The MAX15038 features fixed-frequency PWM mode operation with a switching frequency range of 500kHz to 2MHz set by an external resistor. The MAX15038 provides the option of operating in a pulse-skip mode to improve light-load efficiency. High-frequency operation allows for an all-ceramic capacitor design. The high operating frequency also allows for small-size external components. The low-resistance on-chip nMOS switches ensure high efficiency at heavy loads while minimizing critical inductances, making the layout a much simpler task with respect to discrete solutions. Following a simple layout and footprint ensures first-pass success in new designs. The MAX15038 comes with a high bandwidth (28MHz) voltage-error amplifier. The voltage-mode control architecture and the voltage-error amplifier permit a type III compensation scheme to be utilized to achieve maximum loop bandwidth, up to 20% of the switching frequency. High loop bandwidth provides fast transient response, resulting in less required output capacitance and allowing for all-ceramic-capacitor designs. The MAX15038 provides two three-state logic inputs to select one of nine preset output voltages. The preset output voltages allow customers to achieve ±1% output-voltage accuracy without using expensive 0.1% resistors. In addition, the output voltage can be set to any customer value by either using two external resistors at the feedback with a 0.6V internal reference or applying an external reference voltage to the REFIN input. The MAX15038 offers programmable soft-start time using one capacitor to reduce input inrush current. o Internal 31mΩ RDS(ON) High-Side and 24mΩ RDS(ON) Low-Side MOSFETs o Continuous 4A Output Current Over Temperature o ±1% Output Accuracy Over Load, Line, and Temperature o Operates from 2.9V to 5.5V VIN Supply o Adjustable Output from 0.6V to (0.9 x VIN) o Soft-Start Reduces Inrush Supply Current o 500kHz to 2MHz Adjustable Switching Frequency o Compatible with Ceramic, Polymer, and Electrolytic Output Capacitors o Nine Preset and Adjustable Output Voltages 0.6V, 0.7V, 0.8V, 1.0V, 1.2V, 1.5V, 1.8V, 2.0V, 2.5V, and Adjustable o Monotonic Startup for Safe-Start into Prebiased Outputs o Selectable Forced PWM or Skip Mode for Light Load Efficiency o Overcurrent and Overtemperature Protection o Output Current Sink/Source Capable with Cycleby-Cycle Protection o Open-Drain Power-Good Output o Lead-Free, 4mm x 4mm, 24-Pin Thin QFN Package Ordering Information PART TEMP RANGE PIN-PACKAGE MAX15038ETG+ -40°C to +85°C 24 Thin QFN-EP* +Denotes a lead-free/RoHS-compliant package. *EP = Exposed pad. Typical Operating Circuit INPUT 2.9V TO 5.5V IN EN BST LX VDD OUT Applications Server Power Supplies POLs ASIC/CPU/DSP Core and I/O Voltages DDR Power Supplies Base-Station Power Supplies Telecom and Networking Power Supplies RAID Control Power Supplies OUTPUT 1.8V, 4A MAX15038 PGND CTL2 FB CTL1 FREQ REFIN SS COMP VDD MODE GND PWRGD Pin Configuration appears at end of data sheet. ________________________________________________________________ Maxim Integrated Products For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. 1 MAX15038 General Description MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches ABSOLUTE MAXIMUM RATINGS IN, PWRGD to GND..................................................-0.3V to +6V VDD to GND ..................-0.3V to the lower of +4V or (VIN + 0.3V) COMP, FB, MODE, REFIN, CTL1, CTL2, SS, FREQ to GND ..........................................-0.3V to (VDD + 0.3V) OUT, EN to GND ......................................................-0.3V to +6V BST to LX..................................................................-0.3V to +6V BST to GND ............................................................-0.3V to +12V PGND to GND .......................................................-0.3V to +0.3V LX to PGND ..................-0.3V to the lower of +6V or (VIN + 0.3V) LX to PGND ..........-1V to the lower of +6V or (VIN + 1V) for 50ns ILX(RMS) (Note 1) ......................................................................4A VDD Output Short-Circuit Duration .............................Continuous Converter Output Short-Circuit Duration ....................Continuous Continuous Power Dissipation (TA = +70°C) 24-Pin TQFN (derate 27.8mW/°C above +70°C) ........2222mW Thermal Resistance (Note 2) θJA.................................................................................36°C/W θJC ..................................................................................6°C/W Operating Temperature Range ...........................-40°C to +85°C Junction Temperature ......................................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed the IC’s package power dissipation limits. Note 2: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial. Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = VEN = 5V, CVDD = 2.2µF, TA = TJ = -40°C to +85°C, typical values are at TA = +25°C, circuit of Figure 1, unless otherwise noted.) (Note 3) PARAMETER CONDITIONS MIN TYP MAX UNITS 5.5 V IN IN Voltage Range IN Supply Current Total Shutdown Current from IN 2.9 4.7 8 5 8.5 VIN = 5V, VEN = 0 10 20 VIN = VDD = 3.3V, VEN = 0 45 fS = 1MHz, no load VIN = 3.3V VIN = 5V mA µA 3.3V LDO (VDD) VDD rising VDD Undervoltage Lockout Threshold VDD Output Voltage LX starts/stops switching VDD falling 2.6 2.35 Minimum glitch-width rejection VIN = 5V, IVDD = 0 to 10mA VDD Dropout VIN = 2.9V, IVDD = 10mA VDD Current Limit VIN = 5V, VDD = 0 2.8 2.55 10 V µs 3.1 3.3 3.5 25 40 mA 0.025 µA 20 ns 1 V 0.8 V 0.08 V V BST BST Supply Current VBST = VIN = 5V, VLX = 0 or 5V, VEN = 0 PWM COMPARATOR PWM Comparator Propagation Delay PWM Peak-to-Peak Ramp Amplitude PWM Valley Amplitude 2 10mV overdrive _______________________________________________________________________________________ 4A, 2MHz Step-Down Regulator with Integrated Switches (VIN = VEN = 5V, CVDD = 2.2µF, TA = TJ = -40°C to +85°C, typical values are at TA = +25°C, circuit of Figure 1, unless otherwise noted.) (Note 3) PARAMETER CONDITIONS MIN TYP MAX UNITS ERROR AMPLIFIER COMP Clamp Voltage, High VIN = 2.9V to 5V, VFB = 0.5V, VREFIN = 0.6V 2 V COMP Clamp Voltage, Low VIN = 2.9V to 5V, VFB = 0.7V, VREFIN = 0.6V 0.7 V COMP Slew Rate VFB step from 0.5V to 0.7V in 10ns 1.6 V/µs COMP Shutdown Resistance From COMP to GND, VIN = 3.3V, VCOMP = 100mV, VEN = VSS = 0 6 Ω Internally Preset Output Voltage Accuracy VREFIN = VSS, MODE = GND FB Set Point Value CTL1 = CTL2 = GND, MODE = GND FB to OUT Resistor All VID settings except CTL1 = CTL2 = GND -1 +1 % 0.594 0.6 0.606 V 5.5 8 10.5 kΩ Open-Loop Voltage Gain 115 dB Error-Amplifier Unity-Gain Bandwidth 28 MHz Error-Amplifier Common-Mode Input Range VDD = 2.9V to 3.5V Error-Amplifier Maximum Output Current VCOMP = 1V, VREFIN = 0.6V FB Input Bias Current CTL1 = CTL2 = GND -125 VCTL_ = 0 -7.2 VCTL_ = VDD +7.2 0 VFB = 0.7V, sinking 1 VFB = 0.5V, sourcing -1 VDD - 2 V mA nA CTL_ CTL_ Input Bias Current Low, falling CTL_ Input Threshold Hysteresis µA 0.8 Float VDD/2 High, rising VDD 0.8 All VID transitions V 50 mV REFIN REFIN Input Bias Current VREFIN = 0.6V REFIN Offset Voltage VREFIN = 0.9V, FB shorted to COMP -185 -4.5 nA +4.5 mV LX (All Pins Combined) LX On-Resistance, High-Side ILX = -2A LX On-Resistance, Low-Side ILX = 2A VIN = VBST - VLX = 3.3V 42 VIN = VBST - VLX = 5V 31 VIN = 3.3V 30 VIN = 5V 24 High-side sourcing LX Current-Limit Threshold 5.7 Low-side sinking LX Leakage Current VIN = 5V, VEN = 0 42 mΩ mΩ 7 7 Zero-crossing current threshold, MODE = VDD 54 A 0.2 VLX = 0 -0.01 VLX = 5V +0.01 µA _______________________________________________________________________________________ 3 MAX15038 ELECTRICAL CHARACTERISTICS (continued) MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches ELECTRICAL CHARACTERISTICS (continued) (VIN = VEN = 5V, CVDD = 2.2µF, TA = TJ = -40°C to +85°C, typical values are at TA = +25°C, circuit of Figure 1, unless otherwise noted.) (Note 3) PARAMETER LX Switching Frequency CONDITIONS VIN = 2.9V to 5V MIN TYP MAX RFREQ = 49.9kΩ 0.9 1 1.1 RFREQ = 23.6kΩ 1.8 2 2.2 Switching Frequency Range 500 LX Minimum Off-Time LX Maximum Duty Cycle RFREQ = 49.9kΩ LX Minimum Duty Cycle RFREQ = 49.9kΩ Average Short-Circuit IN Supply Current OUT connected to GND, VIN = 5V RMS LX Output Current 92 MHz 2000 kHz 78 ns 95 5 UNITS % 15 0.15 % A 4 A ENABLE EN Input Logic-Low Threshold EN falling EN Input Logic-High Threshold EN rising EN Input Current VEN = 0 or 5V, VIN = 5V 0.9 1.5 V V 0.01 µA MODE Logic-low, falling 26 MODE Input-Logic Threshold Logic VDD/2 or floating, rising 50 Logic-high, rising 74 MODE Input-Logic Hysteresis MODE falling 5 MODE = GND -5 MODE = VDD 5 MODE Input Bias Current % VDD % VDD µA SS SS Current VSS = 0.45V, VREFIN = 0.6V, sourcing 6.7 8 9.3 µA THERMAL SHUTDOWN Thermal-Shutdown Threshold Rising Thermal-Shutdown Hysteresis 165 °C 25 °C POWER-GOOD (PWRGD) Power-Good Threshold Voltage VFB falling, VREFIN = 0.6V VFB rising, VREFIN = 0.6V 88 90 92 92.5 % VREFIN Clock cycles Power-Good Edge Deglitch VFB rising or falling 48 PWRGD Output Voltage Low IPWRGD = 4mA 0.03 PWRGD Leakage Current VIN = VPWRGD = 5V, VFB = 0.7V, VREFIN = 0.6V 0.01 µA Current-Limit Startup Blanking 112 Clock cycles Autoretry Restart Time 896 Clock cycles 0.1 V HICCUP OVERCURRENT LIMIT 4 _______________________________________________________________________________________ 4A, 2MHz Step-Down Regulator with Integrated Switches (VIN = VEN = 5V, CVDD = 2.2µF, TA = TJ = -40°C to +85°C, typical values are at TA = +25°C, circuit of Figure 1, unless otherwise noted.) (Note 3) PARAMETER CONDITIONS MIN TYP MAX UNITS FB Hiccup Threshold VFB falling 70 % VREFIN Hiccup Threshold Blanking Time VFB falling 28 µs Note 3: Specifications are 100% production tested at TA = +25°C. Limits over the operating temperature range are guaranteed by design. Typical Operating Characteristics (Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9kΩ, IOUT = 4A, TA = +25°C, circuit of Figure 1, unless otherwise noted.) FREQUENCY vs. INPUT VOLTAGE EFFICIENCY vs. OUTPUT CURRENT 90 2.20 MAX15038 toc02 100 MAX15038 toc01 100 90 MAX15038 toc03 EFFICIENCY vs. OUTPUT CURRENT 2.15 70 VOUT = 1.8V 60 VOUT = 2.5V VOUT = 1.8V 70 VOUT = 1.2V 60 VOUT = 1.2V 2.05 2.00 1.95 TA = +85°C 1.90 50 50 40 40 TA = +25°C TA = -40°C 1.85 VIN = 3.3V 10.0 1.0 0.1 FREQUENCY (MHz) 1.05 1.00 TA = +85°C TA = +25°C TA = -40°C 0.85 MAX15038 toc05a 1.10 -0.05 -0.10 -0.15 -0.20 VOUT = 1.2V -0.25 -0.30 VOUT = 1.8V -0.35 VOUT = 2.5V -0.40 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 5.5 5.0 5.5 VOUT = 2.5V -0.02 VOUT = 1.8V -0.04 -0.06 -0.08 -0.10 VOUT = 1.2V -0.12 -0.50 0.80 4.5 0 -0.45 RFREQ = 49.9kΩ 2.5 4.0 LINE REGULATION (LOAD = 4A) 0 OUTPUT-VOLTAGE CHANGE (%) MAX15038 toc04 1.15 3.5 INPUT VOLTAGE (V) LOAD REGULATION 1.20 0.90 3.0 OUTPUT CURRENT (A) FREQUENCY vs. INPUT VOLTAGE 0.95 2.5 10.0 1.0 OUTPUT CURRENT (A) OUTPUT-VOLTAGE CHANGE (%) 0.1 RFREQ = 23.2kΩ 1.80 MAX15038 toc05b VOUT = 2.5V 80 FREQUENCY (MHz) EFFICIENCY (%) EFFICIENCY (%) 2.10 80 0 1 2 LOAD CURRENT (A) 3 4 2.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V) _______________________________________________________________________________________ 5 MAX15038 ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (continued) (Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9kΩ, IOUT = 4A, TA = +25°C, circuit of Figure 1, unless otherwise noted.) SWITCHING WAVEFORMS (FORCED PWM, 2A LOAD) LOAD TRANSIENT SWITCHING WAVEFORMS (SKIP MODE, NO LOAD) MAX15038 toc07 MAX15038 toc06 VOUT AC-COUPLED 100mV/div MAX15038 toc08 AC-COUPLED 50mV/div VOUT AC-COUPLED 100mV/div VOUT 1A/div 2A/div ILX 2A ILX 0A 0A 5V/div IOUT VLX 5V/div 0A VLX 0V 400ns/div 40µs/div 2µs/div SHUTDOWN WAVEFORM (RLOAD = 0.5Ω) SOFT-START WAVEFORM (RLOAD = 0.5Ω) MAX15038 toc10 MAX15038 toc09 VEN 5V/div VEN 5V/div VOUT 1V/div VOUT 1V/div 0V 0V 400µs/div 10µs/div INPUT SHUTDOWN CURRENT vs. INPUT VOLTAGE MAXIMUM OUTPUT CURRENT vs. OUTPUT VOLTAGE 10 9 8 7 6 MAX15038 toc12 11 10 MAXIMUM OUTPUT CURRENT (A) MAX15038 toc11 12 INPUT SHUTDOWN CURRENT (µA) MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches 9 8 7 6 5 4 3 VEN = 0V 2 5 2.5 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 6 5.0 5.5 0.5 1.0 1.5 2.0 OUTPUT VOLTAGE (V) _______________________________________________________________________________________ 2.5 4A, 2MHz Step-Down Regulator with Integrated Switches EXPOSED PAD TEMPERATURE vs. AMBIENT TEMPERATURE RMS INPUT CURRENT DURING SHORT CIRCUIT vs. INPUT VOLTAGE MAX15038 toc14 0.5 RMS INPUT CURRENT (A) 1V/div VOUT 0V 5A/div IOUT 0A IIN 1A/div 0.4 0.3 0.2 0.1 0A 90 80 70 60 50 40 4A LOAD 30 20 10 VOUT = 0V 0 MEASURED ON A MAX15038EVKIT 0 2.5 400µs/div 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V) 0 20 40 60 80 100 AMBIENT TEMPERATURE (°C) FEEDBACK VOLTAGE vs. TEMPERATURE SOFT-START WITH REFIN MAX15038 toc17 MAX15038 toc16 0.64 0.63 FEEDBACK VOLTAGE (V) 100 MAX15038 toc15 MAX15038 toc13 EXPOSED PAD TEMPERATURE (°C) HICCUP CURRENT LIMIT 1A/div IIN 0.62 0A 0.61 0.5V/div VREFIN 0V 0.60 0.59 1V/div VOUT 0V 0.58 VPWRGD 0.57 2V/div 0V 0.56 -40 -15 10 35 60 85 200µs/div TEMPERATURE (°C) STARTING INTO PREBIASED OUTPUT (MODE = VDD/2, VOUT = 2.5V, 2A LOAD) STARTING INTO PREBIASED OUTPUT (MODE = VDD, VOUT = 2.5V, 2A LOAD) MAX15038 toc19 MAX15038 toc18 5V/div VEN 5V/div VEN 0V 0V 1V/div 1V/div VOUT VOUT 0V 0V 2A IOUT 2A IOUT 0A 0A 5V/div VPWRGD 5V/div VPWRGD 0V 0V 200µs/div 200µs/div _______________________________________________________________________________________ 7 MAX15038 Typical Operating Characteristics (continued) (Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9kΩ, IOUT = 4A, TA = +25°C, circuit of Figure 1, unless otherwise noted.) MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches Typical Operating Characteristics (continued) (Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9kΩ, IOUT = 4A, TA = +25°C, circuit of Figure 1, unless otherwise noted.) STARTING INTO PREBIASED OUTPUT (MODE = VDD/2, VOUT = 2.5V, NO LOAD) STARTING INTO PREBIASED OUTPUT (MODE = VDD, VOUT = 2.5V, NO LOAD) MAX15038 toc21 MAX15038 toc20 VEN 2V/div VEN 2V/div 0V 0V VOUT 1V/div VOUT 1V/div 0V 0V VPWRGD 2V/div VPWRGD 2V/div 0V 0V 200µs/div 200µs/div STARTING INTO PREBIASED OUTPUT ABOVE NOMINAL SETPOINT (VOUT = 1.5V) STARTING INTO PREBIASED ABOVE NOMINAL SETPOINT (VOUT = 1.5V) MAX15038 toc22 MAX15038 toc23 VEN 2V/div VEN 2V/div 0V 0V VOUT 1V/div VMODE = VDD, NO LOAD VOUT 1V/div 0V 0V VPWRGD 2V/div VPWRGD 2V/div VMODE = VDD/2, NO LOAD 0V 1ms/div 1ms/div TRANSITION FROM FORCED PWM TO SKIP MODE TRANSITION FROM SKIP MODE TO FORCED PWM MODE MAX15038 toc25 MAX15038 toc24 VMODE 5V/div VMODE 5V/div VLX 5V/div VLX 5V/div VOUT 0.5V/div VOUT 0.5V/div 0V 0V 2ms/div 8 0V 4ms/div _______________________________________________________________________________________ 4A, 2MHz Step-Down Regulator with Integrated Switches PIN NAME 1 MODE FUNCTION 2 VDD 3 CTL1 4 CTL2 5 REFIN 6 SS 7 GND 8 COMP Voltage Error-Amplifier Output. Connect the necessary compensation network from COMP to FB and OUT. COMP is internally pulled to GND when the IC is in shutdown/hiccup mode. 9 FB Feedback Input. Connect FB to the center tap of an external resistive divider from the output to GND to set the output voltage from 0.6V to 90% of VIN. Connect FB through an RC network to the output when using CTL1 and CTL2 to select any of nine preset voltages. 10 OUT Output-Voltage Sense. Connect to the converter output. Leave OUT unconnected when an external resistive divider is used. 11 FREQ Oscillator Frequency Select. Connect a precision resistor from FREQ to GND to select the switching frequency. See the Frequency Select (FREQ) section. Functional MODE Selection Input. See the MODE Selection section for more information. 3.3V LDO Output. Supply input for the internal analog core. Connect a low-ESR, ceramic capacitor with a minimum value of 2.2µF from VDD to GND. Preset Output-Voltage Selection Inputs. CTL1 and CTL2 set the output voltage to one of nine preset voltages. See Table 1 and the Programming the Output Voltage (CTL1, CTL2) section for preset voltages. External Reference Input. Connect REFIN to SS to use the internal 0.6V reference. Connecting REFIN to an external voltage forces FB to regulate to the voltage applied to REFIN. REFIN is internally pulled to GND when the IC is in shutdown/hiccup mode. Soft-Start Input. Connect a capacitor from SS to GND to set the startup time. Use a capacitor with a 1nF minimum value. See the Soft-Start and REFIN section for details on setting the soft-start time. Analog Ground Connection. Connect GND and PGND together at one point near the input bypass capacitor return terminal. Open-Drain, Power-Good Output. PWRGD is high impedance when VFB rises above 92.5% (typ) of VREFIN and VREFIN is above 0.54V. PWRGD is internally pulled low when VFB falls below 90% (typ) of VREFIN or VREFIN is below 0.54V. PWRGD is internally pulled low when the IC is in shutdown mode, VDD is below the internal UVLO threshold, or the IC is in thermal shutdown. 12 PWRGD 13 BST 14, 15, 16 LX 17–20 PGND Power Ground. Connect all PGND pins externally to the power ground plane. Connect all PGND pins together near the IC. 21, 22, 23 IN Input Power Supply. Input supply range is from 2.9V to 5.5V. Bypass IN to PGND with a 22µF ceramic capacitor. 24 EN Enable Input. Logic input to enable/disable the MAX15038. — EP Exposed Pad. Solder EP to a large contiguous copper plane connected to PGND to optimize thermal performance. Do not use EP as a ground connection for the device. High-Side MOSFET Driver Supply. Internally connected to IN through a PMOS switch. Bypass BST to LX with a 0.1µF capacitor. Inductor Connection. All LX pins are internally shorted together. Connect all LX pins to the switched side of the inductor. LX is high impedance when the IC is in shutdown mode. _______________________________________________________________________________________ 9 MAX15038 Pin Description 4A, 2MHz Step-Down Regulator with Integrated Switches MAX15038 Block Diagram VDD MAX15038 3.3V LDO UVLO CIRCUITRY SHUTDOWN CONTROL EN BST CURRENT-LIMIT COMPARATOR BST SWITCH IN BIAS GENERATOR VOLTAGE REFERENCE THERMAL SHUTDOWN CONTROL LOGIC LX IN SS SOFT-START PGND CURRENT-LIMIT COMPARATOR REFIN OUT ERROR AMPLIFIER 8kΩ PWM COMPARATOR MODE FB CTL1 CTL2 VID VOLTAGECONTROL CIRCUITRY FREQ 1VP-P OSCILLATOR COMP PWRGD SHDN FB COMP CLAMPS 0.9 x VREFIN 10 ______________________________________________________________________________________ GND 4A, 2MHz Step-Down Regulator with Integrated Switches 2.2Ω INPUT 2.9V TO 5.5V OPTIONAL IN C6 22µF C7 0.1µF BST C15 1000pF C10 0.1µF L1 0.47µH MAX15038 OUTPUT 1.8V, 4A LX VDD C5 2.2µF OUT C8 22µF C3 560pF CTL2 C9 0.01µF R3 158Ω CTL1 PGND EN FB FREQ C2 1500pF R2 2.67kΩ REFIN R4 49.9kΩ SS C1 33pF C4 0.022µF COMP MODE GND VDD R1 20kΩ PWRGD Figure 1. 1MHz, All-Ceramic-Capacitor Design with VIN = 2.9V to 5.5V and VOUT = 1.8V Detailed Description The MAX15038 high-efficiency, voltage-mode switching regulator delivers up to 4A of output current. The MAX15038 provides output voltages from 0.6V to 0.9 x VIN from 2.9V to 5.5V input supplies, making it ideal for on-board point-of-load applications. The output voltage accuracy is better than ±1% over load, line, and temperature. The MAX15038 features a wide switching frequency range, allowing the user to achieve all-ceramic-capacitor designs and fast transient responses (see Figure 1). The high operating frequency minimizes the size of external components. The MAX15038 is available in a small (4mm x 4mm), lead-free, 24-pin thin QFN package. The REFIN function makes the MAX15038 an ideal candidate for DDR and tracking power supplies. Using internal low-RDS(ON) (24mΩ for the low-side n-channel MOSFET and 31mΩ for the high-side n-channel MOSFET) maintains high efficiency at both heavy-load and high-switching frequencies. The MAX15038 employs voltage-mode control architecture with a high bandwidth (28MHz) error amplifier. The voltage-mode control architecture allows up to 2MHz switching frequency, reducing board area. The op-amp voltage-error amplifier works with type III compensation to fully utilize the bandwidth of the high-frequency switching to obtain fast transient response. Adjustable soft-start time provides flexibilities to minimize input startup inrush current. An open-drain, power-good (PWRGD) output goes high when VFB reaches 92.5% of VREFIN and VREFIN is greater than 0.54V. The MAX15038 provides option for three modes of operation: regular PWM, PWM mode with monotonic startup into prebiased output, or skip mode with monotonic startup into prebiased output. ______________________________________________________________________________________ 11 MAX15038 Typical Application Circuit MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches Controller Function The controller logic block is the central processor that determines the duty cycle of the high-side MOSFET under different line, load, and temperature conditions. Under normal operation, where the current-limit and temperature protection are not triggered, the controller logic block takes the output from the PWM comparator and generates the driver signals for both high-side and low-side MOSFETs. The break-before-make logic and the timing for charging the bootstrap capacitors are calculated by the controller logic block. The error signal from the voltage-error amplifier is compared with the ramp signal generated by the oscillator at the PWM comparator and, thus, the required PWM signal is produced. The high-side switch is turned on at the beginning of the oscillator cycle and turns off when the ramp voltage exceeds the VCOMP signal or the current-limit threshold is exceeded. The low-side switch is then turned on for the remainder of the oscillator cycle. Current Limit The internal, high-side MOSFET has a typical 7A peak current-limit threshold. When current flowing out of LX exceeds this limit, the high-side MOSFET turns off and the synchronous rectifier turns on. The synchronous rectifier remains on until the inductor current falls below the low-side current limit. This lowers the duty cycle and causes the output voltage to droop until the current limit is no longer exceeded. The MAX15038 uses a hiccup mode to prevent overheating during short-circuit output conditions. During current limit, if VFB drops below 70% of REFIN and stays below this level for 12µs or more, the MAX15038 enters hiccup mode. The high-side MOSFET and the synchronous rectifier are turned off and both COMP and REFIN are internally pulled low. If REFIN and SS are connected together, both are pulled low. The part remains in this state for 896 clock cycles and then attempts to restart for 112 clock cycles. If the fault causing current limit has cleared, the part resumes normal operation. Otherwise, the part reenters hiccup mode again. external capacitor from SS to GND. The required capacitance value is determined as: C= where tSS is the required soft-start time in seconds. The MAX15038 also features an external reference input (REFIN). The IC regulates FB to the voltage applied to REFIN. The internal soft-start is not available when using an external reference. A method of soft-start when using an external reference is shown in Figure 2. Connect REFIN to SS to use the internal 0.6V reference. Use a capacitor of 1nF minimum value at SS. Undervoltage Lockout (UVLO) The UVLO circuitry inhibits switching when VDD is below 2.55V (typ). Once VDD rises above 2.6V (typ), UVLO clears and the soft-start function activates. A 50mV hysteresis is built in for glitch immunity. BST The gate-drive voltage for the high-side, n-channel switch is generated by a flying-capacitor boost circuit. The capacitor between BST and LX is charged from the VIN supply while the low-side MOSFET is on. When the low-side MOSFET is switched off, the voltage of the capacitor is stacked above LX to provide the necessary turn-on voltage for the high-side internal MOSFET. Frequency Select (FREQ) The switching frequency is resistor programmable from 500kHz to 2MHz. Set the switching frequency of the IC with a resistor (RFREQ) connected from FREQ to GND. RFREQ is calculated as: 50kΩ 1 RFREQ = × ( − 0.05µs) 0.95µs fS where fS is the desired switching frequency in Hertz. R1 Soft-Start and REFIN The MAX15038 utilizes an adjustable soft-start function to limit inrush current during startup. An 8µA (typ) current source charges an external capacitor connected to SS. The soft-start time is adjusted by the value of the 8µA × tSS 0.6V REFIN R2 C MAX15038 Figure 2. Typical Soft-Start Implementation with External Reference 12 ______________________________________________________________________________________ 4A, 2MHz Step-Down Regulator with Integrated Switches Table 1. CTL1 and CTL2 Output Voltage Selection CTL1 CTL2 VOUT (V) GND GND 0.6 0.7 VDD VDD GND Unconnected 0.8 GND VDD 1.0 Programming the Output Voltage (CTL1, CTL2) Unconnected GND 1.2 Unconnected Unconnected 1.5 As shown in Table 1, the output voltage is pin programmable by the logic states of CTL1 and CTL2. CTL1 and CTL2 are trilevel inputs: VDD, unconnected, and GND. An 8.06kΩ resistor must be connected between OUT and FB when CTL1 and CTL2 are connected to GND. The logic states of CTL1 and CTL2 should be programmed only before power-up. Once the part is enabled, CTL1 and CTL2 should not be changed. If the output voltage needs to be reprogrammed, cycle power or EN and reprogram before enabling. The output voltage can be programmed continuously from 0.6V to 90% of VIN by using a resistor-divider network from OUT to FB to GND as shown in Figure 3a. CTL1 and CTL2 must be connected to GND. Unconnected VDD 1.8 Shutdown Mode Drive EN to GND to shut down the IC and reduce quiescent current to a typical value of 10µA. During shutdown, the LX is high impedance. Drive EN high to enable the MAX15038. Thermal Protection Thermal-overload protection limits total power dissipation in the device. When the junction temperature exceeds TJ = +165°C, a thermal sensor forces the device into shutdown, allowing the die to cool. The thermal sensor turns the device on again after the junction temperature cools by 20°C, causing a pulsed output during continuous overload conditions. The soft-start sequence begins after recovery from a thermal-shutdown condition. Applications Information IN and VDD Decoupling To decrease the noise effects due to the high switching frequency and maximize the output accuracy of the MAX15038, decouple VIN with a 22µF capacitor from VIN to PGND. Also decouple VDD with a 2.2µF VDD GND 2.0 VDD Unconnected 2.5 MAX15038 Power-Good Output (PWRGD) PWRGD is an open-drain output that goes high impedance when VFB is above 0.925 x VREFIN and VREFIN is above 0.54V for at least 48 clock cycles. PWRGD pulls low when V FB is below 90% of V REFIN or V REFIN is below 0.54V for at least 48 clock cycles. PWRGD is low when the IC is in shutdown mode, VDD is below the internal UVLO threshold, or the IC is in thermal shutdown mode. low-ESR ceramic capacitor from VDD to GND. Place these capacitors as close as possible to the IC. Inductor Selection Choose an inductor with the following equation: L= VOUT × (VIN − VOUT ) fS × VIN × LIR × IOUT(MAX) where LIR is the ratio of the inductor ripple current to full load current at the minimum duty cycle. Choose LIR between 20% to 40% for best performance and stability. Use an inductor with the lowest possible DC resistance that fits in the allotted dimensions. Powdered iron ferrite core types are often the best choice for performance. With any core material, the core must be large enough not to saturate at the current limit of the MAX15038. Output-Capacitor Selection The key selection parameters for the output capacitor are capacitance, ESR, ESL, and voltage-rating requirements. These affect the overall stability, output ripple voltage, and transient response of the DC-DC converter. The output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor’s ESR, and the voltage drop due to the capacitor’s ESL. Estimate the output-voltage ripple due to the output capacitance, ESR, and ESL: VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL) ______________________________________________________________________________________ 13 MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches where the output ripple due to output capacitance, ESR, and ESL is: IP − P VRIPPLE(C) = 8 x COUT x fS VRIPPLE(ESR) = IP − P x ESR I VRIPPLE(ESL) = P − P x ESL tON or: I P VRIPPLE(ESL) = P −P x ESL tOFF or whichever is larger. The peak-to-peak inductor current (IP-P) is: V − VOUT V IP−P = IN x OUT fS × L VIN Use these equations for initial output capacitor selection. Determine final values by testing a prototype or an evaluation circuit. A smaller ripple current results in less output-voltage ripple. Since the inductor ripple current is a factor of the inductor value, the output-voltage ripple decreases with larger inductance. Use ceramic capacitors for low ESR and low ESL at the switching frequency of the converter. The ripple voltage due to ESL is negligible when using ceramic capacitors. Load-transient response depends on the selected output capacitance. During a load transient, the output instantly changes by ESR x ∆ILOAD. Before the controller can respond, the output deviates further, depending on the inductor and output capacitor values. After a short time, the controller responds by regulating the output voltage back to its predetermined value. The controller response time depends on the closed-loop bandwidth. A higher bandwidth yields a faster response time, preventing the output from deviating further from its regulating value. See the Compensation Design section for more details. Input-Capacitor Selection The input capacitor reduces the current peaks drawn from the input power supply and reduces switching noise in the IC. The total input capacitance must be equal or greater than the value given by the following equation to keep the input-ripple voltage within 14 specification and minimize the high-frequency ripple current being fed back to the input source: CIN _ MIN = D x TS x IOUT VIN − RIPPLE where VIN-RIPPLE is the maximum allowed input ripple voltage across the input capacitors and is recommended to be less than 2% of the minimum input voltage. D is the duty cycle (VOUT/VIN) and TS is the switching period (1/fS). The impedance of the input capacitor at the switching frequency should be less than that of the input source so high-frequency switching currents do not pass through the input source, but are instead shunted through the input capacitor. The input capacitor must meet the ripple current requirement imposed by the switching currents. The RMS input ripple current is given by: IRIPPLE = ILOAD × VOUT × (VIN − VOUT ) VIN where IRIPPLE is the input RMS ripple current. Compensation Design The power transfer function consists of one double pole and one zero. The double pole is introduced by the inductor L and the output capacitor CO. The ESR of the output capacitor determines the zero. The double pole and zero frequencies are given as follows: fP1_ LC = fP2 _ LC = 1 ⎛ R + ESR ⎞ 2π x L x C O x ⎜ O ⎟ ⎝ RO + RL ⎠ fZ _ ESR = 1 2π x ESR x CO where RL is equal to the sum of the output inductor’s DCR (DC resistance) and the internal switch resistance, RDS(ON). A typical value for RDS(ON) is 24mΩ (low-side MOSFET) and 31mΩ (high-side MOSFET). RO is the output load resistance, which is equal to the rated output voltage divided by the rated output current. ESR is the total equivalent series resistance of the output capacitor. If there is more than one output capacitor of the same type in parallel, the value of the ESR in the above equation is equal to that of the ESR of a single output capacitor divided by the total number of output capacitors. ______________________________________________________________________________________ 4A, 2MHz Step-Down Regulator with Integrated Switches VIN VP − P C1 = R 2 x π x R3 x (1 + L ) × fC RO 2.5 x where VP-P is the ramp peak-to-peak voltage (1V typ). Due to the underdamped nature of the output LC double pole, set the two zero frequencies of the type III compensation less than the LC double-pole frequency to provide adequate phase boost. Set the two zero frequencies to 80% of the LC double-pole frequency. Hence: R1 = C3 = 1 x L x CO x (RO + ESR) x L x CO x (RO + ESR) RL + RO 0.8 x C1 1 RL + RO 0.8 x R 3 1 f Z2 _ EA = 2π × R3 × C3 f P3 _ EA = L 1 2π × R1 × C2 COUT MAX15038 1 f P2 _ EA = 2π × R2 × C3 R3 OUT R2 C3 FB CTL1 The above equations are based on the assumptions that C1 >> C2, and R3 >> R2, which are true in most applications. Placements of these poles and zeros are determined by the frequencies of the double pole and ESR zero of the power transfer function. It is also a function of the desired close-loop bandwidth. The following section outlines the step-by-step design procedure to calculate the required compensation components for the MAX15038. When the output voltage of the MAX15038 is programmed to a preset voltage, R3 is internal to the IC and R4 does not exist (Figure 3b). When externally programming the MAX15038 (Figure 3a), the output voltage is determined by: C1 R1 CTL2 COMP R4 C2 a) EXTERNAL RESISTIVE DIVIDER L VOUT LX COUT MAX15038 R2 OUT R3 8kΩ 0.6×R3 R4 = (for VOUT > 0.6V) (VOUT −0.6) For a 0.6V output, connect an 80kΩ resistor from FB to OUT. The zero-cross frequency of the close-loop, fC should be between 10% and 20% of the switching frequency, fS. A higher zero-cross frequency results in faster transient response. Once fC is chosen, C1 is calculated from the following equation: VOUT LX C3 FB VOLTAGE SELECT CTL1 CTL2 R1 C1 COMP C2 b) INTERNAL PRESET VOLTAGES Figure 3. Type III Compensation Network ______________________________________________________________________________________ 15 MAX15038 The high switching frequency range of the MAX15038 allows the use of ceramic output capacitors. Since the ESR of ceramic capacitors is typically very low, the frequency of the associated transfer function zero is higher than the unity-gain crossover frequency, fC, and the zero cannot be used to compensate for the double pole created by the output filtering inductor and capacitor. The double pole produces a gain drop of 40dB/decade and a phase shift of 180°. The compensation network error amplifier must compensate for this gain drop and phase shift to achieve a stable high-bandwidth closed-loop system. Therefore, use type III compensation as shown in Figures 3 and 4. Type III compensation possesses three poles and two zeros with the first pole, fP1_EA, located at zero frequency (DC). Locations of other poles and zeros of the type III compensation are given by: 1 f Z1_ EA = 2π × R1 × C1 MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches Setting the second compensation pole, f P2_EA , at fZ_ESR yields: R2 = CO x ESR C3 Set the third compensation pole at 1/2 of the switching frequency. Calculate C2 as follows: 1 C2 = π × R1 × ƒ S The above equations provide application compensation when the zero-cross frequency is significantly higher than the double-pole frequency. When the zero-cross frequency is near the double-pole frequency, the actual zero-cross frequency is higher than the calculated frequency. In this case, lowering the value of R1 reduces the zero-cross frequency. Also, set the third pole of the type III compensation close to the switching frequency if the zero-cross frequency is above 200kHz to boost the phase margin. The recommended range for R3 is 2kΩ to 10kΩ. Note that the loop compensation remains unchanged if only R4’s resistance is altered to set different outputs. MODE Selection The MAX15038 features a mode selection input (MODE) that users can select a functional mode for the device (see Table 2). Table 2. Mode Selection MODE CONNECTION GND OPERATION MODE Forced PWM Unconnected or VDD/2 Forced PWM. Soft-startup into a prebiased output (monotonic startup). Skip Mode. Soft-startup into a prebiased output (monotonic startup). VDD COMPENSATION TRANSFER FUNCTION OPEN-LOOP GAIN THIRD POLE DOUBLE POLE GAIN (dB) POWER-STAGE TRANSFER FUNCTION SECOND POLE FIRST AND SECOND ZEROS Figure 4. Type III Compensation Illustration Forced-PWM Mode Connect MODE to GND to select forced-PWM mode. In forced-PWM mode, the MAX15038 operates at a constant switching frequency (set by the resistor at FREQ terminal) with no pulse skipping. PWM operation starts after a brief settling time when EN goes high. The lowside switch turns on first, charging the bootstrap capacitor to provide the gate-drive voltage for the high-side switch. The low-side switch turns off either at the end of the clock period or once the low-side switch sinks 0.875A current (typ), whichever occurs first. If the lowside switch is turned off before the end of the clock period, the high-side switch is turned on for the remaining part of the time interval until the inductor current reaches 0.58A, or the end of clock cycle is encountered. Starting from the first PWM activity, the sink current threshold is increased through an internal 4-step DAC to reach the current limit of 7A after 128 clock periods. This is done to help a smooth recovery of the regulated voltage even in case of accidental prebiased output in spite of the initial forced-PWM mode selection. 16 Soft-Starting into a Prebiased Output Mode (Monotonic Startup) When MODE is left unconnected or biased to VDD/2, the MAX15038 soft-starts into a prebiased output without discharging the output capacitor. This type of operation is also termed monotonic startup. See the Starting into Prebiased Output waveforms in the Typical Operating Characteristics section for an example. In monotonic startup mode, both low-side and highside switches remain off to avoid discharging the prebiased output. PWM operation starts when the FB voltage crosses the SS voltage. As in forced-PWM mode, the PWM activity starts with the low-side switch turning on first to build the bootstrap capacitor charge. The MAX15038 is also able to start into prebiased with the output above the nominal set point without abruptly discharging the output, thanks to the sink current control of the low-side switch through a 4-step DAC in 128 clock cycles. Monotonic startup mode automatically switches to forced-PWM mode 4096 clock cycles delay ______________________________________________________________________________________ 4A, 2MHz Step-Down Regulator with Integrated Switches Changing from skip mode to forced-PWM mode and vice-versa can be done at any time. The output capacitor should be large enough to limit the output-voltage overshoot/undershoot due to the settling times to reach different duty-cycle set points corresponding to forcedPWM mode and skip mode at light loads. Skip Mode Careful PCB layout is critical to achieve clean and stable operation. It is highly recommended to duplicate the MAX15038 EV kit layout for optimum performance. If deviation is necessary, follow these guidelines for good PCB layout: Connect MODE to VDD to select skip mode. In skip mode, the MAX15038 switches only as necessary to maintain the output at light loads (not capable of sinking current from the output), but still operates with fixed-frequency (set by the resistor at FREQ terminal) PWM at medium and heavy loads. This maximizes light-load efficiency and reduces the input quiescent current. In case of prolonged high-side idle activity (beyond eight clock cycles), the low-side switch is turned on briefly to rebuild the charge lost in the bootstrap capacitor before the next on-cycle of the high-side switch. In skip mode, the low-side switch is turned off when the inductor current decreases to 0.2A (typ) to ensure no reverse current flowing from the output capacitor and the best conversion efficiency/minimum supply current. The high-side switch minimum on-time is controlled to guarantee that 0.58A current is reached to avoid high frequency bursts at no load conditions and that might cause a rapid increase of the supply current caused by additional switching losses. Even if skip mode is selected at the device turn-on, the monotonic startup mode is internally selected during soft-start. The transition to skip mode is automatically achieved 4096 clock cycles after the voltage at FB increases above 92.5% of VREFIN. PCB Layout Considerations and Thermal Performance 1) Connect input and output capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 2) Place capacitors on VDD, VIN, and SS as close as possible to the IC and its corresponding pin using direct traces. Keep power ground plane (connected to PGND) and signal ground plane (connected to GND) separate. 3) Keep the high-current paths as short and wide as possible. Keep the path of switching current short and minimize the loop area formed by LX, the output capacitors, and the input capacitors. 4) Connect IN, LX, and PGND separately to a large copper area to help cool the IC to further improve efficiency and long-term reliability. 5) Ensure all feedback connections are short and direct. Place the feedback resistors and compensation components as close as possible to the IC. 6) Route high-speed switching nodes, such as LX, away from sensitive analog areas (FB, COMP). ______________________________________________________________________________________ 17 MAX15038 after the voltage at FB increases above 92.5% of VREFIN. The additional delay prevents an early transition from monotonic startup to forced-PWM mode during soft-start when a prolonged time constant external REFIN voltage is applied. The maximum allowed soft-start time is 2ms when an external reference is applied at REFIN in the case of starting up into prebiased output. Pin Configuration Chip Information PGND LX LX LX BST TOP VIEW PGND PROCESS: BiCMOS 18 17 16 15 14 13 PGND 19 PGND 20 11 FREQ IN 21 For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. 10 OUT MAX15038 IN 22 IN 23 *EP 3 4 5 6 REFIN SS 2 CTL2 1 CTL1 + VDD EN 24 Package Information 12 PWRGD MODE MAX15038 4A, 2MHz Step-Down Regulator with Integrated Switches 9 FB 8 COMP 7 GND PACKAGE TYPE PACKAGE CODE DOCUMENT NO. 24 TQFN T2444-4 21-0139 THIN QFN *EP = EXPOSED PAD. Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 18 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2008 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.