19-4123; Rev 2; 10/08 15A Step-Down Regulator with Internal Switches o Fast Transient Response o Monotonic Power-Up with Precharged Output o Supports Any Output Capacitor No Compensation Required with Polymers/ Tantalum Stable with Ceramic Output Capacitors Using External Compensation o Dynamically Adjustable Output Voltage 0.5% VOUT Accuracy Over Line and Load o Adjustable Valley Current-Limit Protection Thermal Compensation with NTC Supports Foldback Current Limit o Programmable Switching Frequency o Overvoltage Protection o Undervoltage Protection o Voltage Soft-Start and Soft-Shutdown o Power-Good Window Comparator Ordering Information PART MAX15035ETL+ Quick-PWM is a trademark of Maxim Integrated Products, Inc. IN IN IN IN IN IN 20 IN EP3 IN 18 IN AGND REFIN 34 19 IN MAX15035 REF 35 17 N.C. 16 LX EP2 SKIP 36 15 PGND VCC 37 14 PGND LX PGOOD 38 13 PGND N.C. 39 12 PGND 11 PGND 4 5 6 7 8 9 10 PGND PGND 3 PGND 2 PGND 1 PGND + LX N.C. 40 VDD Storage Power Supplies EP1 FB 32 AGND Step-Down Power Supplies N.C. 30 29 28 27 26 25 24 23 22 21 Applications Point-of-Load Applications AGND TOP VIEW TON BST Pin Configuration ILIM 33 DDR Memory—VDDQ or VTT 40 TQFN-EP* *EP = Exposed pad. The MAX15035 is available in a small 40-pin, 6mm x 6mm, TQFN package. GPU Core Supplies PIN-PACKAGE -40°C to +85°C +Denotes a lead-free/RoHS-compliant package. N.C. 31 Server Computers TEMP RANGE EN The MAX15035 includes a voltage-controlled soft-start and soft-shutdown to limit the input surge current, provide a monotonic power-up into a precharged output, and provide a predictable soft-start time. The controller also includes output fault protection—undervoltage and overvoltage protection—as well as thermal-fault protection. o 4.5V to 26V Input Voltage Range N.C. The MAX15035 pulse-width modulation (PWM) controller provides high efficiency, excellent transient response, and high DC-output accuracy. Combined with the internal low on-resistance MOSFETs, the MAX15035 provides a highly efficient and compact solution for small form factor applications that need a high-power density. Maxim’s proprietary Quick-PWM™ quick-response, constant on-time PWM control scheme handles wide input/output voltage ratios (low-duty-cycle applications) with ease and provides 100ns instant-on response to load transients while maintaining a relatively constant switching frequency. The output voltage can be dynamically controlled using the dynamic REFIN, which supports input voltages between 0V to 2V. The REFIN adjustability combined with a resistive voltage-divider on the feedback input allows the MAX15035 to be configured for any output voltage between 0V to 0.9VIN. The controller senses the current across the synchronous rectifier to achieve a low-cost and highly efficient valley current-limit protection. External current-limit control is provided to allow higher current-limit settings for applications with heatsinks and air flow, or for lower current applications that need lower current-limit settings to avoid overdesigning the application circuit. The adjustable current limit provides a high degree of flexibility, allowing thermally compensated protection or foldback current-limit protection using a voltage-divider partially derived from the output. Features THIN QFN (6mm x 6mm) ________________________________________________________________ Maxim Integrated Products 1 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. MAX15035 General Description MAX15035 15A Step-Down Regulator with Internal Switches ABSOLUTE MAXIMUM RATINGS IN to PGND.............................................................-0.3V to +28V TON to GND ...........................................................-0.3V to +28V VDD to GND ..............................................................-0.3V to +6V VCC to GND ................................................-0.3V to (VDD + 0.3V) EN, SKIP, PGOOD to GND.......................................-0.3V to +6V REF, REFIN to GND....................................-0.3V to (VCC + 0.3V) ILIM, FB to GND .........................................-0.3V to (VCC + 0.3V) GND to PGND .......................................................-0.3V to +0.3V LX to PGND ...............................................................-1V to +28V BST to PGND...............................................(VDD - 0.3V) to +34V BST to LX..................................................................-0.3V to +6V BST to VDD .............................................................-0.3V to +28V REF Short Circuit to GND ...........................................Continuous IN RMS Current Rating (continuous)......................................15A PGND RMS Current Rating (continuous) ...............................20A Continuous Power Dissipation (TA = +70°C) 40-Pin, 6mm x 6mm Thin QFN (T4066-MCM) (derate 27mW/°C above +70°C) ................................2162mW Operating Temperature Range ...........................-40°C to +85°C Junction Temperature Range ..........................................+150°C Storage Temperature Range .............................-65°C to +150°C Lead Temperature (soldering, 10s) .................................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise specified. Typical values are at TA = +25°C.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS PWM CONTROLLER Input Voltage Range 26.0 V Quiescent Supply Current (VDD) IDD + ICC FB forced above REFIN 0.7 1.2 mA Shutdown Supply Current (VDD) ISHDN EN = GND, TA = +25°C 0.1 2 µA 3.95 4.2 4.45 V RTON = 97.5kΩ (600kHz) 123 164 205 RTON = 200kΩ (300kHz) 275 303 331 RTON = 302.5kΩ (200kHz) 379 VCC Undervoltage Lockout Threshold VDD-to-VCC Resistance On-Time Minimum Off-Time VIN 4.5 Rising edge, PWM disabled below this VUVLO(VCC) level; hysteresis = 100mV RCC tON tOFF(MIN) TON Shutdown Supply Current VIN = 12V, VFB = 1.0V (Note 3) 442 505 225 350 ns EN = GND, VTON = 26V, VCC = 0V or 5V, TA = +25°C 0.01 1 µA VREF V -50 +50 nA 0 VREF V VREFIN (Note 2) REFIN Input Current IREFIN TA = +25°C, REFIN = 0.5V to 2V FB Voltage Accuracy VFB (Note 2) VFB VREFIN = 0.5V, measured at FB, VIN = 4.5V to 26V, SKIP = VDD VREFIN = 1.0V VREFIN = 2.0V FB Input Bias Current FB Output Low Voltage IFB ns (Note 3) REFIN Voltage Range FB Voltage Range Ω 20 0 TA = +25°C 0.495 TA = 0°C to +85°C 0.493 TA = +25°C 0.995 TA = 0°C to +85°C 0.993 TA = 0°C to +85°C 1.990 0.5 0.505 0.507 V VFB = 0.5V to 2.0V, TA = +25°C 1.0 1.005 1.007 2.0 -0.1 ISINK = 3mA 2.010 +0.1 µA 0.4 V Load-Regulation Error SKIP = VDD 0.1 % Line-Regulation Error VCC = 4.5V to 5.5V, VIN = 4.5V to 26V 0.2 % 2 _______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches (Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise specified. Typical values are at TA = +25°C.) (Note 1) PARAMETER Soft-Start/Soft-Stop Slew Rate Dynamic REFIN Slew Rate SYMBOL SSSR DYNSR CONDITIONS MIN TYP MAX UNITS 0.4 1.2 2.2 mV/µs 3 9.45 18 mV/µs No load 1.990 2.00 2.010 IREF = -10µA to +50µA 1.98 Rising/falling edge on EN Rising edge on REFIN REFERENCE Reference Voltage VREF VCC = 4.5V to 5.5V 2.02 V FAULT DETECTION Output Overvoltage-Protection Trip Threshold With respect to the internal target voltage (error comparator threshold); rising edge; hysteresis = 50mV 250 OVP 300 350 VREF + 0.30 Dynamic transition Minimum OVP threshold mV V 0.7 Output Overvoltage Fault-Propagation Delay tOVP FB forced 25mV above trip threshold Output Undervoltage-Protection Trip Threshold UVP With respect to the internal target voltage (error comparator threshold) falling edge; hysteresis = 50mV -240 -200 -160 mV Output Undervoltage Fault-Propagation Delay tUVP FB forced 25mV below trip threshold 100 200 350 µs 5 UVP falling edge, 25mV overdrive PGOOD Propagation Delay tPGOOD PGOOD Output-Low Voltage ISINK = 3mA PGOOD Leakage Current FB = REFIN (PGOOD high impedance), PGOOD forced to 5V, TA = +25°C IPGOOD Dynamic REFIN Transition Fault Blanking Threshold Thermal-Shutdown Threshold TSHDN 5 OVP rising edge, 25mV overdrive Startup delay µs µs 5 100 200 350 0.4 V 1 µA Fault blanking initiated; REFIN deviation from the internal target voltage (error comparator threshold); hysteresis = 10mV ±50 mV Temperature rising, hysteresis = 15°C 160 °C CURRENT LIMIT ILIM Input Range 0.4 ILIM Input Bias Current TA = +25°C, ILIM = 0.4V to 2V Current-Limit Threshold VILIMIT Current-Limit Threshold (Negative) VINEG Current-Limit Threshold (Zero Crossing) VZX Ultrasonic Frequency -0.1 VREF V +0.1 µA VILIM = 0.4V, VGND - VLX 18 20 22 ILIM = REF (2.0V), VGND - VLX 92 100 108 VILIM = 0.4V, VGND - VLX VILIM = 0.4V, VGND - VLX, SKIP = GND or open SKIP = open (3.3V); VFB = VREFIN + 50mV 18 mV -24 mV 1 mV 30 kHz _______________________________________________________________________________________ 3 MAX15035 ELECTRICAL CHARACTERISTICS (continued) MAX15035 15A Step-Down Regulator with Internal Switches ELECTRICAL CHARACTERISTICS (continued) (Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise specified. Typical values are at TA = +25°C.) (Note 1) PARAMETER SYMBOL MIN SKIP = open (3.3V); VFB = VREFIN + 50mV, VGND - VLX Ultrasonic Current-Limit Threshold Internal BST Switch On-Resistance CONDITIONS TYP MAX -35 RBST IBST = 10mA, VDD = 5V EN Logic-Input Threshold VEN EN rising edge, hysteresis = 450mV (typ) 1.20 EN Logic-Input Current IEN EN forced to GND or VDD, TA = +25°C -0.5 4 UNITS mV 7 Ω INPUTS AND OUTPUTS VSKIP ISKIP V µA V Open (3.3V) 3.0 3.6 Ref (2.0V) 1.7 2.3 Low (GND) SKIP Logic-Input Current 2.20 +0.5 VCC 0.4 High (5V VDD) SKIP Quad-Level Input Logic Levels 1.7 0.4 SKIP forced to GND or VDD, TA = +25°C -2 +2 µA ELECTRICAL CHARACTERISTICS (Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = -40°C to +85°C, unless otherwise specified.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN MAX UNITS PWM CONTROLLER Input Voltage Range Quiescent Supply Current (VDD) On-Time Minimum Off-Time REFIN Voltage Range FB Voltage Range FB Voltage Accuracy 4 VIN IDD + ICC tON 4.5 FB forced above REFIN VIN = 12V, VFB = 1.0V (Note 3) tOFF(MIN) (Note 3) VREFIN (Note 2) VFB (Note 2) VFB Measured at FB, VIN = 4.5V to 26V, SKIP = VDD 26 V 1.2 mA RTON = 97.5kΩ (600kHz) 115 213 RTON = 200kΩ (300kHz) 270 336 RTON = 302.5kΩ (200kHz) 368 516 VREFIN = 0.5V ns 400 ns 0 VREF V 0 VREF V 0.49 0.51 VREFIN = 1.0V 0.99 1.01 VREFIN = 2.0V 1.985 2.015 _______________________________________________________________________________________ V 15A Step-Down Regulator with Internal Switches (Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = -40°C to +85°C, unless otherwise specified.) (Note 1) PARAMETER SYMBOL CONDITIONS MIN MAX UNITS 1.985 2.015 V REFERENCE Reference Voltage VREF VDD = 4.5V to 5.5V Output Overvoltage-Protection Trip Threshold OVP With respect to the internal target voltage (error comparator threshold) rising edge; hysteresis = 50mV 250 350 mV Output Undervoltage-Protection Trip Threshold UVP With respect to the internal target voltage (error comparator threshold); falling edge; hysteresis = 50mV -240 -160 mV Output Undervoltage Fault-Propagation Delay tUVP FB forced 25mV below trip threshold 80 400 µs 0.4 V 3.95 4.45 V V FAULT DETECTION PGOOD Output-Low Voltage VCC Undervoltage Lockout Threshold ISINK = 3mA Rising edge, PWM disabled below this level, VUVLO(VCC) hysteresis = 100mV CURRENT LIMIT ILIM Input Range Current-Limit Threshold VILIMIT Ultrasonic Frequency 0.4 VREF VILIM = 0.4V, VGND = VLX 17 23 ILIM = REF (2.0V), VGND - VLX 90 110 SKIP = open (3.3V), VFB = VREFIN + 50mV 17 mV kHz INPUTS AND OUTPUTS EN Logic-Input Threshold SKIP Quad-Level Input Logic Levels VEN V SKIP EN rising edge hysteresis = 450mV (typ) 1.20 High (5V VDD) VCC 0.4 2.20 V V Mid (3.3V) 3.0 3.6 Ref (2.0V) 1.7 2.3 Low (GND) 0.4 Note 1: Limits are 100% production tested at TA = +25°C. Maximum and minimum limits over temperature are guaranteed by design and characterization. Note 2: The 0 to 0.5V range is guaranteed by design, not production tested. Note 3: On-time and off-time specifications are measured from 50% point to 50% point at the unloaded LX node. The typical 25ns dead time that occurs between the high-side driver falling edge (high-side MOSFET turn-off) and the low-side MOSFET turnon) is included in the on-time measurement. Similarly, the typical 25ns dead time that occurs between the low-side driver falling edge (low-side MOSFET turn-off) and the high-side driver rising edge (high-side MOSFET turn-on) is included in the off-time measurement. _______________________________________________________________________________________ 5 MAX15035 ELECTRICAL CHARACTERISTICS (continued) Typical Operating Characteristics (MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.) 20V 60 50 40 60 50 ULTRASONIC MODE 40 SKIP MODE PWM MODE 30 SKIP MODE PWM MODE 1.495 1.485 20 0.01 0.1 1 100 10 LOAD CURRENT (A) LOAD CURRENT (A) 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 LOAD CURRENT (A) 1.05V OUTPUT EFFICIENCY vs. LOAD CURRENT 1.05V OUTPUT EFFICIENCY vs. LOAD CURRENT 1.05V OUTPUT VOLTAGE vs. LOAD CURRENT 90 100 SKIP MODE 90 EFFICIENCY (%) 12V 60 20V 20V 100 80 12V 50 40 70 PWM MODE 60 50 SKIP MODE PWM MODE PWM MODE 1.04 20 LOAD CURRENT (A) LOAD CURRENT (A) 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 LOAD CURRENT (A) 3.3V OUTPUT EFFICIENCY vs. LOAD CURRENT 3.3V OUTPUT EFFICIENCY vs. LOAD CURRENT 3.3V OUTPUT VOLTAGE vs. LOAD CURRENT 1 100 10 0.01 1 10 100 MAX15035 toc07 90 0.1 SKIP MODE 90 70 20V 60 12V 50 PWM MODE 70 60 50 7V 40 ULTRASONIC MODE 40 30 SKIP MODE PWM MODE 20 1 LOAD CURRENT (A) 10 100 3.365 3.350 3.335 ULTRASONIC MODE 3.320 3.305 3.290 3.275 3.260 30 PWM MODE SKIP MODE 3.245 3.230 20 0.1 3.380 OUTPUT VOLTAGE (V) 80 EFFICIENCY (%) 80 100 MAX15035 toc08 0.1 100 0.01 ULTRASONIC MODE SKIP MODE 30 20 0.01 1.05 ULTRASONIC MODE 40 30 1.06 MAX15035 toc09 70 10 OUTPUT VOLTAGE (V) 80 1 MAX15035 toc05 7V 0.1 MAX15035 toc06 0.01 MAX15035 toc04 100 6 1.505 30 20 EFFICIENCY (%) PWM MODE 70 ULTRASONIC MODE OUTPUT VOLTAGE (V) 80 EFFICIENCY (%) 70 SKIP MODE 90 12V 7V 1.515 MAX15035 toc02 90 EFFICIENCY (%) 100 MAX15035 toc01 100 80 1.5V OUTPUT VOLTAGE vs. LOAD CURRENT 1.5V OUTPUT EFFICIENCY vs. LOAD CURRENT MAX15035 toc03 1.5V OUTPUT EFFICIENCY vs. LOAD CURRENT EFFICIENCY (%) MAX15035 15A Step-Down Regulator with Internal Switches 0.01 0.1 1 LOAD CURRENT (A) 10 100 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 LOAD CURRENT (A) _______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches PWM MODE SWITCHING FREQUENCY vs. INPUT VOLTAGE 250 200 150 100 ULTRASONIC MODE SKIP MODE 0 0.1 0.01 1 NO LOAD MAX15035 toc12 370 360 ILOAD = 5A 6 8 12 10 14 16 18 20 22 -40 24 -20 0 20 40 60 80 LOAD CURRENT (A) INPUT VOLTAGE (V) TEMPERATURE (°C) MAXIMUM OUTPUT CURRENT vs. INPUT VOLTAGE MAXIMUM OUTPUT CURRENT vs. AMBIENT TEMPERATURE NO-LOAD SUPPLY CURRENT (IBIAS) vs. INPUT VOLTAGE MAXIMUM OUTPUT CURRENT (A) 15.60 15 15.40 15.20 15.00 14.80 14.60 14.40 12 0 LFM 100 LFM 13 8 300 LFM 11 6 9 4 ULTRASONIC MODE 7 2 FOUR-LAYER PCB WITH 2oz COPPER USED 14.20 SKIP MODE 5 14.00 9 12 15 18 0 -40 24 21 -20 0 20 40 60 80 100 6 IIN (mA) 14 16 18 20 22 24 ULTRASONIC MODE SKIP MODE MAX15035 toc17 2.005 REF OUTPUT VOLTAGE (V) PWM MODE 12 REF OUTPUT VOLTAGE vs. LOAD CURRENT MAX15035 toc16 100 0.1 10 INPUT VOLTAGE (V) NO-LOAD SUPPLY CURRENT (IIN) vs. INPUT VOLTAGE 1 8 TEMPERATURE (°C) INPUT VOLTAGE (V) 10 PWM MODE 10 100 MAX15035 toc15 MAX15035 toc13 15.80 6 ILOAD = 10A 380 350 10 16.00 MAXIMUM OUTPUT CURRENT (A) ILOAD = 5A IBIAS (mA) 50 390 SWITCHING FREQUENCY (kHz) PWM MODE 300 380 370 360 350 340 330 320 310 300 290 280 270 260 250 240 MAX15035 toc14 SWITCHING FREQUENCY (kHz) 350 SWITCHING FREQUENCY (kHz) MAX15035 toc10 400 SWITCHING FREQUENCY vs. TEMPERATURE MAX15035 toc11 SWITCHING FREQUENCY vs. LOAD CURRENT 2.004 2.003 2.002 2.001 0.01 2.000 6 8 10 12 14 16 18 INPUT VOLTAGE (V) 20 22 24 -10 0 10 20 30 40 50 LOAD CURRENT (µA) _______________________________________________________________________________________ 7 MAX15035 Typical Operating Characteristics (continued) (MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.) MAX15035 15A Step-Down Regulator with Internal Switches Typical Operating Characteristics (continued) (MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.) SOFT-START WAVEFORM (LIGHT LOAD) SOFT-START WAVEFORM (HEAVY LOAD) MAX15035 toc19 MAX15035 toc18 5V A 5V A 0 5V B 0 5V 0 B 1.5V C 0 1.5V C 0 0 D 8A D 1A 0 0 A. EN, 5V/div B. PGOOD, 5V/div IOUT = 8A 200µs/div C. VOUT, 1V/div B. INDUCTOR CURRENT, 10A/div A. EN, 5V/div B. PGOOD, 5V/div IOUT = 1A 200µs/div C. VOUT, 1V/div B. INDUCTOR CURRENT, 10A/div LOAD-TRANSIENT RESPONSE (PWM MODE) SHUTDOWN WAVEFORM MAX15035 toc21 MAX15035 toc20 5V 0 A 5V B 8A A 1A 0 1.5V C 1.5V D 8A B 0 8A C 0 0A A. EN, 5V/div B. PGOOD, 5V/div IOUT = 6A 8 200µs/div C. VOUT, 1V/div B. INDUCTOR CURRENT, 5A/div 20µs/div B. VOUT, 20mV/div C. INDUCTOR CURRENT, IOUT = 1A TO 8A TO 1A 5A/div A. IOUT, 10A/div _______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches LOAD-TRANSIENT RESPONSE (SKIP MODE) OUTPUT OVERVOLTAGE WAVEFORM OUTPUT OVERCURRENT WAVEFORM MAX15035 toc22 MAX15035 toc24 MAX15035 toc23 20A 8A 1A A A 1.5V A 0 B 1.5V MAX15035 Typical Operating Characteristics (continued) (MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.) 0 1.5V B 0 5V 8A C 0A 5V C 0 20µs/div B. VOUT, 20mV/div C. INDUCTOR CURRENT, IOUT = 1A TO 8A TO 1A 5A/div 200µs/div A. INDUCTOR CURRENT, B. VOUT, 1V/div 10A/div C. PGOOD, 5V/div NO-LOAD BIAS CURRENT vs. FREQUENCY OUTPUT CURRENT LIMIT vs. ILIMIT VOLTAGE A. IOUT, 10A/div 26 22 20 18 16 B. PGOOD, 5V/div IOUT = 2A TO 20A 20 18 CURRENT LIMIT (A) IBIAS (mA) 24 200µs/div PREBIAS STARTUP-OUTPUT VOLTAGE MAX15035 toc27 MAX15035 toc26 PWM MODE 28 0 A. VOUT, 1V/div IOUT = 2A TO 20A MAX15035 toc25 30 B 1.5V 1.2V 16 500mV/div 14 12 14 12 10 10 8 8 200 250 300 350 400 450 500 550 600 FREQUENCY (kHz) 500 600 700 800 900 1000 200µs/div ILIMIT VOLTAGE (mV) _______________________________________________________________________________________ 9 MAX15035 15A Step-Down Regulator with Internal Switches Typical Operating Characteristics (continued) (MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.) DYNAMIC OUTPUT-VOLTAGE TRANSITION (PWM MODE) DYNAMIC OUTPUT-VOLTAGE TRANSITION (SKIP MODE) MAX15035 toc28 1.5V MAX15035 toc29 1.5V A A 1.05V 1.05V 1.5V 1.5V B B 1.05V 1.05V 0 -6A 10A C C 0 12V 12V D 0 D 0 A. REFIN, 500mV/div B. VOUT, 200mV/div IOUT = 2A 40µs/div C. INDUCTOR CURRENT, 10A/div D. LX, 10V/div A. REFIN, 500mV/div B. VOUT, 200mV/div IOUT = 2A 40µs/div C. INDUCTOR CURRENT, 10A/div D. LX, 10V/div Pin Description PIN NAME 1, 17, 27, 31, 39, 40 N.C. FUNCTION No Connection. Not internally connected. Shutdown Control Input. Connect to VDD for normal operation. Pull EN low to put the controller into its 2µA (max) shutdown state. The MAX15035 slowly ramps down the target/output voltage to ground and after the target voltage reaches 0.1V, the controller forces LX into a high-impedance state and enters the low-power shutdown state. Toggle EN to clear the fault-protection latch. 2 EN 3, 28 AGND 4 VDD Supply Voltage Input for the DL Gate Driver. Connect to the system supply voltage (+4.5V to +5.5V). Bypass VDD to power ground with a 1µF or greater ceramic capacitor. 5, 16 LX Inductor Connection. Internally connected to EP2. Connect LX to the switched side of the inductor as shown in Figure 1. 6–15 PGND 18–26 IN 29 TON Analog Ground. Internally connected to EP1. Power Ground Power MOSFET Input Power Source. Internally connected to EP3. Switching Frequency-Setting Input. An external resistor between the input power source and TON sets the switching period (tSW = 1/fSW) according to the following equation: ⎛ V ⎞ tSW = CTON (RTON + 6.5kΩ ) ⎜ FB ⎟ V ⎝ OUT ⎠ where CTON = 16.26pF and VFB = VREFIN under normal operating conditions. If the TON current drops below 10µA, the MAX15035 shuts down and enters a high-impedance state. TON is high impedance in shutdown. 30 10 BST Boost Flying Capacitor Connection. Connect to an external 0.1µF capacitor as shown in Figure 1. The MAX15035 contains an internal boost switch/diode (Figure 2). ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches PIN NAME FUNCTION 32 FB Feedback Voltage Sense Connection. Connect directly to the positive terminal of the output capacitors for output voltages less than 2V as shown in Figure 1. For fixed-output voltages greater than 2V, connect REFIN to REF and use a resistive divider to set the output voltage (Figure 6). FB senses the output voltage to determine the on-time for the high-side switching MOSFET. 33 ILIM Current-Limit Threshold Adjustment. The current-limit threshold is 0.05 times (1/20) the voltage at ILIM. Connect ILIM to a resistive divider (from REF) to set the current-limit threshold between 20mV and 100mV (with 0.4V to 2V at ILIM). 34 REFIN External Reference Input. REFIN sets the feedback regulation voltage (VFB = VREFIN) of the MAX15035 using a resistor-divider connected between REF and AGND. The MAX15035 includes an internal window comparator to detect REFIN voltage transitions, allowing the controller to blank PGOOD and the fault protection. 35 REF 2V Reference Voltage. Bypass to analog ground using a 1nF ceramic capacitor. The reference can source up to 50µA for external loads. 36 SKIP Pulse-Skipping Control Input. This four-level input determines the mode of operation under normal steady-state conditions and dynamic output-voltage transitions: VDD (5V) = Forced-PWM operation REF (2V) = Pulse-skipping mode (with forced-PWM during transitions) Open (3.3V) = Ultrasonic mode (without forced-PWM during transitions) GND = Pulse-skipping mode (without forced-PWM during transitions) 37 VCC 5V Analog Supply Voltage. Internally connected to VDD through an internal 20Ω resistor. Bypass VCC to analog ground using a 1µF ceramic capacitor. 38 PGOOD Open-Drain Power-Good Output. PGOOD is low when the output voltage is more than 200mV (typ) below or 300mV (typ) above the target voltage (VREFIN), during soft-start, and soft-shutdown. After the soft-start circuit has terminated, PGOOD becomes high impedance if the output is in regulation. PGOOD is blanked—forced high-impedance state—when a dynamic REFIN transition is detected. EP1 (41) AGND Exposed Pad 1/Analog Ground. Internally connected to the controller’s ground plane and substrate. Connect directly to ground. EP2 (42) LX Exposed Pad 2/Inductor Connection. Internally connected to drain of the low-side MOSFET and source of the high-side MOSFET (Figure 2). Connect LX to the switched side of the inductor as shown in Figure 1. EP3 (43) IN Exposed Pad 3/Power MOSFET Input Power Source. Internally connected to drain of the high-side MOSFET (Figure 2). ______________________________________________________________________________________ 11 MAX15035 Pin Description (continued) MAX15035 15A Step-Down Regulator with Internal Switches 4 5V BIAS SUPPLY C1 1µF C2 1µF BST 36 GND/OPEN/REF/VDD C3 1000pF 35 AGND LO PGND PGOOD EN MAX15035 FB SKIP COUT 6–15 PWR 32 RT 60.4kΩ REF REFIN ILIM AGND 33 R4 40.2kΩ NTC 10kΩ B = 3435 R5 49.4kΩ 3, 28, EP1 PWR AGND OUTPUT 1.05V/1.50V 15A (MAX) PWR R2 54.9kΩ HI PWR L1 5, 16, EP2 R1 49.9kΩ 34 R3 97.6kΩ CIN CBST 0.1µF R10 100kΩ 2 INPUT 7V TO 24V 30 VCC LX OFF AGND 18–26, EP3 PWR 38 ON TON IN 37 AGND VDD RTON 200kΩ 29 SEE TABLE 1 FOR COMPONENT SELECTION. AGND Figure 1. MAX15035 Standard Application Circuit Table 1. Component Selection for Standard Applications VOUT = 1.5V/1.05V AT 15A (FIGURE 1) VOUT = 3.3V AT 6A (FIGURE 6) VOUT = 1.5V/1.05V AT 10A (FIGURE 1) VIN = 7V to 20V Ω (300kHz) TON = 200kΩ VIN = 7V to 20V Ω (300kHz) TON = 332kΩ VIN = 5V to 12V Ω (600kHz) TON = 100kΩ Input Capacitor (3x) 10µF, 25V Taiyo Yuden TMK432BJ106KM (2x) 10µF, 25V Taiyo Yuden TMK432BJ106KM (3x) 10µF, 25V Taiyo Yuden TMK432BJ106KM Output Capacitor (2x) 330µF, 6mΩ, 2V Panasonic EEFSX0D331XR (1x) 330µF, 18mΩ, 4V SANYO 4TPE330MI (1x) 470µF, 7mΩ, 2.5V SANYO 2R5TPLF470M7 Inductor 1.0µH, 5.3mΩ, 27.5A Vishay IHLP4040DZER1R0 1.5µH, 14mΩ, 9A NEC TOKIN MPLC1040L3R3 0.47µH, 3.7mΩ, 15A Cooper FP3-R47-R COMPONENT Table 2. Component Suppliers SUPPLIER SUPPLIER WEBSITE WEBSITE AVX Corp. www.avxcorp.com Pulse Engineering BI Technologies www.bitechnologies.com SANYO NA Corp. www.sanyo.com Cooper Bussmann www.cooperet.com Sumida Corp. www.sumida.com KEMET Corp. www.kemet.com Taiyo Yuden www.t-yuden.com Murata Mfg. Co., Ltd. www.murata.com TDK Corp. www.component.tdk.com NEC TOKIN Corp. www.nec-tokin.com TOKO America, Inc. www.tokoam.com Panasonic Corp. www.panasonic.com Vishay www.vishay.com Würth Electronik GmbH & Co. KG www.we-online.com 12 www.pulseeng.com ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches The MAX15035 standard application circuit (Figure 1) generates a 1.5V or 1.05V output rail for general-purpose use. See Table 1 for component selections. Table 2 lists the component suppliers. Detailed Description The MAX15035 step-down controller is ideal for lowduty-cycle (high-input voltage to low-output voltage) applications. Maxim’s proprietary Quick-PWM pulsewidth modulator in the MAX15035 is specifically designed for handling fast-load steps while maintaining a relatively constant operating frequency and inductor operating point over a wide range of input voltages. The Quick-PWM architecture circumvents the poor load-transient timing problems of fixed-frequency, current-mode PWMs while also avoiding the problems caused by widely varying switching frequencies in conventional constant-on-time (regardless of input voltage) pulse-frequency modulation (PFM) control schemes. +5V Bias Supply (VCC/VDD) The MAX15035 requires an external 5V bias supply in addition to the input. See Figure 6 for an optional 5V bias generation circuit. The 5V bias supply powers both the PWM controller and internal gate-drive power, so the maximum current drawn is determined by: IBIAS = IQ + fSWQG = 2mA to 20mA (typ) The MAX15035 includes a 20Ω resistor between VDD and VCC, simplifying the PCB layout requirement. Free-Running Constant-On-Time PWM Controller with Input Feed-Forward The Quick-PWM control architecture is a pseudo-fixedfrequency, constant on-time, current-mode regulator with voltage feed-forward (Figure 2). This architecture relies on the output filter capacitor’s ESR to act as a current-sense resistor, so the output ripple voltage provides the PWM ramp signal. The control algorithm is simple: the high-side switch on-time is determined solely by a one-shot whose pulse width is inversely proportional to input voltage and directly proportional to output voltage. Another one-shot sets a minimum offtime (200ns typ). The on-time one-shot is triggered if the error comparator is low, the low-side switch current is below the valley current-limit threshold, and the minimum off-time one-shot has timed out. On-Time One-Shot The heart of the PWM core is the one-shot that sets the high-side switch on-time. This fast, low-jitter, adjustable one-shot includes circuitry that varies the on-time in response to input and output voltage. The high-side switch on-time is inversely proportional to the input voltage as sensed by the TON input, and proportional to the feedback voltage as sensed by the FB input: On-Time (tON) = tSW (VFB/VIN) where tSW (switching period) is set by the resistance (RTON) between TON and IN. This algorithm results in a nearly constant switching frequency despite the lack of a fixed-frequency clock generator. Connect a resistor (RTON) between TON and IN to set the switching period tSW = 1/fSW: ⎛ V ⎞ tSW = CTON (RTON + 6.5kΩ ) ⎜ FB ⎟ V ⎝ OUT ⎠ where CTON = 16.26pF. When used with unity-gain feedback (VOUT = VFB), a 96.75kΩ to 303.25kΩ corresponds to switching periods of 167ns (600kHz) to 500ns (200kHz), respectively. High-frequency (600kHz) operation optimizes the application for the smallest component size, trading off efficiency due to higher switching losses. This may be acceptable in ultra-portable devices where the load currents are lower and the controller is powered from a lower voltage supply. Low-frequency (200kHz) operation offers the best overall efficiency at the expense of component size and board space. For continuous conduction operation, the actual switching frequency can be estimated by: VFB + VDIS fSW = tON (VIN − VCHG ) where VDIS is the sum of the parasitic voltage drops in the inductor discharge path, including synchronous rectifier, inductor, and PCB resistances; VCHG is the sum of the resistances in the charging path, including the highside switch, inductor, and PCB resistances; and tON is the on-time calculated by the MAX15035. Power-Up Sequence (POR, UVLO) The MAX15035 is enabled when EN is driven high and the 5V bias supply (V DD) is present. The reference powers up first. Once the reference exceeds its UVLO threshold, the internal analog blocks are turned on and masked by a 50µs one-shot delay in order to allow the bias circuitry and analog blocks enough time to settle to their proper states. With the control circuitry reliably powered up, the PWM controller may begin switching. ______________________________________________________________________________________ 13 MAX15035 Standard Application Circuit MAX15035 15A Step-Down Regulator with Internal Switches TON ON-TIME COMPUTE IN tOFF(MIN) FB ONE-SHOT S tON TRIG BST TRIG Q Q IN Q R LX ONE-SHOT INTEGRATOR (CCV) ERROR AMPLIFIER VDD S Q R PGND FB QUADLEVEL DECODE SKIP FAULT BLANK EA + 0.3V ZERO CROSSING PGOOD AND FAULT PROTECTION VALLEY CURRENT LIMIT ILIM REF EA - 0.2V EN SOFT-START/ SOFT-STOP PGOOD 2V REF REFIN EA BLANK MAX15035 DYNAMIC OUTPUT TRANSITION DETECTION Figure 2. MAX15035 Block Diagram 14 VCC ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches The soft-start circuitry does not use a variable current limit, so full output current is available immediately. PGOOD becomes high impedance approximately 200µs after the target REFIN voltage has been reached. The MAX15035 automatically uses pulse-skipping mode during soft-start and uses forced-PWM mode during soft-shutdown, regardless of the SKIP configuration. For automatic startup, the input voltage should be present before VCC. If the controller attempts to bring the output into regulation without the input voltage present, the fault latch trips. The controller remains shut down until the fault latch is cleared by toggling EN or cycling the VCC power supply below 0.5V. If the VCC voltage drops below 4.25V, the controller assumes that there is not enough supply voltage to make valid decisions. To protect the output from overvoltage faults, the controller shuts down immediately and forces a high impedance on LX. When a fault condition—output UVP or thermal shutdown—activates the shutdown sequence, the protection circuitry sets the fault latch to prevent the controller from restarting. To clear the fault latch and reactivate the controller, toggle EN or cycle VCC power below 0.5V. The MAX15035 automatically uses pulse-skipping mode during soft-start and uses forced-PWM mode during soft-shutdown, regardless of the SKIP configuration. Modes of Operation Ultrasonic Mode (SKIP = Open = 3.3V) Leaving SKIP unconnected activates a unique pulseskipping mode with a minimum switching frequency of 18kHz. This ultrasonic pulse-skipping mode eliminates audio-frequency modulation that would otherwise be present when a lightly loaded controller automatically skips pulses. In ultrasonic mode, the controller automatically transitions to fixed-frequency PWM operation when the load reaches the same critical conduction point (ILOAD(SKIP)) that occurs when normally pulse skipping. An ultrasonic pulse occurs when the controller detects that no switching has occurred within the last 33µs. Once triggered, the ultrasonic controller turns on the low-side MOSFET to induce a negative inductor current (Figure 3). After the inductor current reaches the negative ultrasonic current threshold, the controller turns off the low-side MOSFET and triggers a constant on-time. Shutdown 33µs (typ) When the system pulls EN low, the MAX15035 enters low-power shutdown mode. PGOOD is pulled low immediately, and the output voltage ramps down with a 1.2mV/µs slew rate: VFB VFB t SHDN = = 1.2mV µs 1.2V ms Slowly discharging the output capacitors by slewing the output over a long period of time (typically 0.5ms to 2ms) keeps the average negative inductor current low (damped response), thereby preventing the negative output-voltage excursion that occurs when the controller discharges the output quickly by permanently turning on the low-side MOSFET (underdamped response). This eliminates the need for the Schottky diode normally connected between the output and ground to clamp the negative output-voltage excursion. After the controller reaches the zero target, the MAX15035 shuts down completely—the drivers are disabled (high impedance on LX)—the reference turns off, and the supply currents drop to about 0.1µA (typ). INDUCTOR CURRENT ZERO-CROSSING DETECTION 0 ISONIC ON-TIME (tON) Figure 3. Ultrasonic Waveform ______________________________________________________________________________________ 15 MAX15035 Power-on reset (POR) occurs when VCC rises above approximately 3V, resetting the fault latch and preparing the controller for operation. The VCC UVLO circuitry inhibits switching until VCC rises above 4.25V. The controller powers up the reference once the system enables the controller, VCC exceeds 4.25V, and EN is driven high. With the reference in regulation, the controller ramps the output voltage to the target REFIN voltage with a 1.2mV/µs slew rate: VFB VFB t START = = 1.2mV µs 1.2V ms MAX15035 15A Step-Down Regulator with Internal Switches When the on-time expires, the controller re-enables the low-side MOSFET until the controller detects that the inductor current drops below the zero-crossing threshold. Starting with a negative inductor current pulse greatly reduces the peak output voltage when compared to starting with a positive inductor current pulse. The output voltage at the beginning of the ultrasonic pulse determines the negative ultrasonic current threshold, resulting in the following equation: VISONIC = IL × 0.006 = ( VREFIN − VFB ) × 0.7 where VFB > VREFIN. Forced-PWM Mode (SKIP = VDD) The low-noise, forced-PWM mode (SKIP = VDD) disables the zero-crossing comparator, which controls the low-side switch on-time. This forces the low-side gatedrive waveform to constantly be the complement of the high-side gate-drive waveform, so the inductor current reverses at light loads while LX maintains a duty factor of VOUT/VIN. The benefit of forced-PWM mode is to keep the switching frequency fairly constant. However, forced-PWM operation comes at a cost: the no-load 5V bias current remains between 10mA to 50mA, depending on the switching frequency. Automatic Pulse-Skipping Mode (SKIP = GND or REF) In skip mode (SKIP = GND or 3.3V), an inherent automatic switchover to PFM takes place at light loads. This switchover is affected by a comparator that truncates the low-side switch on-time at the inductor current’s zero crossing. The zero-crossing comparator threshold is set by the differential across LX to PGND. DC output-accuracy specifications refer to the threshold of the error comparator. When the inductor is in continuous conduction, the MAX15035 regulates the valley of the output ripple, so the actual DC output voltage is higher than the trip level by 50% of the output ripple voltage. In discontinuous conduction (SKIP = GND and IOUT < ILOAD(SKIP)), the output voltage has a DC regulation level higher than the error-comparator threshold by approximately 1.5% due to slope compensation. When SKIP is pulled to GND, the MAX15035 remains in pulse-skipping mode. Since the output is not able to sink current, the timing for negative dynamic output-voltage transitions depends on the load current and output capacitance. Letting the output voltage drift down is typically recommended to reduce the potential for audible noise since this eliminates the input current surge during negative output-voltage transitions. See Figures 4 and 5. The MAX15035 automatically always uses forced-PWM operation during shutdown, regardless of the SKIP configuration. DYNAMIC REFIN WINDOW REFIN ACTUAL VOUT OUTPUT VOLTAGE INTERNAL PWM CONTROL LX PGOOD OVP INTERNAL TARGET SKIP NO PULSES: VOUT > VTARGET BLANK HIGH-Z SET TO REF + 300mV BLANK HIGH-Z EA TARGET + 300mV DYNAMIC TRANSITION WHEN SKIP# = GND Figure 4. Dynamic Transition when SKIP = GND 16 ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches MAX15035 DYNAMIC REFIN WINDOW REFIN OUTPUT VOLTAGE INTERNAL PWM CONTROL INTERNAL EA TARGET = ACTUAL VOUT PWM PWM SKIP SKIP LX PGOOD BLANK HIGH-Z OVP SET TO REF + 300mV BLANK HIGH-Z EA TARGET + 300mV EA TARGET + 300mV DYNAMIC TRANSITION WHEN SKIP = REF Figure 5. Dynamic Transition when SKIP = REF Valley Current-Limit Protection The current-limit circuit employs a unique “valley” current-sensing algorithm that senses the inductor current through the low-side MOSFET. If the current through the low-side MOSFET exceeds the valley current-limit threshold, the PWM controller is not allowed to initiate a new cycle. The actual peak current is greater than the valley current-limit threshold by an amount equal to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability are a function of the inductor value and input voltage. When combined with the undervoltage protection circuit, this current-limit method is effective in almost every circumstance. In forced-PWM mode, the MAX15035 also implements a negative current limit to prevent excessive reverse inductor currents when VOUT is sinking current. The negative current-limit threshold is set to approximately 120% of the positive current limit. Integrated Output Voltage The MAX15035 regulates the valley of the output ripple, so the actual DC output voltage is higher than the slopecompensated target by 50% of the output ripple voltage. Under steady-state conditions, the MAX15035’s internal integrator corrects for this 50% output ripple-voltage error, resulting in an output voltage that is dependent only on the offset voltage of the integrator amplifier provided in the Electrical Characteristics table. Dynamic Output Voltages The MAX15035 regulates FB to the voltage set at REFIN. By changing the voltage at REFIN (Figure 1), the MAX15035 can be used in applications that require dynamic output-voltage changes between two set points. For a step-voltage change at REFIN, the rate of change of the output voltage is limited either by the internal 9.45mV/µs slew-rate circuit or by the component selection—inductor current ramp, the total output capacitance, the current limit, and the load during the transition—whichever is slower. The total output capacitance determines how much current is needed to change the output voltage, while the inductor limits the current ramp rate. Additional load current may slow down the output voltage change during a positive REFIN voltage change, and may speed up the output voltage change during a negative REFIN voltage change. ______________________________________________________________________________________ 17 MAX15035 15A Step-Down Regulator with Internal Switches 4 5V BIAS SUPPLY C1 1µF C2 1µF PWR BST LX 2 OFF 36 GND/OPEN/REF/VDD C3 1000pF 18–26, EP3 INPUT 7V TO 24V CIN 30 CBST 0.1µF VCC R10 100kΩ 38 ON TON IN 37 AGND VDD RTON 332kΩ 29 PGND 5, 16, EP2 PWR FB SKIP L1 OUTPUT 3.3V 6–15 PGOOD EN PWR R7 20kΩ REF AGND AGND 34 PWR 32 MAX15035 35 COUT R6 13.0kΩ REFIN ILIM 33 AGND 3, 28, EP1 R4 49.9kΩ REF R5 49.4kΩ 7V TO 15V INPUT 1kΩ AGND PWR AGND 5VBIAS 5.6V OPTIONAL SEE TABLE 1 FOR COMPONENT SELECTION. Figure 6. High Output-Voltage Application Using a Feedback Divider Output Voltages Greater than 2V Although REFIN is limited to a 0 to 2V range, the output-voltage range is unlimited since the MAX15035 utilizes a high-impedance feedback input (FB). By adding a resistive voltage-divider from the output to FB to analog ground (Figure 6), the MAX15035 supports output voltages above 2V. However, the controller also uses FB to determine the on-time, so the voltage-divider influences the actual switching frequency, as detailed in the On-Time One-Shot section. Internal Integration An integrator amplifier forces the DC average of the FB voltage to equal the target voltage. This internal amplifier integrates the feedback voltage and provides a fine adjustment to the regulation voltage (Figure 2), allowing accurate DC output-voltage regulation regardless of the compensated feedback ripple voltage and internal slope-compensation variation. The integrator amplifier has the ability to shift the output voltage by ±55mV (typ). 18 The MAX15035 disables the integrator by connecting the amplifier inputs together at the beginning of all downward REFIN transitions done in pulse-skipping mode. The integrator remains disabled until 20µs after the transition is completed (the internal target settles) and the output is in regulation (edge detected on the error comparator). Power-Good Outputs (PGOOD) and Fault Protection PGOOD is the open-drain output that continuously monitors the output voltage for undervoltage and overvoltage conditions. PGOOD is actively held low in shutdown (EN = GND), and during soft-start and soft-shutdown. Approximately 200µs (typ) after the softstart terminates, PGOOD becomes high impedance as long as the feedback voltage is above the UVP threshold (REFIN - 200mV) and below the OVP threshold (REFIN + 300mV). PGOOD goes low if the feedback voltage drops 200mV below the target voltage (REFIN) or rises 300mV above the target voltage (REFIN), or the SMPS controller is shut down. For a logic-level PGOOD ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches MAX15035 TARGET + 300mV TARGET - 200mV POWER-GOOD AND FAULT PROTECTION FB EN OVP SOFT-START COMPLETE UVP OVP ENABLED ONESHOT 200µs FAULT LATCH FAULT POWER-GOOD IN OUT CLK Figure 7. Power-Good and Fault Protection output voltage, connect an external pullup resistor between PGOOD and VDD. A 100kΩ pullup resistor works well in most applications. Figure 7 shows the power-good and fault-protection circuitry. high impedance on LX. Toggle EN or cycle VCC power below VCC POR to reactivate the controller after the junction temperature cools by 15°C. Overvoltage Protection (OVP) When the internal feedback voltage rises 300mV above the target voltage and OVP is enabled, the OVP comparator immediately forces LX low, pulls PGOOD low, sets the fault latch, and disables the SMPS controller. Toggle EN or cycle VCC power below the VCC POR to clear the fault latch and restart the controller. Firmly establish the input voltage range and maximum load current before choosing a switching frequency and inductor operating point (ripple-current ratio). The primary design trade-off lies in choosing a good switching frequency and inductor operating point, and the following four factors dictate the rest of the design: • Input Voltage Range: The maximum value (V(INMAX)) must accommodate the worst-case input supply voltage. The minimum value (V(INMIN)) must account for the lowest input voltage after drops due to connectors, fuses, and battery selector switches. If there is a choice at all, lower input voltages result in better efficiency. Undervoltage Protection (UVP) When the feedback voltage drops 200mV below the target voltage (REFIN), the controller immediately pulls PGOOD low and triggers a 200µs one-shot timer. If the feedback voltage remains below the undervoltage fault threshold for the entire 200µs, the undervoltage fault latch is set and the SMPS begins the shutdown sequence. When the internal target voltage drops below 0.1V, the MAX15035 forces a high impedance on LX. Toggle EN or cycle VCC power below VCC POR to clear the fault latch and restart the controller. Thermal-Fault Protection (TSHDN) The MAX15035 features a thermal fault-protection circuit. When the junction temperature rises above +160°C, a thermal sensor activates the fault latch, pulls PGOOD low, shuts down the controller, and forces a Quick-PWM Design Procedure • Maximum load current: There are two values to consider. The peak load current (I LOAD(MAX) ) determines the instantaneous component stresses and filtering requirements, and thus drives output capacitor selection, inductor saturation rating, and the design of the current-limit circuit. The continuous load current (ILOAD) determines the thermal stresses and thus drives the selection of input capacitors, MOSFETs, and other critical heat-contributing components. ______________________________________________________________________________________ 19 MAX15035 15A Step-Down Regulator with Internal Switches • • Switching frequency: This choice determines the basic trade-off between size and efficiency. The optimal frequency is largely a function of maximum input voltage due to MOSFET switching losses that are proportional to frequency and VIN2. The optimum frequency is also a moving target, due to rapid improvements in MOSFET technology that are making higher frequencies more practical. Inductor operating point: This choice provides trade-offs between size vs. efficiency and transient response vs. output noise. Low inductor values provide better transient response and smaller physical size, but also result in lower efficiency and higher output noise due to increased ripple current. The minimum practical inductor value is one that causes the circuit to operate at the edge of critical conduction (where the inductor current just touches zero with every cycle at maximum load). Inductor values lower than this grant no further size-reduction benefit. The optimum operating point is usually found between 20% and 50% ripple current. Inductor Selection The switching frequency and operating point (% ripple current or LIR) determine the inductor value as follows: ⎛ ⎞ ⎛ VOUT ⎞ VIN − VOUT L=⎜ ⎟ ⎜⎝ V ⎟⎠ f I LIR ⎝ SW LOAD(MAX) ⎠ IN Find a low-loss inductor having the lowest possible DC resistance that fits in the allotted dimensions. Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well at 200kHz. The core must be large enough not to saturate at the peak inductor current (IPEAK): IPEAK = ILOAD(MAX) + ∆IL 2 Transient Response The inductor ripple current impacts transient-response performance, especially at low VIN - VOUT differentials. Low inductor values allow the inductor current to slew faster, replenishing charge removed from the output filter capacitors by a sudden load step. The amount of output sag is also a function of the maximum duty factor, 20 which can be calculated from the on-time and minimum off-time. The worst-case output sag voltage can be determined by: ( L ∆ILOAD(MAX) VSAG = ⎤ 2 ⎡⎛ VOUT tSW ⎞ ⎢⎜ ⎟ + tOFF(M MIN) ⎥ ⎣⎝ VIN ⎠ ⎦ ) ⎡⎛ ( V − V ⎤ )t ⎞ 2COUT VOUT ⎢⎜⎜ IN OUT SW ⎟⎟ − tOFF(MIN) ⎥ VIN ⎢⎣⎝ ⎥⎦ ⎠ where tOFF(MIN) is the minimum off-time (see the Electrical Characteristics table). The amount of overshoot due to stored inductor energy when the load is removed can be calculated as: VSOAR ≈ (∆ILOAD(MAX) )2L 2COUT VOUT Setting the Valley Current Limit The minimum current-limit threshold must be high enough to support the maximum load current when the current limit is at the minimum tolerance value. The valley of the inductor current occurs at ILOAD(MAX) minus half the inductor ripple current (∆IL); therefore: ILIMIT(LOW) > ILOAD(MAX) − ∆IL 2 where I LIMIT(LOW) equals the minimum current-limit threshold voltage divided by 0.006. The valley current-limit threshold is precisely 1/20 the voltage seen at ILIM. Connect a resistive divider from REF to ILIM to analog ground (AGND) to set a fixed valley current-limit threshold. The external 400mV to 2V adjustment range corresponds to a 20mV to 100mV valley current-limit threshold. When adjusting the currentlimit threshold, use 1% tolerance resistors and a divider current of approximately 5µA to 10µA to prevent significant inaccuracy in the valley current-limit tolerance. The MAX15035 uses the low-side MOSFET’s on-resistance as the current-sense element (R SENSE = RDS(ON)). A good general rule is to allow 0.5% additional resistance for each degree celsius of temperature rise, which must be included in the design margin unless the design includes an NTC thermistor in the ILIM resistive voltage-divider to thermally compensate the current-limit threshold. ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches C1 1µF C2 1µF BST PWR R10 100kΩ 2 OFF 36 GND/OPEN/REF/VDD C3 1000pF AGND 35 AGND PGND 18–26, EP3 R2 54.9kΩ CIN CBST 0.1µF 5, 16, EP2 PWR L1 6–15 PGOOD EN MAX15035 FB SKIP OUTPUT 1.50V 10A 1.05V 7A COUT PWR PWR 32 REF R8 100kΩ REFIN ILIM R4 49.9kΩ 33 REF AGND R5 49.4kΩ 3, 28, EP1 LO INPUT 7V TO 24V 30 R1 49.9kΩ 34 R3 97.6kΩ VCC LX 38 ON TON IN 37 AGND VDD MAX15035 4 5V BIAS SUPPLY RTON 200kΩ 29 HI AGND AGND PWR AGND SEE TABLE 1 FOR COMPONENT SELECTION. Figure 8. Standard Application with Foldback Current-Limit Protection Foldback Current Limit Including an additional resistor between ILIM and the output automatically creates a current-limit threshold that folds back as the output voltage drops (see Figure 8). The foldback current limit helps limit the inductor current under fault conditions, but must be carefully designed to provide reliable performance under normal conditions. The current-limit threshold must not be set too low, or the controller will not reliably power up. To ensure the controller powers up properly, the minimum current-limit threshold (when VOUT = 0V) must always be greater than the maximum load during startup (which at least consists of leakage currents), plus the maximum current required to charge the output capacitors: ISTART = COUT x 1mV/µs + ILOAD(START) Output Capacitor Selection The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and load-transient requirements. Additionally, the ESR impacts stability requirements. Capacitors with a high ESR value (polymers/tantalums) do not need additional external compensation components. In core and chipset converters and other applications where the output is subject to large-load transients, the output capacitor’s size typically depends on how much ESR is needed to prevent the output from dipping too low under a load transient. Ignoring the sag due to finite capacitance: (RESR + RPCB ) ≤ ∆I VSTEP LOAD(MAX) In low-power applications, the output capacitor’s size often depends on how much ESR is needed to maintain an acceptable level of output ripple voltage. The output ripple voltage of a step-down controller equals the total inductor ripple current multiplied by the output capacitor’s ESR. The maximum ESR to meet ripple requirements is: ⎡ V × f ×L ⎤ IN SW ⎥ VRIPPLE RESR ≤ ⎢ ⎢⎣ ( VIN − VOUT ) VOUT ⎥⎦ where fSW is the switching frequency. ______________________________________________________________________________________ 21 MAX15035 15A Step-Down Regulator with Internal Switches With most chemistries (polymer, tantalum, aluminum electrolytic), the actual capacitance value required relates to the physical size needed to achieve low ESR and the chemistry limits of the selected capacitor technology. Ceramic capacitors provide low ESR, but the capacitance and voltage rating (after derating) are determined by the capacity needed to prevent VSAG and VSOAR from causing problems during load transients. Generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem (see the VSAG and VSOAR equations in the Transient Response section). Thus, the output capacitor selection requires carefully balancing capacitor chemistry limitations (capacitance vs. ESR vs. voltage rating) and cost. See Figure 9. For a standard 300kHz application, the effective zero frequency must be well below 95kHz, preferably below 50kHz. With these frequency requirements, standard tantalum and polymer capacitors already commonly used have typical ESR zero frequencies below 50kHz, allowing the stability requirements to be achieved without any additional current-sense compensation. In the standard application circuit (Figure 1), the ESR needed to support a 15mV P-P ripple is 15mV/(10A x 0.3) = 5mΩ. Two 330µF, 9mΩ polymer capacitors in parallel provide 4.5mΩ (max) ESR and 1/(2π x 330µF x 9mΩ) = 53kHz ESR zero frequency. See Figure 10. IN Output Capacitor Stability Considerations For Quick-PWM controllers, stability is determined by the in-phase feedback ripple relative to the switching frequency, which is typically dominated by the output ESR. The boundary of instability is given by the following equation: fSW 1 ≥ π 2πREFFCOUT REFF = RESR + RPCB + RCOMP BST PWR L1 LX IN OUTPUT COUT PGND PWR MAX15035 where COUT is the total output capacitance, RESR is the total ESR of the output capacitors, RPCB is the parasitic board resistance between the output capacitors and feedback sense point, and RCOMP is the effective resistance of the DC- or AC-coupled current-sense compensation (see Figure 11). PWR FB AGND AGND STABILITY REQUIREMENT 1 RESRCOUT ≥ 2fSW Figure 9. Standard Application with Output Polymer or Tantalum INPUT PCB PARASITIC RESISTANCE-SENSE RESISTANCE FOR EVALUATION CIN BST INPUT CIN PWR DH L1 LX OUTPUT COUT PGND CCOMP 0.1µF PWR MAX15035 CLOAD PWR PWR RCOMP 100Ω FB OUTPUT VOLTAGE REMOTELY SENSED NEAR POINT OF LOAD GND AGND PWR STABILITY REQUIREMENT 1 1 RESRCOUT ≥ AND RCOMPCCOMP ≥ 2fSW fSW FEEDBACK RIPPLE IN PHASE WITH INDUCTOR CURRENT Figure 10. Remote-Sense Compensation for Stability and Noise Immunity 22 ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches The DC-coupling requires fewer external compensation capacitors, but this also creates an output load line that depends on the inductor’s DCR (parasitic resistance). Alternatively, the current-sense information may be ACcoupled, allowing stability to be dependent only on the inductance value and compensation components and eliminating the DC load line. OPTION A: DC-COUPLED CURRENT-SENSE COMPENSATION DC COMPENSATION IN <> FEWER COMPENSATION COMPONENTS <> CREATES OUTPUT LOAD LINE <> LESS OUTPUT CAPACITANCE REQUIRED FOR TRANSIENT RESPONSE INPUT CIN BST PWR L LX OUTPUT COUT RSENA PGND RSENB MAX15035 PWR PWR CSEN FB GND AGND STABILITY REQUIREMENT PWR ⎛ ⎞ L RSENBRDCR 1 AND LOAD LINE = ⎜ R ⎟ COUT ≥ 2f RSENA + RSENB SW ⎝ ( SENA || RSENB )CSEN ⎠ FEEDBACK RIPPLE IN PHASE WITH INDUCTOR CURRENT OPTION B: AC-COUPLED CURRENT-SENSE COMPENSATION IN AC COMPENSATION <> NOT DEPENDENT ON ACTUAL DCR VALUE <> NO OUTPUT LOAD LINE INPUT CIN BST PWR L LX OUTPUT COUT RSEN PGND CSEN MAX15035 PWR PWR CCOMP FB RCOMP GND STABILITY REQUIREMENT AGND PWR ⎛ L ⎞ 1 1 AND RCOMPCCOMP ≥ ⎜ ⎟ COUT ≥ ⎝ RSENCSEN ⎠ 2fSW fSW FEEDBACK RIPPLE IN PHASE WITH INDUCTOR CURRENT Figure 11. Feedback Compensation for Ceramic Output Capacitors ______________________________________________________________________________________ 23 MAX15035 Ceramic capacitors have a high-ESR zero frequency, but applications with sufficient current-sense compensation may still take advantage of the small size, low ESR, and high reliability of the ceramic chemistry. Using the inductor DCR, applications using ceramic output capacitors may be compensated using either a DC compensation or AC compensation method (Figure 11). MAX15035 15A Step-Down Regulator with Internal Switches When only using ceramic output capacitors, output overshoot (VSOAR) typically determines the minimum output capacitance requirement. Their relatively low capacitance value may allow significant output overshoot when stepping from full-load to no-load conditions, unless designed with a small inductance value and high switching frequency to minimize the energy transferred from the inductor to the capacitor during load-step recovery. Unstable operation manifests itself in two related but distinctly different ways: double pulsing and feedbackloop instability. Double pulsing occurs due to noise on the output or because the ESR is so low that there is not enough voltage ramp in the output voltage signal. This “fools” the error comparator into triggering a new cycle immediately after the minimum off-time period has expired. Double pulsing is more annoying than harmful, resulting in nothing worse than increased output ripple. However, it can indicate the possible presence of loop instability due to insufficient ESR. Loop instability can result in oscillations at the output after line or load steps. Such perturbations are usually damped, but can cause the output voltage to rise above or fall below the tolerance limits. The easiest method for checking stability is to apply a very fast zero-to-max load transient and carefully observe the output voltage-ripple envelope for overshoot and ringing. It can help to simultaneously monitor the inductor current with an AC current probe. Do not allow more than one cycle of ringing after the initial step-response under/overshoot. Minimum Input-Voltage Requirements and Dropout Performance The output voltage-adjustable range for continuousconduction operation is restricted by the nonadjustable minimum off-time one-shot. For best dropout performance, use the slower (200kHz) on-time settings. When working with low-input voltages, the duty-factor limit must be calculated using worst-case values for on- and off-times. Manufacturing tolerances and internal propagation delays introduce an error to the on-times. This error is greater at higher frequencies. Also, keep in mind that transient response performance of buck regulators operated too close to dropout is poor, and bulk output capacitance must often be added (see the VSAG equation in the Quick-PWM Design Procedure section). The absolute point of dropout is when the inductor current ramps down during the minimum off-time (∆IDOWN) as much as it ramps up during the on-time (∆IUP). The ratio h = ∆IUP/∆IDOWN is an indicator of the ability to slew the inductor current higher in response to increased load, and must always be greater than 1. As h approaches 1, the absolute minimum dropout point, the inductor current cannot increase as much during each switching cycle and V SAG greatly increases unless additional output capacitance is used. A reasonable minimum value for h is 1.5, but adjusting this up or down allows trade-offs between VSAG, output capacitance, and minimum operating voltage. For a given value of h, the minimum operating voltage can be calculated as: ⎛V − VDROOP + VCHG ⎞ ⎟ VIN(MIN) = ⎜ OUT ⎜⎝ 1 − h × tOFF(MIN)fSW ⎟⎠ ( Input Capacitor Selection The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents. The IRMS requirements may be determined by the following equation: ⎛I ⎞ IRMS = ⎜ LOAD ⎟ VOUT (VIN − VOUT ) ⎝ VIN ⎠ The worst-case RMS current requirement occurs when operating with VIN = 2VOUT. At this point, the above equation simplifies to IRMS = 0.5 x ILOAD. For most applications, nontantalum chemistries (ceramic, aluminum, or OS-CON) are preferred due to their resistance to inrush surge currents typical of systems with a mechanical switch or connector in series with the input. If the Quick-PWM controller is operated as the second stage of a two-stage power-conversion system, tantalum input capacitors are acceptable. In either configuration, choose an input capacitor that exhibits less than +10°C temperature rise at the RMS input current for optimal circuit longevity. 24 ) where VDROOP is the voltage-positioning droop, VCHG is the parasitic voltage drop in the charge path, and tOFF(MIN) is from the Electrical Characteristics table. The absolute minimum input voltage is calculated with h = 1. If the calculated VIN(MIN) is greater than the required minimum input voltage, reduce the operating frequency or add output capacitance to obtain an acceptable VSAG. If operation near dropout is anticipated, calculate VSAG to be sure of adequate transient response. Dropout design example: VOUT = 3.3V fSW = 300kHz tOFF(MIN) = 350ns VDROOP = 0V VCHG = 150mV (10A load) h = 1.5 ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches Calculating again with h = 1 gives the absolute limit of dropout: 3.3V − 0V + 150mV ⎡ ⎤ VIN(MIN) = ⎢ ⎥ = 3.52V ⎣1 − (1.0 × 350ns × 300kHz) ⎦ Therefore, VIN must be greater than 3.52V, even with very large output capacitance, and a practical input voltage with reasonable output capacitance would be 3.74V. Applications Information PCB Layout Guidelines Careful PCB layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention. If possible, mount all the power components on the top side of the board with their ground terminals flush against one another. Follow these guidelines for good PCB layout: 1) Keep the high-current paths short, especially at the ground terminals. This is essential for stable, jitterfree operation. 2) Connect all analog grounds to a separate solid copper plane, which connects to the AGND pin of the Quick-PWM controller. This includes the VCC bypass capacitor, REF bypass capacitors, REFIN components, and feedback compensation/dividers. 3) Keep the power traces and load connections short. This is essential for high efficiency. The use of thick copper PCB (2oz vs. 1oz) can enhance full-load efficiency by 1% or more. Correctly routing PCB traces is a difficult task that must be approached in terms of fractions of centimeters, where a single milliohms of excess trace resistance causes a measurable efficiency penalty. 4) Keep the power plane—especially LX—away from sensitive analog areas (REF, REFIN, FB, ILIM). Layout Procedure 1) Place the power components first, with ground terminals adjacent (CIN and COUT). If possible, make all these connections on the top layer with wide, copper-filled areas. 2) Make the DC-DC controller ground connections as shown in Figure 1. This diagram can be viewed as having four separate ground planes: input/output ground, where all the high-power components go; the power ground plane, where the PGND pin and VDD bypass capacitor go; the controller’s analog ground plane where sensitive analog components, the controller’s AGND pin, and VCC bypass capacitor go. The controller’s AGND plane must meet the PGND plane only at a single point directly beneath the IC. This point must also be very close to the output capacitor ground terminal. 3) Connect the output power planes (VCORE and system ground planes) directly to the output filter capacitor positive and negative terminals with multiple vias. Place the entire DC-DC converter circuit as close to the load as is practical. ______________________________________________________________________________________ 25 MAX15035 3.3V − 0V + 150mV ⎡ ⎤ VIN(MIN) = ⎢ ⎥ = 3.74V ( . ) ns kHz − × × 1 1 5 350 300 ⎣ ⎦ MAX15035 15A Step-Down Regulator with Internal Switches Package Information Chip Information TRANSISTOR COUNT: 7169 PROCESS: BiCMOS 26 For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. PACKAGE TYPE PACKAGE CODE DOCUMENT NO. 40 TQFN T4066-MCM 21-0177 ______________________________________________________________________________________ 15A Step-Down Regulator with Internal Switches REVISION NUMBER REVISION DATE DESCRIPTION PAGES CHANGED 0 5/08 Initial release — 1 7/08 Modified Figure 1, Tables 1 and 2. 12 2 10/08 Updated Pin Description, Figure 1, and Detailed Description. 11, 12, 13, 16, 18–21, 24 Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 27 © 2008 Maxim Integrated Products SPRINGER is a registered trademark of Maxim Integrated Products, Inc. MAX15035 Revision History