LTC3406-1.2 1.5MHz, 600mA Synchronous Step-Down Regulator in ThinSOT U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC ®3406-1.2 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation with only 20µA drops <1µA in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3406-1.2 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. PWM pulse skipping mode operation provides very low output ripple voltage for noise sensitive applications. High Efficiency: Up to 90% Very Low Quiescent Current: Only 20µA 600mA Output Current at VIN = 3V 2.5V to 5.5V Input Voltage Range 1.5MHz Constant Frequency Operation No Schottky Diode Required Shutdown Mode Draws < 1µA Supply Current Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Low Profile (1mm) ThinSOTTM Package Switching frequency is internally set at 1.5MHz, allowing the use of small surface mount inductors and capacitors. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. The LTC3406-1.2 is available in a low profile (1mm) ThinSOT package. U APPLICATIO S ■ ■ ■ ■ ■ Cellular Telephones Personal Information Appliances Wireless and DSL Modems Digital Still Cameras MP3 Players Portable Instruments , LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. ThinSOT is a trademark of Linear Technology Corporation. Protected by U.S. Patents including 5481178, 6580258, 6304066, 6127815, 6498466, 6611131. U ■ TYPICAL APPLICATIO Efficiency and Power Loss High Efficiency Step-Down Converter 100 90 2.2µH VIN SW COUT 10µF CER LTC3406-1.2 RUN VOUT GND 340612 TA01a 0.1 80 70 0.01 60 50 POWER LOSS 40 0.001 30 20 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 0 0.1 1 100 10 LOAD CURRENT (mA) POWER LOSS (W) CIN 4.7µF CER VOUT 1.2V 600mA EFFICIENCY (%) VIN 2.7V TO 5.5V 1 EFFICIENCY 0.0001 0.00001 1000 340612 TA01b 340612f 1 LTC3406-1.2 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) Input Supply Voltage .................................. – 0.3V to 6V RUN, VOUT Voltages................................... – 0.3V to VIN SW Voltage (DC) ......................... – 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 800mA N-Channel Switch Sink Current (DC) ................. 800mA Peak SW Sink and Source Current (VIN = 3V)........ 1.3A Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Notes 3, 5) ...................... 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C ORDER PART NUMBER TOP VIEW RUN 1 5 VOUT LTC3406ES5-1.2 GND 2 SW 3 4 VIN S5 PART MARKING S5 PACKAGE 5-LEAD PLASTIC TSOT-23 LTBMQ TJMAX = 125°C, θJA = 250°C/ W, θJC = 90°C/ W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified. SYMBOL PARAMETER CONDITIONS VOUT Regulated Output Voltage IOUT = 100mA ∆VOVL Output Overvoltage Lockout ∆VOVL = VOVL – VOUT ∆VOUT Output Voltage Line Regulation VIN = 2.5V to 5.5V IPK Peak Inductor Current VIN = 3V, VOUT = 1.08V, Duty Cycle < 35% VLOADREG Output Voltage Load Regulation VIN Input Voltage Range IS Input DC Bias Current Active Mode Sleep Mode Shutdown (Note 4) VOUT = 1.08V, ILOAD = 0A VOUT = 1.236V, ILOAD = 0A VRUN = 0V, VIN = 5.5V fOSC Oscillator Frequency VOUT = 1.2V VOUT = 0V RPFET RDS(ON) of P-Channel FET RNFET ● MIN TYP MAX UNITS 1.164 1.2 1.236 V 2.5 6.25 10 % 0.04 0.4 %/V 1 1.25 A ● 0.75 0.5 ● % 5.5 V 300 20 0.1 400 35 1 µA µA µA 1.5 210 1.8 MHz kHz ISW = 100mA 0.4 0.5 Ω RDS(ON) of N-Channel FET ISW = –100mA 0.35 0.45 Ω ILSW SW Leakage VRUN = 0V, VSW = 0V or 5V, VIN = 5V ±0.01 ±1 µA VRUN RUN Threshold ● 1 1.5 V IRUN RUN Leakage Current ● ±0.01 ±1 µA Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3406E-1.2 is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3406-1.2: TJ = TA + (PD)(250°C/W) ● 2.5 1.2 0.3 Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. 340612f 2 LTC3406-1.2 U W TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise specified. (From Figure 1) Efficiency vs Input Voltage Reference Voltage vs Temperature Efficiency and Power Loss 100 1.228 100 VIN = 3.6V 1.218 90 IOUT = 100mA 85 IOUT = 10mA 80 75 70 3 80 70 60 5 4 INPUT VOLTAGE (V) 40 0.1 6 1 10 ILOAD (mA) 340612 G01 1.198 1.188 1.168 –50 –25 1000 100 1.208 1.178 VIN = 2.7V VIN = 3.6V VIN = 4.2V 50 IOUT = 600mA 2 REFERENCE VOLTAGE (V) 90 EFFICIENCY (%) EFFICIENCY (%) 95 50 25 75 0 TEMPERATURE (°C) Oscillator Frequency vs Supply Voltage 1.70 1.8 VIN = 3.6V 125 340612 G03 340612 GO2 Oscillator Frequency vs Temperature 100 Output Voltage vs Load Current 1.225 TA = 25°C 1.55 1.50 1.45 1.40 1.35 1.30 –50 –25 50 25 75 0 TEMPERATURE (°C) 100 1.7 1.6 1.5 1.4 3 4 5 SUPPLY VOLTAGE (V) 340612 G06 RDS(ON) vs Temperature Supply Current vs Supply Voltage 50 VIN = 2.7V 0.6 0.6 0.5 0.5 VIN = 4.2V MAIN SWITCH RDS(ON) (Ω) RDS(ON) (Ω) 0 100 200 300 400 500 600 700 800 900 1000 LOAD CURRENT (mA) 0.7 TA = 25°C 0.3 6 340612 G05 RDS(ON) vs Input Voltage 0.4 1.195 1.175 2 340612 G04 0.7 1.205 1.185 1.3 1.2 125 1.215 SYNCHRONOUS SWITCH VIN = 3.6V 40 0.4 0.3 0.2 0.2 0.1 0.1 ILOAD = 0A 45 SUPPLY CURRENT (µA) FREQUENCY (MHz) 1.60 OUTPUT VOLTAGE (V) OSCILLATOR FREQUENCY (MHz) 1.65 35 30 25 20 15 10 5 MAIN SWITCH SYNCHRONOUS SWITCH 0 0 1 5 4 2 3 INPUT VOLTAGE (V) 6 7 340612 G07 0 –50 –25 50 25 75 0 TEMPERATURE (°C) 0 100 125 340612 G08 2 4 3 5 SUPPLY VOLTAGE (V) 6 340612 G09 340612f 3 LTC3406-1.2 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1) Supply Current vs Temperature 50 Switch Leakage vs Temperature 300 VIN = 3.6V ILOAD = 0A 45 VIN = 5.5V RUN = 0V 250 SWITCH LEAKAGE (nA) SUPPLY CURRENT (µA) 40 35 30 25 20 15 10 200 150 100 MAIN SWITCH 50 5 SYNCHRONOUS SWITCH 0 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 0 –50 –25 50 25 75 0 TEMPERATURE (°C) Discontinuous Operation Switch Leakage vs Input Voltage RUN = 0V TA = 25°C SWITCH LEAKAGE (pA) 100 SW 2V/DIV SYNCHRONOUS SWITCH 80 60 VOUT 50mV/DIV AC COUPLED MAIN SWITCH IL 200mA/DIV 40 20 0 125 340612 G11 340612 G10 120 100 4µs/DIV 3406B12 G13 20µs/DIV VIN = 3.6V ILOAD = 25mA TO 600mA 340612 G15 VIN = 3.6V ILOAD = 25mA 0 1 2 3 4 INPUT VOLTAGE (V) 5 6 340612 G12 (From Figure 1a Except for the Resistive Divider Resistor Values) Load Step Load Step RUN 2V/DIV VOUT 100mV/DIV AC COUPLED VOUT 1V/DIV ILOAD 500mA/DIV IL 500mA/DIV IL 500mA/DIV 40µs/DIV VIN = 3.6V ILOAD = 100mA TO 600mA 340612 G14 340612f 4 LTC3406-1.2 U W TYPICAL PERFOR A CE CHARACTERISTICS (From Figure 1a Except for the Resistive Divider Resistor Values) Load Step Load Step VOUT 100mV/DIV AC COUPLED VOUT 100mV/DIV AC COUPLED ILOAD 500mA/DIV ILOAD 500mA/DIV IL 500mA/DIV IL 500mA/DIV 340612 G16 20µs/DIV 20µs/DIV 340612 G17 VIN = 3.6V ILOAD = 200mA TO 600mA VIN = 3.6V ILOAD = 100mA TO 600mA U U U PI FU CTIO S RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1µA supply current. Do not leave RUN floating. GND (Pin 2): Ground Pin. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2µF or greater ceramic capacitor. VOUT (Pin 5): Output Voltage Feedback Pin. An internal resistive divider divides the output voltage down for comparison to the internal reference voltage. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. W FU CTIO AL DIAGRA U SLOPE COMP OSC 0.65V OSC 4 VIN FREQ SHIFT – VOUT + FB + – 120k S Q R Q RS LATCH VIN – OVDET RUN 1 0.8V REF 0.8V + ∆VOVL SHUTDOWN 5Ω + ICOMP – EA OV SWITCHING LOGIC AND BLANKING CIRCUIT ANTISHOOTTHRU 3 SW + IRCMP – 0.8V 60k + 5 2 GND 3406B12 BD 340612f 5 U LTC3406-1.2 U OPERATIO (Refer to Functional Diagram) VIN 2.7V TO 5.5V 4 CIN** 4.7µF CER VIN SW 3 2.2µH* COUT 10µF CER LTC3406-1.2 1 VOUT RUN 5 † VOUT 1.2V 600mA 340612 F01 GND 2 *MURATA LQH3C2R2M24 **TAIYO YUDEN JMK212BJ475MG † TAIYO YUDEN JMK316BJ106ML Figure 1. Typical Application Main Control Loop The LTC3406-1.2 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. When the load current increases, it causes a slight decrease in the feedback voltage, FB, relative to the 0.8V reference, which in turn causes the EA amplifier’s output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. The comparator OVDET guards against transient overshoots >6.25% by turning the main switch off and keeping it off until the fault is removed. Burst Mode Operation The LTC3406-1.2 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. In Burst Mode operation, the peak current of the inductor is set to approximately 200mA regardless of the output load. Each burst event can last from a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 20µA. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage droops, the EA amplifier’s output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET on. This process repeats at a rate that is dependent on the load demand. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 210kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator’s frequency will progressively increase to 1.5MHz when VOUT rises above 0V. 340612f 6 LTC3406-1.2 U W U U APPLICATIO S I FOR ATIO The basic LTC3406-1.2 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Table 1. Representative Surface Mount Inductors PART NUMBER VALUE (µH) DCR (Ω MAX) Sumida CDRH3D16 1.5 2.2 3.3 4.7 0.043 0.075 0.110 0.162 1.55 1.20 1.10 0.90 3.8 × 3.8 × 1.8 Sumida CMD4D06 2.2 3.3 4.7 0.116 0.174 0.216 0.950 0.770 0.750 3.5 × 4.3 × 0.8 Panasonic ELT5KT 3.3 4.7 0.17 0.20 1.00 0.95 4.5 × 5.4 × 1.2 Murata LQH3C 1.0 2.2 4.7 0.060 0.097 0.150 1.00 0.79 0.65 2.5 × 3.2 × 2.0 Inductor Selection For most applications, the value of the inductor will fall in the range of 1µH to 4.7µH. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is ∆IL = 240mA (40% of 600mA). ∆IL = ⎛ V ⎞ 1 VOUT ⎜ 1 − OUT ⎟ ( f)(L) ⎝ VIN ⎠ (1) The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 720mA rated inductor should be enough for most applications (600mA + 120mA). For better efficiency, choose a low DC-resistance inductor. Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3406-1.2 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3406-1.2 applications. MAX DC SIZE CURRENT (A) W × L × H (mm3) CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: 1/ 2 VOUT (VIN − VOUT )] [ CIN required IRMS ≅ IOMAX VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer’s ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). 340612f 7 LTC3406-1.2 U W U U APPLICATIO S I FOR ATIO ⎛ 1 ⎞ ∆VOUT ≅ ∆IL ⎜ ESR + ⎟ ⎝ 8fC OUT ⎠ where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ∆IL increases with input voltage. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC34061.2’s control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3406-1.2 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 2. 1 0.1 POWER LOSS (W) Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ∆VOUT is determined by: 0.01 0.001 0.0001 0.00001 0.1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 10 100 LOAD CURRENT (mA) 1000 340612 F02 Figure 2. Power Loss vs Load Current 340612f 8 LTC3406-1.2 U W U U APPLICATIO S I FOR ATIO 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is “chopped” between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC) (2) To avoid the LTC3406-1.2 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(θJA) where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3406-1.2 with an input voltage of 2.7V, a load current of 600mA and an ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) at 70°C is approximately 0.52Ω for the P-channel switch and 0.42Ω for the N-channel switch. Using equation (2) to find the series resistance looking into the SW pin gives: RSW = 0.52Ω(0.44) + 0.42Ω(0.56) = 0.46Ω Therefore, power dissipated by the part is: The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. For the SOT-23 package, the θJA is 250°C/ W. Thus, the junction temperature of the regulator is: Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. which is below the maximum junction temperature of 125°C. Thermal Considerations In most applications the LTC3406-1.2 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3406-1.2 is running at high ambient temperature with low supply voltage, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. PD = ILOAD2 • RSW = 165.6mW TJ = 70°C + (0.1656)(250) = 111.4°C Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RSW). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (∆ILOAD • ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. 340612f 9 LTC3406-1.2 U W U U APPLICATIO S I FOR ATIO The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1µF) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 • CLOAD). Thus, a 10µF capacitor charging to 3.3V would require a 250µs rise time, limiting the charging current to about 130mA. Design Example As a design example, assume the LTC3406-1.2 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.6A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. With this information we can calculate L using equation (1), L= When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3406-1.2. These items are also illustrated graphically in Figures 3 and 4. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. (3) Substituting VIN = 4.2V, ∆IL = 240mA and f = 1.5MHz in equation (3) gives: L= PC Board Layout Checklist ⎛ 1.2V ⎞ 1 1.2V⎜ 1 − ( f)(∆IL ) ⎝ VIN ⎟⎠ 1.2V ⎛ 1.2V ⎞ ⎜1 − ⎟ = 2.38 µH 1.5MHz(240mA) ⎝ 4.2V ⎠ A 2.2µH inductor works well for this application. For best efficiency choose a 720mA or greater inductor with less than 0.2Ω series resistance. CIN will require an RMS current rating of at least 0.3A ≅ ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.25Ω. In most cases, a ceramic capacitor will satisfy this requirement. 2. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 3. Keep the (–) plates of CIN and COUT as close as possible. 1 VIA TO VOUT VIA TO VIN RUN LTC3406-1.2 2 – GND VOUT PIN 1 5 + 3 L1 LTC3406-1.2 VOUT COUT VOUT VIN SW VIN 4 L1 CIN VIN 340612 F03 SW COUT CIN GND BOLD LINES INDICATE HIGH CURRENT PATHS Figure 3. LTC3406-1.2 Layout Diagram 340612 F04 Figure 4. LTC3406-1.2 Suggested Layout 340612f 10 LTC3406-1.2 U TYPICAL APPLICATIO S Single Li-Ion 1.2V/600mA Regulator for Lowest Profile, ≤1mm High 4 VIN 2.7V TO 4.2V CIN** 4.7µF CER VIN SW 3 2.2µH† COUT1* 10µF CER LTC3406-1.2 1 RUN VOUT GND VOUT 1.2V 5 340612 TA02 2 *MURATA GRM219R60JI06KE19B **AVX06036D475MAT † FDK MIPW3226D2R2M LTC3406-1.2 Efficiency Load Step 100 90 VOUT 100mV/DIV AC COUPLED EFFICIENCY (%) 80 70 60 ILOAD 500mA/DIV 50 40 IL 500mA/DIV 30 20 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 0 0.1 1 10 LOAD (mA) 100 20µs/DIV VIN = 3.6V ILOAD = 20mA TO 600mA 340612 TA04 1000 340612 TA03 U PACKAGE DESCRIPTIO S5 Package 5-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1635) 0.62 MAX 0.95 REF 2.90 BSC (NOTE 4) 1.22 REF 1.4 MIN 3.85 MAX 2.62 REF 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 TYP 5 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.09 – 0.20 (NOTE 3) 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 1.90 BSC S5 TSOT-23 0302 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 340612f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 11 LTC3406-1.2 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1616 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 3.6V to 25V, VOUT = 1.25V, IQ = 1.9mA, ISD = <1µA, ThinSOT Package LT1676 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 7.4V to 60V, VOUT = 1.24V, IQ = 3.2mA, ISD = 2.5µA, S8 Package LTC1701/LT1701B 750mA (IOUT), 1MHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 2.5V to 5V, VOUT = 1.25V, IQ = 135µA, ISD = <1µA, ThinSOT Package LT1776 500mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 90% Efficiency, VIN = 7.4V to 40V, VOUT = 1.24V, IQ = 3.2mA, ISD = 30µA, N8, S8 Packages LTC1877 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC1878 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC1879 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 15µA, ISD = <1µA, TSSOP-16 Package LTC3403 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = Dynamically Adjustable, IQ = 20µA, ISD = <1µA, DFN Package LTC3404 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10µA, ISD = <1µA, MS8 Package LTC3405/LTC3405A 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 20µA, ISD = <1µA, ThinSOT Package LTC3406 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.6V, IQ = 20µA, ISD = <1µA, ThinSOT Package LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA, MS Package LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD = <1µA, TSSOP-16E Package LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 2.5V, IQ = 25µA, ISD = <1µA, MS Package 340612f 12 Linear Technology Corporation LT/TP 0105 1K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2005