LT3570 1.5A Buck Converter, 1.5A Boost Converter and LDO Controller DESCRIPTION FEATURES n n n n n n n n The LT®3570 is a buck and boost converter with internal power switches and LDO controller. Each converter is designed with a 1.5A current limit and an input range from 2.5V to 36V, making the LT3570 ideal for a wide variety of applications. Switching frequencies up to 2MHz are programmed with an external timing resistor and the oscillator can be synchronized to an external clock up to 2.75MHz. 2.5V to 36V Input Voltage Range Programmable Switching Frequency from 500kHz to 2MHz Synchronizable Up to 2.75MHz VOUT(MIN): 0.8V Independent Soft-Start for Each Converter Separate VIN Supplies for Each Converter Duty Cycle Range: 0% to 90% at 1MHz Available in 24-Lead (4mm × 4mm) QFN and 20-Lead TSSOP Packages The LT3570 features a programmable soft-start function that limits the feedback voltage during start-up helping prevent overshoot and limiting inrush current. The LDO controller is capable of delivering up to 10mA of base current to an external NPN transistor. APPLICATIONS n n n n n Cable and Satellite Set-Top Boxes Automotive Systems Telecom Systems “Dying Gasp” Systems TFT LCD Displays L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION VIN 5V VIN1 VIN2 VIN3 BIAS SHDN1 SHDN2 SHDN3 D1 VOUT1 12V 275mA SHDN1 SHDN2 SHDN3 D3 100nF SW2 D2 6.8μH SW1 143k 10μF FB1 SS1 VC1 10.0k Efficiency BOOST LT3570 100 VOUT2 3.3V 1A 3.3μH 95 32.4k FB2 SS2 VC2 EFFICIENCY (%) 10μF 22μF 22k 10nF 10.2k 90 85 fSW = 1.2MHz VIN = 5V VOUT1 = 12V VOUT2 = 3.3V VOUT3 = 2.5V IOUT1 = 275mA IOUT3 = 100mA 80 1nF 75 22k NPN_DRV Q1 1nF 10nF 22.1k RT SYNC FB3 GND VOUT3 2.5V 100mA 2.2μF 70 0 0.2 0.4 0.6 IOUT2 (A) 0.8 1.0 3570 TA01b 10.2k 15.8k 3570 TA01a 3570fa 1 LT3570 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN1, VIN2, VIN3, VBIAS Voltage ..................................40V BOOST Voltage .........................................................60V BOOST Pin Above SW2 .............................................25V NPN_DRV Voltage .......................................................8V SW1 Voltage .............................................................40V SHDN1, SHDN2, SHDN3 Voltage ..............................40V SYNC, RT Voltage ........................................................3V SS1, SS2 Voltage ........................................................3V FB1, FB2, FB3 Voltage ...............................................10V VC1, VC2 Voltage..........................................................3V Maximum Junction Temperature........................... 125°C Operating Temperature Range (Note 2).. –40°C to 125°C Storage Temperature Range TSSOP ............................................... –65°C to 150°C QFN.................................................... –65°C to 125°C Lead Temperature (Soldering, 10 sec) TSSOP Only ...................................................... 300°C PIN CONFIGURATION FB2 TOP VIEW FB3 NPN_DRV VIN3 BIAS BOOST TOP VIEW 24 23 22 21 20 19 VIN2 1 18 VC2 VIN2 2 17 SS2 SW2 3 16 GND 25 SW1 4 15 RT FB1 1 20 VC1 SHDN1 2 19 SS1 SHDN2 3 18 VIN1 SHDN3 4 17 GND SYNC 5 RT 6 21 16 SW1 15 SW2 GND 5 14 SYNC SS2 7 14 VIN2 GND 6 13 SHDN3 VC2 8 13 BOOST FB2 9 12 VIN3 FB3 10 SHDN2 SHDN1 FB1 9 10 11 12 VC1 SS1 8 VIN1 7 11 NPN_DRV FE PACKAGE 20-LEAD PLASTIC TSSOP UF PACKAGE 24-LEAD (4mm s 4mm) PLASTIC QFN TJMAX = 125°C, θJA = 37°C/W EXPOSED PAD (PIN 25) IS GND, MUST BE SOLDERED TO PCB TJMAX = 125°C, θJA = 38°C/W EXPOSED PAD (PIN 25) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3570EUF#PBF LT3570EUF#TRPBF 3570 24-Lead (4mm × 4mm) Plastic QFN –40°C to 125°C LT3570IUF#PBF LT3570IUF#TRPBF 3570 24-Lead (4mm × 4mm) Plastic QFN –40°C to 125°C LT3570EFE#PBF LT3570EFE#TRPBF LT3570FE 20-Lead Plastic TSSOP –40°C to 125°C LT3570IFE#PBF LT3570IFE#TRPBF LT3570FE 20-Lead Plastic TSSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3570fa 2 LT3570 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN1,2,3 = 12V, VSHDN1,2,3 = 12V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS 2.1 2.5 V 2.1 2.5 V 0 1.5 μA Minimum Operating Voltage (VIN1) (Note 3) l Minimum Operating Voltage (VIN2) (Note 3) l Shutdown Current (Note 4) VSHDN1,2,3 = 0V VIN1 Quiescent Current VSHDN1 = 12V, VSHDN2,3 = 0V, VC1 = 0.4V (Not Switching) VSHDN1 = 0V, VSHDN2,3 = 12V 3.2 65 4.5 150 mA μA VIN2 Quiescent Current VSHDN1,3 = 0V, VSHDN2 = 12V, VC2 = 0.4V (Not Switching) VSHDN1,3 = 12V, VSHDN2 = 0V 3.5 3.5 4.5 4.5 mA mA VIN3 Quiescent Current VSHDN1,2 = 0V, VSHDN3 = 12V VSHDN1,2 = 12V, VSHDN3 = 0V 700 0 950 1.5 μA μA Bias Quiescent Current VBIAS = 2.5V 2.3 3.1 mA VSHDN1,2,3 Pin Threshold IVIN2 > 100μA 1.4 V 1.25 1.4 V 30 0.1 50 1.5 μA μA 500 2100 550 2300 kHz kHz 0.3 l VSHDN1,2,3 Pin UVLO 1.1 VSHDNX Pin Current VSHDNX = 12V, VSHDNY,Z = 0V (Note 5) VSHDN1,2,3 = 0V Switching Frequency RT = 44.2k RT = 7.87k 450 1900 Maximum Duty Cycle RT = 44.2k RT = 7.87k 95 80 Synchronous Frequency Threshold Synchronous Frequency Ratio, fSYN/fOSC % % 0.3 1.5 RT = 44.2k RT = 7.87k 1.3 1.3 Synchronous Frequency Minimum On/Off Time 50 l FB1,2,3 Pin Voltage FB1,2,3 Pin Voltage Line Regulation VVIN1,2,3 = 2.5V to 40V, VC1,2 = 1V FB1,2 Pin Bias Current VFB1,2 = 800mV, VC1,2 = 1V (Note 6) V 772 ns 788 804 0.01 mV %/V 30 200 200 nA FB3 Pin Bias Current VFB3 = 800mV (Note 6) 30 SS1,2 Pin Source Current VSS1,2 = 500mV 4.5 μA nA VC1,2 Pin Source Current VFB1,2 = 600mV 12 μA VC1,2 Pin Sink Current VFB1,2 = 1V 12 μA Error Amplifier 1 Transconductance 190 μMho Error Amplifier 1 Voltage Gain 100 V/V VC1 Pin Switching Threshold 750 mV VC1 to SW1 Current Gain 5.9 A/V SW1 SW1 Current Limit (Note 7) 1.5 2.4 SW1 VCESAT ISW1 = 1A (Note 7) 240 SW1 Leakage Current SW1 = 40V, VSHDN1 = 0V 0.2 3.1 A mV 5 μA 3570fa 3 LT3570 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN1,2,3 = 12V, VSHDN1,2,3 = 12V unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS SW2 Error Amplifier 2 Transconductance 195 μMho Error Amplifier 2 Voltage Gain 100 V/V VC2 Pin Switching Threshold 700 mV VC2 to SW2 Current Gain 5.4 SW2 Current Limit (Note 7) 1.5 2.4 SW2 VCESAT ISW2 = 1A (Note 7) 240 SW2 Leakage Current SW2 = 0V, VIN2 = 40V, VSHDN2 = 0V 0.2 BOOST Pin Current ISW2 = 0.5A ISW2 = 1.5A 15 30 A/V 3.1 A mV 5 μA mA mA LDO LDO Maximum Output Current VFB3 = 600mV 10 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3570E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3570I is guaranteed over the full –40°C to 125°C operating junction temperature range. Note 3: VIN2 supplies power for the part. VIN1 supplies power only to the boost converter. VIN3 supplies power only to the LDO Controller. 20 mA Note 4: Shutdown current is for each individual input current. Note 5: Current flows into the pin. Note 6: Current flows out of the pin. Note 7: Switch current limit and switch VCESAT guaranteed by design and/or correlation to static test. Note 8: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed the maximum operating junction temperature range when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. TYPICAL PERFORMANCE CHARACTERISTICS Frequency vs Temperature Feedback Voltage vs Temperature RT = 7.87k 0.795 4.8 2000 0.790 0.785 0.780 4.6 CURRENT (μA) FREQUENCY (kHz) VOLTAGE (V) Soft-Start Current vs Temperature 5.0 2500 0.800 1500 RT = 20k 1000 0.770 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3570 G01 0 –50 –25 4.2 4.0 RT = 44.2k 500 0.775 4.4 3.8 0 25 50 75 100 125 150 TEMPERATURE (°C) 3570 G02 3.6 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3570 G03 3570fa 4 LT3570 TYPICAL PERFORMANCE CHARACTERISTICS VIN1 Quiescent Current vs Temperature VIN2 Quiescent Current vs Temperature 3.5 3.0 4.0 900 3.5 800 700 CURRENT (mA) 2.0 1.5 1.0 0.5 0 –50 –25 0 CURRENT (μA) 3.0 2.5 CURRENT (mA) VIN3 Quiescent Current vs Temperature 2.5 2.0 1.5 200 0.5 100 0 1.25 25 50 75 100 125 150 TEMPERATURE (°C) 75 100 125 150 1.00 0.75 0.50 0 –50 –25 SHDN Pin Current vs Voltage 40 30 20 10 0.25 0 50 50 0 0 25 50 75 100 125 150 TEMPERATURE (°C) 3570 G07 0 5 10 15 20 25 VOLTAGE (V) 30 35 40 3570 G09 3570 G08 SW1 Saturation Voltage vs SW1 Current SW1 Current Limit vs Duty Cycle 3.0 350 2.5 300 250 2.0 VOLTAGE (mV) 0 –50 –25 25 3570 G06 CURRENT (μA) SHDN PIN VOLTAGE (V) 2.5 CURRENT (A) CURRENT (mA) 1.50 0.5 0 TEMPERATURE (°C) SHDN Pin UVLO vs Temperature Bias Pin Current vs Temperature 1.0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 3570 G05 3.0 1.5 400 300 3570 G04 2.0 500 1.0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 600 1.5 1.0 200 150 100 TJ = 125°C TJ = 25°C TJ = –40°C 0.5 0 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 3570 G10 50 0 TJ = 125°C TJ = 25°C TJ = –40°C 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 CURRENT (A) 3570 G11 3570fa 5 LT3570 TYPICAL PERFORMANCE CHARACTERISTICS SW2 Saturation Voltage vs SW2 Current 3.0 350 2.5 300 250 2.0 VOLTAGE (mV) CURRENT (A) SW2 Current Limit vs Duty Cycle 1.5 1.0 200 150 100 TJ = 125°C TJ = 25°C TJ = –40°C 0.5 0 0 TJ = 125°C TJ = 25°C TJ = –40°C 50 0 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 CURRENT (A) 3570 G12 3570 G13 BOOST Pin Current vs Switch Current NPN_DRV Output Current vs VIN3 30 18 16 25 CURRENT (mA) CURRENT (mA) 20 TJ = –40°C 15 TJ = 25°C 14 TJ = 25°C TJ = 125°C 10 TJ = 125°C 12 TJ = –40°C 10 8 6 4 5 2 0 0 0.2 0.4 0.6 CURRENT (A) 0.8 1.0 3570 G14 0 1 5 10 15 20 25 VOLTAGE (V) 30 35 40 3570 G15 3570fa 6 LT3570 PIN FUNCTIONS (QFN/TSSOP) VIN2 (Pins 1,2/Pin 14): Input Voltage for the Buck Regulator. This pin also supplies the current to the internal circuitry of the LT3570. This pin must be locally bypassed with a capacitor. SW2 (Pin 3/Pin15): Switch Node. This pin connects to the emitter of an internal NPN power switch. Connect a diode, inductor and boost capacitor to this pin to form the buck regulator SW1 (Pin 4/Pin16): Switch Node. This pin connects to the collector of an internal NPN power switch. Connect a diode and inductor to this pin to form the boost regulator GND (Pins 5, 6, 16, 25/Pins 17, 21): Ground. The Exposed Pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The Exposed Pad must be soldered to the circuit board for proper operation. VIN1 (Pin 7/Pin18): Input Voltage for the Boost Regulator. This pin supplies current to drive the boost NPN transistor of the LT3570. This pin must be locally bypassed with a capacitor. SS1 (Pin 8/Pin 19): Soft-Start Pin. Place a soft-start capacitor here. Upon start-up, a current charges the capacitor to 2V. This pin ramps the reference voltage of the boost switcher. VC1 (Pin 9/Pin 20): Control Voltage and Compensation Pin for the Internal Error Amplifier. Connect a series RC from this pin to ground to compensate the switching regulator loop for the boost regulator. FB1 (Pin 10/Pin 1): Feedback Pin. The LT3570 regulates this pin to 788mV. Connect the feedback resistors to this pin to set the output voltage for the boost switching regulator. SHDN1 (Pin 11/Pin 2): Shutdown Pin. Tie to 1.5V or more to enable the boost switcher. Ground to shutdown the part. SHDN2 (Pin 12/Pin 3): Shutdown Pin. Tie to 1.5V or more to enable the buck switcher. Ground to shutdown the part. SHDN3 (Pin13/Pin 4): Shutdown Pin. Tie to 1.5V or more to enable the NPN LDO. Ground to shut down the part. SYNC (Pin 14/Pin 5): Synchronization Pin. The SYNC pin is used to synchronize the internal oscillator to an external signal. The synchronizing range is equal to the initial operating frequency set by the RT pin up to 1.3 times the initial operating frequency. RT (Pin 15/Pin 6): Frequency Set Pin. Place a resistor to GND to set the internal frequency. The range of oscillation is 500kHz to 2MHz. SS2 (Pin 17/Pin 7): Soft-Start Pin. Place a soft-start capacitor here. Upon start-up, a current charges the capacitor to 2V. This pin ramps the reference voltage of the buck switcher. VC2 (Pin 18/Pin 8): Control Voltage and Compensation Pin for the Internal Error Amplifier. Connect a series RC from this pin to ground to compensate the switching regulator loop for the buck regulator. FB2 (Pin 19/Pin 9): Feedback Pin. The LT3570 regulates this pin to 788mV. Connect the feedback resistors to this pin to set the output voltage for the buck switching regulator. FB3 (Pin 20/Pin 10): Feedback Pin. The LT3570 regulates this pin to 788mV. Connect the feedback resistors to this pin to set the output voltage for the LDO controller. NPN_DRIVE (Pin 21/Pin 11): Base Drive for the External NPN. This pin provides a bias current to drive the base of the NPN. This base current is driven from the IN3 supply voltage. VIN3 (Pin 22/Pin 12): Input Voltage for the NPN LDO. This pin supplies current to drive the base of the NPN. This pin must be locally bypassed with a capacitor. BIAS (Pin 23): QFN Package Only. This pin supplies current to the internal circuitry of the LT3570 if greater than 2.5V. This pin must be locally bypassed with a capacitor. BOOST (Pin 24/Pin 13): Bias for the Base Drive of the NPN Switch for the Buck Regulator. This pin provides a bias voltage higher than VIN2. The voltage on this pin is charged up through an external Schottky diode. 3570fa 7 8 C2 R2B R1B D3 L2 C6B D2 C5 R4 FB2 SS2 SW2 Q2 BOOST RT SYNC R5 VIN2 788mV VDD R A8 Q S + + – – + A5 VDD A7 A6 VDD – + Figure 1. Block Diagram C4B C4A R3A – + A3 VDD SHDN2 R3B SHDN1 VC1 A9 OSCILLATOR A10 REGULATOR VDD VC2 BIAS Q A4 R S 788mV SHDN3 A11 A1 VDD A2 VDD – + + + – – + VDD FB1 SS1 GND 788mV VDD R6 Q1 SW1 VIN1 FB3 NPN_DRV VIN3 3570 BD C6A R2A R1A D1 L1 C1 R2C R1C Q3 C3 LT3570 BLOCK DIAGRAM 3570fa LT3570 OPERATION The LT3570 is a constant frequency, current mode, buck converter and boost converter with an NPN LDO regulator. Operation can be best understood by referring to the Block Diagram. If all of the SHDN pins are held low, the LT3570 is shut down and draws zero quiescent current. When any of the pins exceed 1.4V the internal bias circuits turn on. Each regulator will only begin regulating when its corresponding SHDN pin is pulled high. Each switching regulator controls the output voltage in a similar manner. The operation of the switchers can be understood by looking at the boost regulator. A pulse from the oscillator sets the RS flip-flop A4 and turns on the internal NPN bipolar power switch Q1. Current in Q1 and the external inductor L1 begins to increase. When this current exceeds a level determined by the voltage at VC1, comparator A3 resets A4, turning off Q1. The current in L1 flows through the external Schottky diode D1 and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC1 pin controls the current through the inductor to the output. The internal error amplifier A1 regulates the output voltage by continually adjusting the VC1 pin voltage. The threshold for switching on the VC1 pin is approximately 750mV and an active clamp of 1.15V limits the output current. The soft-start capacitor C6A allows the part to slowly start up by ramping the internal reference. The driver for the buck regulator can operate from either VIN2 or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. The driver for the boost regulator is operated from VIN1. The BIAS pin allows the internal circuitry to draw its current from a lower voltage supply than the input. This reduces power dissipation and increases efficiency. If the voltage on the BIAS pin falls below 2.5V, then the LT3570 quiescent current will flow from VIN2. 3570fa 9 LT3570 APPLICATIONS INFORMATION FB Resistor Network The output voltage is programmed with a resistor divider (refer to the Block Diagram) between the output and the FB pin. Choose the resistors according to: ⎞ ⎛ V R1= R2 ⎜ OUT – 1⎟ ⎝ 788mV ⎠ Buck Inductor Selection and Maximum Output Current A good first choice for the inductor value is L= VOUT2 + VF for SW2 0.75 • f where VF is the voltage drop of the catch diode (~0.4V) and f is the switching frequency. With this inductance value or greater, the maximum load current will be 1A, independent of input voltage. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.1Ω. Table 1 lists several vendors and types that are suitable. Table 1. Inductors PART NUMBER Sumida CDRH4D28-3R3 CDRH4D28-4R7 CDC5D23-2R2 CR43-3R3 CDRH5D28-100 Coilcraft DO1608C-332 DO1608C-472 MOS6020-332 D03314-103 D03314-222 Toko (D62F)847FY-2R4M (D73LF)817FY-2R2M Coiltronics TP3-4R7 TP1-2R2 TP4-100 VALUE (μH) ISAT (A) DCR (Ω) HEIGHT (mm) 3.3 4.7 2.2 3.3 10 1.57 1.32 2.50 1.44 1.3 0.049 0.072 0.03 0.086 0.048 3.0 3.0 2.5 3.5 3.0 3.3 4.7 3.3 10 2.2 2.00 1.50 1.8 0.8 1.6 0.080 0.090 0.046 0.520 0.200 2.9 2.9 2.0 1.4 1.4 The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a slightly higher maximum load current and will reduce the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (VOUT2/VIN2 > 0.5) a minimum inductance is required to avoid subharmonic oscillations, see Application Note 19. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3570 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3570 will deliver depends on the switch current limit, the inductor value and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor: ΔIL2 = (1– DC2)( VOUT2 + VF ) L•f where DC2 is the duty cycle and is defined as: DC2 = VOUT2 VIN2 The peak inductor and switch current is: 2.4 2.2 2.5 2.7 0.037 0.03 2.7 3.0 4.7 2.2 10 1.5 1.3 1.5 0.181 0.188 0.146 2.2 1.8 3.0 ISWPK2 =ILPK2 =IOUT2 + ΔIL2 2 To maintain output regulation, this peak current must be less than the LT3570’s switch current limit ILIM2. ILIM2 is at least 1.5A at low duty cycles and decreases linearly 3570fa 10 LT3570 APPLICATIONS INFORMATION to 1.2A at DC2 = 0.8. The maximum output current is a function of the chosen inductor value: IOUT2(MAX) =ILIM2 – ΔIL2 2 = 1.5 • (1– 0.25 • DC2) – ΔIL2 2 Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors and choose one to meet cost or space goals. Then use these equations to check that the LT3570 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT2 is less than ΔIL2/2. Boost Inductor Selection For most applications the inductor will fall in the range of 2.2μH to 22μH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the power switch, which has a 1.5A current limit. Higher values also reduce input ripple voltage and reduce core loss. The following procedure is suggested as a way of choosing a more optimum inductor. Assume that the average inductor current for a boost converter is equal to the load current times VOUT1/VIN1 and decide whether or not the inductor must withstand continuous overload conditions. If average inductor current at maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 1.5A overload condition. Also be aware that boost converters are not short-circuit protected, and that under short conditions, inductor current is limited only by the available current of the input supply Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core materials fall somewhere in between. The following formula assumes continuous mode operation but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. IPEAK1 = IOUT1 • VOUT1 VIN1 ( VOUT1 – VIN1 ) + VIN1 2 • f • L • VOUT1 Make sure that IPEAK1 is less than the switch current ILIM1. ILIM1 is at least 1.5A at low duty cycles and decreases linearly to 1.2A at DC1 = 0.8. The maximum switch current limit can be calculated by the following formula: ILIM1 = 1.5 • (1 – 0.25 • DC1) where DC1 is the duty cycle and is defined as: DC1= 1– VIN1 VOUT1 Remember also that inductance can drop significantly with DC current and manufacturing tolerance. Consideration should also be given to the DC resistance of the inductor as this contributes directly to the efficiency losses in the overall converter. Table 1 lists several inductor vendors and types that are suitable. Buck Output Capacitor Selection For 5V and 3.3V outputs, a 10μF, 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. For lower voltages, 10μF is adequate for ripple requirements but increasing COUT will improve transient performance. Other types and values will also work; the following discusses tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilize the LT3570’s control loop. Because the LT3570 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple 3570fa 11 LT3570 APPLICATIONS INFORMATION and small circuit size, are therefore an option. You can estimate output ripple with the following equations: VRIPPLE = ΔIL2 for ceramic capacitors 8 • f • COUT and VRIPPLE = ΔIL2 • ESR for electrolytic capacitors (tantalum and aluminum) The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC(RMS) = ΔIL2 Table 2. Low ESR Surface Mount Capacitors VENDOR TYPE SERIES Taiyo Yuden Ceramic X5R, X7R AVX Ceramic Tantalum X5R, X7R TPS Kemet Tantalum Ta Organic Al Organic T491, T494, T495 T520 A700 Sanyo Ta or Al Organic POSCAP Panasonic Al Organic SP CAP TDK Ceramic X5R, X7R Boost Output Capacitor Selection 12 Another constraint on the output capacitor is that it must have greater energy storage than the inductor; if the stored energy in the inductor transfers to the output, the resulting voltage step should be small compared to the regulation voltage. For a 5% overshoot, this requirement indicates: ⎛ I ⎞ COUT > 10 • L • ⎜ LIM2 ⎟ ⎝ VOUT2 ⎠ response for large changes in load current. Table 2 lists several capacitor vendors. 2 The low ESR and small size of ceramic capacitors make them the preferred type for LT3570 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Electrolytic capacitors are also an option. The ESRs of most aluminum electrolytic capacitors are too large to deliver low output ripple. Tantalum, as well as newer, lower ESR organic electrolytic capacitors intended for power supply use are suitable. Chose a capacitor with a low enough ESR for the required output ripple. Because the volume of the capacitor determines its ESR, both the size and the value will be larger than a ceramic capacitor that would give similar ripple performance. One benefit is that the larger capacitance may give better transient Low ESR capacitors should be used at the output to minimize the output ripple voltage. Multilayer ceramic capacitors are the best choice, as they have a very low ESR and are available in very small packages. Always use a capacitor with a sufficient voltage rating. Boost regulators have large RMS ripple current in the output capacitor, which must be rated to handle the current. The formula to calculate this is: IRIPPLE(RMS) =IOUT V –V DC1 = IOUT1 OUT1 IN1 VIN1 1– DC1 and is largest when VIN1 is at its minimum value if VOUT1 and IOUT1 are constant. With a 1.5A current limit, the maximum that the output current ripple can be is ~0.75A. Table 2 lists several capacitor vendors. Buck Input Capacitor Selection Bypass the input of the LT3570 circuit with a 10μF or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors, or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3570 input and to force this switching current 3570fa 12 LT3570 APPLICATIONS INFORMATION into a tight local loop, minimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively and it must have an adequate ripple current rating. The RMS input current is: IIN2(RMS) =IOUT2 • VOUT2 ( VIN2 – VOUT2 ) VIN2 < IOUT2 2 and is largest when VIN2 = 2 • VOUT2 (50% duty cycle). Considering that the maximum load current is ~1.5A, RMS ripple current will always be less than 0.75A. The high frequency of the LT3570 reduces the energy storage requirements of the input capacitor, so that the capacitance required is often less than 10μF. The combination of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types. An alternative to a high value ceramic capacitor is a lower value along with a larger electrolytic capacitor, for example a 1μF ceramic capacitor in parallel with a low ESR tantalum capacitor. For the electrolytic capacitor, a value larger than 10μF will be required to meet the ESR and ripple current requirements. Because the input capacitor is likely to see high surge currents when the input source is applied, tantalum capacitors should be surge rated. The manufacturer may also recommend operation below the rated voltage of the capacitor. Be sure to place the 1μF ceramic as close as possible to the VIN2 and GND pins on the IC for optimal noise immunity. A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source), this tank can ring, doubling the input voltage and damaging the LT3570. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Boost Input Capacitor Selection The capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular and does not contain large squarewave currents as found in the output capacitor. Capacitors in the range of 10μF to 100μF with an ESR of 0.3Ω or less work well up to the full 1.5A switch current. Higher ESR capacitors may be acceptable at low switch currents. Input capacitor ripple current for boost converters is: IRIPPLE = 0.3 • VIN1 • VOUT1 – VIN1 f • L • VOUT1 Buck Diode Selection The catch diode (D2 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID(AVG) =IOUT1 • VIN1 – VOUT1 VIN1 The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. Table 3 lists several Schottky diodes and their manufacturers. Table 3. Schottky Diodes PART NUMBER VR (V) IAVE (A) VF AT 1A (mV) MBRM120E 20 1 530 MBRM140 40 1 550 B120 20 1 500 B130 30 1 500 30 1 420 On Semiconductor Diodes Inc. International Rectifier 10BQ030 3570fa 13 LT3570 APPLICATIONS INFORMATION Boost Diode Selection D3 A Schottky diode is recommended for use with the LT3570 inverter/boost regulator. The Microsemi UPS120 is a very good choice. Where the input to output voltage differential exceeds 20V, use the UPS140 (a 40V diode). These diodes are rated to handle an average forward current of 1A. For applications where the average forward current of the diode is less than 0.5A, use an ON Semiconductor MBR0520L diode. BOOST VIN The minimum operating voltage of an LT3570 application is limited by the undervoltage lockout (2.5V) and by the maximum duty cycle. The boost circuit also limits the minimum input voltage for proper start-up. If the input voltage ramps slowly, or the LT3570 turns on when the output is already in regulation, the boost capacitor may not be fully charged. Because the boost capacitor charges with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load current generally goes to zero SW D2 GND C2 (2a) D3 BOOST Pin Considerations The capacitor and diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a 0.1μF capacitor and fast switching diode (such as the CMDSH-3 or MMSD914LT1) will work well. Figure 2 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for full efficiency. For outputs of 3.3V and higher, the standard circuit (Figure 2a) is best. For outputs between 2.8V and 3.3V, use a small Schottky diode (such as the BAT-54). For lower output voltages, the boost diode can be tied to the input (Figure 2b). The circuit in Figure 2a is more efficient because the BOOST pin current comes from a lower voltage source. Finally, as shown in Figure 2c, the anode of the boost diode can be tied to another source that is at least 3V. For example, if you are generating 3.3V and 1.8V and the 3.3V is on whenever the 1.8V is on, the 1.8V boost diode can be connected to the 3.3V output. In any case, be sure that the maximum voltage at the BOOST pin is less than 60V and the voltage difference between the BOOST and SW2 pins is less than 25V. C3 LT3570 BOOST C5 LT3570 VIN SW D2 GND C2 (2b) D3 VEXT BOOST C5 LT3570 VIN SW D2 GND (2c) C2 3570 F02 Figure 2. Boost Pin Configurations once the circuit has started. Even without an output load current, in many cases the discharged output capacitor will present a load to the switcher that will allow it to start. Switcher Frequency Compensation The LT3570 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT3570 does not depend on the ESR of the output capacitor for stability so you are free to use ceramic capacitors to achieve low output ripple and small circuit size. To compensate the feedback loop of the LT3570, a series resistor-capacitor network should be connected from the VC pin to GND. For most applications, a capacitor in the range of 500pF to 4.7nF will suffice. A good starting value for the compensation capacitor, CC, is 1nF. The 3570fa 14 LT3570 APPLICATIONS INFORMATION compensation resistor, RC, is usually in the range of 5k to 50k. A good technique to compensate a new application is to use a 50k potentiometer in place of RC, and use a 1nF capacitor for CC. By adjusting the potentiometer while observing the transient response, the optimum value for RC can be found. Figures 3a to 3c illustrate this process for the circuit of Figure 1 with load current stepped from 100mA to 500mA for the buck converter. Figure 3a shows the transient response with RC equal to 1.6k. The phase margin is poor as evidenced by the excessive ringing in the output voltage and inductor current. In Figure 3b, the value of RC is increased to 5.75k, which results in a more damped response. Figure 3c shows the result when RC is increased further to 25k. The transient response is nicely damped and the compensation procedure is complete. The same procedure is used to compensate the boost converter. Soft-Start The Soft-start time is programmed with an external capacitor to ground on SS. An internal current source charges it with a nominal 4.5μA. The voltage on the soft-start pin is used to control the feedback voltage. The soft-start time is determined by the equation: tSS = 0.2 • CSS where CSS is in nF and tSS is in ms. In the event of a commanded shutdown, ULVO on the input or a thermal shutdown, the capacitor is discharged automatically. The soft-start will remain low and only charge back up after the fault goes away and the voltage on SS is less than approximately 100mV. IOUT 500mA/DIV IOUT 500mA/DIV VOUT VOUT 3570 F03a 200μs/DIV 200μs/DIV Figure 3a. Transient Response Shows Excessive Ringing 3570 F03b Figure 3b. Transient Response is Better IOUT 500mA/DIV VOUT 200μs/DIV 3570 F03c Figure 3c. Transient Response Well Damped 3570fa 15 LT3570 APPLICATIONS INFORMATION 25 The free-running frequency is set through a resistor from the RT pin to ground. The oscillator frequency vs RT can be seen in Figure 4. The oscillator can be synchronized with an external clock applied to the SYNC pin. When synchronizing the oscillator, the free running frequency must be set approximately 10% lower than the desired synchronized frequency. 2250 MINIMUM DUTY CYCLE (%) Oscillator 20 15 10 5 0 500 2000 750 1000 1250 1500 FREQUENCY (kHz) 1750 FREQUENCY (kHz) 1750 2000 3570 F05 Figure 5. Minimum Duty Cycle vs Frequency 1500 1250 LDO Regulator 1000 750 500 250 5 10 15 20 25 30 35 RESISTANCE (kΩ) 40 45 3570 F04 Figure 4. Frequency vs RT Resistance Buck Regulator Minimum On-Time As the input voltage is increased, the LT3570 is required to turn on for shorter periods of time. Delays associated with turning off the power switch determine the minimum on-time that can be achieved and limit the minimum duty cycle. Figure 5 shows the minimum duty cycle versus frequency for the LT3570. When the required on-time has decreased below the minimum on-time of the LT3570 the inductor current will increase, exceeding the current limit. If the current through the inductor exceeds the current limit of the LT3570, the switch is prevented from turning on for 10μs allowing the inductor current to decrease. The 10μs off-time limits the average current that can be delivered to the load. To return to normal switching frequency either the input voltage or load current must decrease. The LT3570 LDO regulator is capable of delivering up to 10mA of base drive for an external NPN transistor. For stable operation the total output capacitance can be from 1μF up to 100μF. The regulator has its own independent supply voltage which allows for the base of the NPN to be driven from a higher voltage than its collector. This allows for the NPN regulator to run more efficiently. The power Dissipated in the external NPN is equal to: PDISS = (VCOL – VOUT3) • ILOAD where VCOL is the collector voltage of the NPN. The maximum output voltage is limited to: VIN3 – 1.4V and VCOL – 0.2V or 8V The short-circuit protection of the NPN regulator is set by the max output current of the NPN_DRV pin multiplied by the beta of the NPN. Thermal Shutdown An internal temperature monitor will turn off the internal circuitry and prevent the switches from turning on when the die temperature reaches approximately 160°C. When the die temperature has dropped below this value the part 3570fa 16 LT3570 APPLICATIONS INFORMATION will be enabled again going through a soft-start cycle. Note: Overtemperature protection is intended to protect the device during momentary overload conditions. Continuous operation above the specified maximum operating junction temperature may result in device degradation or failure. to system ground at one location. Additionally, keep the SW and BOOST nodes as small as possible. This is implemented in the suggested layout of Figure 8 for the QFN package which shows the topside metal from the DC1106A demonstration board. PCB Layout Thermal Considerations For proper operation and minimum EMI, care must be taken during printed circuit board (PCB) layout. Figure 6 shows the high current paths in the step-down regulator circuit. Note that in the step-down regulator, large switched currents flow in the power switch, the catch diode and the input capacitor. To deliver the power that the LT3570 is capable of, it is imperative that a good thermal path be provided to dissipate the heat generated within the package. This can be accomplished by taking advantage of the large thermal pad on the underside of the IC. It is recommended that multiple vias in the printed circuit board be used to conduct heat away from the IC and into a copper plane with as much area as possible. Figure 7 shows the high current paths in the step-up regulator. In the boost regulator, large switched currents flow through the power switch, the switching diode, and the output capacitor. The loop formed by these large switched current components should be as small as possible. Place these components on the same side of the circuit board and connect them on that layer. Place a local, unbroken ground plane below these components and tie this ground plane Related Linear Technology Publications Application notes 19, 35, 44, 76 and 88 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1375 data sheet has a more extensive discussion of output ripple, loop compensation, and stability testing. LT3570 L2 HIGH FREQUENCY CIRCULATING PATH CIN D1 COUT LOAD 3570 F06 Figure 6. Buck High Speed Switching Path L2 LT3570 CIN D1 HIGH FREQUENCY SWITCHING PATH COUT LOAD 3750 F07 Figure 7. Boost High Speed Switching Path Figure 8. Suggested Layout 3570fa 17 LT3570 TYPICAL APPLICATIONS DSL Modem VIN 8V TO 28V C9 10μF VIN1 VIN2 VIN3 BIAS SHDN1 SHDN2 SHDN3 SHDN1 SHDN2 SHDN3 D3 BOOST SW2 SW1 C1 10μF L2 10μH D2 L1 D1 4.7μH VOUT1 8V 250mA C8 100nF R1 105k LT3570 FB1 SS1 VC1 R2 11.5k FB2 SS2 VC2 VOUT2 5V R3 118k C2 22μF C5 10nF R8 25k R4 22.1k C7 1nF R7 25k NPN_DRV C6 1nF Q1 10nF VOUT3 3.3V 500mA R5 34.0k RT FB3 GND SYNC C3 2.2μF R6 10.7k R9 20.0k 3570 TA02 “Dying Gasp” System VIN 12V C9 10μF C10 0.1μF VIN1 VIN2 VIN3 BIAS SHDN1 SHDN2 SHDN3 BOOST SW2 VOUT1 34V SW1 R1 442k FB1 SS1 VC1 R2 10.5k LT3570 FB2 SS2 VC2 L2 22μH VOUT2 3.3V 200mA R3 205k C2 22μF R8 51k C5 10nF R4 64.9k C7 1nF R7 20k C4 10nF C8 100nF D2 L1 D1 47μH C1 10μF D3 NPN_DRV C6 1nF Q1 R5 137k RT SYNC R9 44.2k FB3 GND R6 64.9k VOUT3 2.5V 200mA C3 2.2μF 3570 TA03 3570fa 18 LT3570 PACKAGE DESCRIPTION UF Package 24-Lead Plastic QFN (4mm × 4mm) (Reference LTC DWG # 05-08-1697) BOTTOM VIEW—EXPOSED PAD 4.00 ± 0.10 (4 SIDES) 0.70 ±0.05 R = 0.115 TYP 0.75 ± 0.05 PIN 1 TOP MARK (NOTE 6) PIN 1 NOTCH R = 0.20 TYP OR 0.35 × 45° CHAMFER 23 24 0.40 ± 0.10 1 2 4.50 ± 0.05 2.45 ± 0.05 3.10 ± 0.05 (4 SIDES) 2.45 ± 0.10 (4-SIDES) PACKAGE OUTLINE (UF24) QFN 0105 0.200 REF 0.25 ±0.05 0.50 BSC 0.25 ± 0.05 0.00 – 0.05 0.50 BSC NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS FE Package 20-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663) Exposed Pad Variation CB 6.40 – 6.60* (.252 – .260) 3.86 (.152) 3.86 (.152) 20 1918 17 16 15 14 13 12 11 6.60 ±0.10 2.74 (.108) 4.50 ±0.10 6.40 2.74 (.252) (.108) BSC SEE NOTE 4 0.45 ±0.05 1.05 ±0.10 0.65 BSC 1 2 3 4 5 6 7 8 9 10 RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.25 REF 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 1.20 (.047) MAX 0° – 8° 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE20 (CB) TSSOP 0204 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3570fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT3570 TYPICAL APPLICATION PDA Core VIN 4V TO 12V C9 10μF VIN1 VIN2 VIN3 BIAS SHDN1 SHDN2 SHDN3 D1 VOUT1 15V 200mA SHDN1 SHDN2 SHDN3 C8 100nF SW2 R1 191k LT3570 FB1 SS1 VC1 R2 10.7k FB2 SS2 VC2 VOUT2 3.3V 500mA R3 34k C2 22μF R8 25k C5 10nF R4 10.7k C7 1nF R7 25k NPN_DRV C6 1nF C4 10nF L2 8.2μH D2 L1 12μH SW1 C1 10μF D3 BOOST Q1 R5 13.7k SYNC RT FB3 GND R6 10.7k R9 20k VOUT3 1.8V 500mA C3 4.7μF 3570 TA04 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1767 1.5A, 1.25MHz Step-Down Switching Regulator 3V to 25V Input, VREF = 1.2V, Synchronizable up to 2MHz, MSOP Package LT1930/LT1930A 1A (ISW), 1.2MHz/2.2MHz, High Efficiency Step-Up DC/DC Converter VIN: 2.6V to 16V, VOUT(MAX) = 34V, IQ = 4.2mA/5.5mA, ISD < 1μA, ThinSOTTM Package LT1939 25V, 2.4MHz Step-Down DC/DC Converter and LDO Controller VIN: 3V to 40V, VOUT(MIN) = 0.8V, IQ = 2mA, ISD < 1μA, 3mm × 3mm DFN LT1943 Quad Output, 2.6A Buck, 2.6A Boost, 0.3A Boost, 0.4A Inverter 1.2MHz TFT DC/DC Converter VIN: 4.5V to 22V, VOUT(MAX) = 40V, IQ = 10mA, ISD < 35μA, TSSOP28E Package LT1945 Dual Output Pos/Neg 350mA (ISW), Constant Off-Time, High Efficiency Step-Up DC/DC Converter VIN: 1.2V to 15V, VOUT(MAX) = ±34V, IQ = 20μA, ISD < 1μA, 10-Pin MS Package LT3463 Dual Output Pos/Neg 250mA (ISW), Constant Off-Time, High Efficiency Step-Up DC/DC Converter with Integrated Schottkys VIN: 2.4V to 15V, VOUT(MAX) = ±40V, IQ = 40μA, ISD < 1μA, 3mm × 3mm DFN10 Package LT3467 1.1A, 1.3MHz Step Up DC/DC Converter with Integrated Soft-Start VIN: 2.4V to 16V, VOUT(MAX) = 40V, ISD < 1 μA, Low profile (1mm) SOT-23 Package LT3500 40V, 2A, 2.4MHz Step-Down DC/DC Converter and LDO Controller VIN: 3V to 40V, VOUT(MIN) = 0.8V, IQ = 2mA, ISD < 1μA, 3mm × 3mm DFN LT3507 36V, 2.5MHz Triple (2.4A, 1.5A, 1.5A) Step-Down DC/DC Converter VIN: 4V to 36V, VOUT(MIN) = 0.8V, IQ = 7mA, ISD < 1μA, 5mm × 7mm QFN38 Package and LDO Controller ThinSOT is a trademark of Linear Technology Corporation. 3570fa 20 Linear Technology Corporation LT 1108 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2008