LTC3600 15V, 1.5A Synchronous Rail-to-Rail Single Resistor Step-Down Regulator FEATURES DESCRIPTION n The LTC3600 is a high efficiency, monolithic synchronous buck regulator whose output is programmed with just one external resistor. The accurate internally generated 50μA current source on the ISET pin allows the use of a single external resistor to program an output voltage that ranges from 0V to 0.5V below VIN. The VOUT voltage feeds directly back to the error amplifier in unity gain fashion and equals the ISET voltage. The operating supply voltage range is 4V to 15V, making it suitable for dual lithium-ion battery and 5V or 12V input point-of-load power supply applications. n n n n n n n n n n n Single Resistor Programmable VOUT ±1% ISET Accuracy Tight VOUT Regulation Independent of VOUT Voltage Easy to Parallel for Higher Current and Heat Spreading Wide VOUT Range 0V to VIN – 0.5V High Efficiency: Up to 96% 1.5A Output Current Adjustable Frequency: 200kHz to 4MHz 4V to 15V VIN Range Current Mode Operation for Excellent Line and Load Transient Response <1μA Supply Current in Shutdown Available in Thermally Enhanced 12-Pin (3mm × 3mm) DFN and MSOP Packages The operating frequency is synchronizable to an external clock or programmable from 200kHz to 4MHz with an external resistor. High switching frequency allows the use of small surface mount inductors. The unique constant on-time architecture is ideal for operating at high frequency in high step-down ratio applications that also demand fast load transient response. APPLICATIONS n n n n Voltage Tracking Supplies Point-of-Load Power Supplies Portable Instruments Distributed Power Systems L, LT, LTC, LTM, Linear Technology, the Linear logo and OPTI-LOOP are registered trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 5705919, 5847554, 6580258. TYPICAL APPLICATION High Efficiency, 1MHz, 1.5A Step-Down Converter 9 LTC3600 VIN 100 BOOST 0.1μF 8 RUN 50μA 5 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW 7 VOUT 11 ISET 1 0.1μF MODE/ SYNC INTVCC RT 6 49.9k 3 10 1μF GND PGFB 13 4 ITH PGOOD 2 12 3600 TA01a VOUT 2.5V 22μF 90 VIN = 12V VOUT = 2.5V 80 DCM 1.0 0.9 0.8 POWER LOSS 70 0.7 60 50 0.6 0.5 CCM 40 0.4 30 0.3 20 0.2 CCM 10 0 0.001 POWER LOSS (W) + EFFICIENCY (%) VIN 12V Efficiency and Power Loss vs Output Current 0.1 DCM 0.01 0.1 1 LOAD CURRENT (A) 10 0 3600 TA01b 3600fb 1 LTC3600 ABSOLUTE MAXIMUM RATINGS (Notes 1, 5) VIN, SW Voltage ......................................... –0.3V to 16V SW Transient Voltage (Note 6) .......................–2V to 21V VOUT, ISET Voltage ............................................0V to VIN BOOST Voltage ............................–0.3V to VIN + INTVCC RUN Voltage................................................–0.3V to 12V INTVCC Voltage ............................................ –0.3V to 7V ITH, RT Voltage ..................................... –0.3V to INTVCC MODE/SYNC, PGFB, PGOOD Voltage .... –0.3V to INTVCC Operating Junction Temperature Range (Note 2).................................................. –40°C to 125°C MSE Package Lead Temperature (Soldering, 10 sec) ................................................ 300°C PIN CONFIGURATION TOP VIEW TOP VIEW ISET 1 12 PGOOD ITH 2 11 VOUT RT 3 PGFB 4 RUN 5 MODE/SYNC 6 13 GND ISET ITH RT PGFB RUN MODE/SYNC 10 INTVCC 9 BOOST 8 VIN 7 SW DD PACKAGE 12-LEAD (3mm w 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 55°C/W EXPOSED PAD (PIN 13) IS GND, MUST BE SOLDERED TO PCB 1 2 3 4 5 6 13 GND 12 11 10 9 8 7 PGOOD VOUT INTVCC BOOST VIN SW MSE PACKAGE 12-LEAD PLASTIC MSOP TJMAX = 125°C, θJA = 43°C/W EXPOSED PAD (PIN 13) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3600EDD#PBF LTC3600EDD#TRPBF LFXB 12-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3600IDD#PBF LTC3600IDD#TRPBF LFXB 12-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3600EMSE#PBF LTC3600EMSE#TRPBF 3600 12-Lead Plastic MSOP –40°C to 125°C LTC3600IMSE#PBF LTC3600IMSE#TRPBF 3600 12-Lead Plastic MSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3600fb 2 LTC3600 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the specified operating junction temperature range, otherwise specifications are at TA = 25°C (Note 2). SYMBOL PARAMETER VIN VIN Supply Range ISET ISET Reference Current CONDITIONS l ISET > 45μA, VIN – VSET ISET Load Regulation IOUT = 0 to 1.5A Error Amp Input Offset (Note 4) MAX V 50 50 50.5 51 μA μA 0.02 0.05 %/V mV 0.25 –3 0.05 VISET = 0, ROUT = 0 UNITS 15 340 l Error Amp Load Regulation Error Amplifier Transconductance 49.5 49.3 l ISET DROP_OUT Voltage gm(EA) TYP 4 ISET Line Regulation Minimum VOUT Voltage MIN μA 3 mV 0.1 % 10 0.63 mV 0.9 mS tON(MIN) Minimum On-Time 30 ns tOFF(MIN) Minimum Off-Time 130 ns ILIM Current Limit l 1.6 2 2.4 A Negative Current Limit –0.9 A RTOP Top Power NMOS On-Resistance 200 mΩ RBOTTOM Bottom Power NMOS On-Resistance VUVLO INTVCC Undervoltage Lockout Threshold INTVCC Rising 3.45 UVLO Hysteresis INTVCC Falling 150 Run Threshold Run Hysteresis RUN Rising RUN Falling RUN Pin Leakage RUN = 12V Internal VCC Voltage 5.5V < VIN < 15V INTVCC Load Regulation ILOAD = 0mA to 20mA OV Output Overvoltage PGOOD Upper Threshold PGFB Rising 0.620 0.645 0.680 V UV Output Undervoltage PGOOD Lower Threshold PGFB Falling 0.520 0.555 0.590 V PGOOD Hysteresis PGFB Returning 10 mV PGOOD Pull-Down Resistance 1mA Load 200 Ω PGOOD Leakage Current PGOOD = 5V MODE/SYNC Threshold MODE VIL(MAX) MODE VIH(MIN) SYNC VIH(MIN) SYNC VIL(MAX) VRUN VINTVCC VMODE/SYNC MODE/SYNC Pin Current fOSC Switching Frequency 100 l 4.8 RT = 36.1k 1.55 0.13 1.8 V V 0 2 μA 5 5.4 V % 1 μA 0.4 V V V V 4.3 2.5 0.4 9.5 l 0.92 V mV 0.3 MODE = 5V VOUT Pin Resistance to Ground mΩ 3.7 1 μA 1.06 MHz 600 kΩ V V VINOV VIN Overvoltage Lockout VIN Rising VIN Falling 17.5 16 IQ Input DC Supply Current Discontinuous Shutdown (Note 3) Mode = 0, RT = 36.1k VIN = 12V, Run = 0 700 0 1100 1.5 μA μA 3600fb 3 LTC3600 ELECTRICAL CHARACTERISTICS Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LT3600 is tested under pulsed load conditions such that TJ ≈ TA. The LT3600E is guaranteed to meet performance specifications from 0°C to 85°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LT3600I is guaranteed over the full –40°C to 125°C operating junction temperature range. Note that the maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. The junction temperature (TJ, in °C) is calculated from the ambient temperature (TA, in °C) and power dissipation (PD, in watts) according to the formula: TJ = TA + (PD • θJA), where θJA (in °C/W) is the package thermal impedance. Note 3: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency. Note 4: The LTC3600 is tested in a feedback loop that adjusts VOUT to achieve a specified error amplifier output voltage (ITH). Note 5: This IC includes overtemperature protection that is intended protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 6: Duration of voltage transient is less than 20ns for each switching cycle. 3600fb 4 LTC3600 TYPICAL PERFORMANCE CHARACTERISTICS 50.5 51 VIN =15V 50 VOUT 99 50.3 VISET 49 97 50.1 ISET (μA) 98 ISET (μA) NORMALIZED VISET AND VOUT (%) ISET Current vs VISET ISET Current vs Temperature Load Regulation 100 49.9 48 47 46 96 49.7 45 VIN = 12V VOUT = 3.3V 95 49.5 –50 –25 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 IOUT (A) 0 0 44 25 50 75 100 125 150 TEMPERATURE (°C) 3600 G02 0 2 4 6 8 10 VISET 12 3600 G01 Quiescent Current vs Temperature 50.4 3.5 ISET (μA) 49.6 49.4 ISET (VISET = 0V) ISET (VISET = 2.5V) 49.2 0 2 4 6 8 10 12 14 16 18 0.8 3.0 0.7 2.5 2.0 0.5 0.3 1.0 DCM 0.2 0.5 0.1 0 –100 0 –50 0 50 100 TEMPERATURE (°C) 3600 G04 150 0 200 4 6 8 VIN 10 12 14 16 3600 G06 Transient Response CCM Operation, External Compensation MTOP 2 3600 G05 RDS(ON) vs Temperature 20 0 –50 0.6 0.4 1.5 VIN 260 240 220 200 180 160 140 120 100 80 60 40 VRUN = 0 0.9 CCM IQ (μA) QUIESCENT CURRENT (mA) 50.2 RDS(ON) (mΩ) Shutdown IQ vs VIN 1.0 4.0 49.8 16 3600 G03 ISET Current Line Regulation 50.0 14 Transient Response CCM Operation, Internal Compensation VOUT 100mV/DIV ACCOUPLED VOUT 100mV/DIV ACCOUPLED IL 1A/DIV IL 1A/DIV MBOT 3600 G08 0 50 100 TEMPERATURE (°C) 150 VIN = 12V VOUT = 3.3V IOUT = 0A TO 1A L = 4.7μH 20μs/DIV fSW = 1MHz RITH = 27.5kΩ, CITH = 250pF MODE = INTVCC COUT = 47μF VIN = 12V VOUT = 3.3V IOUT = 0A TO 1A L = 4.7μH 20μs/DIV fSW = 1MHz ITH = INTVCC MODE = INTVCC COUT = 47μF 3600 G09 3600 G07 3600fb 5 LTC3600 TYPICAL PERFORMANCE CHARACTERISTICS Transient Response DCM, Operation, Internal Compensation Transient Response DCM, Operation, External Compensation VOUT 100mV/DIV ACCOUPLED VOUT 100mV/DIV ACCOUPLED IL 1A/DIV IL 1A/DIV 20μs/DIV ISET VOLTAGE VOUT 2V/DIV VOUT VIN = 12V VOUT = 3.3V IOUT = 0.1A TO 1A L = 4.7μH Discontinuous Conduction Mode Operation 20μs/DIV fSW =1MHz ITH = INTVCC MODE = 0 COUT = 47μF 3600 G11 Switching Frequency/Period vs RT VSW 5V/DIV 3600 G14 3600 G13 VIN = 15V VOUT = 2.5V MODE = INTVCC L = 2.2μH VIN = 15V VOUT = 2.5V MODE = 0 L = 2.2μH 3.5 6 3.0 5 2.5 4 2.0 3 tSW 1.5 PERIOD (μs) FREQUENCY (MHz) IL 1A/DIV VSW 5V/DIV 3600 G12 1ms/DIV Continuous Conduction Mode Operation IL 1A/DIV ISET VOLTAGE IINDUCTOR CURRENT 500mA/DIV 3600 G10 fSW = 1MHz RITH = 27.5kΩ, CITH = 250pF MODE = 0 COUT = 47μF VIN = 12V VOUT = 3.3V IOUT = 0.1A TO 1A L = 4.7μH Output Tracking 2 1.0 0.5 1 fSW 0 0 0 50 100 RT (kΩ) 150 200 3600 G15 Switch Leakage Current 10 INTVCC Load Regulation 5.00 VIN = 12V 4.98 6 MBOT 4 MTOP 2 INTVCC VOLTAGE (V) LEAKAGE CURRENT (μA) 8 4.96 4.94 4.92 4.90 4.88 0 –50 –30 –10 10 30 50 70 90 110 130 TEMPERATURE (°C) 3600 G16 4.86 0 10 20 30 40 50 60 70 80 90 100 INTVCC CURRENT (mA) 3600 G17 3600fb 6 LTC3600 TYPICAL PERFORMANCE CHARACTERISTICS Rising RUN Threshold vs Temperature EFFICIENCY (%) 1.55 1.50 1.45 1.40 –60 –40 –20 0 20 40 60 80 100 120 140 TEMPERATURE (°C) 100 100 90 90 80 80 70 EFFICIENCY (%) 1.60 RUN THRESHOLD (V) Efficiency vs Load Current VOUT = 1.2V, VIN = 12V Efficiency vs Load Current VOUT = 3.3V, VIN = 12V DCM 60 50 CCM 40 70 50 30 30 20 10 10 0.01 0.1 1 LOAD CURRENT (A) CCM 40 20 0 0.001 DCM 60 0 0.001 10 0.01 0.1 1 LOAD CURRENT (A) 10 3600 G20 3600 G19 3600 G18 Start-Up Waveform Start-Up Waveform Start-Up Waveform RUN 5V/DIV RUN 5V/DIV RUN 5V/DIV VOUT 2V/DIV VOUT 2V/DIV VOUT 2V/DIV IL IL 1A/DIV IL 1A/DIV 1ms/DIV 3600 G21 MODE = CCM NO PREBIASED VOUT VIN = 12V VOUT = 3.3V 3600 G22 1ms/DIV MODE = DCM NO PREBIASED VOUT VIN = 12V VOUT = 3.3V 1ms/DIV 3600 G23 MODE = CCM VOUT IS PREBIASED TO 2V VIN = 12V VOUT = 3.3V Start-Up Waveform VIN Overvoltage VIN 5V/DIV RUN 5V/DIV VOUT 1V/DIV VOUT 2V/DIV IL 1A/DIV SW 10V/DIV 1ms/DIV MODE = DCM VOUT IS PREBIASED TO 2V VIN = 12V VOUT = 3.3V 3600 G24 20ms/DIV VIN = 12V TO 18V TO 12V VOUT = 3.3V IOUT = 1A VIN RESISTOR = 30Ω MODE = CCM 3600 G25 3600fb 7 LTC3600 PIN FUNCTIONS ISET (Pin 1): Accurate 50μA Current Source. Positive input to the error amplifier. Connect an external resistor from this pin to signal GND to program the VOUT voltage. Connecting an external capacitor from ISET to ground will soft start the output voltage and reduce current inrush when turning on. VOUT can also be programmed by driving ISET directly with an external supply from 0 to VIN, in which case the external supply would be sinking the provided 50μA. Do not drive VISET above VIN or below GND. Do not float ISET. ITH (Pin 2): Error Amplifier Output and Switching Regulator Compensation Point. The internal current comparator’s trip threshold is linearly proportional to this voltage, whose normal range is from 0.3V to 2.4V. For external compensation, tie a resistor (RITH) in series with a capacitor (CITH) to signal GND. A separate 10pF high frequency filtering capacitor can also be placed from ITH to signal GND. Tying ITH to INTVCC activates internal compensation. RT (Pin 3): Switching Frequency Programming Pin. Connect an external resistor (between 100k to 10k) from RT to SGND to program the frequency from 400kHz to 4MHz. Tying the RT pin to INTVCC programs 1MHz operation. Do not float the RT pin. PGFB (Pin 4): Power Good Feedback. Place a resistor divider on VOUT to detect power good level. If PGFB is more than 0.645V, or less than 0.555V, PGOOD will be pulled down. Tie PGFB to INTVCC to disable the PGOOD function. Tying PGFB to a voltage greater than 0.64V and less than 4V will force continuous synchronous operation regardless of the MODE/SYNC state. RUN (Pin 5): Run Control Input. Enables chip operation by tying RUN above 1.5V. Tying it below 1V shuts down the switching regulator. Tying it below 0.4V shuts off the entire chip. When tying RUN to more than 12V, place a resistor (100k to 500k) between RUN and the voltage source. MODE/SYNC (Pin 6): Operation Mode Select. Tie this pin to INTVCC to force continuous synchronous operation at all output loads. Tying it to GND enables discontinuous mode operation at light loads. Applying an external clock signal to this pin will synchronize switching frequency to the external clock. During external clock synchronization, RT value should be set up such that the free running frequency is within 30% of the external clock frequency. SW (Pin 7): Switch Node Connection to External Inductor. Voltage swing of SW is from a diode voltage drop below ground to VIN. VIN (Pin 8): Input voltage. Must decouple to GND with a capacitor close to the VIN pin. BOOST (Pin 9): Boosted Floating Driver Supply for Internal Top Power MOSFET. The (+) terminal of the bootstrap capacitor connects here. This pin swings from a diode voltage drop below INTVCC up to VIN + INTVCC. INTVCC (Pin 10): Internal 5V Regulator Output. The internal power drivers and control circuits are powered from this voltage. Decouple this pin to GND with a minimum of 1μF low ESR ceramic capacitor. VOUT (Pin 11): Output Voltage Pin. Output of the LTC3600 voltage regulator. Also the negative input of the error amplifier which is driven to be the same voltage as ISET. PGOOD (Pin 12): Output Power Good with Open-Drain Logic. PGOOD is pulled to ground when the PGFB pin is more than 0.645V or less than 0.555V. PGOOD open-drain logic will be disabled if PGFB is tied to INTVCC. GND (Exposed Pad Pin 13): Ground. Return path of internal power MOSFETs. Connect the exposed pad to the negative terminal of the input capacitor and output capacitor. 3600fb 8 LTC3600 FUNCTIONAL DIAGRAM 200k 100k VON 400k GND 2pF VOUT 0.2V 100pF 4V VIN 8 ION PLL-SYNC (±30%) IION = VIN × INTVCC INTVCC 10 R 6 SWITCH LOGIC AND ANTISHOOT-THROUGH M1 SW 7 CB L1 IREV VOUT SENSE+ – – 600k TG ON Q + ICMP RT 3 20k + CVCC BOOST 9 S OSC CIN t7IN RT V tON = VON (1pF) IION MODE/SYNC VIN 5V REG VON BUFFER COUT RT ENABLE BG –3.3μA TO 6.7μA M2 PGB SENSE– GND 13 PGOOD 12 3.3μA 0μA TO 10μA VOUT 11 1 240k – – + 0.645V OV INTVCC RPG2 PGFB + 4 2 ITH 100k RPG1 50pF – UV + VIN – 50μA RUN 0.555V + EA + – 1.5V ISET 3600 BD 1 5 RSET RITH CITH RUN 3600fb 9 LTC3600 OPERATION Main Control Loop The LTC3600 is a current mode monolithic step down regulator. The accurate 50μA current source on the ISET pin allows the user to use just one external resistor to program the output voltage in a unity gain buffer fashion. In normal operation, the internal top power MOSFET is turned on for a fixed interval determined by a fixed one-shot timer OST. When the top power MOSFET turns off, the bottom power MOSFET turns on until the current comparator ICMP trips, restarting the one-shot timer and initiating the next cycle. Inductor current is determined by sensing the voltage drop across the SW and PGND nodes of the bottom power MOSFET. The voltage on the ITH pin sets the comparator threshold corresponding to inductor valley current. The error amplifier, EA, adjusts this ITH voltage by comparing the VOUT voltage with the voltage on ISET. If the load current increases, it causes a drop in the VOUT voltage relative to VISET. The ITH voltage then rises until the average inductor current matches that of the load current. At low load current, the inductor current can drop to zero and become negative. This is detected by current reversal comparator, IREV , which then shuts off the bottom power MOSFET, resulting in discontinuous operation. Both power MOSFETs will remain off with the output capacitor supplying the load current until the ITH voltage rises above the zero current level (0.8V) to initiate another cycle. Discontinuous mode operation is disabled by tying the MODE pin to INTVCC, which forces continuous synchronous operation regardless of output load. The operating frequency is determined by the value of the RT resistor, which programs the current for the internal oscillator as well as the current for the internal one-shot timer. An internal phase-locked loop servos the switching regulator on-time to track the internal oscillator to force constant switching frequency. If an external synchronization clock is present on the MODE/SYNC pin, the regulator on-time and switching frequency would then track the external clock. Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output power good feedback voltage VPGFB exits a 7.5% window around the regulation point. Continuous operation is forced during an OV condition. To defeat the PGOOD function, simply tie PGFB to INTVCC. Pulling the RUN pin to ground forces the LTC3600 into its shutdown state, turning off both power MOSFETs and all of its internal control circuitry. Bringing the RUN pin above 0.7V turns on the internal reference only, while still keeping the power MOSFETs off. Further increasing the RUN voltage above 1.5V turns on the entire chip. INTVCC Regulator An internal low drop out (LDO) regulator produces the 5V supply that powers the drivers and the internal bias circuitry. The INTVCC can supply up to 50mA RMS and must be bypassed to ground with a minimum of 1μF ceramic capacitor. Good bypassing is necessary to supply the high transient currents required by the power MOSFET gate drivers. Applications with high input voltage and high switching frequency will increase die temperature because of the higher power dissipation across the LDO. Connecting a load to the INTVCC pin is not recommended since it will further push the LDO into its RMS current rating while increasing power dissipation and die temperature. VIN Overvoltage Protection In order to protect the internal power MOSFET devices against transient voltage spikes, the LTC3600 constantly monitors the VIN pin for an overvoltage condition. When VIN rises above 16V, the regulator suspends operation by shutting off both power MOSFETs and discharges the ISET pin voltage to ground. Once VIN drops below 15V, the regulator immediately resumes normal switching operation by first charging up the ISET pin to its programmed voltage. Programming Switching Frequency Connecting a resistor from the RT pin to GND programs the switching frequency from 200kHz to 4MHz according to the following formula: Frequency (Hz) = 3.6 • 1010 (1/ F) RT (Ω) For ease of use, the RT pin can be connected directly to the INTVCC pin for 1MHz operation. Do not float the RT pin. The internal on-time phase-locked loop has a synchronization range of 30% around its programmed frequency. Therefore, during external clock synchronization, the proper 3600fb 10 LTC3600 OPERATION RT value should be selected such that the external clock frequency is within this 30% range of the RT programmed frequency. Output Voltage Tracking and Soft Start The LTC3600 allows the user to program its output voltage ramp rate by means of the ISET pin. Since VOUT servos its voltage to that of the ISET pin, placing an external capacitor CSET on the ISET pin will program the ramp-up rate of the ISET pin and thus the VOUT voltage. t − ⎡ ⎤ VOUT(t) = I ISET s R SET ⎢1− e R SET s C SET ⎥ ⎣ ⎦ from 0 to 90% VOUT t SS ≅ − R SET s CSET s Cn(1− 0.9) t SS 2.3R SET s C SET The soft-start time tSS (from 0% to 90% VOUT) is 2.3 times of time constant (RSET • CSET). The ISET pin can also be driven by an external voltage supply capable of sinking 50μA. When starting up into a pre-biased VOUT, the LTC3600 will stay in discontinuous mode and keep the power switches off until the voltage on ISET has ramped up to be equal to VOUT, at which point the switcher will begin switching and VOUT will ramp up with ISET. Output Power Good When the LTC3600’s output voltage is within the 7.5% window of the regulation point, which is reflected back as a VPGFB voltage in the range of 0.555V to 0.645V, the output voltage is in regulation and the PGOOD pin is pulled high with an external resistor connected to INTVCC or another voltage rail. Otherwise, an internal open-drain pull-down device (200Ω) will pull the PGOOD pin low. To prevent unwanted PGOOD glitches during transients or dynamic VOUT changes, the LTC3600’s PGOOD falling edge includes a blanking delay of approximately 20μs. Internal/External ITH Compensation For ease of use, the user can simplify the loop compensation by tying the ITH pin to INTVCC to enable internal compensation. This connects an internal 100k resistor in series with a 50pF capacitor to the output of the error amplifier (internal ITH compensation point). This is a trade-off for simplicity instead of OPTI-LOOP® optimization, where ITH components are external and are selected to optimize the loop transient response with minimum output capacitance. Minimum Off-Time Considerations The minimum off-time, tOFF(MIN), is the smallest amount of time that the LTC3600 is capable of turning on the bottom power MOSFET, tripping the current comparator and turning the power MOSFET back off. This time is generally about 50ns. The minimum off-time limit imposes a maximum duty cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached, due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: t ON + t OFF(MIN) VIN(MIN) = VOUT • t ON Conversely, the minimum on-time is the smallest duration of time in which the top power MOSFET can be in its “on” state. This time is typically 20ns. In continuous mode operation, the minimum on-time limit imposes a minimum duty cycle of: DMIN = fSW • tON(MIN) Where tON(MIN) is the minimum on-time. As the equation shows, reducing the operating frequency will alleviate the minimum duty cycle constraint. In the rare cases where the minimum duty cycle is surpassed, the output voltage will still remain in regulation, but the switching frequency will decrease from its programmed value. This is an acceptable result in many applications, so this constraint may not be of critical importance in most cases. High switching frequencies may be used in the design without any fear of severe consequences. As the sections on inductor and capacitor selection show, high switching frequencies allow the use of smaller board components, thus reducing the size of the application circuit. 3600fb 11 LTC3600 APPLICATIONS INFORMATION CIN and COUT Selection The input capacitance, CIN, is needed to filter the trapezoidal wave current at the drain of the top power MOSFET. To prevent large voltage transients from occurring, a low ESR input capacitor sized for the maximum RMS current should be used. The maximum RMS current is given by: ⎛V ⎞ IRMS = IOUT(MAX) ⎜ OUT ⎟ ⎝ VIN ⎠ VIN VOUT significantly higher ESR, but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have excellent low ESR characteristics and small footprints. Their relatively low value of bulk capacitance may require multiples in parallel. Using Ceramic Input and Output Capacitors −1 This formula has a maximum at VIN = 2VOUT , where IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. For low input voltage applications, sufficient bulk input capacitance is needed to minimize transient effects during output load changes. The selection of COUT is determined by the effective series resistance (ESR) that is required to minimize voltage ripple and load step transients as well as the amount of bulk capacitance that is necessary to ensure that the control loop is stable. Loop stability can be checked by viewing the load transient response. The output ripple, ΔVOUT , is determined by: ΔIL ΔVOUT ≈ + ΔIL • RESR 8 • fSW • COUT The output ripple is highest at maximum input voltage since ΔIL increases with input voltage. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Special polymer capacitors are very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the VIN input. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R and X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Since the ESR of a ceramic capacitor is so low, the input and output capacitor must instead fulfill a charge storage requirement. During a load step, the output capacitor must instantaneously supply the current to support the load until the feedback loop raises the switch current enough to support the load. The time required for the feedback loop to respond is dependent on the compensation and the output capacitor size. Typically, three to four cycles are required to respond to a load step, but only in the first cycle does the output drop linearly. The output droop, VDROOP , is usually about two to three times the linear drop of the first cycle. Thus, a good place to start with the output capacitor value is approximately: COUT 2.5 • ΔIOUT fSW • VDROOP More capacitance may be required depending on the duty cycle and load step requirements. 3600fb 12 LTC3600 APPLICATIONS INFORMATION In most applications, the input capacitor is merely required to supply high frequency bypassing, since the impedance to the supply is very low. A 22μF ceramic capacitor is usually enough for these conditions. Place this input capacitor as close to VIN pin as possible. Inductor Selection Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current: ⎛ V ⎞⎛ V ⎞ ΔIL = ⎜ OUT ⎟ ⎜1− OUT ⎟ ⎝ f SW t L ⎠ ⎝ VIN ⎠ Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors, and output voltage ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a trade-off between component size, efficiency and operating frequency. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). Note that the largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to: ⎛ ⎞⎛ VOUT ⎞ VOUT L=⎜ ⎜1− ⎟ ⎟ ⎝ f SW t ΔIL(MAX) ⎠ ⎝ VIN(MAX)⎠ Table 1. Inductor Selection Table INDUCTOR IHLP-1616BZ-11 Series IHLP-2020BZ-01 Series FDV0620 Series MPLC0525L Series HCP0703 Series RLF7030 Series WE-TPC 4828 Series INDUCTANCE (μH) 1.0 2.2 4.7 1 2.2 3.3 4.7 5.6 6.8 1 2.2 3.3 4.7 1 1.5 2.2 1 1.5 2.2 3.3 4.7 6.8 8.2 1 1.5 2.2 3.3 4.7 6.8 1.2 1.8 2.2 2.7 3.3 3.9 4.7 DCR (mΩ) 24 61 95 18.9 45.6 79.2 108 113 139 18 37 51 68 16 24 40 9 14 18 28 37 54 64 8.8 9.6 12 20 31 45 17 20 23 27 30 47 52 MAX CURRENT (A) 4.5 3.25 1.7 7 4.2 3.3 2.8 2.5 2.4 5.7 4 3.2 2.8 6.4 5.2 4.1 11 9 8 6 5.5 4.5 4 6.4 6.1 5.4 4.1 3.4 2.8 3.1 2.7 2.5 2.35 2.15 1.72 1.55 DIMENSIONS (mm) 4.3 × 4.7 4.3 × 4.7 4.3 × 4.7 5.4 × 5.7 5.4 × 5.7 5.4 × 5.7 5.4 × 5.7 5.4 × 5.7 5.4 × 5.7 6.7 × 7.4 6.7 × 7.4 6.7 × 7.4 6.7 × 7.4 6.2 × 5.4 6.2 × 5.4 6.2 × 5.4 7 × 7.3 7 × 7.3 7 × 7.3 7 × 7.3 7 × 7.3 7 × 7.3 7 × 7.3 6.9 × 7.3 6.9 × 7.3 6.9 × 7.3 6.9 × 7.3 6.9 × 7.3 6.9 × 7.3 4.8 × 4.8 4.8 × 4.8 4.8 × 4.8 4.8 × 4.8 4.8 × 4.8 4.8 × 4.8 4.8 × 4.8 HEIGHT (mm) 2 2 2 2 2 2 2 2 2 2 2 2 2 2.5 2.5 2.5 3 3 3 3 3 3 3 3.2 3.2 3.2 3.2 3.2 3.2 2.8 2.8 2.8 2.8 2.8 2.8 2.8 MANUFACTURER Vishay www.vishay.com Toko www.toko.com NEC/Tokin www.nec-tokin.com Cooper Bussmann www.cooperbussmann.com TDK www.tdk.com Würth Elektronik www.we-online.com 3600fb 13 LTC3600 APPLICATIONS INFORMATION Once the value for L is known, the type of inductor must be selected. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard”, which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly depends on the price versus size requirements and any radiated field/EMI requirements. New designs for surface mount inductors are available from Toko, Vishay, NEC/Tokin, Cooper, TDK, and Würth Elektronik. Refer to Table 1 for more details. Checking Transient Response The OPTI-LOOP compensation allows the transient response to be optimized for a wide range of loads and output capacitors. The availability of the ITH pin not only allows optimization of the control loop behavior but also provides a DC coupled and AC filtered closed loop response test point. The DC step, rise time and settling at this test point truly reflects the closed loop response. Assuming a predominantly second order system, phase margin and/ or damping factor can be estimated using the percentage of overshoot seen at this pin. The ITH external components shown in the Figure 1 circuit will provide an adequate starting point for most applications. The series R-C filter sets the dominant pole-zero loop compensation. The values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because their various types and values determine the loop feedback factor gain and phase. An output current pulse of 20% to 100% of full load current having a rise time of 1μs to 10μs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ΔILOAD • ESR, where ESR is the effective series resistance of COUT . ΔILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine phase margin. The gain of the loop increases with the RITH and the bandwidth of the loop increases with decreasing CITH. If RITH is increased by the same factor that CITH is decreased, the zero frequency will be kept the same, thereby keeping the phase the same in the most critical frequency range of the feedback loop. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Linear Technology Application Note 76. In some applications, a more severe transient can be caused by switching in loads with large (>10μF) load capacitors. The discharged load capacitors are effectively put in parallel with COUT , causing a rapid drop in VOUT . No regulator can deliver enough current to prevent this problem, if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot Swap™ controller is designed specifically for this purpose and usually incorporates current limit, short-circuit protection, and soft-start. 3600fb 14 LTC3600 APPLICATIONS INFORMATION Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: % Efficiency = 100% – (L1 + L2 + L3 + …) where L1, L2, etc., are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3600 circuits: 1) I2R losses, 2) transition losses, 3) switching losses, 4) other losses. 1. I2R losses are calculated from the DC resistances of the internal switches, RSW , the external inductor, RL, and board trace resistance, Rb. In continuous mode, the average output current flows through inductor L but is “chopped” between the internal top and bottom power MOSFETs. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (D) as follows: RSW = (RDS(ON)TOP)(D) + (RDS(ON)BOT)(1-D) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT2 (RSW + RL + Rb) 2. Transition loss arises from the brief amount of time the top power MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, internal power MOSFET gate capacitance, internal driver strength, and switching frequency. 3. The INTVCC current is the sum of the power MOSFET driver and control currents. The power MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a power MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the DC control bias current. In continuous mode, IGATECHG = fSW (QT + QB), where QT and QB are the gate charges of the internal top and bottom power MOSFETs and fSW is the switching frequency. Since INTVCC is a low dropout regulator output powered by VIN, the INTVCC current also shows up as VIN current, unless a separate voltage supply (>5V and <6V) is used to drive INTVCC. 4. Other “hidden” losses such as copper trace and internal load resistances can account for additional efficiency degradations in the overall power system. It is very important to include these system level losses in the design of a system. Other losses including diode conduction losses during dead-time and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In a majority of applications, the LTC3600 does not dissipate much heat due to its high efficiency and low thermal resistance of its exposed pad DFN or MSOP package. However, in applications where the LTC3600 is running at high ambient temperature, high VIN, high switching frequency and maximum output current load, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 160°C, both power switches will be turned off until temperature is about 15°C cooler. To avoid the LTC3600 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TRISE = PD • θJA 3600fb 15 LTC3600 APPLICATIONS INFORMATION As an example, consider the case when the LTC3600 is used in application where VIN = 12V, IOUT = 1.5A, frequency = 4MHz, VOUT = 1.8V. The equivalent power MOSFET resistance RSW is: VOUT + RDS(ON)BOT VIN 1.8 10.2 = 0.2 • + 0.1 • 12 12 RSW = RDS(ON)TOP • VIN − VOUT VIN = 0.115Ω • The VIN current during 4MHz forced continuous operation with no load is about 11mA, which includes switching and internal biasing current loss, transition loss, inductor core loss and other losses in the application. Therefore, the total power dissipated by the part is: PD = IOUT2 • RSW + VIN • IVIN (No Load) = 2.25A2 • 0.115Ω + 12V • 11mA = 0.39W The DFN 3mm × 3mm package junction-to-ambient thermal resistance, θJA, is around 55°C/W. Therefore, the junction temperature of the regulator operating in a 50°C ambient temperature is approximately: °C TJ = 0.39W t 55 + 50°C = 71°C W Remembering that the above junction temperature is obtained from an RDS(ON) at 25°C, we might recalculate the junction temperature based on a higher RDS(ON) since it increases with temperature. Redoing the calculation assuming that RSW increased 25% at 71°C yields a new junction temperature of 75°C, which is still very far away from thermal shutdown or maximum allowed junction temperature rating. Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3600. Check the following in your layout: 1. Do the capacitors CIN connect to the power VIN and power GND as close as possible? These capacitors provide the AC current to the internal power MOSFETs and their drivers. 2. Are COUT and inductor closely connected? The (–) plate of COUT returns current to PGND and the (–) plate of CIN. 3. The ground terminal of the ISET resistor must be connected to the other quiet signal GND and together connect to the power GND on only one point. The ISET resistor should be placed and routed away from noisy components and traces, such as the SW line, and its trace should be minimized. 4. Keep sensitive components away from the SW pin. The ISET resistor, RT resistor, the compensation components CITH and RITH, and the INTVCC bypass capacitor, should be routed away from the SW trace and the inductor. 5. A ground plane is preferred, but if not available, keep the signal and power grounds segregated with small signal components returning to the signal GND at one point which is then connected to the power GND at the exposed pad with minimal resistance. Flood all unused areas on all layers with copper, which reduces the temperature rise of power components. These copper areas should be connected to one of the input supplies: VIN or GND. 3600fb 16 LTC3600 APPLICATIONS INFORMATION Design Example As a design example, consider using the LTC3600 in an application with the following specifications: VIN = 10.8V to 13.2V VOUT = 1.8V IOUT(MAX) = 1.5A IOUT(MIN) = 500mA fSW = 2MHz First, RSET is selected based on: RSET = VOUT 50μA = 1.8V = 36kΩ 50μA For best accuracy, a 0.1% 36.1k resistor is selected. Because efficiency is important at both high and low load current, discontinuous mode operation will be utilized. Select from the characteristic curves the correct RT resis- tor value for 2MHz switching frequency. Based on that, RT should be 18.2k. Then calculate the inductor value for about 40% ripple current at maximum VIN: 1.8V 1.8V ⎞ ⎛ ⎞⎛ L=⎜ 1− ⎜ ⎟ ⎟ = 1.3μH ⎝ 2MHz t 0.6A ⎠ ⎝ 13.2V ⎠ The nearest standard value inductor would be 1.2μH. COUT will be selected based on the ESR that is required to satisfy the output voltage ripple requirement and the bulk capacitance needed for loop stability. For this design, one 47μF ceramic capacitor will be used. CIN should be sized for a maximum current rating of: ⎛ 1.8V ⎞ ⎛ 13.2V ⎞ IRMS = 1.5A ⎜ − 1⎟ ⎝ 13.2V ⎟⎠ ⎜⎝ 1.8V ⎠ 1/ 2 = 0.51A Decoupling the VIN pin with one 22μF ceramic capacitor is adequate for most applications. 3600fb 17 LTC3600 TYPICAL APPLICATIONS 9 LTC3600 VIN VIN 4V TO 15V BOOST 8 RUN 100k 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW 7 VOUT 3.3V 22μF VOUT 11 ISET 1 MODE/ SYNC INTVCC 6 10 66.5k 0.1μF RT GND PGFB 13 3 1μF ITH PGOOD 4 2 36.5k 12 3600 F01 56k 10pF 68pF Figure 1. 12V to 3.3V 1MHz Buck Regulator 12V to 1.2V 2MHz Buck Regulator 9 LTC3600 VIN VIN 4V TO 15V BOOST 8 RUN 100k 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 0.47μH SW 7 VOUT 1.2V 22μF VOUT 11 ISET 1 MODE/ SYNC INTVCC 6 10 RT 3 GND PGFB 13 ITH PGOOD 4 2 12 3600 TA02 100k 0.1μF 24k 100k 1μF 18.7k 100k 3600fb 18 LTC3600 TYPICAL APPLICATIONS 0.9V FPGA Power Supply VIN 4V TO 12V 9 LTC3600 VIN 9 BOOST BOOST 50μA 5 0.1μF + PWM CONTROL AND SWITCH DRIVER ERROR AMP 10μF – VIN 1.1μH SW 1.1μH 7 ISET 1 MODE/ SYNC INTVCC 6 10 RT PGFB ITH PGOOD 3 4 2 12 1μF 22μF ERROR AMP 10μF – GND 13 VOUT 11 PGOOD 12 ITH PGFB RT INTVCC MODE/ SYNC 2 4 3 10 6 VOUT (0.9V, 6A) 10pF 9 LTC3600 VIN SW 5 + PWM CONTROL AND SWITCH DRIVER 7 22μF RUN 50μA 0.1μF GND 13 VOUT 11 10pF 9 FPGA ISET 1 1μF BOOST BOOST 8 LTC3600 VIN 8 RUN 50μA 5 0.1μF + PWM CONTROL AND SWITCH DRIVER ERROR AMP 10μF – 1.1μH SW 1.1μH 7 22μF 22μF MODE/ SYNC INTVCC 1 4.53k 6 10 1μF RT PGFB ITH PGOOD 3 4 2 12 ERROR AMP 10μF – GND 13 VOUT 11 11 ISET SW 7 5 + PWM CONTROL AND SWITCH DRIVER VIN RUN 50μA 0.1μF GND 13 VOUT 0.1μF VIN 8 RUN VIN LTC3600 8 PGOOD 12 10pF ITH PGFB RT INTVCC MODE/ SYNC 2 4 3 10 6 10pF 15k 1μF ISET 1 0.1μF 3600 TA03 330pF INTVCC 5k LTC6902* SET MOD DIV V+ PH GND OUT1 OUT3 OUT2 OUT4 *EXTERNAL CLOCK FOR FREQUENCY SYNCHRONIZATION IS RECOMMENDED 3600fb 19 LTC3600 TYPICAL APPLICATIONS High Efficiency Fast Load Response Power Supply 4V TO 15V 9 LTC3600 VIN BOOST 8 RUN 50μA 100k 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW VOUT 2.52V 1.5A 7 GND 13 VOUT 22μF 11 MODE/ SYNC INTVCC ISET 1 6 10 RT PGFB ITH PGOOD 3 4 2 12 56k 1μF 402Ω 68pF IN LT3083 3.3V VCONTROL 50μA SET OUT 10μF 0.1μF 3600 TA04 24.9k 3600fb 20 LTC3600 TYPICAL APPLICATIONS LED Driver with Programmable Dimming Control 15V 9 LTC3600 VIN BOOST 8 RUN 100k 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 10μH* SW 0.1Ω 7 GND 13 VOUT 22μF IOUT 11 MODE/ SYNC INTVCC ISET 1 6 10 (LED CURRENT: 20mA TO 500mA) RT PGFB ITH PGOOD 3 4 2 12 ** 1μF 0k TO 1k 3600 TA05 * TDK LTF5022T-100M1R4-LC ** LUXEON LXML-PWN1-0100 High Efficiency 12V Audio Driver 12V 9 LTC3600 VIN BOOST 8 RUN 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – SW 1 MODE/ SYNC INTVCC 6 10 10μF 4.7μH 7 GND 13 VOUT 11 ISET 8Ω SPEAKER RT PGFB ITH PGOOD 3 4 2 12 4.7μF 10μF 3600 TA06 10nF 1μF AUDIO SIGNAL 120k 3k 220pF 3600fb 21 LTC3600 TYPICAL APPLICATIONS Programmable 1.5A Current Source 12V 9 LTC3600 VIN BOOST 8 RUN 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW 0.1Ω IOUT = 0A TO 1.5A 7 GND 13 VOUT 22μF 11 MODE/ SYNC INTVCC ISET 1 6 10 0k TO 3k RT PGFB ITH PGOOD 3 4 2 12 1μF 3600 TA07 12V Fan Speed Controller INTVCC 80.6k 12V DC FAN * VIN V+ + LT1784 16.2k – 49.9k 15V 9 LTC3600 VIN BOOST 8 RUN 100k 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW 7 GND 13 VOUT 22μF 11 MODE/ SYNC INTVCC ISET 1 6 10 RT PGFB ITH PGOOD 3 4 2 12 113k 100k 1μF 6.04k *10k NTC THERMISTOR MURATA NCP18XH103F03RB ALARM: LOGIC 1 IF TEMP > 85°C 3600 TA08 3600fb 22 LTC3600 TYPICAL APPLICATIONS 15V, 3A Dual Phase Single-Output Regulator 9 LTC3600 VIN VIN 4V TO 15V BOOST 8 RUN 100k 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW 7 22μF VOUT 11 ISET 1 MODE/ SYNC INTVCC 6 10 RT GND PGFB 3 13 4 ITH PGOOD 2 10pF 1μF VOUT = 2.5V 3A 9 LTC3600 VIN VIN 4V TO 15V 12 27k 150pF BOOST 8 RUN 100k 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW 7 22μF VOUT 11 ISET MODE/ SYNC INTVCC 6 1 10 24.9k 0.1μF RT 3 1μF 100k GND SET 4 ITH PGOOD 2 12 3600 TA09 10pF OUT1 V+ INTVCC GND PGFB 13 LTC6908-1* OUT2 MOD *EXTERNAL CLOCK FOR FREQUENCY SYNCHRONIZATION IS RECOMMENDED 3600fb 23 LTC3600 TYPICAL APPLICATIONS 1.5A Lab Supply with Programmable Current Limit 15V 9 LTC3600 VIN BOOST 8 RUN 100k 50μA 5 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW GND 13 VOUT 11 MODE/ SYNC INTVCC ISET 1 6 10 0.1Ω 7 RT PGFB ITH PGOOD 3 4 2 12 IOUT = 0A TO 1.5A 22μF 1μF 0k TO 3k 9 LTC3600 VIN BOOST 8 RUN 100k 5 50μA 0.1μF + 10μF PWM CONTROL AND SWITCH DRIVER ERROR AMP – 2.2μH SW VOUT = 0V TO 12V 7 GND 13 VOUT 22μF 11 MODE/ SYNC INTVCC ISET 6 10 1 0k TO 240k RT PGFB ITH PGOOD 3 4 2 12 3600 TA10 1μF 3600fb 24 LTC3600 PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. DD Package 12-Lead Plastic DFN (3mm w 3mm) (Reference LTC DWG # 05-08-1725 Rev A) 0.70 ±0.05 3.50 ±0.05 2.10 ±0.05 2.38 ±0.05 1.65 ±0.05 PACKAGE OUTLINE 0.25 ±0.05 0.45 BSC 2.25 REF RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 3.00 ±0.10 (4 SIDES) R = 0.115 TYP 7 0.40 ±0.10 12 2.38 ±0.10 1.65 ±0.10 PIN 1 NOTCH R = 0.20 OR 0.25 w 45° CHAMFER PIN 1 TOP MARK (SEE NOTE 6) 6 0.200 REF 1 0.23 ±0.05 0.45 BSC 0.75 ±0.05 2.25 REF (DD12) DFN 0106 REV A 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD AND TIE BARS SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3600fb 25 LTC3600 PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. MSE Package 12-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1666 Rev F) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 t 0.102 (.112 t .004) 5.23 (.206) MIN 2.845 t 0.102 (.112 t .004) 0.889 t 0.127 (.035 t .005) 6 1 1.651 t 0.102 (.065 t .004) 1.651 t 0.102 3.20 – 3.45 (.065 t .004) (.126 – .136) 12 0.65 0.42 t 0.038 (.0256) (.0165 t .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 4.039 t 0.102 (.159 t .004) (NOTE 3) 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 7 NO MEASUREMENT PURPOSE 0.406 t 0.076 (.016 t .003) REF 12 11 10 9 8 7 DETAIL “A” 0s – 6s TYP 3.00 t 0.102 (.118 t .004) (NOTE 4) 4.90 t 0.152 (.193 t .006) GAUGE PLANE 0.53 t 0.152 (.021 t .006) 1 2 3 4 5 6 DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 0.650 NOTE: (.0256) 1. DIMENSIONS IN MILLIMETER/(INCH) BSC 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.86 (.034) REF 0.1016 t 0.0508 (.004 t .002) MSOP (MSE12) 0911 REV F 3600fb 26 LTC3600 REVISION HISTORY REV DATE DESCRIPTION A 03/12 Clarified Feature and Description 1 Clarified Electrical Characteristics 3 Clarified ISET (Pin 1) Description 8 Clarified Functional Diagram 9 Modified Application Circuit 28 Changed MODE/SYNC Threshold SYNC VIH(MIN) from 1V to 2.5V 3 B 04/12 PAGE NUMBER 3600fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LTC3600 TYPICAL APPLICATION High Efficiency, Low Noise 1A Supply 9 LTC3600 VIN VIN 8V TO 15V BOOST 8 0.1μF RUN 100k 50μA 5 + 10μF 3.3μH SW PWM CONTROL AND SWITCH DRIVER ERROR AMP 7 – VTRACK = VOUT + 0.5V 22μF VOUT 11 ISET MODE/ SYNC INTVCC RT 6 1 10 3 GND PGFB 13 4 2 12 56k 1μF 10k ITH PGOOD 68pF 3600 TA11 IN LT3080 VCONTROL 10μA LT3080 SET OUT 10μF 0.1μF VOUT = 0V TO 5V 1mA TO 1A 0k to 499k RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC3601 15V, 1.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4.5V to 15V, VOUT(MIN) = 0.6V, IQ = 300μA, ISD < 1μA, 4mm × 4mm QFN-20 and MSOP-16E Packages LTC3603 15V, 2.5A (IOUT), 3MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4.5V to 15V, VOUT(MIN) = 0.6V, IQ = 75μA, ISD < 1μA, 4mm × 4mm QFN-20 and MSOP-16E Packages LTC3633 15V, Dual 3A (IOUT), 4MHz, Synchronous Step-Down DC/ DC Converter 95% Efficiency, VIN: 3.6V to 15V, VOUT(MIN) = 0.6V, IQ = 500μA, ISD < 15μA, 4mm × 5mm QFN-28 and TSSOP-28E Packages LTC3605 15V, 5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 15V, VOUT(MIN) = 0.6V, IQ = 2mA, ISD < 15μA, 4mm × 4mm QFN-24 and MSOP-16E Packages LTC3604 15V, 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 3.6V to 15V, VOUT(MIN) = 0.6V, IQ = 300μA, ISD < 14μA, 3mm × 3mm QFN-16 and MSOP-16E Packages LT3080 1.1A, Parallelable, Low Noise, Low Dropout Linear Regulator 300mV Dropout Voltage (2 Supply Operation), Low Noise = 40μVRMS VIN: 1.2V to 36V, VOUT: 0V to 35.7V, MSOP-8, 3mm × 3mm DFN Packages LT3083 Adjustable 3A Single Resistor Low Dropout Regulator 310mV Dropout Voltage, Low Noise 40μVRMS VIN: 1.2V to 23V, VOUT: 0V to 22.7V, 4mm × 4mm DFN, TSSOP-16E Packages 3600fb 28 Linear Technology Corporation LT 0412 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2011