19-6043; Rev 1; 3/12 EVALUATION KIT AVAILABLE MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications General Description Benefits and Features The MAX17498A/MAX17498B/MAX17498C devices are current-mode fixed-frequency flyback/boost converters with a minimum number of external components. They contain all the control circuitry required to design wide input voltage isolated and nonisolated power supplies. The MAX17498A has its rising/falling undervoltage lockout (UVLO) thresholds optimized for universal offline (85V AC to 265V AC) applications, while the MAX17498B/ MAX17498C support UVLO thresholds suitable to lowvoltage DC-DC applications. S Peak Current-Mode Flyback/Boost Converter The switching frequency of the MAX17498A/MAX17498C flyback converters is 250kHz, while that of the MAX17498B flyback/boost converter is 500kHz. These frequencies allow the use of tiny magnetic and filter components, resulting in compact, cost-effective power supplies. An EN/UVLO input allows the user to start the power supply precisely at the desired input voltage, while also functioning as an on/off pin. The OVI pin enables implementation of an input overvoltage-protection scheme that ensures that the converter shuts down when the DC input voltage exceeds the desired maximum value. S Programmable Soft-Start to Reduce Input Inrush Current The devices incorporate a flexible error amplifier and an accurate reference voltage (REF) to enable the end user to regulate both positive and negative outputs. Programmable current limit allows proper sizing and protection of the primary switching FET. The MAX17498B supports a maximum duty cycle of 92% and provides programmable slope compensation to allow optimization of control-loop performance. The MAX17498A/MAX17498C support a maximum duty cycle of 49%, and have fixed internal slope compensation for optimum control-loop performance. The devices provide an open-drain PGOOD pin that serves as a power-good indicator and enters the high-impedance state to indicate that the flyback /boost converter is in regulation. An SS pin allows programmable soft-start time for the flyback/boost converter. Hiccup-mode overcurrent protection and thermal shutdown are provided to minimize dissipation under overcurrent and overtemperature fault conditions. The devices are available in a space-saving, 16-pin (3mm x 3mm) TQFN package with 0.5mm lead spacing. Ordering Information appears at end of data sheet. Typical Application Circuits appears at end of data sheet. S Current-Mode Control Provides Excellent Transient Response S Fixed Switching Frequency 250kHz (MAX17498A/MAX17498C) 500kHz (MAX17498B) S Flexible Error Amplifier to Regulate Both Positive and Negative Outputs S Programmable Voltage or Current Soft-Start S Power-Good Signal (PGOOD) S Reduced Power Dissipation Under Fault Hiccup-Mode Overcurrent Protection Thermal Shutdown with Hysteresis S Robust Protection Features Flyback/Boost Programmable Current Limit Input Overvoltage Protection S Optimized Loop Performance Programmable Slope Compensation for Flyback /Boost Maximizes Obtainable Phase Margin S High Efficiency 175mI, 65V Rated n-Channel MOSFET Offers Typical Efficiency Greater Than 80% No Current-Sense Resistor S Optional Spread Spectrum S Space-Saving, 16-Pin (3mm x 3mm) TQFN Package Applications Front-End AC-DC Power Supplies for Industrial Applications (Isolated and Nonisolated) Telecom Power Supplies Wide Input Range DC Input Flyback /Boost Industrial Power Supplies For related parts and recommended products to use with this part, refer to www.maxim-ic.com/MAX17498A.related. ���������������������������������������������������������������� Maxim Integrated Products 1 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications ABSOLUTE MAXIMUM RATINGS IN to SGND.............................................................-0.3V to +40V EN/UVLO to SGND.......................................... -0.3V to IN + 0.3V OVI to SGND............................................... -0.3V to VCC + 0.3V VCC to SGND...........................................................-0.3V to +6V SS, LIM, EA-, EA+, COMP, SLOPE, REF to SGND.........................................-0.3V to (VCC + 0.3V) LX to SGND............................................................-0.3V to +70V PGOOD to SGND.....................................................-0.3V to +6V PGND to SGND.....................................................-0.3V to +0.3V Continuous Power Dissipation (Single-Layer Board) TQFN (derate 20.8mW/°C above +70°C)..................1700mW Operating Temperature Range......................... -40°C to +125°C Storage Temperature Range............................. -65°C to +160°C Junction Temperature (continuous).................................+150°C Lead Temperature (soldering, 10s).................................+300°C Soldering Temperature (reflow).......................................+260°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (VIN = +15V, VEN/UVLO = +2V, COMP = open, CIN = 1µF, CVCC = 1µF, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER CONDITIONS MIN TYP MAX UNITS INPUT SUPPLY (VIN) IN Voltage Range (VIN) IN Supply Startup Current Under UVLO MAX17498A 4.5 29 MAX17498B/MAX17498C 4.5 36 V IINSTARTUP, VIN < UVLO or EN/UVLO = SGND 22 36 Switching, fSW = 250kHz (MAX17498A/MAX17498C) 1.8 3 2 3.25 19 20.5 22 3.85 4.15 4.4 3.65 3.95 4.25 V EN/UVLO = SGND, IIN = 1mA (MAX17498A) (Note 2) 31 33.5 36 V VCC Output Voltage Range 6V < VIN < 29V, 0mA < IVCC < 50mA 4.8 5 5.2 V VCC Dropout Voltage VIN = 4.5V, IVCC = 20mA 160 300 VCC Current Limit VCC = 0V, VIN = 6V IN Supply Current (IIN) IN Boostrap UVLO Rising Threshold Switching, fSW = 500kHz (MAX17498B) MAX17498A MAX17498B/MAX17498C IN Bootstrap UVLO Falling Threshold IN Clamp Voltage µA mA V LINEAR REGULATOR (VCC) 50 100 Rising 1.18 1.23 1.28 Falling 1.11 1.17 1.21 0V < VEN/UVLO < 1.5V, TA = +25NC -100 0 +100 mV mA ENABLE (EN/UVLO) EN/UVLO Threshold EN/UVLO Input Leakage Current V nA ���������������������������������������������������������������� Maxim Integrated Products 2 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications ELECTRICAL CHARACTERISTICS (continued) (VIN = +15V, VEN/UVLO = +2V, COMP = open, CIN = 1µF, CVCC = 1µF, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER CONDITIONS MIN TYP MAX Rising 1.18 1.23 1.28 Falling 1.11 1.17 1.21 0V < VOVI < 1.5V, TA = +25NC -100 0 +100 235 250 265 470 500 530 47.5 48.75 50 90 92 94 UNITS OVERVOLTAGE PROTECTION (OVI) OVI Threshold OVI Masking Delay OVI Input Leakage Current 2 SWITCHING FREQUENCY AND MAXIMUM DUTY CYCLE (fSW and DMAX) MAX17498A/MAX17498C Switching Frequency MAX17498B Maximum Duty Cycle Minimum Controllable On Time MAX17498A/MAX17498C MAX17498B tONMIN V µs 110 nA kHz % ns SOFT-START (SS) SS Set-Point Voltage SS Pullup Current VSS = 400mV SS Peak Current-Limit-Enable Threshold 1.2 1.22 1.24 V 9 10 11 µA 1.11 1.17 1.21 V -100 +100 nA -100 +100 nA ERROR AMPLIFIER (EA+, EA-, and COMP) EA+ Input Bias Current EA- Input Bias Current VEA+ = 1.5V, TA = +25NC VEA- = 1.5V, TA = +25NC Error-Amplifier Open-Loop Voltage Gain 90 dB Error-Amplifier Transconductance VCOMP = 2V, VLIM = 1V 1.5 1.8 2.1 mS Error-Amplifier Source Current VCOMP = 2V, EA- < EA+ 80 120 210 µA Error-Amplifier Sink Current VCOMP = 2V, EA- > EA+ 80 120 210 µA 0.45 0.5 0.55 I 175 380 mI A Current-Sense Transresistance INTERNAL SWITCH DMOS Switch On-Resistance (RDSON) ILX = 200mA DMOS Peak Current Limit LIM = 100K 1.62 1.9 2.23 DMOS Runaway Current Limit LIM = 100K 1.9 2.3 2.6 A LX Leakage Current VLX = 65V, TA = +25NC 0.1 1 µA 9 10 11 µA Peak Switch Current Limit with LIM Open 0.39 0.45 0.54 A Runaway Switch Current Limit with LIM Open 0.39 0.5 0.6 A CURRENT LIMIT (LIM) LIM Reference Current ���������������������������������������������������������������� Maxim Integrated Products 3 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications ELECTRICAL CHARACTERISTICS (continued) (VIN = +15V, VEN/UVLO = +2V, COMP = open, CIN = 1µF, CVCC = 1µF, TA = TJ = -40°C to +125°C, unless otherwise noted. Typical values are at TA = +25°C.) (Note 1) PARAMETER CONDITIONS MIN TYP MAX UNITS Number of Peak Current-Limit Hits Before Hiccup Timeout 8 # Number of Runaway CurrentLimit Hits Before Hiccup Timeout 1 # Overcurrent Hiccup Timeout 32 ms SLOPE COMPENSATION (SLOPE) SLOPE Pullup Current 9 SLOPE-Compensation Resistor Range MAX17498B Default SLOPE-Compensation Ramp SLOPE = open 10 30 11 µA 150 kI 60 mV/µs POWER-GOOD SIGNAL (PGOOD) PGOOD Output-Leakage Current (Off State) VPGOOD = 5V, TA = +25NC -1 +1 µA PGOOD Output Voltage (On State) IPGOOD = 10mA 0 0.4 V PGOOD Higher Threshold EA- rising 93.5 95 96.5 % PGOOD Lower Threshold EA- falling 90.5 92 93.5 % PGOOD Delay After EA- Reaches 95% Regulation 4 ms +160 NC 20 NC THERMAL SHUTDOWN Thermal-Shutdown Threshold Thermal-Shutdown Hysteresis Temperature rising Note 1: All devices are 100% production tested at TA = +25NC. Limits over temperature are guaranteed by design. Note 2: The MAX17498A is intended for use in universal input power supplies. The internal clamp circuit at IN is used to prevent the bootstrap capacitor from changing to a voltage beyond the absolute maximum rating of the device when EN/UVLO is low (shutdown mode). Externally limit the maximum current to IN (hence to clamp) to 2mA (max) when EN/UVLO is low. ���������������������������������������������������������������� Maxim Integrated Products 4 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Typical Operating Characteristics (VIN = +15V, VEN/UVLO = +2V, COMP = open, CIN = 1µF, CVCC = 1µF, TA = TJ = -40°C to +125°C, unless otherwise noted.) BOOTSTRAP UVLO WAKE-UP LEVEL vs. TEMPERATURE (MAX17498A) IN UVLO WAKE-UP LEVEL vs. TEMPERATURE (MAX17498B/MAX17498C) 20.22 20.20 20.18 20.16 4.10 4.05 4.00 3.95 3.90 20.14 -40 -20 0 20 40 60 80 -40 -20 100 120 0 20 40 60 80 TEMPERATURE (°C) TEMPERATURE (°C) IN UVLO SHUTDOWN LEVEL vs. TEMPERATURE EN/UVLO RISING LEVEL vs. TEMPERATURE 4.005 4.000 3.995 3.990 3.985 MAX17498 toc04 4.010 100 120 1.235 EN/UVLO RISING LEVEL (V) MAX17498 toc03 4.015 IN UVLO SHUTDOWN LEVEL (V) MAX17498 toc02 20.24 4.15 IN UVLO WAKE-UP LEVEL (V) MAX17498 toc01 BOOTSTRAP UVLO WAKE-UP LEVEL (V) 20.26 1.230 1.225 1.220 1.215 3.980 1.210 3.975 0 20 40 60 80 0 20 40 60 80 TEMPERATURE (°C) TEMPERATURE (°C) EN/UVLO FALLING LEVEL vs. TEMPERATURE OVI RISING LEVEL vs. TEMPERATURE OVI RISING LEVEL (V) 1.165 1.160 1.155 1.150 100 120 1.225 MAX17498 toc05 1.170 EN/UVLO FALLING LEVEL (V) -40 -20 100 120 MAX17498 toc06 -40 -20 1.220 1.215 1.145 1.140 -40 -20 0 20 40 60 TEMPERATURE (°C) 80 100 120 1.210 -40 -20 0 20 40 60 80 100 120 TEMPERATURE (°C) ����������������������������������������������������������������� Maxim Integrated Products 5 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Typical Operating Characteristics (continued) (VIN = +15V, VEN/UVLO = +2V, COMP = open, CIN = 1µF, CVCC = 1µF, TA = TJ = -40°C to +125°C, unless otherwise noted.) OVI FALLING LEVEL vs. TEMPERATURE IN CURRENT UNDER UVLO vs. TEMPERATURE 1.150 1.145 1.140 1.135 28 26 24 22 20 -40 -20 0 20 40 60 80 100 120 -40 -20 0 20 40 60 80 100 120 TEMPERATURE (°C) TEMPERATURE (°C) IN CURRENT DURING SWITCHING vs. TEMPERATURE LX AND PRIMARY CURRENT WAVEFORM MAX17498 toc10 MAX17498 toc09 2.6 IN CURRENT DURING SWITCHING (mA) MAX17498 toc08 IN CURRENT UNDER UVLO (µA) 1.155 OVI FALLING LEVEL (V) 30 MAX17498 toc07 1.160 2.4 VLX 20V/div 2.2 2.0 IPRI 0.5A/div 1.8 1.6 1.4 -40 -20 0 20 40 60 80 100 120 1µs/div TEMPERATURE (°C) EN STARTUP WAVEFORM EN SHUTDOWN WAVEFORM MAX17498 toc11 MAX17498 toc12 EN/UVLO 5V/div EN/UVLO 5V/div VOUT 5V/div VOUT 5V/div VCOMP 1V/div VCOMP 1V/div 400µs/div 400µs/div ����������������������������������������������������������������� Maxim Integrated Products 6 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Typical Operating Characteristics (continued) (VIN = +15V, VEN/UVLO = +2V, COMP = open, CIN = 1µF, CVCC = 1µF, TA = TJ = -40°C to +125°C, unless otherwise noted.) PEAK CURRENT LIMIT (ILIM) vs. RLIM AT ROOM TEMPERATURE PEAK CURRENT LIMIT AT RLIM = 100kI vs. TEMPERATURE 1200 1000 800 600 400 200 0 0 10 20 30 40 50 60 70 MAX17498 toc14 1400 PEAK CURRENT LIMIT AT RLIM (A) 1600 1.99 1.98 1.97 1.96 1.95 1.94 80 -40 -20 0 20 40 60 80 100 120 RLIM AT ROOM TEMPERATURE (kI) TEMPERATURE AT GIVEN RLIM (°C) TRANSIENT RESPONSE FOR 50% LOAD STEP ON FLYBACK OUTPUT (5V) SHORT-CIRCUIT PROTECTION MAX17498 toc15 MAX17498 toc16 VLX 50V/div ILOAD 500mA/div VOUT 500mV/div VOUT 200mV/div 2ms/div 10ms/div BODE PLOT - (5V OUTPUT AT 24V INPUT) EFFICIENCY GRAPH AT 24V INPUT (FLYBACK REGULATOR) MAX17498 toc17 100 90 80 PHASE 36°/div VIN = 24V MAX17498 toc18 IPRI 2A/div EFFICIENCY (%) PEAK CURRENT LIMIT (mA) 2.00 MAX17498 toc13 1800 70 60 50 40 30 GAIN 10dB/div BW = 8.3kHz PM = 63° 20 10 0 LOG (F) 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 LOAD CURRENT (A) ����������������������������������������������������������������� Maxim Integrated Products 7 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications REF N.C. EA+ TOP VIEW N.C. Pin Configuration 12 11 10 9 PGOOD 13 PGND 14 MAX17498A MAX17498B MAX17498C LX 15 EP (SGND) 2 3 4 LIM EN/UVLO 1 OVI + VCC IN 16 8 SS 7 COMP 6 EA- 5 SLOPE TQFN-EP Pin Description PIN NAME FUNCTION 1 EN/UVLO Enable/Undervoltage-Lockout Pin. Drive to > 1.23V to start the devices. To externally program the UVLO threshold of the input supply, connect a resistor-divider between input supply EN/UVLO and SGND. 2 VCC Linear Regulator Output. Connect input bypass capacitor of at least 1µF from VCC to SGND as close as possible to the IC. 3 OVI Overvoltage Comparator Input. Connect a resistor-divider between the input supply (OVI) and SGND to set the input overvoltage threshold. 4 LIM Current-Limit Setting Pin. Connect a resistor between LIM and SGND to set the peak-current limit for nonisolated flyback converter. Peak-current limit defaults to 500mA if unconnected. 5 SLOPE Slope Compensation Input Pin. Connect a resistor between SLOPE and SGND to set slopecompensation ramp. Connect to VCC for minimum slope compensation. See the Programming Slope Compensation (SLOPE) section. 6 EA- Inverting Input of the Flexible Error Amplifier. Connect to mid-point of resistor-divider from the positive terminal output to SGND. 7 COMP Flexible Error-Amplifier Output. Connect the frequency-compensation network between COMP and SGND. 8 SS 9 EA+ Soft-Start Pin. Connect a capacitor from SS to SGND to set the soft-start time interval. Noninverting Input of the Flexible Error Amplifier. Connect to SS to use 1.22V as the reference. 10, 12 N.C. No Connection 11 REF Internal 1.22V Reference Output Pin. Connect a 100pF capacitor from REF to SGND. ���������������������������������������������������������������� Maxim Integrated Products 8 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Pin Description (continued) PIN NAME FUNCTION 13 PGOOD 14 PGND 15 LX External Transformer/Inductor Connection for the Converter 16 IN Internal Linear Regulator Input. Connect IN to the input-voltage source. Bypass IN to PGND with a 1µF (min) ceramic capacitor. — EP (SGND) Exposed Pad. Internally connected to SGND. Connect EP to a large copper plane at SGND potential to provide adequate thermal dissipation. Connect EP (SGND) to PGND at a single point. Open-Drain Output. PGOOD goes high when EA- is within 5% of the set point. PGOOD pulls low when EA- falls below 92% of its set-point value. Power Ground for Converter Detailed Description The MAX17498A offers a bootstrap UVLO wakeup level of 20V with a wide hysteresis of 15V (min) optimized for implementing an isolated and nonisolated universal (85V AC to 265V AC) offline single-switch flyback converter or telecom (36V to 72V) power supplies. The MAX17498B/MAX17498C offer a UVLO wakeup level of 4.4V and are well suited for low-voltage DC-DC flyback/ boost power supplies. An internal reference (1.22V) can be used to regulate the output down to 1.23V in nonisolated flyback and boost applications. Additional semi-regulated outputs, if needed, can be generated by using additional secondary windings on the flyback converter transformer. A flexible error amplifier and REF allow the end-user selection between regulating positive and negative outputs. The devices utilize peak current-mode control and external compensation for optimizing the loop performance for various inductors and capacitors. The devices include a cycle-by-cycle peak current limit and eight consecutive occurrences of current-limit event trigger hiccup mode, that protect external components by halting switching for a period of time (32ms). The devices also include voltage soft-start for nonisolated designs and current soft-start for isolated designs to allow monotonic rise of the output voltage. The voltage or current soft-start can be selected using the SLOPE pin. See the Block Diagram for more information. Input Voltage Range The MAX17498A has different rising and falling UVLO thresholds on the IN pin than those of the MAX17498B/ MAX17498C. The thresholds for the MAX17498A are optimized for implementing power-supply startup schemes typically used for offline AC-DC power supplies. The MAX17498A is therefore well suited for operation from the rectified DC bus in AC-DC power-supply applications typically encountered in front-end industrial power-supply applications. As such, the MAX17498A has no limitation on the maximum input voltage as long as the external components are rated suitably and the maximum operating voltages of the MAX17498A are respected. The MAX17498A can successfully be used in universal input-rectified (85V to 265V AC) bus applications, rectified 3-phase DC bus applications, and telecom (36V to 72V DC) applications. The MAX17498B/MAX17498C are intended for implementing a flyback (isolated and nonisolated) and boost converter with an on-board 65V rated n-channel MOSFET. The IN pin of the MAX17498B/MAX17498C has a maximum operating voltage of 36V. The MAX17498B/ MAX17498C implement rising and falling thresholds on the IN pin that assume power-supply startup schemes, typical of lower voltage DC-DC applications, down to an input voltage of 4.5V DC. Therefore, flyback converters with a 4.5V to 36V supply voltage range can be implemented with the MAX17498B/MAX17498C. Internal Linear Regulator (VCC) The internal functions and driver circuits are designed to operate from a 5V Q5% power-supply voltage. The devices have an internal linear regulator that is powered from the IN pin and generates a 5V power rail. The output of the linear regulator is connected to the VCC pin and should be decoupled with a 2.2µF capacitor to ground for stable operation. The VCC converter output supplies the operating current for the devices. The maximum operating voltage of the IN pin is 29V for the MAX17498A and 36V for the MAX17498B/MAX17498C. ���������������������������������������������������������������� Maxim Integrated Products 9 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Configuring the Power Stage (LX) Maximum Duty Cycle The devices use an internal n-channel MOSFET to implement internal current sensing for current-mode control and overcurrent protection of the flyback/boost converter. To facilitate this, the drain of the internal nMOSFET is connected to the source of the external MOSFET in the MAX17498A high-input-voltage applications. The gate of the external MOSFET is connected to the IN pin. Ensure by design that the IN pin voltage does not exceed the maximum operating gate-voltage rating of the external MOSFET. The external MOSFET gate-source voltage is controlled by the switching action of the internal nMOSFET, while also sensing the source current of the external MOSFET. In the MAX17498B/MAX17498C-based applications, the LX pin is directly connected to either the flyback transformer primary winding or to the boostconverter inductor. IN The MAX17498A/MAX17498C operate at a maximum duty cycle of 49%. The MAX17498B offers a maximum duty cycle of 92% to implement both flyback and boost converters involving large input-to-output voltage ratios in DC-DC applications. Power-Good Signal (PGOOD) The devices include a PGOOD signal that serves as a power-good signal to the system. PGOOD is an open-drain signal and requires a pullup resistor to the preferred supply voltage. The PGOOD signal monitors EA- and pulls high when EA- is 95% (typ) of its regulation value (1.22V). For isolated power supplies, PGOOD cannot serve as a power-good signal. REF CHIPEN VCC HICCUP 5V, 50mA LDO SS 33V CLAMP (MAX17498A ONLY) 10µA MAX17498A MAX17498B MAX17498C POK BG SSDONEF 1.17V EN/UVLO CHIPEN VSLOPE OSC 1.23V OVI RUNAWAY 1.23V 8 PEAK OR 1 RUNAWAY PEAK 10µA LIMINT LIM VSUM 1.23V SLOPE LX CLK VCS 10µA VCS VSUM 250mV DECODER CONTROL LOGIC AND DRIVER PGND PGOOD PWM PGOOD COMP COMP EA- FIXED SLOPE BLOCK VARIABLE SLOPE VOLTAGE SS CURRENT SS SSDONE EA+ EA- CHIPEN Figure 1. MAX17498A/MAX17498B/MAX17498C Block Diagram ��������������������������������������������������������������� Maxim Integrated Products 10 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Soft-Start The devices implement soft-start operation for the flyback /boost converter. A capacitor connected to the SS pin programs the soft-start period for the flyback/ boost converter. The soft-start feature reduces the input inrush current. These devices allow the end user to select between voltage soft-start usually preferred in nonisolated applications and current soft-start, which is useful in isolated applications to get a monotonic rise in the output voltage. See the Programming Soft-Start of the Flyback/ Boost Converter (SS) section. Spread-Spectrum Factory Option and input overvoltage-protection voltage (VOVI), the resistor values for the divider can be calculated as follows, assuming a 24.9kI resistor for ROVI: V R EN= R OVI × OVI − 1 kΩ VSTART where ROVI is in kI while VSTART and VOVI are in volts. V = R OVI + R EN × START − 1 kΩ R SUM 1.23 For EMI-sensitive applications, a spread-spectrumenabled version of the device can be requested from the factory. The frequency-dithering feature modulates the switching frequency by Q10% at a rate of 4kHz. This spread-spectrum-modulation technique spreads the energy of switching-frequency harmonics over a wider band while reducing their peaks, helping to meet stringent EMI goals. where REN and ROVI are in kI. In universal AC input applications, RSUM might need to be implemented as equal resistors in series (RDC1, RDC2, RDC3) so that voltage across each resistor is limited to its maximum operation voltage. Applications Information For low-voltage DC-DC applications based on the MAX17498B/MAX17498C, a single resistor can be used in the place of RSUM, as the voltage across it is approximately 40V. Startup Voltage and Input OvervoltageProtection Setting (EN/UVLO, OVI) The devices’ EN /UVLO pin serves as an enable /disable input, as well as an accurate programmable input UVLO pin. The devices do not commence startup operation unless the EN/UVLO pin voltage exceeds 1.23V (typ). The devices turn off if the EN/UVLO pin voltage falls below 1.17V (typ). A resistor-divider from the input DC bus to ground can be used to divide down and apply a fraction of the input DC voltage (VDC) to the EN/UVLO pin. The values of the resistor-divider can be selected so that the EN/UVLO pin voltage exceeds the 1.23V (typ) turn-on threshold at the desired input DC bus voltage. The same resistor-divider can be modified with an additional resistor (ROVI) to implement input overvoltage protection in addition to the EN/UVLO functionality as shown in Figure 2. When voltage at the OVI pin exceeds 1.23V (typ), the devices stop switching and resume switching operations only if voltage at the OVI pin falls below 1.17V (typ). For given values of startup DC input voltage (VSTART), R= DC1 R= DC1 R= DC1 R SUM kΩ 3 VDC RDC1 RSUM RDC2 RDC3 EN/UVLO REN OVI MAX17498A MAX17498B MAX17498C ROVI Figure 2. Programming EN/UVLO and OVI ��������������������������������������������������������������� Maxim Integrated Products 11 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Startup Operation The MAX17498A is optimized for implementing an offline single-switch flyback converter and has a 20V IN UVLO wake-up level with hysteresis of 15V (min). In offline applications, a simple cost-effective RC startup circuit is used. When the input DC voltage is applied, the startup resistor (RSTART) charges the startup capacitor (CSTART), causing the voltage at the IN pin to increase towards the wake-up IN UVLO threshold (20V typ). During this time, the MAX17498A draws a low startup current of 20µA (typ) through RSTART. When the voltage at IN reaches the wake-up IN UVLO threshold, the MAX17498A commences switching operations and drives the internal n-channel MOSFET whose drain is connected to the LX pin. In this condition, the MAX17998A draws 1.8mA current from CSTART, in addition to the current required to switch the gate of the external nMOSFET. Since this current cannot be supported by the current through RSTART, the voltage on CSTART starts to drop. When suitably configured, as shown in Figure 10, the external nMOSFET is switched by the LX pin and the flyback/forward converter generates an output voltage (VOUT) bootstrapped to the IN pin through the diode (D2). If VOUT exceeds the sum of 5V and the drop across D2 before the voltage on CSTART falls below 5V, then the IN voltage is sustained by VOUT, allowing the MAX17498A to continue operating with energy from VOUT. The large hysteresis (15V typ) of the MAX17498A allows for a small startup capacitor (CSTART). The low startup curent (20µA typ) allows the use of a large start resistor (RSTART), thus reducing power dissipation at higher DC bus voltages. Figure 3 shows the typical RC startup scheme for the MAX17498A. RSTART might need to be implemented as equal, multiple resistors in series (RIN1, RIN2, and RIN3) to share the VDC VOUT applied high DC voltage in offline applications so that the voltage across each resistor is limited to the maximum continuous operating-voltage rating. RSTART and CSTART can be calculated as: Q GATE × fsw t SS C START = µF IIN + × 10 6 10 where IIN is the supply current drawn at the IN pin in mA, QGATE is the gate charge of the external nMOSFET used in nC, fSW is the switching frequency of the converter in Hz, and tSS is the soft-start time programmed for the flyback/forward converter in ms. See the Programming Soft-Start of the Flyback /Boost Converter (SS) section. R START = (VSTART − 10) × 50 kΩ 1 + C START where CSTART is the startup capacitor in µF. For designs that cannot accept power dissipation in the startup resistors at high DC input voltages in offline applications, the startup circuit can be set up with a current source instead of a startup resistor as shown in Figure 4. VDC RIN1 VDC VOUT D1 RSTART RIN2 D1 COUT VDC RIN3 COUT RIN1 RSTART IN RIN2 VOUT VOUT RIN3 MAX17498A IN D2 CSTART LX LDO VCC MAX17498A RISRC D2 IN CSTART LX LDO VCC CVCC CVCC Figure 3. MAX17498A RC-Based Startup Circuit Figure 4. MAX17498A Current Source-Based Startup Circuit ��������������������������������������������������������������� Maxim Integrated Products 12 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications VDC Resistors RSUM and RISRC can be calculated as: VOUT VSTART MΩ 10 VBEQ1 = RISRC MΩ 70 = R SUM D1 IN IN VCC LDO CIN CVCC MAX17498B MAX17498C LX COUT Np Ns Figure 5. MAX17498B/MAX17498C Typical Startup Circuit with IN Connected Directly to DC Input VDC RZ = 9 × (VINMIN − 6.3) kΩ where VINMIN is the minimum input DC voltage. VOUT D2 RZ Programming Soft-Start of the Flyback/Boost Converter (SS) D1 Q1 ZD1 6.3V NB IN IN CIN MAX17498B MAX17498C LDO COUT LX Np The IN UVLO wakeup threshold of the MAX17498B/ MAX17498C is set to 3.9V (typ) with a 200mV hysteresis, optimized for low-voltage DC-DC applications down to 4.5V. For applications where the input DC voltage is low enough (e.g., 4.5V to 5.5V DC) that the power loss incurred to supply the operating current of the MAX17498B/MAX17498C can be tolerated, the IN pin is directly connected to the DC input, as shown in Figure 5. In the case of higher DC input voltages (e.g., 16V to 32V DC), a startup circuit, such as that shown in Figure 6, can be used to minimize power dissipation in the startup circuit. In this startup scheme, the transistor (Q1) supplies the switching current until a bias winding NB comes up. The resistor (RZ) can be calculated as: Ns VCC CVCC Figure 6. MAX17498B/MAX17498C Typical Startup Circuit with Bias Winding to Turn Off Q1 and Reduce Power Dissipation The startup capacitor (CSTART) can be calculated as: Q GATE × fSW t SS C START = µF IIN + × 10 6 10 where IIN is the supply current drawn at the IN pin in mA, QGATE is the gate charge of the external MOSFET used in nC, fSW is the switching frequency of the converter in kHz, and tSS is the soft-start time programmed for the flyback converter in ms. The soft-start period in the voltage soft-start scheme of the devices can be programmed by selecting the value of the capacitor connected from the SS pin to GND. The capacitor CSS can be calculated as: C= SS 8.13 × t SS nF where tSS is expressed in ms. The soft-start period in the current soft-start scheme depends on the load at the output and the soft-start capacitor. Programming Output Voltage The devices incorporate a flexible error amplifier that allows regulating to both the positive and negative outputs. The positive output voltage of the converter can be programmed by selecting the correct values for the resistor-divider connected from VOUT, the flyback /boost output to ground, with the midpoint of the divider connected to the EA- pin (Figure 7). With RB selected in the range of 20kI to 50kI, RU can be calculated as: V RU = RB × OUT − 1 kΩ 1.22 where RB is in kI. ��������������������������������������������������������������� Maxim Integrated Products 13 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications The negative output voltage of the converter can be programmed by selecting the correct values for the resistor-divider connected from VOUT, the flyback /boost output to REF with the midpoint of the divider connected to the EA+ pin (Figure 8). With R1 selected in the range of 20kI to 50kI, R2 can be calculated as: VOUT RU EARB V R2 = R1× OUT kΩ 1.22 where R1 is in kI. MAX17498A MAX17498B MAX17498C Figure 7. Programming the Positive Output Voltage Current-Limit Programming (LIM) The devices include a robust overcurrent-protection scheme that protects the device under overload and short-circuit conditions. For the flyback/boost converter, the devices include a cycle-by-cycle peak current limit that turns off the driver whenever the current into the LX pin exceeds an internal limit that is programmed by the resistor connected from the LIM pin to GND. The devices include a runaway current limit that protects the device under high-input-voltage shortcircuit conditions when there is insufficient output voltage available to restore the inductor current built up during the on period of the flyback/boost converter. Either eight consecutive occurrences of the peak currentlimit event or one occurrence of the runaway current limit trigger a hiccup mode that protects the converter by immediately suspending switching for a period of time (tRSTART). This allows the overload current to decay due to power loss in the converter resistances, load, and the output diode of the flyback/boost converter before soft-start is attempted again. The resistor at the LIM pin for a desired current limit (IPK) can be calculated as: R LIM =50 × IPK kΩ where IPK is expressed in amperes. For a given peak current-limit setting, the runaway current limit is typically 20% higher. The peak currentlimit-triggered hiccup operation is disabled until the end of soft-start, while the runaway current-limit-triggered hiccup operation is always enabled. Programming Slope Compensation (SLOPE) Since the MAX17498A/MAX17498C operate at a maximum duty cycle of 49%, in theory they do not require slope compensation for preventing subharmonic instability that occurs naturally in continuous-mode peak current-modecontrolled converters operating at duty cycles greater than 50%. In practice, the MAX17498A/MAX17498C require a minimum amount of slope compensation to provide stable, jitter-free operation. The MAX17498A/ VOUT EAREA- MAX17498A MAX17498B MAX17498C REF R1 R2 EA+ Figure 8. Programming the Negative Output Voltage MAX17498C allow the user to program this default value of slope compensation simply by connecting the SLOPE pin to VCC. It is recommended that discontinuous-mode designs also use this minimum amount of slope compensation to provide noise immunity and jitter-free operation. The MAX17498B flyback/boost converter can be designed to operate in either discontinuous mode or to enter into the continuous-conduction mode at a specific heavy-load condition for a given DC input voltage. In the continuous-conduction mode, the flyback/boost converter needs slope compensation to avoid subharmonic instability that occurs naturally over all specified load and line conditions in peak current-mode-controlled converters operating at duty cycles greater than 50%. A minimum amount of slope signal is added to the sensed current signal even for converters operating below 50% duty to provide stable, jitter-free operation. The SLOPE pin allows the user to program the necessary slope compensation by setting the value of the resistor (RSLOPE) connected from SLOPE pin to ground. R SLOPE =0.5 × S E kΩ where the slope (SE) is expressed in millivolts per microsecond. ��������������������������������������������������������������� Maxim Integrated Products 14 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Error Amplifier, Loop Compensation, and Power-Stage Design of the Flyback/Boost Converter The flyback/boost converter requires proper loop compensation to be applied to the error-amplifier output to achieve stable operation. The goal of the compensator design is to achieve the desired closed-loop bandwidth and sufficient phase margin at the crossover frequency of the open-loop gain-transfer function of the converter. The error amplifier provided in the devices is a transconductance amplifier. The compensation network used to apply the necessary loop compensation is shown in Figure 9. The flyback/boost converter can be used to implement the following converters and operating modes: • Nonisolated flyback converter in discontinuousconduction mode (DCM flyback) • Nonisolated flyback converter conduction mode (CCM flyback) in continuous- • Boost converter in discontinuous-conduction mode (DCM boost) • Boost converter in continuous-conduction mode (CCM boost) Calculations for loop-compensation values (RZ, CZ, and CP) for these converter types and design procedures for powerstage components are detailed in the following sections. DCM Flyback Primary-Inductance Selection In a DCM flyback converter, the energy stored in the primary inductance of the flyback transformer is ideally delivered entirely to the output. The maximum primaryinductance value for which the converter remains in discontinuous mode at all operating conditions can be calculated as: L PRIMAX ≤ (VINMIN × D MAX ) 2 × 0.4 (VOUT + VD ) × IOUT × fSW where DMAX is 0.35 for the MAX17498A/MAX17498C and 0.7 for the MAX17498B, VD is the voltage drop of the output rectifier diode on the secondary winding, and fSW is the switching frequency of the power converter. Choose the primary inductance value to be less than LPRIMAX. Duty-Cycle Calculation The accurate value of the duty cycle (DNEW) for the selected primary inductance (LPRI) can be calculated using the following equation: D NEW = 2.5 × L PRI × (VOUT + VD ) × IOUT × fSW VINMIN Turns-Ratio Calculation (Ns /Np) Transformer turns ratio (K = Ns/Np) can be calculated as: K= (VOUT + VD ) × (1 − D MAX ) VINMIN × D MAX Peak /RMS-Current Calculation The transformer manufacturer needs RMS current values in the primary and secondary to design the wire diameter for the different windings. Peak current calculations are useful in setting the current limit. Use the following equations to calculate the primary and secondary peak and RMS currents. Maximum primary peak current: V × D NEW IPRIPEAK = INMIN L PRI × fSW Maximum primary RMS current: I= PRIRMS IPRIPEAK × Maximum secondary peak current: I I SECPEAK = PRIPEAK K Maximum secondary RMS current: I SECRMS = IPRIPEAK × COMP RZ CZ CP MAX17498A MAX17498B MAX17498C Figure 9. Error-Amplifier Compensation Network D NEW 3 I SECPEAK × L PRI × fSW 3 (VOUTF + VD ) For current-limit setting, the peak current can be calculated as: = ILIM IPRIPEAK × 1.2 Primary RCD Snubber Selection Ideally, the external n-channel MOSFET experiences a drain-source voltage stress equal to the sum of the input voltage and reflected voltage across the primary ��������������������������������������������������������������� Maxim Integrated Products 15 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications winding during the off period of the nMOSFET. In practice, parasitic inductances and capacitors in the circuit, such as leakage inductance of the flyback transformer, cause voltage overshoot and ringing. Snubber circuits are used to limit the voltage overshoots to safe levels within the voltage rating of the external nMOSFET. The snubber capacitor can be calculated using the following equation: C SNUB = 2 × L LK × IPRIPEAK 2 × K 2 VOUT 2 where LLK is the leakage inductance that can be obtained from the transformer specifications (usually 1% to 2% of the primary inductance). The power to be dissipated in the snubber resistor is calculated using the following formula: PSNUB = 0.833 × L LK × IPRIPEAK 2 × fSW The snubber resistor can be calculated based on the following equation: R SNUB = 6.25 × VOUT 2 PSNUB × K 2 The voltage rating of the snubber diode is: V = VINMAX + 2.5 × OUT VDSNUB K Output-Capacitor Selection X7R ceramic output capacitors are preferred in industrial applications due to their stability over temperature. The output capacitor is usually sized to support a step load of 50% of the maximum output current in the application so that the output-voltage deviation is contained to 3% of the output-voltage change. The output capacitance can be calculated as: ×t I C OUT = STEP RESPONSE ∆VOUT t RESPONSE ≅ ( 0.33 1 + ) fC fSW where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable outputvoltage deviation, and fC is the target closed-loop crossover frequency. fC is chosen to be 1/10 the switching frequency (fSW). For the flyback converter, the output capacitor supplies the load current when the main switch is on, and therefore, output-voltage ripple is a function of load current and duty cycle. Use the following equation to calculate the output-capacitor ripple: D NEW × IPRIPEAK − K × IOUT ∆VCOUT = 2 × IPRIPEAK × fSW × C OUT 2 where IOUT is load current and DNEW is the duty cycle at minimum input voltage. Input-Capacitor Selection The MAX17498A is optimized to implement offline AC-DC converters. In such applications, the input capacitor must be selected based on either the ripple due to the rectified line voltage, or based on holdup-time requirements. Holdup time can be defined as the time period over which the power supply should regulate its output voltage from the instant the AC power fails. The MAX17498B /MAX17498C are useful in implementing low-voltage DC-DC applications where the switchingfrequency ripple must be used to calculate the input capacitor. In both cases, the capacitor must be sized to meet RMS current requirements for reliable operation. Capacitor Selection Based on Switching Ripple (MAX17498B/MAX17498C): For DC-DC applications, X7R ceramic capacitors are recommended due to their stability over the operating temperature range. The ESR and ESL of a ceramic capacitor are relatively low, so the ripple voltage is dominated by the capacitive component. For the flyback converter, the input capacitor supplies the current when the main switch is on. Use the following equation to calculate the input capacitor for a specified peak-to-peak input switching ripple (VIN_RIP): CIN = D NEW × IPRIPEAK 1 − (0.5 × D NEW ) 2 2 × fSW × VIN_RIP Capacitor Selection Based on Rectified Line-Voltage Ripple (MAX17498A): For the flyback converter, the input capacitor supplies the input current when the diode rectifier is off. The voltage discharge (VIN_RIP), due to the input average current, should be within the limits specified: CIN = 0.5 × IPRIPEAK × D NEW fRIPPLE × VIN_RIP where fRIPPLE, the input AC ripple frequency equal to the supply frequency for half-wave rectification, is two times the AC supply frequency for full-wave rectification. ��������������������������������������������������������������� Maxim Integrated Products 16 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Capacitor Selection Based on Hold-Up Time Requirements (MAX17498A): For a given output power (PHOLDUP) that needs to be delivered during hold-up time (tHOLDUP), DC bus voltage at which the AC supply fails (VINFAIL), and the minimum DC bus voltage at which the converter can regulate the output voltages (VINMIN), the input capacitor (CIN) is estimated as: CIN = to 3 x IOUT. Select fast-recovery diodes with a recovery time less than 50ns, or Schottky diodes with low junction capacitance. Error-Amplifier Compensation Design The loop compensation values are calculated as: 2 1 + 0.1× fSW × VOUT × IOUT fP 3 × PHOLDUP × t HOLDUP (VINFAIL 2 − VINMIN 2 ) = R Z 450 × 2 × L PRI × fSW The input capacitor RMS current can be calculated as: IINCRMS = 0.6 × VINMIN × (D MAX ) 2 where: fSW × L PRI External MOSFET Selection MOSFET selection criteria includes the maximum drain voltage, peak /RMS current in the primary, and the maximum allowable power dissipation of the package without exceeding the junction temperature limits. The voltage seen by the MOSFET drain is the sum of the input voltage, the reflected secondary voltage on the transformer primary, and the leakage inductance spike. The MOSFET’s absolute maximum VDS rating must be higher than the worst-case drain voltage: V + VD × 2.5 VDSMAX = VINMAX + OUT K The drain current rating of the external MOSFET is selected to be greater than the worst-case peak current-limit setting. Secondary-Diode Selection Secondary-diode-selection criteria includes the maximum reverse voltage, average current in the secondary, reverse recovery time, junction capacitance, and the maximum allowable power dissipation of the package. The voltage stress on the diode is the sum of the output voltage and the reflected primary voltage. The maximum operating reverse-voltage rating must be higher than the worst-case reverse voltage: VSECDIODE= 1.25 × (K × VINMAX + VOUT ) The current rating of the secondary diode should be selected so that the power loss in the diode (given as the product of forward-voltage drop and the average diode current) should be low enough to ensure that the junction temperature is within limits. This necessitates that the diode current rating be in the order of 2 x IOUT fP = IOUT π × VOUT × C OUT CZ = CP = 1 π × R Z × fP 1 π × R Z × fSW fSW is the switching frequency of the devices and can be obtained from the Electrical Characteristics section. CCM Flyback Transformer Turns-Ratio Calculation (K = Ns /Np) The transformer turns ratio can be calculated using the following formula: K= (VOUT + VD ) × (1 − D MAX ) VINMIN × D MAX where DMAX is the duty cycle assumed at minimum input (0.35 for MAX17498A/MAX17498C and 0.7 for MAX17498B). Primary-Inductance Calculation Calculate the primary inductance based on the ripple: L PRI = (VOUT + VD ) × (1 − D NOM) × K 2 × IOUT × β × fSW where DNOM, the nominal duty cycle at nominal operating DC input voltage (VINNOM), is given as: D NOM = (VOUT + VD ) × K VINNOM + (VOUT + VD ) × K The output current, down to which the flyback converter should operate in CCM, is determined by selection of the fraction A in the above primary inductance formula. For example, A should be selected as 0.15 so that the converter operates in CCM down to 15% of the maximum ��������������������������������������������������������������� Maxim Integrated Products 17 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications output load current. Since the ripple in the primary current waveform is a function of duty cycle and is maximum-atmaximum DC input voltage, the maximum (worst-case) load current, down to which the converter operates in CCM, occurs at maximum operating DC input voltage. VD is the forward drop of the selected output diode at maximum output current. Peak/RMS-Current Calculation RMS current values in the primary and secondary are needed by the transformer manufacturer to design the wire diameter for the different windings. Peak current calculations are useful in setting the current limit. Use the following equations to calculate the primary and secondary peak and RMS currents. Maximum primary peak current: I × K VINMIN × D MAX IPRIPEAK OUT = + 1 D − MAX 2 × L PRI × fSW Maximum primary RMS current: = IPRIRMS IPRIPEAK 2 + ∆IPRI 2 − (IPRIPEAK × ∆IPRI) 3 × D MAX where DIPRI is the ripple current in the primary current waveform, and is given by: VINMIN × D MAX ∆IPRI = L PRI × fSW Maximum secondary peak current: I I SECPEAK = PRIPEAK K Primary RCD Snubber Selection The design procedure for primary RCD snubber selection is identical to that outlined in the DCM Flyback section. Output-Capacitor Selection X7R ceramic output capacitors are preferred in industrial applications due to their stability over temperature. The output capacitor is usually sized to support a step load of 50% of the maximum output current in the application so that the output-voltage deviation is contained to 3% of the output-voltage change. The output capacitance can be calculated as: ×t I C OUT = STEP RESPONSE ∆VOUT t RESPONSE ≅ ( 0.33 1 + ) fC fSW where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable outputvoltage deviation, and fC is the target closed-loop crossover frequency. fC is chosen to be less than 1/5 the worst-case (lowest) RHP zero frequency (fRHP). The right half-plane zero frequency is calculated as: fZRHP = (1 − D MAX ) 2 × VOUT 2 × π × D MAX × L PRI × IOUT × K 2 For the CCM flyback converter, the output capacitor supplies the load current when the main switch is on, and therefore, the output-voltage ripple is a function of load current and duty cycle. Use the following equation to estimate the output-voltage ripple: IOUT × D MAX ∆VCOUT = fSW × C OUT Maximum secondary RMS current: = I SECRMS 2 ∆I I SECPEAK 2 + SEC − (I SECPEAK × ∆I SEC ) 3 × 1 − D MAX where DISEC is the ripple current in the secondary current waveform, and is given by: VINMIN × D MAX ∆I SEC = L PRI × fSW × K Input-Capacitor Selection The design procedure for input-capacitor selection is identical to that outlined in the DCM Flyback section. External MOSFET Selection The design procedure for external MOSFET selection is identical to that outlined in the DCM Flyback section. Secondary-Diode Selection The design procedure for secondary-diode selection is identical to that outlined in the DCM Flyback section. Current-limit setting the peak current can be calculated as: = ILIM IPRIPEAK × 1.2 ��������������������������������������������������������������� Maxim Integrated Products 18 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Error-Amplifier Compensation Design In the CCM flyback converter, the primary inductance and the equivalent load resistance introduces a right half-plane zero at the following frequency: fZRHP = (1 − D MAX ) 2 × VOUT 2 × π × D MAX × L PRI × IOUT × K 2 f 200 × I OUT × 1 + RHP (1 − D MAX ) 5 × fP 2 where fP, the pole due to output capacitor and load, is given by: fP = (1 + D MAX ) × IOUT 2 × π × C OUT × VOUT The above selection sets the loop-gain crossover frequency (fC, where the loop gain equals 1) equal to 1/5 the right half-plane zero frequency: f fC ≤ ZRHP 5 With the control-loop zero placed at the load pole frequency: 1 CZ = 2π × R Z × fP With the high-frequency pole placed at 1/2 the switching frequency: CP = ILIM = IPK × 1.2 where IPK is given by: The loop-compensation values are calculated as: = RZ Peak /RMS-Current Calculation To set the current limit, the peak current in the inductor can be calculated as: 1 π × R Z × fSW DCM Boost In a DCM boost converter, the inductor current returns to zero in every switching cycle. Energy stored during the on time of the main switch is delivered entirely to the load in each switching cycle. 2 × (VOUT − VINMIN) × IOUT IPK = L INMIN × fSWMIN LINMIN is the minimum value of the input inductor, taking into account tolerance and saturation effects. fSWMIN is the minimum switching frequency for the MAX17498B from the Electrical Characteristics section. Output-Capacitor Selection X7R ceramic output capacitors are preferred in industrial applications due to their stability over temperature. The output capacitor is usually sized to support a step load of 50% of the maximum output current in the application so that the output-voltage deviation is contained to 3% of the output-voltage change. The output capacitance can be calculated as: ×t I C OUT = STEP RESPONSE ∆VOUT t RESPONSE ≅ ( 0.33 1 + ) fC fSW where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable outputvoltage deviation, and fC is the target closed-loop crossover frequency. fC is chosen to be 1/10 the switching frequency (fSW). For the boost converter, the output capacitor supplies the load current when the main switch is on, and therefore, the output-voltage ripple is a function of duty cycle and load current. Use the following equation to calculate the output-capacitor ripple: IOUT × L IN × IPK ∆VCOUT = VINMIN × C OUT Inductance Selection The design procedure starts with calculating the boost converter’s input inductor so that it operates in DCM at all operating line and load conditions. The critical inductance required to maintain DCM operation is calculated as: 2 (V OUT − VINMIN ) × VINMIN × 0.4 L IN ≤ IOUT × VOUT 2 × fSW where VINMIN is the minimum input voltage. ��������������������������������������������������������������� Maxim Integrated Products 19 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Input-Capacitor Selection The value of the required input ceramic capacitor can be calculated based on the ripple allowed on the input DC bus. The input capacitor should be sized based on the RMS value of the AC current handled by it. The calculations are: 3.75 × IOUT CIN = VINMIN × fSWMIN × (1 − D MAX ) The capacitor RMS can be calculated as: I I CIN_RMS = PK 2× 3 Error-Amplifier Compensation Design The loop-compensation values for the error amplifier can now be calculated as: = CZ G DC × G M × 10 = 2 × π × fSW (GDC × 10) nF where GDC, the DC gain of the power stage, is given as: G DC = 8 × (VOUT − VINMIN) × fSW × VOUT 2 × L IN (2VOUT − VINMIN )2 ×I OUT × C OUT × (VOUT − VINMIN) V R Z = OUT IOUT × C Z × (2VOUT − VINMIN) where VINMIN is the minimum operating input voltage and IOUT is the maximum load current: CP = C OUT × ESR RZ Slope Compensation In theory, the DCM boost converter does not require slope compensation for stable operation. In practice, the converter needs a minimum amount of slope for good noise immunity at very light loads. The minimum slope is set for the devices by connecting the SLOPE pin to the VCC pin. Output-Diode Selection The voltage rating of the output diode for the boost converter ideally equals the output voltage of the boost converter. In practice, parasitic inductances and capacitances in the circuit interact to produce voltage overshoot during the turn-off transition of the diode that occurs when the main switch turns on. The diode rating should therefore be selected with the necessary margin to accommodate this extra voltage stress. A voltage rating of 1.3 x VOUT provides the necessary design margin in most cases. The current rating of the output diode should be selected so that the power loss in the diode (given as the product of forward-voltage drop and the average diode current) is low enough to ensure that the junction temperature is within limits. This necessitates that the diode current rating be in the order of 2 x IOUT to 3 x IOUT. Select fastrecovery diodes with a recovery time less than 50ns or Schottky diodes with low junction capacitance. Internal MOSFET RMS Current Calculation The voltage stress on the internal MOSFET, whose drain is connected to LX, ideally equals the sum of the output voltage and the forward drop of the output diode. In practice, voltage overshoot and ringing occur due to the action of circuit parasitic elements during the turn-off transition. The maximum rating of the devices’ internal n-channel MOSFET is 65V, making it possible to design boost converters with output voltages up to 48V and sufficient margin for voltage overshoot and ringing. The RMS current into LX is useful in estimating the conduction loss in the internal nMOSFET, and is given as: ILX_RMS = IPK 3 × L INS × fSW 3 × VINMIN where IPK is the peak current calculated at the lowest operating input voltage (VINMIN). CCM Boost In a CCM boost converter, the inductor current does not return to zero during a switching cycle. Since the MAX17498B implements a nonsynchronous boost converter, the inductor current enters DCM operation at load currents below a critical value equal to 1/2 the peak-topeak ripple in the inductor current. Inductor Selection The design procedure starts with calculating the boost converter’s input inductor at nominal input voltage for a ripple in the inductor current equal to 30% of the maximum input current: V × D × (1 − D) L IN = IN 0.3 × IOUT × fSW where D is the duty cycle calculated as: V + VD − VIN D = OUT VOUT + VD VD is the voltage drop across the output diode of the boost converter at maximum output current. ��������������������������������������������������������������� Maxim Integrated Products 20 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Peak/RMS-Current Calculation To set the current limit, the peak current in the inductor and internal nMOSFET can be calculated as: IPK VOUT × D MAX × (1 − D MAX ) IOUT + L INMIN × fSWMIN (1 − D) × 1.2 for D MAX ≥ 0.5 0.25 × VOUT I IPK = + OUT × 1.2 for D MAX ≥ 0.5 L INMIN × fSWMIN (1 − D) Input-Capacitor Selection The input ceramic capacitor value required can be calculated based on the ripple allowed on the input DC bus. The input capacitor should be sized based on the RMS value of the AC current handled by it. The calculations are: 3.75 × IOUT CIN = VINMIN × fSW × (1 − D MAX ) The input-capacitor RMS current can be calculated as: I CIN_RMS = ∆ILIN 2× 3 DMAX, the maximum duty cycle, is obtained by substitutwhere: ing the minimum input operating voltage (VINMIN) in the equation above for duty cycle. LINMIN is the minimum V × D MAX × (1 − D MAX ) value of the input inductor taking into account tolerance = ∆ILIN OUT for D MAX < 0.5 L INMIN × fSWMIN and saturation effects. fSWMIN is the minimum switch ing frequency for the MAX17498B from the Electrical 0.25 × VOUT = ∆ILIN Characteristics section. for D MAX ≥ 0.5 L INMIN × fSWMIN Output-Capacitor Selection X7R ceramic output capacitors are preferred in industrial Error-Amplifier Compensation Design applications due to their stability over temperature. The The loop-compensation values for the error amplifier can output capacitor is usually sized to support a step load now be calculated as: of 50% of the maximum output current in the application, such that the output-voltage deviation is contained to 3% 203 × VOUT 2 × C OUT × (1 − D MAX ) RZ = of the output-voltage change. The output capacitance IOUTMAX × L IN can be calculated as: ×t I C OUTF = STEP RESPONSE ∆VOUT where IOUTMAX is the maximum load current: CZ = 0.33 1 + t RESPONSE ≅ ( ) fC fSW where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable output-voltage deviation, and fC is the target closedloop crossover frequency. fC is chosen to be 1/10 the switching frequency (fSW). For the boost converter, the output capacitor supplies the load current when the main switch is on, and therefore, the output-voltage ripple is a function of duty cycle and load current. Use the following equation to calculate the output-capacitor ripple: IOUT × D MAX ∆VCOUT = C OUT × fSW VOUT × C OUT 2 × I OUTMAX × R Z CP = 1 π × fSW × R Z Slope-Compensation Ramp The slope required to stabilize the converter at duty cycles greater than 50% can be calculated as: SE = 0.41(VOUT − VINMIN ) V per µs L IN where LIN is in µH. ��������������������������������������������������������������� Maxim Integrated Products 21 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Output-Diode Selection The design procedure for output-diode selection is identical to that outlined in the DCM Boost section. Internal MOSFET RMS Current Calculation The voltage stress on the internal MOSFET, whose drain is connected to LX, ideally equals the sum of the output voltage and the forward drop of the output diode. In practice, voltage overshoot and ringing occur due to the action of circuit parasitic elements during the turn-off transition. The maximum rating of the internal n-channel MOSFET of the devices is 65V, making it possible to design boost converters with output voltages up to 48V and sufficient margin for voltage overshoot and ringing. The RMS current into LX is useful in estimating the conduction loss in the internal nMOSFET, and is given as: I × D MAX ILXRMS = OUT (1 − D MAX ) where DMAX is the duty cycle at the lowest operating input voltage and IOUT is the maximum load current. Thermal Considerations It should be ensured that the junction temperature of the devices does not exceed +125°C under the operating conditions specified for the power supply. The power dissipated in the devices to operate can be calculated using the following equation: P= IN VIN × IIN where VIN is the voltage applied at the IN pin and IIN is operating supply current. The internal n-channel MOSFET experiences conduction loss and transition loss when switching between on and off states. These losses are calculated as: PCONDUCTION = ILXRMS 2 × R DSONLX PTRANSITION = 0.5 × VINMAX × IPK × (t R + t F ) × fSW where tR and tF are the rise and fall times of the internal nMOSFET in CCM operation. In DCM operation, since the switch current starts from zero, only tF exists and the transition-loss equation changes to: PTRANSITION = 0.5 × VINMAX × IPK × t F × fSW Additional loss occurs in the system in every switching cycle due to energy stored in the drain-source capacitance of the internal MOSFET being lost when the MOSFET turns on and discharges the drain-source capacitance voltage to zero. This loss is estimated as: PCAP =0.5 × C DS × VDSMAX 2 × fSW The total power loss in the devices can be calculated from the following equation: PLOSS = PIN + PCONDUCTION + PTRANSITION + PCAP The maximum power that can be dissipated in the devices is 1666mW at +70°C temperature. The powerdissipation capability should be derated as the temperature rises above +70°C at 21mW/°C. For a multilayer board, the thermal-performance metrics for the package are given below: θ JA = 48°C / W θ JC = 10°C / W The junction-temperature rise of the devices can be estimated at any given maximum ambient temperature (TAMAX) from the following equation: TJMAX = T AMAX + (θ JA × PLOSS ) If the application has a thermal-management system that ensures that the exposed pad of the devices is maintained at a given temperature (TEPMAX) by using proper heatsinks, then the junction-temperature rise of the devices can be estimated at any given maximum ambient temperature from the following equation: T= JMAX TEPMAX + (θ JC × PLOSS ) ��������������������������������������������������������������� Maxim Integrated Products 22 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Layout, Grounding and Bypassing All connections carrying pulsed currents must be very short and as wide as possible. The inductance of these connections must be kept to an absolute minimum due to the high di/dt of the currents in high-frequency switching power converters. This implies that the loop areas for forward and return pulsed currents in various parts of the circuit should be minimized. Additionally, small-current loop areas reduce radiated EMI. Similarly, the heatsink of the main MOSFET presents a dV/dt source, and therefore, the surface area of the MOSFET heatsink should be minimized as much as possible. Ground planes must be kept as intact as possible. The ground plane for the power section of the converter should be kept separate from the analog ground plane, except for a connection at the least noisy section of the power ground plane, typically the return of the input filter capacitor. The negative terminal of the filter capacitor, ground return of the power switch, and current-sensing resistor must be close together. PCB layout also affects the thermal performance of the design. A number of thermal vias that connect to a large ground plane should be provided under the exposed pad of the part for efficient heat dissipation. For a sample layout that ensures first-pass success, refer to the MAX17498B evaluation kit layout available at www.maxim-ic.com. For universal AC input designs, follow all applicable safety regulations. Offline power supplies can require UL, VDE, and other similar agency approvals. ��������������������������������������������������������������� Maxim Integrated Products 23 NEUTRAL 85V AC TO 265V AC LINE D1 S5KC-13-F C1 0.1µF, 630V R1 10I VIN VOUT2 C2 100µF D2 RB160M-60TR R8 1.2MI R7 1.2MI L1 1µH C7 2.2µF, 50V R15 3MI R14 3MI 3MI R12 3MI R6 20.5kI R5 82kI R4 2.2MI R3 2.2MI R2 2.2MI VIN Q1 BC849CW IN C9 22nF C6 0.47µF, 35V R23 10kI N2 FQT1N80TF R9 15kI C11 47pF R17 1kI C4 2.2µF VCC R11 49.9kI C12 47nF IN OVI EN/UVLO PGND COMP EA- SLOPE VCC LX EP EA+ N.C. REF N.C. PGOOD MAX17498A LIM SS IN REF R22 49.3kI R10 133kI C3 100pF C8 0.1µF, 25V IN REF VOUT1 D5 BZT52C18-7F R20 10I R16 100kI, 0.5W N1 FQD1N80TM D3 US1K-TP C10 2.2nF, 250V T1 D6 D4 RF101L2STE25 C18 141µF, 6.3V C14 10µF, 16V C15 10µF, 16V VOUT1 C16 OPEN VOUT2 VOUT1 -3.3V, 2A PGND VOUT2 8.7V, 0.3A MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Typical Application Circuits Figure 10. MAX17498A Nonisolated Multiple-Output AC-DC Power Supply ��������������������������������������������������������������� Maxim Integrated Products 24 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications VIN VOUT D2 T1 VIN C1 18V TO 36V 10µF, INPUT 63V C4 OPEN C2 4.7µF, 50V VOUT C12 22µF, 16V C3 3.3nF R1 7.5kI C5 0.22µF, 50V C13 22µF, 16V C14 22µF, 16V 5V, 1.5A OUTPUT GND D1 PGND IN LX SS C9 47nF REF EA+ U1 R6 75kI PGOOD PGOOD R12 10kI MAX17498B LIM VCC VOUT VCC R9 10kI C6 2.2µF, 16V VFB R15 1kI EAR11 15kI COMP VIN PGND C10 100pF REF R13 511I EN/UVLO EN/UVLO OVI OVI R16 OPEN C15 4.7nF R18 20kI U2 SLOPE 2 R7 OPEN R4 20kI R20 30.3kI C18 OPEN VFB R3 348kI PGND R4 OPEN VCC VCC R17 OPEN U3 3 C16 33pF 1 R19 10kI R5 10kI Figure 11. MAX17498B Isolated DC-DC Power Supply ��������������������������������������������������������������� Maxim Integrated Products 25 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications VIN VIN 10.8V TO 13.2V DC EP IN C1 10µF C3 47nF C2 0.1µF SS SS IN R1 75kI LIM PGND C4 2.2µF L1 15µH VOUT 24V, 0.2A D1 VCC LX SS26-TP VCC R2 12kI C7 4.7µF, 35V MAX17498B SLOPE PGOOD PGOOD R9 10kI R3 9.92kI VCC EA- R4 184kI N.C. VOUT REF R5 15kI COMP C5 10nF REF C6 47pF VIN R6 481kI C8 100pF N.C. PGND SS EN/UVLO EA+ R7 25kI OVI R8 49.9kI Figure 12. MAX17498B Boost Power Supply ��������������������������������������������������������������� Maxim Integrated Products 26 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Ordering Information PART TEMP RANGE MAX17498AATE+ -40°C to +125°C PIN-PACKAGE 16 TQFN-EP* 250kHz, Offline Flyback Converter DESCRIPTION MAX17498BATE+ -40°C to +125°C 16 TQFN-EP* 500kHz, Low-Voltage DC-DC Flyback/Boost Converter MAX17498CATE+ -40°C to +125°C 16 TQFN-EP* 250kHz, Low-Voltage DC-DC Flyback Converter +Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad. Package Information For the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages. Note that a “+”, “#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. PACKAGE TYPE PACKAGE CODE OUTLINE NO. LAND PATTERN NO. 16 TQFN-EP T1633+5 21-0136 90-0032 ��������������������������������������������������������������� Maxim Integrated Products 27 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications Revision History REVISION NUMBER REVISION DATE 0 9/11 Initial release — 1 3/12 Removed future product references for MAX17498B and MAX17498C 27 DESCRIPTION PAGES CHANGED Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2012 Maxim Integrated Products 28 Maxim is a registered trademark of Maxim Integrated Products, Inc. MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications ��������������������������������������������������������������� Maxim Integrated Products 29 MAX17498A/MAX17498B/MAX17498C AC-DC and DC-DC Peak Current-Mode Converters for Flyback/Boost Applications ��������������������������������������������������������������� Maxim Integrated Products 30