MP155, MP156 - Monolithic Power Systems

AN066
PRIMARY-SIDE REGULATOR
The Future of Analog IC Technology
Design Guidelines for Buck Regulator
using MP15X
Application Note
Prepared by Hommy Ding
June 07, 2012
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ABSTRACT
This application note provides design guidelines for a buck regulator with current-mode control using
MPS’ MP15X series of regulators, including step-by-step instructions and experimental results using a
design prototype.
Figure 1: Typical Buck Regulator Using the MP15X
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INDEX
Abstract ................................................................................................................................................. 2
Design Procedure .................................................................................................................................. 5
Determine the Input and Output Specifications ............................................................................... 5
IC Part Selection............................................................................................................................. 6
Inductor Design .............................................................................................................................. 7
Freewheeling Diode...................................................................................................................... 11
Output Capacitor Design............................................................................................................... 12
Dummy Load Selection................................................................................................................. 12
Feedback Circuit........................................................................................................................... 12
a. Sample Diode Selection..................................................................................................... 12
b. Feedback Resistors ........................................................................................................... 12
c. Sample and Hold Capacitor ............................................................................................... 12
Thermal Check ............................................................................................................................. 13
Auxiliary VCC Supply.................................................................................................................... 15
Design Flow......................................................................................................................................... 17
DESIGN SUMMARY............................................................................................................................ 18
example verification ............................................................................................................................. 18
References: ......................................................................................................................................... 23
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AN INTRODUCTION TO THE MP15X
The MP15X is a series of primary-side regulators that provide accurate constant voltage (CV)
regulation without an opto-coupler, and can support buck, buck-boost, and flyback topologies.
Applications for the MP15X include home appliances, white goods, and consumer electronics. It has
multiple integrated protection features, such as internal VCC under-voltage lockout (UVLO), overload
protection (OLP), short-current protection (SCP), open-loop protection, and over-temperature protection,
thus minimizing the number of external components. This application note also includes a step-by-step
design procedure for a buck converter, which also applies to other various offline applications.
The MP15X is a fully-integrated switching regulator. Figure 2 shows the device’s operation as a buck
regulator (as per Figure 1) in CCM. The integrated MOSFET turns ON at the beginning of each cycle
when the feedback voltage drops below the 2.5V reference voltage, which indicates insufficient output
voltage. The internal MOSFET turns OFF when its current reaches the internal-peak–current limit. The
freewheeling diode (D1) remains OFF until the inductor current charges the sampling capacitor (C4) to
the output voltage. Then the sampling capacitor voltage follows the output voltage to sample and hold
the output voltage to regulate the output voltage. The sampling capacitor voltage will decrease when
the inductor current falls below the output current. When the feedback voltage falls below the 2.5V
reference voltage, the internal switch turns ON to begin another switching cycle.
MOS
Diode
IL
Ipeak
Io
Vo
VFB
2.5V
Figure 2: CCM Buck Converter Using the MP15X
By monitoring the sampled output voltage across C4 regulates the output voltage as per the following
equation:
Vo = 2.5V ⋅
R1 + R 2
R2
(1)
The MP15X features an internal error amplifier (EA) and ramp compensation (shown in Figure 3) to
ensure accurate CV regulation.
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FB
Comparator
+
EA
VFB
-
+
+
∑
Vramp
+
+
-
Vramp
Vref
2.5V
Ipeak
Figure 3: EA and Ramp Compensation
The MP15X samples the feedback voltage 6µs after the internal MOSFET turns OFF. The buck
converter voltage ripple changes with the load condition. If the FB voltage equals the fixed voltage
reference, the converter will have poor load regulation. Under this condition, the MP15X has an internal
EA to change the FB comparator reference to achieve good output regulation. When the sampled
voltage differs from the 2.5V reference, the EA contributes an error signal to the 2.5V reference voltage,
thus changing the effective reference as shown in Figure 3. The EA’s high DC gain minimizes the
steady-state output voltage error. At the same time, an exponential voltage sinking source pulls down
the reference voltage. The ramp compensation changes the FB comparator’s reference voltage based
on the load condition. This ramp compensation results in a kind of feed-forward compensation: As the
load current increases, the sinking current decreases exponentially, which means the comparator
reference increases slightly, resulting in better load regulation. Under maximum load condition, the
compensation is about the 1mV/µs.
DESIGN PROCEDURE
Determine the Input and Output Specifications
-Input AC voltage range: Vac(min), Vac(max), for example 85VAC to 265VAC RMS
-DC bus voltage range: Vin(max), Vin(min)
-Output: Vout, Iout(min), Iout(max), Pout
-Estimated efficiency: η. Estimates the power conversion efficiency to calculate the maximum input
power. Generally, η is set to be 0.7.
Then the maximum input power can be given as:
Pin =
Pout
η
(2)
The MP15X can output power ≤3W. Normally, a half-wave rectifier supplies the DC input voltage when
the output power is less than 2W, and a full-wave rectifier supplies the DC input voltage when the
output power exceeds 2W. This application note describes the converter using a half-wave rectifier as
an example. When using a half-wave rectifier, the DC input capacitor (Cin) is usually 3µF/W. Choose an
input capacitor with a minimum DC voltage ≥70V; a very low DC input voltage will cause the MP15X to
enter thermal shutdown. Figure 4 shows the typical DC bus voltage waveform using a half-wave
rectifier.
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Figure 4: Input Voltage Waveform
From the waveform above, the AC input voltage VAC and DC input voltage VDC are then:
VAC (Vac ,t) = 2 ⋅ Vac ⋅ sin(2 ⋅ π ⋅ f ⋅ t),2kπ < 2πft < (2k + 1)π,k = 0,1,2...
VDC (Vac ,t) = 2 ⋅ Vac 2 −
2 ⋅ Pin
π
⋅ (t − )
Cin
2
(3)
(4)
When VAC=VDC (t1), the DC input voltage reaches its minimum (VDC(min)), calculated as:
VDC(min) = VDC (Vac(min) ,t1 )
(5)
Then, the minimum average DC input voltage (Vin(min)) is:
Vin(min) =
2 ⋅ Vac(min) + VDC(min)
(6)
2
The maximum average DC input voltage (Vin(max)) is then:
Vin(max) = 2 ⋅ Vac(max)
(7)
IC Part Selection
The MP15X family includes three parts: MP150, MP155, and MP156. They each have different internal
IC consumption values when the MOSFETs do no switch. This consumption value determines the noload power consumption each part can achieve. Select an appropriate part initially based on the noload power. Table 1 shows a brief selection guideline.
Table 1: MP15X Selection—No-Load Power Consumption
P/N
Internal IC Consumption
(No Switching)
No-Load Power Loss
85VAC to 265VAC
MP150
300µA
≤ 150mW
MP155
250µA
≤ 100mW
MP156
165µA
≤ 30mW (7V ≤ VO ≤ 30V)
The parts have different peak current limits and ON-state resistance. A part with a higher peak current
limit and smaller ON-state resistance can deliver more power and higher output current. Table 2 lists the
parts according to their output power and current.
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Table 2: MP15X Selection—Maximum Output Power, 85VAC to 265VAC
Part Number
Adapter
MP150
PO≤2W, IO≤200mA
MP155, MP156
PO≤3W, IO≤220mA
After selecting the converter’s components design, calculate the IC loss and perform a thermal check to
ensure that the converter functions within desired specifications using the selected part. If the OTP
triggers with the rated output power, select a part with a higher power rating and recalculate the
parameters based on the following design procedure.
Inductor Design
The inductance determines the maximum converter output power, so selecting an inductor with the
desired output power is very important. The MP15X’s integrated MOSFET turns ON when the load
causes the FB voltage to drop below 2.5V. Under heavy loads, the output drops very fast and the
MOSFET turn-off time decreases. The operating frequency increases as the load increases. The
MP15X has a minimum off-time limit that determines a maximum switching frequency, and limits the
maximum power. The principle of inductor design is to choose an inductor with a maximum power limit
bigger than the desired maximum output power. Calculate the maximum output power capability as per
the following instructions:
After determining the remaining converter parameters, different inductance values will lead to different
operating modes. Figure 5 shows the different operating conditions when the converter outputs
maximum power.
a. SCP
b. DCM, Ip > Ipk, toff < tminoff
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c. DCM, Ip = Ipk, toff < tminoff
d. CCM, Ip = Ipk, toff > tminoff
e. CCM, Ip = Ipk, toff = tminoff
Figure 5: Maximum Power Under Different Conditions
Condition a: The converter inductor is very small (tens of µH), which makes the current slew rate very
fast. Within the SCP’s leading-edge blanking time (tLEB2—avoids premature switching pulse termination
due to the parasitic capacitance), the MOSFET current exceeds the SCP threshold. Then the SCP
triggers and the converter cannot work normally. Avoid this condition.
Condition b: Uses a larger inductor than Condition a. The internal MOSFET current is less than the
SCP threshold within tLEB2, so SCP does not trigger and the converter works normally. However, the
small inductor value leads to a peak current that exceeds the peak current limit (IPK) within IPK’s leadingedge blanking time (tLEB1). Then the peak current under this condition is:
Ip =
(Vin − Vo ) ⋅ tLEB1
L
(8)
1 2
1
LIp
2
tLEB1 + t min off
(9)
And the maximum power is calculated as:
Pmax =
Though the converter can work normally under this condition, the inductor is so small that the peak
current is not controlled by the peak current limit under full load. Avoid this condition.
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Condition c: The converter works in DCM at the maximum power output. The peak current limit and
the inductor determine the turn-on time (ton). The inductor current slew rate, which is bigger than tLEB1, is
then:
t on =
L ⋅ Ipk
Vin − Vo
(10)
The converter reaches maximum power when the off-time equals the minimum off time (tminoff). The
maximum power is then:
Pmax =
1 2
1
LIpk
2
t on + t min off
(11)
Condition d: The converter works in CCM the output reaches maximum power. tLEB1 determines the
current ripple. This condition occurs at low output voltages.
Δimin =
Vin − Vo
⋅ tLEB1
L
(12)
1
⋅ Δimin
2
(13)
Then the average output current is:
Io _ max = Ipk −
So the maximum power under this condition is:
Pmax = Vo ⋅ Io _ max
(14)
Condition e: This converter works in CCM when it reaches the maximum power. tminoff determines the
current ripple.
Δimin =
Vo
⋅ t min off
L
(15)
The average output current under this condition is:
Io _ max = Ipk −
1
⋅ Δimin
2
(16)
And the maximum power is then:
Pmax = Vo ⋅ Io _ max
(17)
The operation mode where the converter outputs the maximum power changes with Vin and Vo. By
analyzing different maximum power conditions, we get an inductor vs. maximum power curve. Figure 6
shows the curve for 5V and 12V (Ipk = 290mA, tminoff = 18µs, Vin = 375VDC). The green zones provide
the safest converter working regions.
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a) Vo = 5V
b) Vo = 12V
Figure 6: Inductor Value vs. Maximum Output Power
The peak current limit, minimum off time and the inductance affect the maximum power output: These
parameters’ tolerances affect the maximum output power capability. Normally, the peak current limit
tolerance is ±10%, the minimum off time tolerance is ±17% and the inductance tolerance is ±20%. We
can obtain a maximum value (Po_max) and a minimum value (Po_min) of maximum power considering the
tolerance of the parameters. Po_min is used to design the minimum inductance. So the converter we
designed can output the required maximum output power considering the tolerance of the parameters.
Figure 7 compares the minimum value and typical value of the maximum output power for 5V (a) and
12V (b).
a) Vo = 5V
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b) Vo = 12V
Figure 7: Minimum Value of the Maximum Output Power
We can select a minimum inductance by calculating the maximum output power,
Po _ min (L) ≥ Pout
(18)
Accounting for costs, use a standard off-the-shelf inductor—use a standard inductor value greater than
or equal to the calculated values.
Freewheeling Diode
Select a diode with a maximum reverse block voltage rating that exceeds the maximum input voltage.
For universal voltage applications, use a diode with a 600V reverse block voltage. Determine the diode
current rating from the RMS current as follows:
IrmsDCM = Ipk ⋅
V
1 Io
⋅ 2 ⋅ (1 − o ) for DCM
3 Ipk
Vin
IrmsCCM = (Io2 +
V
Δi2
) ⋅ (1 − o ) for CCM
3
Vin
(19)
(20)
Where Δi is the current ripple of inductor, and is equal to 2(Ipk - Io).
The reverse recovery of freewheeling diode affects the efficiency and the circuit operation, so use an
ultrafast diode. For DCM, select a diode with a reverse recovery time of less than 75ns, such as
EGC10JH from ZOWIE. For CCM, select an ultrafast diode with a reverse recovery time of less than
35ns, such as UGC10JH.
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Output Capacitor Design
The output capacitor maintains the DC output voltage. Estimate the output voltage ripple as:
VDCM _ ripple
I
= o
fsCo
VCCM _ ripple =
2
⎛I −I ⎞
⋅ ⎜ pk o ⎟ + Ipk ⋅ RESR for DCM
⎜ I
⎟
⎝ pk ⎠
(21)
Δi
+ Δi ⋅ RESR for CCM
8fsCo
(22)
Where fs is switching frequency, and RESR is ESR of output capacitor.
To lower the output voltage ripple, use ceramic, tantalum or low-ESR electrolytic capacitors.
Dummy Load Selection
The output requires a dummy load to maintain the load regulation under no-load condition. This can
ensure sufficient inductor energy to charge the sample-and-hold capacitor to detect the output voltage.
Most applications can use a 3mA dummy load, and this load can be adjusted according the regulation.
Increasing the dummy load adversely affects the efficiency and no-load consumption. If the user does
not care about no-load regulation, use a Zener diode.
Feedback Circuit
a. Sample Diode Selection
The diode should have the same or higher voltage rating as the freewheeling diode. The current
through the diode is very small, so use fast and slow diodes such as FR10X and 1N400X. However, the
sample diode and freewheeling diode should have the same forward voltage drop for better regulation.
b. Feedback Resistors
The MP15X provides accurate constant voltage (CV) regulation, and the resistor divider determines the
output voltage as:
Vo = 2.5V ×
R1 + R 2
R2
(23)
Choose appropriate R1 and R2 to maintain the FB voltage at 2.5V. R2 is typically between 5kΩ and
10kΩ.
c. Sample and Hold Capacitor
The feedback capacitor provides sample-and-hold function. Design this capacitor for good output
voltage regulation. Figure 8 shows the detailed operation waveforms under DCM. Figure 2 shows the
detailed operation waveforms under CCM.
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Figure 8: Detailed Operation in DCM
When the MOSFET turns off and the freewheeling diode turns on, if the feedback capacitor voltage is
less than the output voltage, the inductor charges the capacitor until the feedback capacitor voltage
equals the output voltage. This makes the feedback capacitor (C4) sample the output voltage. But if the
feedback capacitor voltage exceeds the output voltage, the capacitor is only discharged by the
feedback resistors. So The feedback capacitor’s discharge rate should exceed that of the output
capacitor by the load Then the voltage of feedback capacitor cannot exceed the output voltage.
Otherwise the converter may work abnormally.
In CCM, when the feedback capacitor’s discharge rate exceeds that of the output capacitor, the
feedback capacitor’s voltage will equal the output voltage. In DCM, when the freewheeling diode turns
off and the converter works in the discontinuous area, the feedback capacitor’s voltage remains below
output voltage. This results in a higher output voltage than the rated output voltage under light load and
result in loose output voltage regulation.
From the previous analysis, we can find that a fast feedback capacitor discharge rate causes poor lightload regulation, and a slow discharge rate affects circuit operation.
To estimate the capacitance:
C
Vo
C
1 Vo
⋅ o ≤ CFB ≤
⋅ o⋅
2 R1 + R 2 Io
R1 + R 2 Io
(24)
Where Co is the output capacitance.
We can obtain a rough value of the feedback capacitance, and then choose an appropriate value for
practical applications.
Thermal Check
The MP15X has an internal OTP function that triggers when the IC junction temperature increases to
150°C. The part will not resume function unless the Vcc voltage drops below 2.4V. The part
temperature increases as the output power increases, thus perform a thermal check and choose an
appropriate part after designing the converter.
To ensure a stated margin, the maximum junction thermal shutdown temperature is Tb (normally
125°C). Let Ta represent the maximum ambient temperature for normal MP15X applications. The
maximum temperature rise (ΔT) is then Tb - Ta. Given that the junction-to-ambient thermal resistance
θJA is 100°C/W, the maximum IC power loss is:
Pmax_ loss =
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Tb − Ta
θJA
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(25)
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Two factors contribute to the MP15X’s power loss: power loss of the Integrated MOSFET, and internal
IC consumption. The power loss of the integrated MOSFET can be divided into conduction loss and
switching loss. The internal IC consumption includes the MOSFET driving loss.
Figure 9 shows the MOSFET current under DCM and CCM.
Ip
0
t on
t sw
t
DCM
CCM
Figure 9: MOSFET Current Under DCM and CCM
Calculate the duty cycle as:
DDCM =
2Io Vo
⋅
for DCM
Ipk Vin
(26)
Vo
for CCM
Vin
(27)
DCCM =
Estimate the MOSFET RMS current as:
IMOS _ DCM = Ipk ⋅
IMOS _ CCM
DDCM
for DCM
3
Δi2
= (I +
) ⋅ DCCM for CCM
3
2
o
(28)
(29)
The MOSFET conduction loss is then:
PMOS _ con = IMOS 2 ⋅ Rds _ on
(30)
When the converter operates in DCM, the MOSFET turns on at zero current. The MOSFET turn-on
power loss is very small and can be ignored. So for DCM mode, calculate the turn-off loss.
However, CCM requires calculations for both turn-on and turn-off losses. The integrated MOSFET’s
turn-on and turn-off times are very small (~50ns), so use the simplified model shown in Figure 10[1] to
calculate the power loss during turn on and turn.off
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a) Turn On
b) Turn Off
Figure 10: MOSFET Switching Process
The MOSFET switching loss is then:
PMOSFET _ on =
1
⋅ Vin ⋅ Ip ⋅ (t d(on) + t r ) ⋅ fs
2
(31)
PMOSFET _ off =
1
⋅ Vin ⋅ Ip ⋅ (t d(off ) + t f ) ⋅ fs
2
(32)
Where td(on) is the turn-on delay time, tr is the rise time, td(off) is the turn-off delay time, and tf is the fall
time. Internal IC consumption power loss can be calculated as:
PIC = Vin ⋅ ICC
(33)
Where ICC is the operation current under a full load.
Normally conduction loss is the primary contributor to IC power loss, and the lower the input voltage,
the greater the conduction loss. So we only need to do a thermal check when Vin equals Vinmin.
Auxiliary VCC Supply
MP155 and MP156 have a function of auxiliary Vcc supply. When the output voltage exceeds VCC
(typically 5.5V), we can use an auxiliary VCC supply by connecting a diode and a resistor between C3
and C4 as shown in Figure 11.
Figure 11: Auxiliary VCC Supply
Then VCC can be clamped to 5.8V, and the internal regulator is forced off at all times. This can
eliminate IC power consumption due to charging the VCC capacitor from the Drain pin. We can lower
the no-load consumption through an auxiliary VCC supply. As this can cause additional power loss, and
select an appropriate resistor value as per:
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R6 ≈
Vo − 5.8V
IC
(34)
IC power consumption (No switching) is different for different parts. For instance, the MP155 requires
250µA. In addition, we recommend adding a 1N4148 diode. When the output voltage is 12V under
these conditions, use a 25kΩ resistor.
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DESIGN FLOW
Start
Pin ≥ 2W
N
Input capacitor Design
Y
Full-bridge
rectifier
Pno_load ≥
100mW
IC part choose
Half-bridge
rectifier
Pno_load ≤
100mW
Pno_load ≤
30mW
N
Po ≤ 2W
N
Y
Po ≤ 3W
Y
Io ≤ 200mA
N
Po ≤ 3W
N
Fail to output the
power
Y
N
Io ≤ 220mA
Io ≤ 220mA
Y
Y
Y
MP150
MP155
MP156
N
Parameter design
IC loss calculation
Tj = Ta + ΔT
Choose a higher
IC part
Thermal check
Tj ≤ 125℃
N
Y
End
Figure 12: Design Flow for the MP15X
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DESIGN SUMMARY
MPS’ design tool makes designing with the MP15X easier. The tools can calculate all key
parameters to build a reliable design with excellent performance.
Figure 13 shows a detailed buck converter reference design for the MP155. The inductor value is
the most important component for this converter. Poor inductor selection may not deliver the
desired rated power.
Choose a sample-and-hold capacitor with an appropriate value to achieve good regulation.
Determine a dummy load to regulate the voltage under no-load condition. However, very large
dummy loads will deteriorate the efficiency and increase no-load consumption.
Perform a thermal check after designing the parameters, especially for applications with high
ambient temperatures.
z
z
z
z
D1
D1
1N4007
1N4148
R1
16.2K
R5
24.9K
C1
U1
5
Drain
Vcc
FB
L1
RF4
10
2
C2
C7
2.2uF
470pF
R2
4.3K
12V/150mA
L2
D2
4
L
220nF
1
1mH
1N4007
Source
Source
Vout
3
1.8mH
MP155
C3
C4
4.7uF/400V
4.7uF/400V
D3
C5
WUGC10JH
100uF/16V
C6
R5
1uF
6.04K
D4
N
GND
1N4007
GND
Figure 13: Buck Converter Application Using the MP155
EXAMPLE VERIFICATION
The following is a buck converter using the MP155 as a design example that has been built and tested
(Input: 85VAC to 265VAC; Output: 12V/0.15A). MPS’ design tool can calculate the values of key
components. The following describes the design procedure using MPS’ design tool:
1. Input the system specifications, including input voltage, output requirements, efficiency and etc.
1. System Spec
Input Spec
Minimum Line Voltage
Maximum Line Voltage
Line Voltage Frequency
Output Voltage
Output Current
Estimated Efficiency
No Load Power Consumption
Output Voltage Ripple ratio
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Vac_low
Vac_high
flne
Vo
Io
η
Pnoload
λ.vrp
85
265
50
12
0.15
0.70
<=100
1.00
V
V
Hz
V
A
mW
%
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2. After determining the specifications, select the input rectifier topology. Use a half-wave rectifier for
output power less than 2W. This example (1.8W, 12V 0.15A) uses the half-wave rectifier topology. The
tool can then calculate the minimum input capacitance and obtain the input DC voltage.
2. Input Capacitor
Rectifier Selection
Input Capacitor
Calculate the Minimum DC Voltage
The minimum DC input voltage
The Minimum mean DC input voltage
The maximum mean DC input voltage
Half-Wave
Cin
9.40
uF
Vin_min
VDC_min
VDC_max
71.76
95.98
367.70
V
V
V
3. The tool will recommend an IC part for the user according the output specifications. However, the
tool will notify the user if the output specification exceeds the capability of the MP15X.
3. IC Selection
IC Selection
Peak Current Limitation
On-State Resistance
Maximum DCM Current
Maximum CCM Current
Ipeak
Ron
IDCM_max
ICCM_max
MP155
290.00
mA
ohm
20.00
130.00
mA
220.00
mA
4. The tool will suggest and inductance value based on an analysis of the inductor design. The user can
choose a standard off-the-shelf inductor, but must choose a value greater than or equal to the
suggested value.
4. Inductor Parameters
Suggested Inductance Value
Inductance Value
L
L1
1.40
1.80
mH
mH
5. The tool will suggest an output capacitor value based on the output voltage ripple and capacitor ESR,
and recommend a dummy load value. The tool will calculate the resistor value based on the output
voltage. The user can also adjust the value according the regulation and no-load consumption.
5 Output Design
Output Cap ESR
Output Cap
Output Dummy Load
Cesr
Cout
Rdummy
0.30
100.00
6.00
ohm
uF
kohm
6. The user must choose a diode with a maximum DC blocking voltage higher than the maximum DC
input voltage. For typical applications, use a 600V/1A diode.
6 Output Diode Voltage
Output Voltage of Output Diode
V_d
374.77
V
7. With the analysis of the feedback circuit, the tool can calculate the feedback resistors and the sample
capacitor given an R2 value.
7 Feedback Circuit
Lowside Feedback Resistor
Highside Feedback Resistor
Maximum Feedback Capacitor
AN066 Rev. 1.0
12/30/2013
R2
R1
C_FB
4.30
16.34
0.33
kohm
kohm
uF
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AN066– PRIMARY-SIDE REGULATOR
8. At the end of the design process, the tool will calculate the junction temperature of MP15X according
the power loss of IC part and ambient temperature. If the temperature exceeds 125°C, the tool will
notify the user to lower the output specification or choose a part with a higher power rating.
8 Thermal Check
Ambient Temperature
Junction Temperature
Ta
Tj
60.00
83.89
degree
degree
9. When the output voltage exceeds VCC (Typical value is 5.5V), add an auxiliary VCC supply by
connecting a diode and a resistor to decrease the no-load power consumption. The tool will
recommend a 1N4148 diode and calculate a resistor value.
9 Auxiliary VCC Supply
Diode
Resistor Value
D3
R3
1N4148
24.80
kohm
Figure 14 shows the drain-source voltage waveform (Vds) and inductor current (IL) under full load and no
load. The MP155 has a frequency foldback feature. At the light-load or no-load conditions, the output
drops very slowly. This increases the MOSFET turn-on time. The frequency decreases as the load
decreases. At the same time, the peak current limit starts to decrease from 0.3A as the OFF-time
increases.
Figure 15 and Figure 16 show the measured efficiency versus load and no load consumption at different
input voltage. A Buck converter based on MP155 has a high efficiency above 75% under full load
conditions. The addition of the auxiliary VCC supply, the no load consumption is about 70mW with 2mA
dummy load.
Figure 17 and Figure 18 show the load regulation and line regulation. The MP155 has an internal EA and
ramp compensation that improve the regulation. The load regulation is about ±3.4%.
AN066 Rev. 1.0
12/30/2013
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© 2013 MPS. All Rights Reserved.
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AN066– PRIMARY-SIDE REGULATOR
a) Full load
b) No load
Figure 14: Vds and IL Waveforms (115VAC)
Efficiency Vs Load
85
Efficiency(%)
80
75
70
115Vac
230Vac
65
60
55
50
0.01
0.03
0.05
0.07
0.09
0.11
0.13
0.15
Load(A)
Figure 15: Efficiency
AN066 Rev. 1.0
12/30/2013
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© 2013 MPS. All Rights Reserved.
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AN066– PRIMARY-SIDE REGULATOR
No load consumption Vs Input voltage
No load consumption(mW
80
75
70
65
60
55
50
85
135
185
235
Input voltage(Vac)
Figure 16: No-Load Consumption
Load regulation
12.8
Output voltage(V)
12.7
12.6
12.5
12.4
12.3
115Vac
12.2
230Vac
12.1
12
11.9
11.8
0
0.05
0.1
0.15
Load(A)
Figure 17: Load Regulation
Line regulation
12.8
Output voltage(V)
12.7
12.6
12.5
12.4
Full load
No load
12.3
12.2
12.1
12
11.9
11.8
85
135
185
235
Input Voltage(Vac)
Figure 18: Line Regulation
AN066 Rev. 1.0
12/30/2013
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© 2013 MPS. All Rights Reserved.
22
AN066– PRIMARY-SIDE REGULATOR
REFERENCES:
[1] Weixun Lin, “Technology of Modern Power Electronics” Zhejiang University Book Concern, 2002-07.
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
AN066 Rev. 1.0
12/30/2013
www.MonolithicPower.com
MPS Proprietary Information. Patent Protected. Unauthorized Photocopy and Duplication Prohibited.
© 2013 MPS. All Rights Reserved.
23