Why TinySwitch Technology

Designing Low Power Switchers with
LinkSwitch and TinySwitch-II
1
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• The focus of this presentation is power supplies of 20 W or less
Agenda
2
•
Introduction
•
LinkSwitch
– Operation
– Performance
– Designing with LinkSwitch
– Hints and Tips
– Application Examples
– LinkSwitch Summary
•
TinySwitch and TinySwitch-II
– Why TinySwitch Technology
– Choosing TinySwitch-II vs TinySwitch
– Operation
– Designing with TinySwitch Technology
– Application Examples
– Hints and Tips
– TinySwitch Technology Summary
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Introduction
3
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Company Overview
4
•
Leader in high voltage monolithic power conversion ICs
•
> One billion devices shipped
•
Revolutionary products
•
Proven quality and delivery performance
– 3 µ CMOS not capacity limited
•
Pioneer in energy efficiency (EcoSmart®)
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• Power Integrations was the world’s first semiconductor company to introduce highly
energy efficient products by using EcoSmart technology.
• TinySwitch received the 1999 Discover award for the best technological innovation in
the environment category for its EcoSmart features.
• 10% of the world’s electrical energy is wasted by products that are in standby.
• EcoSmart technology practically eliminates standby waste.
Technology Leadership
5
•
Integrated high-voltage, high frequency MOSFET
•
Patented device structure
•
Uses industry standard 3 µ CMOS process
•
Widely available capacity
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Discrete PWM Circuit
Start-up
Feedback
Compensation
PWM Controller
Thermal
Shutdown
Oscillator
High Voltage
MOSFET
Gate Drive
Current Limit
6
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• In addition to the high voltage MOSFET and controller, Power Integrations’ ICs
integrate:
– start-up circuit
– lossless current limit
– oscillator timing components
– feedback compensation
– thermal shutdown
– gate driver circuit
Equivalent Power Integrations Solution
20 to 50 components eliminated
7
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• Newer PI products also integrate functions such as:
– soft start
– frequency jittering for low EMI
– line OV/UV protection
– programmable lossless current limit
– remote ON/OFF
– very low standby/no-load power consumption
Continuous Innovation
TIME
8
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• PI is on the leading edge of innovation in power conversion, continuously introducing
breakthrough topologies and technologies.
Cost Effective Over Wide Power Range
DPA-Switch 0 W - 100 W
LinkSwitch 0 W - 4 W
TinySwitch-II
TinySwitch
2 W - 20 W
TOPSwitch-GX
10 W - 250 W
Output Power (Watts)
9
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• Power Integrations’ products cost effectively cover:
– 95% of all AC-DC power supplies with product families ranging
from 0 W to 250 W
LinkSwitch
0 W to 3 W
TinySwitch
2 W to 20 W
TOPSwitch-GX
10 W to 250 W
– High volume 24/48 V DC-DC converter applications ranging
from 0 W to 100 W with DPA-Switch
• This graph only approximates the power capabilities of each product family. For
more accurate data, see the output power table on each product family data sheet.
Comprehensive Design Support
•
Design Accelerator Kits
– Fully tested power supply
– Product samples
– Complete design documentation
•
PI Expert design software
•
Technical documents on website
10
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• PI has the most comprehensive design tools in the industry
Global Applications Support
Fully Equipped Applications Labs
11
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Fully equipped PI applications labs are located worldwide:
• United States
– San Jose, California
Chicago, Illinois
Atlanta, Georgia
– London, UK
Munich, Germany
Milano, Italy
– Taipei, Taiwan
Seoul, South Korea
Shenzhen, PRC
– Shanghai, PRC
Yokohama, Japan
Bangalore, India
• Europe
• Asia
Wide Customer Acceptance
12
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• Virtually every major OEM worldwide uses Power Integrations’ ICs in their products.
Low Power (<20 W) Applications
Chargers / Adapters
PC & Monitor
standby
13
TV standby
Industrial
White Goods
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• >50% of the AC to DC power supply unit volume is under 20 W, covering a wide range
of end products and applications.
LinkSwitch
– 0 W to 3 W
– Replaces linear transformer
solutions at equal or lower cost
– Regulation by PWM control
– Primary sensed approximate
CV/CC output
TinySwitch-II
– 2 W to 20 W
– Replaces regulated linear, RCC and
other solutions at equal or lower cost
– Regulation by ON/OFF control
– Secondary sensed feedback for
accurate CV or CV/CC outputs
Both meet all worldwide energy efficiency standards
14
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• RCC: Ringing choke converter (this is a self oscillating converter)
• CV: Constant Voltage
• CC: Constant Current
Energy Efficiency Standards
Will Make Linear Solutions Obsolete
• No-load EC requirement for external power supplies
– <300 mW by 2005
• Energy Star requirement for consumer audio and
DVD products
– < 1 W stand-by, by January 1, 2003
• US Presidential Executive Order
– < 1 W stand-by now on all Federal Government purchases
• Japanese “Top Runner” program
– Promotes lowest standby in consumer products
• Many other standards and programs worldwide
– Blue Angel, China Sustainable Energy Program, etc.
Linears will not be able to cost effectively meet many of these standards
15
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Introducing LinkSwitch®
The Linear Killer Switch
providing
Switcher Benefits at Linear
Cost
16
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• LinkSwitch based solutions are cost competitive, even when compared to low-end,
unregulated linear trickle chargers
• Almost 1 Billion low-power (0.5 to 3 W) linear-transformer-based power supplies are
produced worldwide, each year
• Driven by energy efficiency requirements, these will convert to switchers
• LinkSwitch has enabled cost effective conversion to begin NOW!
LinkSwitch: Breakthrough Technology
17
•
Extremely simple circuit configuration - easy to design
•
Only 14 components - low cost
•
Primary side controlled constant current charging - high efficiency
– No primary or secondary side current sense resistor required
•
Fully protected for thermal, short circuit and open loop faults
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• Bridge rectifier is counted as a single component
• An extra resistor is allowed for pre-loading (explained later)
Linear vs LinkSwitch
•
Bulky and heavy
– Higher shipping costs
– Covers adjacent outlets
•
Smaller and lighter
– Lower shipping costs
– Occupies single outlet
•
Requires multiple designs
– Higher inventory costs
•
One design works worldwide
– Lower inventory costs
•
Energy inefficient
– Will not meet most future standards
– Annual energy waste exceeds cost
of power supply
•
Extremely Energy efficient
– Meets all worldwide standards
– Saves enough energy to pay for
complete power supply in 1 year
18
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• Standby energy loss is reduced by almost an order of magnitude
• Unregulated linear shown, regulated linear would typically have higher zero load
consumption
• (LinkSwitch is more cost effective than RCC solutions in replacing linears and
requires 30-60 fewer components. Therefore RCC comparisons are not included in
this presentation.)
LinkSwitch Operation
19
Seminar_lowpower_100102_screen_102102
Flyback Fundamentals
•
LinkSwitch is designed for discontinuous mode Flyback operation
– All energy in transformer transferred to secondary during switch off-time
•
During diode conduction VO is transformed to primary as VOR
– VOR ≈ VO × NP/NS
20
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• VOR on the primary side is a close representation of the output voltage for flyback
converters
• Unstable operation may result if a LinkSwitch device is used in the continuous
conduction mode (CCM). Therefore, CCM operation is not recommended.
High-side MOSFET allows direct VOR sensing
High-side Switch Reference
Low-side
Switch
Reference
•
Sensing VOR is difficult with low
side MOSFET
– can only sense VOR + VIN with
respect to source
21
• (≈VO as output diode drop neglected)
•
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Sensing VOR is easy with high
side MOSFET
– can sense VOR directly with respect
to source
High-side MOSFET Waveforms
SOURCE to RTN
voltage
Leakage
inductance spike
RTN to SOURCE
voltage
DRAIN to
SOURCE voltage
PI-3299-091002
22
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• Referenced to Source VFB can be sensed directly
• (Leakage inductance spike causes an error in VFB (above VOR))
• (For illustration, ripple on VFB exaggerated)
VFB ≈ VOR
LinkSwitch Indirectly Senses VO from VOR
VFB ≈ VOR = VO ×
DCLAMP
CCLAMP
NP
NS
Diode drop neglected
•
CCLAMP samples and holds VFB≈VOR
•
RFB converts VFB into feedback control current IC
•
Clamp circuit (DCLAMP, CCLAMP, RFB) also:
– Limits voltage across MOSFET due to leakage inductance
– Provides supply current (IC) to power LinkSwitch
23
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• (Leakage inductance energy introduces an error in the feedback voltage meaning that
the VFB is not a perfect representation of VO)
• (Electrically, the secondary diode may be placed in upper or lower end of secondary
but EMI may be improved by connecting as shown)
Start-up: Charging CONTROL Pin Capacitor
CONTROL pin capacitor is charged
to 5.75 V from DRAIN via internal
high voltage current source
•
No external start-up resistor required
24
• (Same principal as TOPSwitch)
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Start-up: Drain Starts Switching
Output voltage
begins to rise
When CONTROL pin
reaches 5.6 V, the internal
current source is turned off
IC
As output voltage rises
current into CONTROL
pin rises
25
Stored energy powers LinkSwitch,
discharging capacitor
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• CONTROL pin is a current fed pin with an internal voltage clamp
• (Same principal as TOPSwitch)
Start-up Waveforms
CONTROL pin
voltage
LinkSwitch powered from CONTROL
pin capacitor, output voltage rises
Charging
CONTROL pin
26
Output in regulation, LinkSwitch
powered from VOR
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Normal start-up: CONTROL pin and SOURCE pin node switching waveforms
• At start-up, the CONTROL pin capacitor is charged to 5.6 V, by the internal, highvoltage current source (from the DRAIN)
• At 5.6 V, the internal current source turns off, and MOSFET switching is enabled
• Energy in the CONTROL pin capacitor powers the LinkSwitch device
• The output voltage rises, and reaches its regulation value
• When VFB exceeds 5.75 V, current flows into the CONTROL pin providing feedback
• The MOSFET duty cycle is modulated to control the CV portion of the output VI curve,
the internal current limit is adjusted to maintain the CC portion of the output VI curve
(explained in more detail later)
• Due to the (approximate) 100 Ω impedance of the CONTROL pin, feedback (control)
current raises the CONTROL pin voltage from 5.6 V to 5.75 V
• The CONTROL pin voltage is set by an internal shunt regulator, making it a current
driven input. Any in-circuit testing performed at Incoming Inspection must limit the
current supplied to the CONTROL pin to the range specified in the device data sheet;
which also has recommended test circuits.
Auto-restart Waveforms
Feedback current <LinkSwitch supply
current causes CONTROL pin capacitor to
discharge to 4.7 V, initiating auto-restart
•
27
Some feedback current,
<LinkSwitch supply current,
increases switching time
Auto-restart limits average output current to 8% of the nominal CC
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• Abnormal start-up: CONTROL pin and SOURCE pin node switching waveforms
(during an output overload, short-circuit or an open feedback loop condition)
• Once the CONTROL pin reaches 5.6 V, MOSFET switching is enabled
• Energy in the CONTROL pin capacitor powers the LinkSwitch device
• Feedback current <~1 mA (the LinkSwitch supply current) allows the CONTROL pin
capacitor to discharge. When the CONTROL pin reaches 4.6 V, auto-restart is
initiated
• With the MOSFET disabled, the CONTROL pin capacitor is charged and discharged
for 7 cycles
• MOSFET switching is enabled after the 7th charge/discharge cycle, and the overall
sequence repeats (if the overload, short-circuit or open feedback condition still
exists)
• While MOSFET switching is occurring, there is usually some feedback current, even
under most fault conditions. This slows the discharge rate of the CONTROL pin
capacitor, which increases the length of time that switching occurs for, during the
start-up attempt
Primary Based CC/CV Output Regulation
typical peak power
point at 85 VAC
CC
•
CC regulated by internal current
limit control
•
CV regulated by duty cycle control
Duty cycle control
Peak
power
point
CV
typical peak
power point at
85 VAC
CC
CV
42 kHz to
30 kHz
Current
limit
control
Autorestart
PI-3090-081302
28
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• IC ∝ VOR ∝ VO
• Load <peak power: LinkSwitch duty cycle is reduced to maintain an approximate CV
output (PWM control)
• At peak power point internal current limit is at maximum
• Loads >peak power: VO falls, reducing IC. LinkSwitch internal current limit is reduced
to maintain an approximate CC output down to ~30% of VO
• Below ~30% of VO (IC <~1mA) LinkSwitch enters auto-restart
• At no-load: switching frequency switches from 42 kHz to 30 kHz, reducing no-load
power consumption
LinkSwitch Block Diagram
Feedback Control / Supply pin
Short Circuit / Fault Protection
Current limit
adjusted to
maintain CC
Lossless
Current
Sense
uses
RDS(ON)
42 kHz
switching
frequency for
low EMI and 3 W
from EE13 core
Integrated
700 V
MOSFET
PWM for CV
Low Frequency
Standby
29
High Voltage
Startup
On-chip hysteretic
thermal shutdown
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• LinkSwitch integrates all of the switcher complexity into just three terminals, making
the switching solution as simple as a linear regulator circuit
Operation Summary
30
•
Cost equivalent to linears
•
Provides CV/CC output
•
Simple transformer
– no bias winding
– powered from primary leakage
clamp
– works to zero output voltage
•
Fault protection
– output short circuit
– hysteretic thermal protection
– broken feedback loop
•
Low component count
– simplest CV/CC solution
– low manufacturing cost
•
No optocoupler
– simple layout
– low cost
•
Low standby consumption
– meets US <1 W and EC <300 mW
specifications
•
No current sense resistor
– higher efficiency
– simple design
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LinkSwitch Performance
31
Seminar_lowpower_100102_screen_102102
Output CV/CC Tolerances
•
Tolerances achievable in low cost, high volume
manufacturing
– ±10% estimated CV tolerance at peak power point
– ±20% estimated CC tolerance*
(dominated by transformer inductance tolerance)
– Includes LinkSwitch and other component variations
*with ±10% primary inductance tolerance
32
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• Tighter primary inductance tolerances produce tighter CC tolerances.
With no primary inductance variation, the CC tolerance is about +12%
• From full load to no-load, the output voltage typical increases about +40%
2.7 W, 9 V Linear vs LinkSwitch: CV
2.7 W Unregulated Linear
output envelope (98-132 VAC)
Linear does not meet rated
output power (9 V, 300 mA)
2.7 W LinkSwitch output below 120 VAC
envelope (85-265 VAC)
Output Voltage (V)
LinkSwitch ± 4% at
rated output
(85 VAC to 265 VAC)
PI-3502-051303
Unregulated linear
± 28% at rated output
(98 VAC to 132 VAC)
Output Current (mA)
33
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• This linear didn’t meet its full specification; at 98 VAC in, it only delivered about 2 W
• In a single unit-to-unit comparison to a typical unregulated linear design…
• The LinkSwitch had better regulation
– Linear regulation: load (0-300 mA) –13%, line ±28%
– LinkSwitch regulation: load (0-300 mA) –12%, line ±4%
• The LinkSwitch provided full power over the entire input range (85-265 VAC)
– The Linear provided rated power only at 120 VAC and above
• The LinkSwitch had a significantly tighter output characteristic
– Having a tighter peak power point tolerance reduces charging time
2.7 W, 9 V Linear vs LinkSwitch: CC/Overload
15
Output voltage (V)
2.7 W Linear
LinkSwitch
2.7 W LinkSwitch
Specified
adapter output
power
12
9
6
Input power 22 W,
protected by onetime thermal fuse.
3
0
0
500
1000
1500
2000
2500
Output Current (mA)
PI-3503-051303
Auto-restart limits overload current:
- Average 50 mA, Pk 1 A
- Input power 200 mW
- Protects both supply and load
34
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• This comparison was made with a typical unregulated linear design
• The linear may be damaged by an overload or a short circuit on its output
• The LinkSwitch CC output characteristic, plus its auto-restart and thermal shut-down
functions protect it and the load from damage. Additionally, it will resume normal
operation after the fault is removed
Linear vs LinkSwitch: Output Ripple
808 mV pk-pk
Unregulated Linear, 9 V, 2.7 W
adapter 115 VAC, Full Load
50 mV, 2 ms/div
LinkSwitch, 9 V, 2.7 W adapter
115 VAC, Full Load
Measured with resistive load, at end of cable
35
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• LinkSwitch has <20% the output ripple of a typical unregulated linear
PI-3505-051303
PI-3504-051303
200 mV, 2 ms/div
162 mV pk-pk
1.5 x The Efficiency of Unregulated Linear
100%
90%
LinkSwitch: 75 %
80%
Linear: 53 %
60%
50%
40%
98 VAC
30%
115 VAC
20%
132 VAC
10%
265 VAC
85 VAC
0%
0
100
200
300
PI-3506-051303
Efficiency
70%
400
Output Current (mA)
• Regulated linear has much poorer efficiency (<25%)
36
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• LinkSwitch efficiency is high, even at light loads (2X the linear’s efficiency at 100 mA)
Linear vs LinkSwitch: No-load Consumption
Measurements were made at 115 VAC
(The no-load consumption at 265 VAC is only 250 mW!)
• The no-load energy savings alone, can pay for the
cost of the entire power supply, in less than 1 year
37
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• The unloaded unregulated linear dissipates 1.65 W at 115 VAC,
The unloaded LinkSwitch only consumes 200 mW at 115 VAC
Linear vs LinkSwitch: Comparison Summary
PARAMETER
LINEAR
LinkSwitch
Output Specification
2.7 W, 9 V
2.7 W, 9 V
BOM Cost
1×
1×
Input Voltage
98 to 132 VAC
85 to 265 VAC
Full Load Efficiency (115 VAC)
53%
75%
No Load Input Power (115 VAC)
1.6 W
200 mW
Annual Energy Cost (2.7 W load)
$ 5.34
$ 3.8
Annual Energy Cost (no-load)
$ 1.68
$ 0.22
Short-circuit Current
2.3 A
50 mA
Short-circuit Protection
One time thermal
fuse
Self-resetting
Auto-restart
Weight
9.4 oz / 267 g
2 oz / 56 g
Volume
11 in / 176 cm
Shipping Cost by Sea (per unit)
1 × (reference)
0.4 ×
Shipping Cost by Air (per unit)
10 ×
4×
3
3
3
2.45 in / 40 cm
3
PI-3214-092202
38
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• The annual LinkSwitch Energy savings, at either full or no-load, exceed the cost of
the entire power supply
• A significant portion of overall linear adapter cost is involved in shipping it. The
lighter-weight and smaller size of LinkSwitch based adapters reduces shipping costs.
• The cost comparisons are referenced to that of shipping a linear adapter by sea (1X)
Improving CV Tolerance with Opto Feedback
•
Tolerances achievable in low cost, high volume
manufacturing
– +10% voltage tolerance with Zener reference (VR1)
– < +5% voltage tolerance with IC reference (TL431)
– +20% current limit tolerance
(dominated by transformer inductance tolerance*)
– Includes the variations of the LinkSwitch, other
components and the operating temperature range
*Primary inductance tolerances must be ≤ ±10% for these figures to be valid
39
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• Ideal for replacing regulated linears or discrete switching supplies (RCCs)
• CC tolerances can be improved by reducing the primary inductance tolerances.
(See the Application Example section, for tips on how to improve CC tolerance)
• R5 is only required for output voltages > 6 V, to limit opto-LED current. For outputs
<6 V, the slope resistance of VR1 is typically sufficient to perform this function.
Designing with LinkSwitch
40
Seminar_lowpower_100102_screen_102102
Specifying a LinkSwitch Design
•
•
•
41
A CV/CC (charger) supply is
specified at the typical
constant output current
Maximum output
current
A CV (adapter or auxiliary)
supply is specified to deliver a
minimum full load output
current
Typical output
current
Minimum output
current
LinkSwitch design procedure
assumes CV/CC
– To design for a CV adapter
increase full load output
current by 20% to ensure full
load current delivery with
worst case design
PI-3090-081302
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Step by Step Design Process
1. Select VOR
2. Calculate secondary component voltage drops
3. Calculate transformer turns ratio
4. Calculate output power
5. Calculate primary inductance
6. Design transformer
7. Select component values
8. Build prototype
9. Refine design
42
•
All covered by Application Note AN-35 LinkSwitch Design Guide
•
Supported by Design Spreadsheet as part of PI Expert
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Definition of Components & Parameters
Secondary side loss
components
•
All secondary side voltage drops and power losses must be accounted for
43
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• RLF is the leakage inductance filter resistor - improves CV characteristics
• RLF ~100 Ω works well for typical transformer design
Step 1: Select a Value for the
Reflected Output Voltage (VOR)
44
•
VOR determines the feedback voltage (VFB)
– For no-load consumption <300 mW, VOR should be between 40 – 60 V
– VOR > 60 V may be used, if higher no-load is consumption is acceptable
– Higher VOR also increases the output power capability of the design
•
For initial design set VOR = 50 V
– Default value in design spreadsheet
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• For a universal input supply, setting VOR to 50 V usually gives the best compromise
between the no-load power consumption and the maximum available output power
• A low value of VOR keeps the peak drain voltage of the LinkSwitch at a value that is
lower than that of a standard switching power supply. If the voltage rating of the
input capacitor is sufficient, a LinkSwitch design can operate safely during an input
over-voltage condition, such as a line surge or voltage swell
Step 2: Calculate Secondary Voltage (VSEC)
VSEC = VO + VRCABLE + VDOUT + VRSEC
•
VDOUT and VSEC defined at peak secondary current
•
If no better measurements available use estimates shown
VRSEC = RSEC × ISEC(PEAK)
VRCABLE = IO × RCABLE
0.15 Ω
0.7 V/ 1.1 V
0.3 Ω
ISEC(PEAK) ≈ 4 × IO
ISEC(RMS) ≈ 2 × IO
VO at nominal peak
output power point
PI-3095-090402
45
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• The peak and RMS secondary current estimates are valid for output voltages near 5 V.
Lower output voltages will require higher values.
• The peak VOR determines the feedback voltage: VDOUT and VSEC are determined at the
peak secondary current
• VDOUT (at a peak output current of roughly four times the rated IO):
– A typical Schottky diode forward voltage drop is about 0.7 V
– A typical ultra-fast diode forward voltage drop is about 1.1 V
Step 3: Calculate Transformer Turns Ratio
NP
V
= OR
NS VSEC
=
46
50
VSEC
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Step 4: Calculate Power Processed by
Transformer PO(EFF)
PO(EFF) = PO + PCABLE + PDIODE + PBIAS + PS(CU) + (PCORE/2)
PCORE = 0.1 W
PS(CU) = I2SEC(RMS) × RSEC
PBIAS = IDCT × VOR
0.15 Ω 0.7/1.1 V
= 2.3 mA × 50 V
PCABLE = IO2 × RCABLE
0.3 Ω
= 0.115 W
ISEC(PEAK) ≈ 4 × IO
ISEC(RMS) ≈ 2 × IO
47
PDIODE = VDOUT × IO
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• In PI-Expert, the design spreadsheet calculates all of the above parameters, including
accurate core losses, which are based on specific core part numbers and geometries
• PCORE is divided by two, since only the core loss that occurs during the transfer of
energy to the secondary needs to be considered
• Power loss in RLF is negligible, and can be ignored
• For a more accurate PDIODE calculation, use an average voltage drop, if it is known
Step 5: Calculate Primary Inductance
•
LP is the transformer primary inductance
LP ( NOM ) =
=
2 × PO ( EFF )
(I
2
P
× fs
)
2 × PO ( EFF )
2710
× ∆L
× 1.03
– LP tolerance ≤ +10% to meet ≤ +20% CC tolerance
48
•
Use the I2f parameter (specified in the LinkSwitch datasheet)
– Combines the tolerances of both the current limit and the switching frequency
– 2710 A2Hz specifies the nominal primary inductance at the peak power point
•
The term ∆L compensates for non-ideal ferrite material
– Inductance falls slightly a flux density increases
– ∆L values of 1 to 1.05 are typical for low-cost ferrite materials
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• The I2f coefficient is specified in the LinkSwitch datasheet with a tolerance of ±6.2%
• I2f is a useful parameter since LinkSwitch based supplies are designed to always
operate in the discontinuous conduction mode. Therefore, output power is directly
proportional to this term (P = 0.5 • L • I2f)
Step 6: Design the Transformer
•
Secondary turns Ns
– For an initial estimate, use 2.5 turns per volt (of output voltage)
– If the flux density is too high, increase the number of turns (both NP & NS)
NS ≈ VSEC × 2.5
•
Primary turns Np
NP ≈
49
VOR
× NS
VSEC
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• The flux density calculation is covered on the next slide
Step 6: Design the Transformer (cont.)
•
Calculate flux density
– Flux density < 3300 gauss (330 mT)
BM (gauss ) =
•
100 × IP ( A ) × LP (µH)
NP × A e (cm 2 )
Calculate gap size
– Transformer manufacturer can calculate Lg more accurately for a given core
material
2
2
A L (nH / t 2 ) × L e (cm) L (mm) =  4π × NP × A e (cm ) − L e (cm) × 10 
µr =
g


µr
4π × A e (cm 2 )

 LP (µH) × 100
•
Gap limits required to maintain a primary inductance tolerance of < ±10%
– Single (center) leg gap >0.08 mm (accomplished by grinding down the center leg)
– A gap in all legs >0.05 mm (all 3 legs of an EE core are separated by plastic film)
50
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• Film gapping may provide tighter primary inductance (LP) tolerances (+7%)
– check with your magnetics vendor
• If an EE13 core is used, the guideline in Step 5 will allow these requirements to be
met
• (Lp is primary inductance in µH, Lg is core center leg gap in mm, Le is core effective
path length in cm and Ae is core effective area in cm2)
• When film gapping, use film of ½ the gap length: Example, if 0.05 mm is the total gap
length, 0.025 mm thick film is inserted between all legs of the core
Step 7: Clamp, Bias and Feedback Components
CCLAMP:
0.1 µF, 100 V, 20% - FILM
Ceramic not recommended
CCP:
Battery load: 0.22 µF, 10 V
Resistive load: 1 µF, 10 V
R FB =
VFB − 5.75 V
2.3 mA
VFB = VOR + VLEAK
PRFB ≈ 0.1 W, use 1/4 W, 1%
≈ VOR + 5 V
DCLAMP:
1N4937 or UF4005
1 A, 600 V, trr<200 ns
1N400x not recommended
VLEAK≈ 5 V
RLF: 100 Ω, 1/4 W, 5%
51
Lowpwr 022404
• CCP: To allow sufficient time for start-up into a resistive load a 1 µF capacitor should
be used
• CCLAMP: The value of low cost ceramic capacitors vary with temperature and applied
voltage and may cause output oscillation
• (RLF = Leakage Filter Resistor)
• IDCT = 2.3 mA at peak power point
• VLEAK: Note that this is not a real circuit component. It represents the voltage error in
the value of VFB due to the transformer leakage energy at full load / output peak power
point.
Step 8: Select Input Components
Σ(C1,C2): 85 to 265 VAC input = 3 µF/W, 400 V
195 to 265 VAC input = 1 µF/W, 400 V
L1: Low cost discrete
inductor for EMI
filtering. A resistor can
be used at ≤1.5 W for
lower cost
RF1: Flame proof
fusible resistor,
wire wound,
10 Ω, 1-2 W or
fuse.
Half wave rectification
can be used at <1.5 W
for lower cost
52
Lowpwr 022404
• RF1 should be a fusible, flameproof type (during failure it must not emit incandescent
material that may damage transformer insulation)
• Metal film resistor not recommended due to insufficient instantaneous power
capability (repeated inrush at high line causes failure)
• A resistor substituted for L1 in the EMI filter should be fusible and flameproof type
• Typically C1 and C2 have the same value
• Input capacitor values below 4.7 µF will typically reduce differential surge capability
from 2.5 kV. Verify required surge withstand before selecting small values of C1 and
C2.
Step 9: Refine Design
53
•
Build prototype using nominal primary inductance
•
Verify output VI characteristic
– If necessary adjust RLF and RFB to give desired output voltage at peak
power point
•
If nominal CC is different from design target recalculate LP based
on measured parameters on prototype:
– Secondary winding resistance
– Actual secondary peak and RMS currents
– Diode forward voltage at peak secondary current
– Output cable resistance
– Feedback voltage VFB
•
Build next iteration and verify
Lowpwr 022404
Design Tools
•
AN-35
– LinkSwitch Design Guide
•
DAK-16A
– Includes tested EP-16A board
– Engineering Report (EPR-16A)
– Data sheet and device samples
– Blank PC Board
•
Design Ideas
– DI-18, DI-19, DI-58, and DI-59
•
PI Expert
– Version 5.0 for PIXls design
spreadsheet
54
Lowpwr 022404
LinkSwitch Hints and Tips
55
Seminar_lowpower_100102_screen_102102
Design Hints and Tips Contents
•
•
•
56
Optimizing output CV/CC
characteristics
– Effect of output diode choice
– Compensating for transformer
leakage inductance
– Limiting no-load output voltage
– Effect of output cable resistance
Transformer design
considerations
– Minimum gap size
– Gap Uniformity
•
Layout considerations
•
Measurement techniques for
switching waveform and VLEAK
•
Output filter selection
•
Using larger VOR for higher power
•
Tighter CV tolerance with opto
feedback
•
Specifying a LinkSwitch Design
•
Estimated Manufacturing
Tolerances
Minimizing no-load consumption
– Minimizing transformer capacitance
– Minimizing external capacitance
– Selecting lower VOR
Lowpwr 022404
Effect of Output Diode on CC Linearity
8V
1 A, 60 V Schottky
(11DQ06)
6V
1 A, 100 V Ultra Fast
(UF4002)
1 A, 100 V Fast
(1N4934)
4V
PI-3507-051303
2V
400 mA
57
600 mA
500 mA
•
Schottky and ultra fast (trr=50 ns) diodes give the best CC linearity
•
Fast recovery PN diodes (trr=150 ns) cause CC region to bend outwards
– Caused by slower diode forward recovery
– Designs using fast diodes may not meet +/-20% CC tolerance
Lowpwr 022404
• The primary feedback resistor has been adjusted for each diode type to achieve
similar peak power output voltage
• The increased forward voltage drop, during the forward recovery time of the fast
diode, increases the primary clamp feedback voltage at a given output voltage. The
LinkSwitch internal current limit is therefore higher for a given output voltage and the
CC characteristic bends outwards.
Effect of High Leakage Inductance LLEAK
Specified peak
power point
Lower peak power point
due to high leakage
58
•
Poorer CV regulation
•
Moves the CV/CC transition down the peak power curve
•
Causes a slight “bowing out” of the CC region
Lowpwr 022404
• (Leakage energy degrades the tracking of VOR (VFB) with VO)
Increasing RFB to Compensate for High LLEAK
Peak power point
after increasing RFB
Peak power point
before increasing RFB
59
•
Moves CV/CC transition up peak power curve
•
Does not improve CV or CC regulation
•
Increases no-load consumption
Lowpwr 022404
• The peak power curve shown corresponds to the product of the maximum output
current and voltage of a discontinuous mode flyback. As the output voltage changes,
the output current changes to maintain this product constant.
Benefits of Using RLF: Better CV/CC
Characteristics
Actual Peak
Power Point
(9.2 V, 290 mA)
Output Voltage (V)
12
9
100 Ω RLF
Specified Peak
Power Point
(9 V, 300 mA)
Without RLF
Without RLF
meets power
curve at lower
voltage
6
PI-3508-051303
3
0
0
100
200
300
400
Output Current (mA)
•
RLF filters leakage voltage, improving CV/CC characteristic and decreases
zero load voltage and consumption
60
Lowpwr 022404
• RLF also reduces EMI caused by DCLAMP
• Chart shows that without RLF, RFB would need to be increased for the output to meet
the specified peak power point. This would cause both the no-load voltage and
consumption to increase.
Effect of High Cable Resistance
High output cable resistance causes larger output drop with load
Power loss in cable
reduces effective peak
power curve at end of
cable
CC point unchanged
61
•
Poorer CV regulation
•
Lower Overall Efficiency
Lowpwr 022404
• Since the output current of the LinkSwitch is limited, once the device transitions from
the CV portion of the output VI curve, past the peak power point (onto the CC portion
of the output VI curve), the more power that is dissipated in the cable resistance, the
lower the voltage will be, at the end of the cable
Small Pre-load Reduces No-load Voltage
15
No pre-load
1 mA pre-load
2 mA pre-load
(B)
12
(C)
Pre-load
resistor
No-load at 265 VAC
(A) 250 mW
(B) 268 mW
(C) 279 mW
PI-3509-051303
Output voltage (V)
(A)
9
0
4
8
12
Output Current (mA)
•
62
Secondary peak charging causes output voltage to rise at no-load
– Small pre-load reduces no-load output voltage by > 1 V
– Minimal (~20 mW) increase in no-load consumption
Lowpwr 022404
•
Minimum gap size recommendation
– Recommendations based on ±10% LP tolerance
– Center leg gapping: ≥ 0.08 mm
– Film gapping: ≥ 0.05 mm
– Verify with magnetics vendor
•
Ensure gap is uniform
– Uneven gapping makes CC portion non-linear
– Verify by measuring di/dt of transformer
current waveform
Uneven gapping
PI-2961-073102
Transformer Gapping
changes primary
current gradient
63
Lowpwr 022404
• The increase in transformer di/dt only occurs when the current is near the peak. This
phenomena is due to the crowding of magnetic lines of flux, in the core, near the
narrowest part of the uneven gap, and means that saturation is being approached.
• To measure di/dt of transformer current waveform, feed power supply from DC source
or use large input capacitor (100 µF). Monitor current using a current probe.
• 0.05 mm is total gap size i.e. the tape or spacer thickness is 0.025 mm between all
legs of the EE cores
Minimizing No-load Consumption
64
•
40 V ≤ VOR ≤ 60 V
– 40 V will give the lowest consumption
•
Minimize switching node capacitance
– Remove snubbers on LinkSwitch and output diode
– Use double coated/heavy nyleze/L2 magnet wire for primary winding
•
Do not vacuum impregnate transformer
– Varnish increases primary capacitance ~ 5x
– Dip varnishing does not increase capacitance significantly
Lowpwr 022404
• VOR below 40 V limits output power capability
Battery Loads Do Not Require Output π Filter
Output Voltage Ripple
Output Voltage Ripple
162 mV pk-pk
With Battery or Battery Model Load
With Resistive Load
•
65
5 mV, 2 ms/div
PI-3510-051303
50 mV, 2 ms/div
PI-3505-051303
12 mV pk-pk
Battery acts as a filter capacitor
Lowpwr 022404
• Measured with x1 probe at end of output cable with parallel 0.1 µF and 1 µF capacitors
and 20 MHz bandwidth
• (Apparent high frequency modulation on falling slope of waveforms is due to digital
oscilloscope aliasing)
Effect of Output π Filter
•
Without π filter
146 mV pk-pk
(line+switching ripple)
50 mV, 2 ms/div
PI-3511-051403
•
With π filter
84 mV pk-pk
(line+switching ripple)
50 mV, 2 ms/div
PI-3513-051403
66
132 mV pk-pk
(switching ripple)
50 mV, 20 µs/div
PI-3512-051403
46 mV pk-pk
(switching ripple)
50 mV, 20 µs/div
PI-3514-051403
Lowpwr 022404
• π filter reduces switching ripple
• (Results taken from EP-16, C4: 470 µF / 10 V, L1: ferrite bead, C5: 100 µ F / 10 V)
Improving CV Tolerance with Optocoupler
•
±2% reference including temperature provides ± 5% CV tolerance
– The sense voltage (VOPTO+VREF) sets the nominal specified output voltage
67
Lowpwr 022404
• Typically R1=R3=RFB/2
• Increasing R3, while keeping R1+R3=RFB, increases loop gain & improves CV
regulation.
• Maximum value of R3 limited by opto transistor dissipation
• For typical transformer leakage inductance values R2 (RLF) is 100 Ω
• C2, C3 typically 0.1 µF, 50 V. C3 provides DC voltage for optocoupler
• R4 biases VR1 close to its specified test current; a value of 200 Ω provides ~5 mA.
• R5 may be required for Zener voltages above 5 V and for TL431 designs, to limit LED
current and ensure stability. Values in the range 22-68 Ω are typical.
• High CTR optocoupler (200-400%) improves CV regulation, if required.
• See Application Examples section for more information.
• Optocoupler is connected to primary return (non-switching side of D1), to reduce
common mode EMI, which would result if connected to the switching side of D1.
• Swapping the positions of D1 and R2 will improve EMI, as R2 would no longer see the
switching waveform at the cathode of D1.
Designing for Optocoupler Feedback
Peak output power curve
Inherent output
characteristic without
opto coupler feedback
Inherent CC to CV
transition point:
VO(NO_OPTO)
±5%
Output characteristic
with opto coupler
feedback: VO(OPTO)
Tolerance
envelope
without opto
– The Linkswitch circuit should be designed for a nominal inherent (without opto)
peak power point voltage that is 5% above the nominal specified voltage
– Example: 5 V output specification,
VF(OPTO)+VREF=VO(OPTO)=5V, VO(NO_OPTO) for LinkSwitch design = 5.25 V
68
Lowpwr 022404
Opto CC Behavior during Bench Test
Peak output power curve
Inherent characteristic
without opto feedback
CV Control
Bench Testing:
When load is increased,
CC operation is only
entered into after
reaching peak power
curve.
Normal
Operation:
As battery voltage
rises, output
current does not
exceed CC value
•
69
CC Control
During charging, only rising CC characteristic is followed
– ±20% Output CC tolerance is still maintained with opto-coupler feedback
– Falling characteristic only seen during lab testing
Lowpwr 022404
Operation during Normal Battery Charging:
• As the battery charges, IO is under CC control, as the output voltage rises
• When VO reaches the feedback threshold (set by the secondary sense circuit), the
opto provides feedback, and the LinkSwitch transitions to CV mode (PWM) control
Operation observed in laboratory bench testing:
• As the load is increased, the output voltage falls when the peak power point is
reached. This reduces the current through the secondary sense circuit, which
reduces the CONTROL pin current. This reduces the internal current limit of the
LinkSwitch, which further reduces the output voltage (positive feedback) and
transitions the output into CC control mode
• Therefore, a slight overshoot in IO may be observed in bench testing (as depicted in
the slide), as the load is increased [This will not occur in normal battery charging]
• This effect can be eliminated by setting the sense voltage to 10% above the inherent
peak power point voltage
• See the LinkSwitch data sheet for more information
Estimated Manufacturing Tolerances
•
Complete analysis of tolerance calculations is provided in AN-35
LinkSwitch Design Guide
•
±20% Overall estimated CC tolerance for a 3 W design
– Includes all device, external component and temp. variations (Tj: 25°C to 65°C)
•
Transformer tolerance dominates CC variation
– I2f coefficient tolerance ±6% is the second most dominant
•
At lower power CC tolerance is slightly higher (~± 22% at 1.5 W)
•
± 10% CV tolerance due to the following variables
– Finite gain of LinkSwitch
– Feedback resistor tolerance
– CONTROL pin voltage tolerance
– Output diode forward drop variation
70
Lowpwr 022404
Higher VOR for Higher Output Power
•
VOR>60 V increases power capability for open frame designs
– 4.5 W (Universal) with 100 VOR, no-load ~500 mW at 265 VAC (see Note 1)
– 5 W (230 VAC ±15%) with 80 VOR, no-load ~450 mW at 265 VAC (see Note 1)
– Output power above these levels limited by thermal dissipation constraints
•
Design must still remain fully discontinuous
– Continuous mode operation with LinkSwitch can cause instability
•
Useful for designs that can accommodate the increased no-load consumption
Note 1: Ambient temperature must be maintained at a temperature that assures that the device
case temperatures do not trip thermal shutdown
71
Lowpwr 022404
• Higher VOR allows higher duty cycle, increasing power capability
Single Point Failure Safety Testing
CCP
Shorted: LinkSwitch stops, PASS
Open: Auto-restart, PASS
LinkSwitch
DRAIN Open: LinkSwitch stops, PASS
CONTROL Open: LinkSwitch stops, PASS
Low cost 0.01 µF, 100 V ceramic capacitor
added in parallel to CCLAMP
DOUT or Secondary winding
Shorted: Auto-restart, PASS
Open: No output, PASS
RFB
Shorted: VO low, PASS
Open: Auto-restart, PASS
COUT
Shorted: Auto-restart, PASS
Open: Poor CV, PASS
DCLAMP
Shorted: Input fuse opens, PASS
Open: Auto-restart or fuse opens, PASS
RLF
Shorted: Poor CV, PASS
Open: Auto-restart, PASS
CCLAMP with 2nd capacitor fitted
Shorted: VO low, PASS
Open: No-load Vo increases, PASS
•
Primary Winding
Shorted: No effect or input fuse opens, PASS
Open: Supply stops, PASS
LinkSwitch meets single point failure testing with one additional capacitor
72
Lowpwr 022404
• Shorting of DRAIN to SOURCE pin not required as creepage and clearance of 2.9 mm
between pins meets agency requirement (>2.5 mm) with correctly designed PC board.
Correct Scope Drain Voltage Measurement
•
Connect scope ground to the DRAIN pin / high voltage DC rail
– Do not connect scope ground to SOURCE pin: excess capacitance falsely
triggers current limit
– Invert scope input to display normal VDS waveform
– Unit under test must be powered from an isolation transformer
Isolation
Transformer
73
Lowpwr 022404
Measuring VFB
•
Connect battery powered DVM directly across CCLAMP
– Sufficient common mode rejection of source switching node to measure VFB directly
57.5
VDC
74
Lowpwr 022404
PC Board Layout Considerations
Place CONTROL pin
capacitor close to device
SOURCE is the switching node only use sufficient copper area
for heat sinking to minimize
radiated EMI
Missing pin
maximizes board
creepage distance
Input capacitor
placed to shield
input filter inductor
(not shown)
Small secondary
loop minimizes
leakage
inductance
75
Primary Return used as
electrostatic shield to reduce EMI
Keep secondary components
away from primary side to
reduce EMI
Lowpwr 022404
LinkSwitch Applications Examples
76
Seminar_lowpower_100102_screen_102102
Applications Examples
•
2.75 W, Universal input charger (DI-18)
– 5.5 V / 500 mA, CV/CC
•
1.5 W, Universal input charger (DI-19)
– 5.5 V / 270 mA, CV/CC
•
2.7 W, Universal input adapter
– 9 V / 300 mA, CV
•
1 W, Universal input, portable audio charger
– 1.5 V / 700 mA, CV/CC
•
2.6 W, Universal input with opto feedback (DI-44)
– 5.2 V / 500 mA, CV/CC
•
4.8 W, 230-375 VDC input, standby / auxiliary supply
– 12 V / 400 mA, CV
DI=Design Idea
77
Lowpwr 022404
• The latest Design ideas from Power Integrations can be found at
www.powerint.com/appcircuits.htm
2.75 W Charger Specification (DI-18)
Input Voltage
Output CV/CC Specification
VALUE
10
85-265 VAC
9
Output Voltage
5.5 V
Output Current
500 mA
Output Power
2.75 W
Efficiency
>70%
3
<300 mW
2
No load
8
VO (V)
7
6
5
4
PI-3516-051403
DESCRIPTION
1
Conducted EMI
Surge
CISPR22B/
EN55022B
0
0
200
300
400
IO (mA)
EN1000-4-5
Class 3
PI-3212-091802
78
100
Lowpwr 022404
500
600
700
2.75 W Charger Schematic
Full wave
rectification cost
effective >~1.5W
Meets EN55022B/CISPR22B with no Y
capacitor. Lower cost resistive π filter
possible with lower efficiency
PN diode possible
for lower cost with
lower efficiency
3.3uF can save cost but with lower
differential surge withstand rating (<2.5 kV)
79
Lowpwr 022404
• Resistive π filter reduces efficiency ~10%
• Half wave rectification above ~1.5 W output powers requires larger input capacitors
• (Primary is split as part of primary winding is configured as a shield. This reduces
primary to secondary common mode currents and therefore conducted EMI)
2.75 W Charger CV/CC Output Characteristic*
10
Output Voltage (V)
9
Vin=85V
Vin=115V
Vin=185V
Vin=265
8
7
6
5
4
3
PI-3517-051403
2
1
0
0
*Measured
at the end
of the output cable
80
100 200 300 400 500 600 700
Output Current (mA)
Lowpwr 022404
Output Characteristic* of 100 Randomly
Selected (2.75 W) Charger Samples
265 VAC
132 VAC
Auto-restart
IOUT (100 mA/div)
PI-3518-051403
VO (2 V/div)
85 VAC
*Measured at the end of the output cable
• These results show that the CC portion of the output curve could be
better “centered” to optimize the manufacturing yield
81
Lowpwr 022404
• “Centering” would require lowering the transformer primary winding inductance
2.75 W Charger Efficiency
80
70
INPUT VOLTAGE
NO-LOAD
INPUT POWER
85 VAC
193 mW
40
115 VAC
210 mW
30
185 VAC
219 mW
20
230 VAC
251 mW
265 VAC
274 mW
Vin=85V
Vin=115V
Vin=185V
Vin=265V
50
PI-3519-051403
Efficiency (%)
60
10
0
0
100
200
300
400
500
PI-3249-091802
600
Output Current (mA)
82
•
High efficiency (71%) due to no current sense losses
•
EcoSmart: easily meets <300 mW no-load consumption
Lowpwr 022404
EP-16A PC Board Layout
1.7 x 1.1 inches (43 x 28 mm)
•
83
Low cost (CEM1) single sided board
– No surface mount components required
Lowpwr 022404
2.75 W Charger Thermal Performance
RFB
LinkSwitch
Output Diode
•
High efficiency operation reduces the dissipation of the LinkSwitch
– The absence of a secondary current sense resistor reduces the power,
that has to be processed by the transformer, by up to ~1 W
– This also reduces the temperature rise within the charger/adapter enclosure
(the enclosure’s ambient temperature only rose 15°C above the external ambient)
•
Minimal SOURCE copper-area is needed to heatsink the LinkSwitch
– The LinkSwitch temperature rose <25°C above the ambient within the enclosure
– Minimizing the area of copper connected to the switching node reduces EMI
84
Lowpwr 022404
• A typical discrete switching supply’s sense resistors drop a total of 1.3 secondaryside volts in the process of driving an NPN transistor and an opto-coupler LED. With
an operating efficiency of 70% and at an output current of 0.5 A, that represents a loss
of 0.65 W of output power, which requires an additional 0.93 W of input power
2.75 W Charger EMI Performance
QP
AV
QP
PI-3520-051403
AV
QP
CISPR22-B / EN55022 B
FCC B
Measured with artificial hand connected to output return
85
Lowpwr 022404
• (EMI shown with output return connected to artificial hand connection of LISN. This
degrades EMI results by providing a capacitive current path to earth ground. EMI
results without artificial hand connected are better than shown above).
PI-3522-051403
QP
2.75 W Charger Summary
86
•
Cost competitive even with
unregulated linear transformer
based chargers with much better
performance (CV/CC)
•
A low parts count solution
•
Small size and light weight
•
High efficiency 71%
•
Meets worldwide standby energy
requirements
•
Meets worldwide EMI standards
•
Fully fault protected from…
– short circuits or open feedback loops
(by its integrated auto-restart function)
– over heating (by its auto-recovering,
hysteretic thermal shutdown function)
Lowpwr 022404
• This 2.75 W charger is available in Design Accelerator Kit DAK-16A. The DAK
includes device samples, a second (blank) PCB, and full design documentation
1.5 W, 5.5 V Charger Schematic (DI-19)
Low cost resistive π filter
meets EN55022B/CISPR22B
Half wave rectification
for low cost, two
diodes used for EMI
gating and surge
withstand
87
2.2µF for low cost but
lower differential surge
withstand (~1 kV)
Lowpwr 022404
Only 1 A diode required
due to secondary CC PN diode for lower cost
1.5 W, 5.5 V Low Cost Charger Performance
(DI-19)
•
Half wave input rectification
•
Low cost resistive π filter
•
Efficiency > 62 %
•
No-load consumption
– 219 mW at 115 VAC
– 282 mW at 265 VAC
88
10
85 VAC
265 VAC
9
8
7
6
5
4
3
PI-3523-051403
Universal Input, 5.5 V, 270 mA
output
– 100/115 VAC only design can
lower input capacitor costs
Output voltage (V)
•
2
1
0
0
50
100
150
200
250
Output Current (mA)
Lowpwr 022404
300
350
2.7 W, 9 V Adapter Schematic
1 µF electrolytic CONTROL
pin capacitor for start-up
into resistive loads
2.2 uF can save cost but with lower
differential surge withstand (~1 kV)
89
Lowpwr 022404
2 mA pre-load to reduce
no-load output voltage
Only 1 A diode required due
to secondary CC
2.7 W, 9 V Adapter Performance
•
Efficiency >73%
•
No-load input power
– 222 mW at 85 VAC
– 280 mW at 265 VAC
15
85 VAC
265 VAC
12
9
6
PI-3523-051403
Universal Input, 9 V, 300 mA
nominal output
– 100/115 VAC only design can
lower input capacitor costs
Output voltage (V)
•
3
0
0
100
200
300
Output Current (mA)
90
Lowpwr 022404
400
1 W, 1.5 V Portable Audio Charger Schematic
Low cost resistive π filter
meets EN55022B/CISPR22B
Pre-load to reduce
no-load voltage
2.2 µF for low cost but lower
differential surge withstand (~1 kV)
Half wave rectification for
low cost, 2 diodes for EMI
gating and surge withstand
91
Schottky diode used for high
efficiency with low output voltage
Lowpwr 022404
1 W, 1.5 V Portable Audio Charger Performance
No-load input power
– 235 mW at 110 VAC
– 264 mW at 265 VAC
3
85 VAC
265 VAC
2.5
2
1.5
1
PI-3524-051403
•
Universal Input, 1.5 V, 700 mA
output
– 100/115 VAC only design can
lower input capacitor costs
Output voltage (V)
•
0.5
0
0
100 200 300 400 500 600 700 800
Output Current (mA)
92
Lowpwr 022404
2.6 W, 5.2 V Accurate CV Charger
(with Opto-coupled Feedback)
R5 biases VR1 at
its test current
RFB was split
into R1 and R4
A Schottky diode was
used for high efficiency
Increasing R4 can improve
regulation, but is limited
by the opto-transistor’s
dissipation rating
93
The opto-coupler regulates the output
voltage by setting the voltage across R4 and
C5, which adjusts the CONTROL pin current
Lowpwr 022404
• For higher output voltages (i.e., lower Zener impedance) or when using a reference IC
(such as a TL431), a series resistor may be required to limit the opto-LED current
• C5 can be a ceramic capacitor, to keep costs low
2.6 W, 5.2 V Accurate CV Charger Performance
6
•
Efficiency >68%
•
No-load input power
– 167 mW at 85 VAC
– 220 mW at 265 VAC
5
4
85 VAC
3
265 VAC
2
PI-3525-051403
Universal Input, 5.2 V, 500 mA output
– 5.2 V ±7% at terminals
– 5.2 V ±8% at end of cable
– 100/115 VAC only design can lower
input capacitor costs
Output voltage (V)
•
1
0
0
100
200
300
400
500
600
Output Current (mA)
(Measured at end of 0.2 Ω cable)
94
Lowpwr 022404
• Output voltage regulation figures include line regulation (+1%), load regulation
(+2.3%), Zener tolerance (+2%) and Zener temperature coefficient for 50 °C temperature
range (+1.7%).
4.8 W, 12 V Auxiliary Supply Schematic
1 µF electrolytic CONTROL
pin capacitor, for starting
up into a resistive load
Low-pass output filter reduces
(resistive load) output ripple
Pre-load reduces
no-load output voltage
95
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• This circuit is ideal for auxiliary supplies in white goods and home appliances
4.8 W, 12 V, Auxiliary Supply Performance
20
230 VDC to 375 VDC input, 12 V,
400 mA nominal output
•
80 VOR
•
Efficiency >78%
•
No-load input power
– 390 mW at 230 VDC
– 456 mW at 375 VDC
250 VDC
375 VDC
18
Output voltage (V)
•
16
14
12
10
8
6
PI-3526-051403
4
2
0
0
100
200
300
400
500
Output Current (mA)
96
Lowpwr 022404
600
LinkSwitch: Switcher Benefits at Linear Cost
•
Universal input, CV/CC regulated output operation
– Higher performance than unregulated linear supplies
– A single design works worldwide, which simplifies inventory logistics
•
Smaller Size and Weight
– Lower shipping costs for both supplier and OEM
– High tolerance to mechanical shock – easily passes drop testing
– The power supply matches the state-of-the-art product it powers
– End user convenience – doesn’t block multiple outlets
•
EcoSmart
– High operating efficiency
– Low standby power consumption
– Meets all worldwide standards
•
Self-Resetting Fault Protections
– Fully protected from over heating,
short-circuits and open feedback loops
97
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Linears Will Be Converted
LinkSwitch
Enables Cost Effective Conversion of
Up to 1 Billion Linears Built Annually
Today!
98
Seminar_lowpower_100102_screen_102102
Designing Low Power EcoSmart
Switchers using
TinySwitch and TinySwitch-II
99
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Agenda
100
•
Why TinySwitch Technology?
•
Choosing TinySwitch-II vs TinySwitch
•
Operation
•
Designing with TinySwitch Technology
•
Application Examples
•
Hints and Tips
•
Summary
•
Questions and Answers
Lowpwr 022404
Why TinySwitch Technology
Most energy efficient
– <10 mW no-load consumption at 230 VAC
•
FREQUENCY
JITTER
LINE UV
DETECTION
4W
6W
132
Y
Y
Y
TNY266 P or G
10 W
15 W
6W
9.5 W
132
Y
Y
Y
TNY267 P or G
13 W
19 W
8W
12 W
132
Y
Y
Y
TNY268 P or G
16 W
23 W
10 W
15 W
132
Y
Y
Y
TinySwitch
230 VAC ±15%
230 VAC ±15%
OPEN
FRAME
9W
OPEN
FRAME
5.5 W
TinySwitch-II
ADAPTER
TNY264 P or G
PRODUCT
AUTO RESTART
CONTINUOUS
OUTPUT POWER
Very simple low cost circuit
– ON/OFF regulation
SWITCHING
FREQUENCY
(kHz)
CONTINUOUS
OUTPUT POWER
ADAPTER
•
85-265 VAC
85-265 VAC
TNY253 P or G
4W
2W
44
TNY254 P or G
5W
4W
44
TNY255 P or G
10 W
6.5 W
130
PI-3238-082902
101
Lowpwr 022404
Continuous Output Power Rating Terms Defined:
• ADAPTER – the power supply is in a non-ventilated, close-quarters enclosure, and is
delivering a continuous output power [the rating] while the enclosure is in an ambient
environment that is at 50 °C (outside the enclosure)
• OPEN FRAME – the power supply has adequate heat sinking on the PI device and is
subject to some convective air flow, and is delivering a continuous output power
[the rating]
• All power ratings in the above table and on the PI device data sheets are for
continuous power delivery (peak power or periodic pulsed power ratings are not
given, nor dealt with on this slide). Short-term peak power capabilities will be higher,
and will be limited only by the maximum current limit of the device in question
• See the PI datasheets for more details
• The TNY253, TNY254, and TNY255 devices target specific very-low-power
applications, and are therefore not rated for open frame designs
Choosing TinySwitch-II vs TinySwitch
•
TinySwitch-II is the best choice for most applications
– Enhanced features lower system cost
– Applications up to 23 W (230 VAC), 15 W (85-265 VAC)
– <30 mW no-load consumption at 230 VAC (with bias winding)
– <300 mW no-load consumption at 230 VAC (without bias winding)
•
TinySwitch is the recommended choice for applications requiring:
– <10 mW no-load consumption at 230 VAC (using bias winding)
– <100 mW no-load consumption at 230 VAC (without bias winding)
– Low video noise, such as analog TV Standby circuits: the 44 kHz (versus the
132 kHz of TinySwitch-II) switching frequency allows the MOSFET Drain
node to be heavily snubbed, to suppress EMI noise generation
102
Lowpwr 022404
• The TinySwitch-II will still offer superior system cost benefits in TV standby circuits, if
heavy Drain-Source snubbing is not necessary to meet EMI noise requirements
• Both TinySwitch-II and TOPSwitch-GX based circuits can be configured for no-load
power consumption of under 100 mW
Operation
103
Lowpwr 022404
TinySwitch Regulates by ON/OFF Control
Enable signal sampled each cycle
•
The MOSFET drain current ramps to a fixed current limit every ON cycle
– Each ON cycle processes a fixed (maximum) amount of energy
– Cycles are disabled (OFF cycles) as necessary, to maintain output regulation
– The effective switching frequency reduces proportionally with load reduction
– Requires transformer gluing to minimize audible noise at light load conditions
•
Maximum energy per cycle ensures lowest no-load frequency/consumption
104
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• The INTERNAL ENABLE LOGIC SIGNAL shown in the above timing diagram is not a
signal (nor a voltage) on the IC (package) ENABLE pin
• The ENABLE pin is “current driven,” and is internally fed from a current limited,
constant (DC) voltage source. Therefore, the voltage across the collector-emitter of
the external opto-coupled transistor–and the current through it–are both virtually
constant. This means that the TinySwitch responds very quickly to any change in the
ENABLE pin current (which renders the supply very responsive to load transients)
• While the value of current being drawn from the ENABLE pin remains below the
threshold value (50 µA for the TinySwitch, and 250 µA for the TinySwitch-II), the
INTERNAL ENABLE LOGIC SIGNAL stays at a logic high. Whenever the current being
drawn from the ENABLE pin exceeds the threshold value, the INTERNAL ENABLE
LOGIC SIGNAL goes to a logic low
• The INTERNAL ENABLE LOGIC SIGNAL is sampled, before the start of each switching
cycle. If it is low, MOSFET switching is disabled (OFF) for that next cycle. If it is high,
MOSFET switching is enabled (ON) for that next cycle
TinySwitch Technology Benefits
Built-in current
limit and thermal
protection
No Bias winding required
No control loop
compensation
components
are required!
TinySwitch is self biasing.
The BYPASS (BP) pin capacitor
is supplied from an internal
high-voltage current source
•
Can be used in continuous and discontinuous conduction modes
•
High bandwidth: excellent transient response, no start-up overshoot
•
Using an optional bias winding can further reduce the no-load/standby
power consumption
– <10 mW of no-load power consumption is achievable, even at 265 VAC !
105
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• The output voltage is effectively being sampled each clock cycle. If the output
voltage is above the regulation set-point value, switching is disallowed. If the output
voltage is below the regulation set-point value, switching is allowed
• This regulation scheme has extremely high bandwidth (half the clock frequency), and
therefore requires no control loop compensation
• A low-voltage bias winding on the transformer can be used to supply current into the
BYPASS pin, which disables the internal high-voltage current source, further
reducing the amount of no-load power the device will consume
• Even without supplemental current from a bias winding, the no-load power
consumption of a typical application circuit is usually <100 mW, for a TinySwitch,
and <300 mW, for a TinySwitch-II
TinySwitch Technology Benefits
•
Overall +7% Vo tolerance with simple Zener diode feedback (saves cost)
– The TinySwitch feedback current (IFB) is independent of load current
• The change in Zener voltage (∆VZ) is almost zero over the range of ∆IFB
•
Typical PWM controllers have >1 mA ∆IFB and therefore large ∆VZ
IZ
IBIAS
IFB
IZ = IFB + IBIAS
106
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• A low-current Zener diode can be used to get optimum regulation, while keeping the
Zener bias current low. This will help to minimize the no-load power consumption
• The sample Zener diode I-V curve shown highlights the difference between the
TinySwitch technology and conventional PWM operation. For optimum regulation,
IBIAS should be chosen from the Zener diode manufacturer’s data sheet
• Less than ±6% output voltage tolerance may be possible, if a 1% Zener diode is used.
This assumes that the operating temperature range will be 0–50 °C, and that the
output voltage is about 5 V
TinySwitch-II Additional Features/Benefits
•
Integrated auto-restart fault protection lowers system cost
– Output diode needs only be rated to the overload current just prior to auto-restart
– Open feedback loops and output short circuits are fully protected against
•
Programmable line under-voltage detection prevents turn off glitches
•
Frequency jittering lowers EMI filter costs
– Fully specified, independent of line or load
•
Multi-level current limit practically eliminates audible noise
– Standard varnished transformers can be used - no gluing required
•
132 kHz operation reduces transformer size
•
Tighter current limit/frequency tolerances lower system cost
•
Increased DRAIN pin creepage, for high pollution environments
•
Built-in Zener clamp on the BYPASS pin
– A simple resistor feed from a low-voltage bias winding enables lower no-load
power consumption
107
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• TinySwitch does not have a built-in Zener clamp on its BYPASS pin. Therefore, it
requires an external Zener clamp diode, when it is fed current from a low-voltage bias
winding
• Without auto-restart, the output diode needs to be rated for the full short-circuit
current
Designing with
TinySwitch Technology
108
Lowpwr 022404
Designing with TinySwitch-II
•
Design Concept
– Choose a transformer inductance value that will deliver full load power, at full
frequency and the device current limit
– Leave margin for tolerances, losses and transient load requirements
•
PI Expert
– PI Expert automatically calculates all power-train component values, with the
above concerns adequately considered
109
Lowpwr 022404
• The design tools mentioned on this slide are specific to TinySwitch-II
• TinySwitch has separate design tools, that are covered on the next slide
• PI Expert provides a full optimization function for TinySwitch-II designs. This means
that the software fully optimizes the design automatically, without requiring
numerous manual reiterations
• Note: within PI Expert, efficiency is either a user supplied input value or a software
determined estimation. Actual efficiency should always be verified on an early
prototype, then that measured efficiency should be entered into PI Expert, for the final
iterations of the design process
• The names of the parameters PI Expert uses are defined in the software’s help system
Designing with TinySwitch
110
•
PI Expert has a spreadsheet dedicated to TinySwitch designs
•
Application Note AN-23: ‘TinySwitch Flyback Design Methodology’
– AN-23 provides a detailed, step-by-step, flow-charted design procedure
•
Application Note AN-24: ‘Audio Noise Suppression Techniques’
– AN-24 provides techniques for reducing audible noise from Flyback
transformers that will be used in an application that may reside in a
low-power or standby power mode most of the time
– Topologies that use single-piece core inductors, such as Buck and BuckBoost converters, do not require audible noise reduction measures
Lowpwr 022404
• The design tools mentioned on this slide are specific to TinySwitch
• The TinySwitch-II design tools were covered on the previous slide
TinySwitch and TinySwitch-II
Application Examples
Exceeding
Worldwide Energy Efficiency Standards
111
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Adapter/Charger Applications with
Low No-Load Power Consumption
•
3 W Adapter with:
<300 mW no-load consumption (DI-13)
– 9 V output, 85-265 VAC input
•
3 W Cell Phone Charger with: <30 mW no-load consumption
– 5 V, 600 mA CC output, 85-265 VAC input
(DI-28)
•
3 W Adapter with:
<10 mW no-load consumption
– 12 V output, 85-265 VAC input
(DI-27)
DI: Design Idea
112
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• These applications specifically demonstrate the techniques required to meet global
no-load consumption standards
• These techniques are not limited to the specific cases presented here
• All of the Design Ideas referred to above, and the newest Design Ideas from Power
Integrations are available at www.powerint.com/appcircuits.htm
Applications Requiring High Standby
Efficiency
•
10 W Standby Power Supply: POUT >600 mW with PIN <1 Watt
– 5 V, 15 V outputs, 140-375 VDC input
•
15 W Standby Power Supply: POUT >600 mW with PIN <1 Watt
– 5 V, 15 V outputs, 140-375 VDC input
•
1.3 W TV Standby Power Supply: POUT >650 mW with PIN <1 Watt (DI-7)
– 7.5 V output, 120-375 VDC input
•
1.2 W Non-Isolated Aux Supply:
– 12 V output, 85-265 VAC input
•
11 W Multiple Output DVD Supply: POUT >650 mW with PIN <1 Watt (DI-33)
– 3.3 V, 5 V, 12 V, -12 V outputs, 85-265 VAC input
POUT >600 mW with PIN <1 Watt (DI-42)
DI: Design Idea
113
Lowpwr 022404
• The POUT versus 1 W PIN data in the above slide is the actual performance of the
Design Idea circuits
• These applications specifically demonstrate techniques required to convert power
very efficiently, at 1 W of input power and below
• These techniques are not limited to the specific cases presented here
• All of the Design Ideas referred to above, and the newest Design Ideas from Power
Integrations are available at www.powerint.com/appcircuits.htm
Typical No-Load Consumption Curves
1000
Input Power (mW)
300
EUROPEAN STANDARD
TinySwitch-II
100
*
TinySwitch
TinySwitch-II Bias winding
10
PI-3527-051403
TinySwitch Bias winding
*
1
50
100
150
200
Input Voltage (VAC)
250
300
•
Charger applications with secondary CC circuit add 3 mW to 5 mW
*
Some OEMs require these limits at 100 VAC
114
Lowpwr 022404
• TinySwitch technology currently provides solutions that exceed all existing and
proposed future global energy efficiency standards
• These solutions use simple techniques that add very little cost to that of standard
TinySwitch and TinySwitch-II designs
• (Charger applications with CV/CC output characteristics normally require additional
secondary-side bias current, resulting in slightly higher input power consumption)
3 W Adapter: <300 mW No-Load (DI-13)
Specification Table
DESCRIPTION
Input Voltage
VALUE
85-265 VAC
Output Voltage
9 V ±7%
Output Current
330 mA
Output Power
Efficiency
No-load
•
3W
>70%
<300 mW
PI-3248-082702
115
Lowpwr 022404
Device Choice:
– Standard TinySwitch-II circuit will
meet no-load target
– TNY264 is the correct choice based
on device power table for adapter
applications (enclosed nonventilated)
3 W Adapter: <300 mW No-Load (DI-13)
Shield winding
reduces EMI
•
No transformer bias winding required
– Device powered entirely from DRAIN (D) pin voltage
•
Measured no-load consumption: 110/210 mW at 115/230 VAC
•
Measured full load efficiency: 74/72% at 115/230 VAC
116
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• Addition of Zener bias current improves regulation without exceeding 300 mW
• (A VOR of 96 V was used to maximize efficiency)
3 W Cell Phone Charger: <30 mW No-Load
(DI-28)
Specification Table
DESCRIPTION
Input Voltage
VALUE
85-265 VAC
Output Voltage
5 V ±10%
Output Current
600 mA
Output Power
Efficiency
No-load
Conducted EMI
50/60 Hz Leakage
Current
3W
>60%
•
Device Choice:
– TinySwitch-II with bias winding will
meet no-load target
– Secondary CC circuit losses increase
effective power delivered by the
transformer to approx 4 W
– TNY264 is correct choice based on
device power table for adapter
applications (enclosed non-ventilated)
<30 mW
CISPR22B
EN55022B
<5 µA
PI-3239-091302
117
Lowpwr 022404
• It is important to keep the 50/60 Hz leakage current low in chargers for applications
such as cell phones, which may have metallic casings. 50/60 Hz leakage current must
be limited to prevent customers from “feeling” the current when touching the unit
being charged
3 W Cell Phone Charger: <30 mW No-Load (DI-28)
The bias voltage is only required
at no-load: TinySwitch-II will selfbias, if voltage drops with output
C3 reduces the leakage
spike, which improves EMI
The bias voltage supplies
>500 µA (the max TinySwitch-II
consumption) at no-load
R3 provides VR3
with bias current
Q1 lets the
VR3 anode
connect to
the load side
of sense
resistor R5
High value bias capacitor retains charge
at the low no-load switching frequency
•
118
Low cost current sense circuit.
Meets EMI without a Y capacitor
– The bias winding was designed to work as an electromagnetic shield
– The AC leakage current is very low (<5 µA)
Lowpwr 022404
• Many CV/CC circuits require a Forward (versus a Flyback) bias winding, to ensure
that the bias supply voltage does not collapse if the output voltage drops (when over
loaded). However, the TinySwitch-II will automatically turn its internal high-voltage
current source back on, if the bias winding voltage collapses. Therefore, a simple
Flyback winding can be used, since that winding only needs to supply bias current at
no-load, to minimize the no-load power consumption
• The built-in Zener clamp on the BYPASS pin of the TinySwitch-II eliminates the need
for an external Zener diode, as is required in an equivalent TinySwitch circuit
3 W Cell Phone Charger: <30 mW No-Load (DI-28)
Measured Output Characteristics
•
119
Measured no-load consumption : 20/25 mW at 115/230 VAC
Lowpwr 022404
• Simple secondary CC circuitry provides output current regulation to zero output volts
3 W Adapter: <10 mW No-Load (DI-27)
Specification Table
DESCRIPTION
Input Voltage
VALUE
85-265 VAC
Output Voltage
12 V ±7%
Output Current
250 mA
Output Power
Efficiency
No-load
•
Device Choice:
– Very low no-load target requires
TinySwitch with bias winding
– TNY254 correct choice based on device
power table for adapter applications
(enclosed non-ventilated)
3W
>70%
<10 mW
PI-3240-091302
120
Lowpwr 022404
3 W Adapter: <10 mW No-Load (DI-27)
The bias circuit supplies >200 µA
(the max TinySwitch consumption),
at no-load
A simple RC snubber effectively
attenuates EMI. The no-load target
is still achieved due to the very low
no-load switching frequency of the
TinySwitch
No extra Zener diode
bias current keeps
no-load power down.
Using a low current
Zener minimizes the
unit-to-unit output
voltage variance
An external Zener clamp is required to
protect the TinySwitch BYPASS (BP) pin
High value bias capacitor retains charge
at the low no-load switching frequency
• Meets <10 mW no-load, with only 24 components!!
• Measured no-load consumption: 6/8 mW at 115/230 VAC
121
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• Transformer wire gauges were selected to completely fill each winding layer, and the
bias winding was used as an electromagnetic shield, to minimize EMI and to eliminate
the need for a Y capacitor
• Unlike the TinySwitch-II, the TinySwitch requires an external Zener diode clamp on
the BYPASS pin, whenever an external bias current is fed into the BYPASS pin
• Setting VOR to 60 V limits the output short circuit current to 1 A
3 W Adapter: <10 mW No-Load (DI-27)
•
Influence of the external BYPASS current value on no-load consumption
– The bias winding voltage, and the values of C5 and R3 should be calculated to
ensure that at no-load, the current into the BYPASS pin is >225 µA, but <250 µA,
to minimize the no-load power consumption
Optimal External
bias current
µ
122
Lowpwr 022404
• Insufficient external bias current (<200 µA) significantly increases the no-load power
consumption, since the internal high-voltage current source must provide the rest of
the supply current
• Excessive external bias current (>250 µA) may increase the no-load consumption, as
the dissipation of the Zener clamp diode (VR3 in this circuit) increases
• In circuits that are designed around a TinySwitch-II, the optimum external bias current
is higher (typically >500 µA), since the internal power consumption of the device is
slightly higher
10 W Standby Power Supply
Specification Table
DESCRIPTION
VALUE
Input Voltage
140-375 VDC
•
Device Choice:
– TNY266 correct choice for a wide
input range, open frame power supply
Output Voltage
V1
5 V ±5%
V2
15 V +6/-20% Primary
Output Current
I1
2A
I2
50 mA
Output Power
10 W
POUT at PIN = 1 W
>600 mW
PI-3241-091302
123
Lowpwr 022404
10 W Standby Power Supply
Zener clamp reduces losses over RC
snubber or RCD clamp to maximize
circuit efficiency
The TinySwitch
constant feedback
current enables ±7% *
output regulation
from a simple Zener
diode reference
R2 chosen to provide >500 µA
(max TinySwitch-II consumption)
maximizing circuit efficiency
15 V primary output powers main power
supply controller IC
• Measured Performance: >600 mW output with < 1 W input power
• Easily meets President Bush’s 1 Watt Executive Order
124
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• The VOR was set to 130 V, to maximize the power capability of the TinySwitch-II
• * ≤ ±5% output voltage tolerance can be obtained by using a TL431 reference in place
of the Zener diode
15 W Standby Power Supply
•
Specification Table
DESCRIPTION
Input Voltage
VALUE
Device Choice:
– TNY268 correct choice for a wide
input range, open frame power supply
140-375 VDC
Output Voltage
V1
5 V ±5%
V2
15 V +6/-20% primary
Output Current
I1
3A
I2
50 mA
Output Power
POUT at PIN = 1 W
15 W
>600 mW
PI-3242-091302
125
Lowpwr 022404
15 W Standby Power Supply
Zener clamp reduces losses over RC
snubber or RCD clamp to maximize
circuit efficiency
The TinySwitch
constant feedback
current enables ±7% *
output regulation with
a simple Zener diode
reference
15 V primary output powers main power
supply controller IC
R2 chosen to provide >500 µA
(max TinySwitch-II consumption)
maximizing circuit efficiency
• Measured Performance: >600 mW output with < 1 W input power
• Easily meets President Bush’s 1 Watt Executive Order
126
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• The VOR was set to 125 V, to maximize the power capability of the TinySwitch-II
• * ≤ ±5% output voltage tolerance can be obtained by using a TL431 reference instead
of the Zener diode
1.3 W TV Standby Supply (DI-7)
Specification Table
DESCRIPTION
Input Voltage
•
VALUE
120-375 VDC
Output Voltage
7.5 V ±5%
Output Current
173 mA
Output Power
1.3 W
Efficiency
>70%
No-load
<100 mW
POUT at PIN = 1 W
>600 mW
Device Choice:
– TinySwitch allows RC Drain
snubbing to reduce video noise.
– TNY253 correct choice for power
level
PI-3245-091302
127
Lowpwr 022404
• TinySwitch-II could also be used if lowest video noise is not a requirement - e.g. in
digital TVs.
1.3 W TV Standby Supply (DI-7)
Simple RC snubber reduces video noise.
Targets for low no-load consumption and
high standby efficiency achieved with low
TinySwitch switching frequency
No transformer bias winding: still achieves
<100 mW no-load, 70% standby efficiency
Fast diode for reduced
radiated noise
May not be necessary depending
on location of main TV power
supply Y capacitor
•
Measured Performance
– <100 mW no-load consumption at 375 VDC
– >650 mW output power with <1 W input power
•
Complete standby supply with as few as 13 components!!
128
Lowpwr 022404
• (A VOR of 50 V was used to limit output short circuit current <1 A)
1.2 W Non-Isolated Aux Power Supply (DI-42)
Specification Table
DESCRIPTION
Input Voltage
VALUE
85-265 VAC
Output Voltage
12 V ±7%
Output Current
100 mA
Output Power
1.2 W
Efficiency
>60%
POUT at PIN = 1 W
Surge Rating
•
Device Choice:
– Single piece core inductor allows use
of TinySwitch without audible noise
considerations
– TNY254 chosen (See DI-42 BuckBoost converter)
>600 mW
2 kV
IEC1000-4-5
PI-3246-091302
129
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1.2 W Non-Isolated Aux Power Supply: (DI-42)
Simple input stage
meets 2 kV IEC1000-4-5
surge requirements
Output reference to Line.
Typically required for Triacs
and associated control
circuitry in appliance and
industrial motor drives
•
•
130
Measured Performance
– >650 mW output power with <1 W input power
Complete auxiliary supply with as few as 11 components !!
Lowpwr 022404
• L1 should be rated for more than the TNY254 current limit (300 mA is a good choice)
• Two diodes (1N4007s, with PIV ratings of 1000 V) are required, to meet a 2 kV surge
voltage withstand rating
• A simple modification to the input circuitry can provide 6 kV of surge voltage
withstand rating ( see DI-42, for a description of that circuit modification)
• This circuit is available for evaluation as a Design Accelerator Kit (DAK- 7)
• Many other non-isolated configurations can be designed with the TinySwitch, the
TinySwitch-II or the new LinkSwitch-TN (Example: DI-11 Buck converter). The latest
new application circuits are available, at www.powerint.com/appcircuits.htm
11 W DVD Supply: <50 mW No-Load (DI-33)
Specification Table
DESCRIPTION
VALUE
Input Voltage
85-265 VAC
Output
V1
V2
V3
V4
3.3 V ±5%
5 V ±5%
12 V ±10%
-12 V ±10%
Output
I1
I2
I3
I4
300-700 mA
300-1600 mA
400 mA
100 mA
•
Device choice:
– TinySwitch-II with bias winding will
meet no-load target
– TNY268 is correct choice for peak
power capability
•
Alternative Device choice:
– At this power level, also consider
TOPSwitch-GX for additional features
– For higher power levels use
TOPSwitch-GX (DI-39)
11 W Cont
Output Power
17 W Peak
Efficiency
>75%
No-Load
<100 mW
POUT at PIN = 1 W
>600 mW
PI-3247-091302
131
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11 W DVD Supply: <50 mW No-Load (DI-33)
The bias circuit supplies
>500 µA at no-load
High value
bias capacitor
retains charge
at the low noload switching
frequency
C2, R5 and R7 snub
the leakage spike, to
reduce EMI
Dual feedback
improves output
cross regulation
Shield windings reduce EMI
•
132
Measured Performance:
– no-load: 30/41 mW at 115/230 VAC, minimum full load efficiency: 77%
– >650 mW of output power at 1 W of input power
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• The standby power consumption was measured at 115 VAC and 230 VAC, with equal
loading on both the 3.3 V and 5 V outputs. The other outputs were at zero load
• The transformer shield windings significantly reduce the amount of EMI generated,
allowing a simple pi filter (C1, C4 and L1) to adequately attenuate the conducted EMI
• Simple transformer construction (without shield windings) can be used together with
a common mode (input) choke. Choices should be made, based on the relative cost
of implementing these two options
• The VOR was set to 120 V in this design. This value allows the power capability of the
TinySwitch-II to meet the maximum output power requirement while maintaining good
cross regulation between the two main outputs and the other two outputs
• The optional line under-voltage lockout function of the TinySwitch-II can be activated
by simply connecting a resistor between the rectified DC input rail and the EN/UV pin.
A 2 MΩ resistor sets a low under-voltage lockout threshold (UVLO) at 100 VDC. This
prevents the power down process from producing any glitches on the outputs, as the
supply shuts off. The line under-voltage function increases the no-load consumption
by approx 50 mW at 230 VAC. However, the circuit can still meet a 100 mW no-load
power consumption target
11 W DVD Supply: Cross Regulation (DI-33)
OUTPUT
VOLTAGE
VOLTAGE
RANGE
(VAC)
LOAD
RANGE
+3.3 V
85-265
40-100%
+5 V
85-265
20-100%
+12 V
85-265
100%
-12 V
85-265
100%
REGULATION (%)
-6
-5
-4
-3
-2
-1
0
1
2
3
4
5
6
PI-3255-082702
This table summarizes the worst-case variations of each output voltage.
The measurements were recorded across the full input line voltage
range, and over the specified load range of each output
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11 W DVD Supply: Conducted EMI (DI-33)
•
Test Conditions
– 11 W output
– Output grounded through artificial hand (EMI reduced further with floating output)
•
> 10 dB margin (AV and QP) at all frequencies
115 VAC
230 VAC
QP
AV
QP
AV
Quasi peak
Quasi peak
Average
Average
PI-3528-051403
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• Meets international standards without requiring a common mode choke
PI-3529-051403
Hints and Tips
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Optimizing Efficiency & No-Load Performance
•
Using Transformer Bias Winding (most significant)
– Designed to supply max device current under specified conditions e.g. no-load
– Use large enough bias capacitor to retain charge at standby or no-load frequency
– Other load conditions non-critical - devices will self bias if external supply is lost
– 230 VAC power dissipation reduced by up to 65/160 mW TinySwitch/TinySwitch-II
•
Other Transformer Considerations
– Reduce capacitance - tape between primary layers
– Design with low VOR - reduces clamp losses
– Reduce leakage inductance - reduces clamp losses
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Optimizing Efficiency & No-Load Performance
•
Minimize Bias Currents in Secondary Circuits
– CV only circuits (adapters/standby), Zeners should be left unbiased if regulation is
acceptable - best performance with low current Zeners
– CV/CC designs (chargers) bias currents should be minimized
•
Choice of primary clamp circuits
– Zener clamp for lowest dissipation - dissipates power only during leakage spike
– RCD clamps often provide acceptable performance with resistor value >200 kΩ
– RC snubber typically used only with TinySwitch - switching frequency low at full
load and very low at no-load
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Other Hints and Tips
•
Minimizing audible noise in TinySwitch designs
– Design the transformer for low flux density, <2000 gauss (200 mT), at full load
– Glue the transformer core halves together, according to the guidelines in AN-24
– Only dip-varnishing the transformer does not usually produce acceptable results
– Dip-varnishing the transformer (in addition to gluing) is not necessary
– Use low-cost Film capacitors in the clamp circuit, as Ceramic capacitors can
generate audible noise
•
TinySwitch-II practically eliminates audible noise generation
– A standard dip-varnished transformer works fine, no gluing is required!
– Gluing the transformer core halves together (if preferred) also works well
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• Varnishing tends to increase transformer capacitance, which results in higher
switching losses. This will influence full load efficiency but have only a small effect
on standby/no-load consumption, due to the low switching frequency at light loads
PCB Layout Guidelines
Y capacitor returned to DC rail.
Routes common mode surge
currents away from TinySwitch
Maintain tight
clamp current loop
to reduce EMI
Notches force high
frequency current
through capacitor
Power currents in
SOURCE trace
Maintain tight output
current loop to reduce
EMI and secondary
impedance
Position BP pin
capacitor to avoid
power currents in
SOURCE traces
Position EN/UV trace
away from DRAIN
node to avoid noise
pick-up
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Position EN/UV resistor
close to device to
minimize noise pick-up
Maintain tight loop from
opto to device EN pin to
avoid noise pick-up
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PI-2707-012901
Summary
•
TinySwitch and TinySwitch-II based power supplies exceed the
requirements of all existing and proposed energy efficiency standards
•
TinySwitch-II is the best choice for most applications
– Enhanced features lower system cost
– Applications up to 23 W (230 VAC), 15 W (85-265 VAC)
– <30 mW no-load consumption at 230 VAC (with bias winding)
– <300 mW no-load consumption at 230 VAC (without bias winding)
•
TinySwitch is a better choice for applications requiring:
– <10 mW no-load consumption at 230 VAC (using bias winding)
– <100 mW no-load consumption at 230 VAC (without bias winding)
– Video noise sensitive applications if RC snubbers are required
•
TinySwitch Technology provides simple, cost effective, and energy
efficient replacements for RCC & Linear solutions in the 2-20 W range
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