Designing Wide Range Power Supplies for Three Phase Industrial

Designing Wide Range
Power Supplies for
Three Phase Industrial
Three-phase regulated supplies operate
under demanding conditions
With the volatile nature of a three-phase source, special design considerations and techniques need to
be applied. The StackFET configuration provides a design solution at significantly lower cost.
By Rahul Joshi, Power Integrations
ndustrial equipment operating from a
three-phase AC supply often requires
an auxiliary power stage that supplies
regulated, low-voltage DC to the control
electronics. Specifications for these
supplies are much more demanding
than for the typical single-phase supply. The nominal input voltage is higher,
and equipment designed for threephase input has to tolerate larger input
supply-voltage variations. Line surges,
extended sags and sub-cycle drop-outs
often occur in an industrial environment
as a result of large loads being switched
off and on, or as a result of fuses being
cleared for fault conditions elsewhere on
the line. Three-phase applications can
occasionally lose a phase or a neutral
connection. Industrial equipment is
expected to handle all of these conditions without malfunctioning. Applications such as energy meters must work
reliably over these extreme conditions.
This article looks at the challenges
of designing switched-mode power
supplies for three-phase applications,
and presents a compact, cost-effective
design that operates over a very wide
input voltage range.
• Easy-to-create multiple output voltages
• Very low component count and cost
Design Goal
A three-phase input, off-line switching power supply that has wide input
voltage range, high overall operating
efficiency, and good immunity to input
voltage perturbations.
Most switching power supplies can
operate over the universal input voltage
range to provide worldwide coverage.
For three-phase applications such as
energy meters, the power supply must
work from 57 to 580 VAC, from all three
phases and with the occasional loss of
a phase or a neutral connection.
For auxiliary power supply designs,
the flyback topology is best-suited, and
offers these advantages:
• Use of a single active switch that simplifies circuit design
• Use of a single-wound component
in the topology (eliminates large filter
chokes on the output)
A flyback converter typically requires
a minimum MOSFET breakdown voltage of 1.6 times the rectified peak of
the maximum AC input voltage. For 580
VAC, a 1200 V MOSFET would be required, adding cost and (normally) ruling
out the use of an integrated switching
IC that could dramatically simplify the
solution (when compared to a discrete
An IC such as the LinkSwitch®-TN
from Power Integrations incorporates
a 700 V MOSFET and controller into a
single device, and can eliminate 20 to
30 external components when compared to a circuit using a discrete MOSFET and external control IC. The 700V
rating of this IC would normally limit use
to single-phase applications. However,
by adding an external MOSFET in a cascode or StackFETTM configuration, it is
possible to distribute the voltage stress
across two devices, resulting in an over-
Power Systems Design Europe
November 2006
Note: A standard
PMW controller
would suffer from
poor efficiency
under high-line and
light-load conditions, due to the
short duty cycle relative to the operating
frequency. ON/OFF
control eliminates
this problem.
Figure 1. Circuit Schematic.
all voltage rating equal to the sum of the
individual MOSFET voltages.
Design Solution
The circuit in Figure 1 is a 12 V, 250
mA wide-range flyback power supply
that operates from a single-phase or a
three-phase input. Using the StackFET
technique with a low-cost 600 V MOSFET results in an overall voltage rating of
1300 V and allows supply operation over
the desired wide input voltage range
of 57 to 580 VAC. The supply will work
from 47–63 Hz, single- or three-phase
110 VAC, 220 VAC or 440 VAC. This
supply comfortably handles the loss of
one or more phases or the neutral, as
well as extended sags and surges.
Circuit Operation
The circuit in Figure 1 is based on a
LinkSwitch-TN IC, the LNK304P (U1)
that is configured as a flyback, to leverage its 66 kHz switching frequency.
This reduces switching losses and
improves efficiency. The IC’s ON/OFF
control regulates the output by skipping
switching cycles. As the load is reduced, the effective switching frequency
decreases, scaling the switching losses
and maximizing the operating efficiency.
The AC input is full-wave-rectified by
diodes D1 through D8. Resistors R1
through R4 provide in-rush current limiting and protection against catastrophic
circuit failure. Capacitors C5 through C8
are used to filter the rectified AC supply.
To meet maximum bus voltage of 820
VDC, 450 V input capacitors C5, C7 and
C6, C8 are connected in series with
ancing resistors R13 to R16 to equalize
the voltage. The C5/C7 and C6/C8 capacitor sets are used in conjunction with
L1 to form a p filter for EMI reduction.
Capacitor C9, which is placed very close
to U1 and T1, shunts switching induced
noise currents, to minimize differential
mode EMI generation. Combining this
EMI reduction technique with 1) the jittering of the switching frequency of U1,
2) E-ShieldTM winding in the transformer,
and 3) the safety Y-rated capacitor C1
across the isolation barrier, allows the
design to easily meet conducted EMI
limits (as specified in EN55022-B).
The high-voltage DC is applied to
one end of the transformer primary, and
the other end driven by MOSFET Q1.
MOSFET Q1 and the MOSFET inside
the LNK304P effectively form a cascode
arrangement. When the internal MOSFET of U1 turns on, the source of Q1 is
pulled low, which allows gate current to
flow through the resistor string R6, R7
and R8 from the junction capacitance of
VR1, VR2 and VR3, to turn on Q1. Zener
VR4 limits the gate-source voltage applied to Q1. When turned OFF, VR1 to
VR3 (connected in series) form a 450 V
clamping network that ensures the drain
voltage of U1 remains close to 450 V;
any input voltage above 450 V will be
dropped across Q1. This arrangement
distributes the sum total of flyback voltage and DC bus voltage across Q1 and
the internal MOSFET within U1. Resistor R9 limits high frequency ringing that
occurs when VR1 to VR3 conduct. The
clamping network, VR5, D9 and R10,
limits the peak voltage that appears
across Q1 and U1 (due to leakage inductance) during the flyback interval.
The circuit on the secondary of
transformer T1 provides output rectification, filtering and feedback. Diode D10
rectifies the output. Capacitor C2 filters
the rectified output. Inductor L2 and
capacitor C3 form a second-stage filter,
which helps to reduce the high-frequency switching ripple in the output. Zener
diode VR6 conducts when the voltage
at the output exceeds the total drop of
VR6 and the optocoupler diode inside
U2. A change in output voltage results in
a change in the current through the optocoupler diode. This, in turn, increases the
current through the transistor inside U2B.
When this current exceeds the FEEDBACK (FB) pin threshold current, the
next switching cycle is disabled. Output
regulation is maintained by adjusting the
number of enabled and disabled switching cycles. Once a switching cycle of
U1 is enabled, the current ramps to the
internal current limit of U1. Resistor R11
limits the optocoupler current during
transient loads, and sets the gain of the
feedback loop. Resistor R12 provides
bias current to the Zener diode, VR6.
If the FEEDBACK pin is not pulled
high for a period of 50 ms, the internal
power MOSFET switch in U1 is disabled
for 800 ms. Alternately enabling and
disabling the switch protects the circuit
against output overload, an output short
circuit, or an open feedback loop.
No auxiliary winding or bias wind51
20V Solid-State photovoltaic relays have 50 percent lower
on-state resistance
Figure 2. Trace 1 - U1 Drain Voltage (200 V / div) and
Trace 2 - Q1 Drain Voltage (100 V / div).
ing on the transformer is required to
power U1, as it is self-powered from the
DRAIN (D) pin. At start-up and during
the off-time of the internal MOSFET, the
local decoupling capacitor (C4) is kept
charged via an internal high-voltage current source.
Figure 4. Conducted EMI at 230 V.
VR2 and VR3. This clamping ensures
safe operation of U1. Trace 1 shows the
voltage on the drain of Q1 referenced
to primary ground (negative of C8). The
actual voltage across the MOSFET Q1
in the OFF state (trace 1) is the difference between the two traces, in this
case 170 V.
Circuit Test Results
The oscilloscope plot shown in Figure
2 was captured at an input voltage of
312 VAC (440 VDC bus voltage). At
turn off the drain voltage of U1 (trace
2) is clamped to a voltage of 450 V,
which is the total voltage across VR1,
As the AC input voltage is increased
to 580 VAC (820 VDC), the voltage drop
across the MOSFET Q1 in the OFF state
is less than 550 V, which allows the use
of a low cost 600 V to 800 V external
The efficiency characteristic of this
design is shown in Figure 3. The curve
reveals that the efficiency drops at
higher input voltage due to increased
switching losses and dissipation in the
cascade connected power stage (Q1
and the internal MOSFET within U1).
However this is still significantly higher
than a regulated linear transformer
The circuit meets conducted EMI
requirements with a comfortable margin
when tested at 230 VAC, as shown in
Figure 4. The blue and red upper lines
represent the quasi-peak and average
limits, perEN55022 B. The lower lines
represent the corresponding quasi-peak
and average test results.
The StackFET technique provides
a cost-effective solution for auxiliary
power supplies in industrial applications.
This technique allows the designer to
benefit from the simplicity afforded by
an integrated switching IC when used
for high input voltages required by threephase AC input.
Figure 3. Efficiency vs. Input Voltage.
Power Systems Design Europe
November 2006
International Rectifier has introduced
a series of photovoltaic relays for applications including power supplies, power
distribution, audio equipment, and instrumentation, computers and computer
Compared to its predecessor, the
PVN012A family offers a 50 percent
reduction of AC/DC on-state resistance
(RDD-on) and 37.5 percent increase of
maximum AC/DC load current rating at
full (100%) duty cycle. The new series is
also rated for maximum pulsed (surge)
load current.
With extremely low on-resistance and
high volumetric load current density in
such a small package, the PVN012A
family exceeds the performance capabilities of traditional electromechanical
relays. In comparison, the PVN012A
family offers a smaller footprint, high
input-to-output isolation, bounce-free
operation, solid-state reliability, stable
on-resistance over life, and greater input
These new 20V single-pole, normally
open, solid state relays utilize a
HEXFET® MOSFET output switch,
driven by a unique integrated photovoltaic generator circuit. The output switch
is controlled by radiation from a GaAlAs
light-emitting diode (LED) that is optically
isolated from the photovoltaic generator.
The new series is available in 6-pin DIP,
6-pin SMT, and in tape and reel.
Dual DC/DC Converter Delivers 1.6A
per Channel from a DFN Package
Linear Technology Corporation announces the LT3506 and LT3506A, dual
current mode PWM step-down DC/DC
converters with two 2A power switches
packaged in a 16-lead 5mm x 4mm
DFN package. Each channel is capable
of delivering up to 1.6A of output current. Their wide input range of 3.6V to
25V makes them suitable for regulating
power from a wide variety of sources,
including four cell batteries, 5V and
12V rails, unregulated wall transformers, lead acid batteries and distributed
power supplies. The LT3506 switches
at 575kHz while the LT3506A switches
at 1.1MHz enabling the use of tiny, low
cost inductors and ceramic capacitors,
while delivering low, predictable output
The LT3506 and LT3506A’s low VCESAT (210mV @1A) internal switches
offer efficiencies of up 88%, minimizing thermal constraints and maximizing battery run-time. Low voltage
outputs are easily attended due to an
internal reference of 0.80V. Each channel has independent shutdown and
soft-start pins as well as independent
power good indicators to ease power
sequencing. The channels
switch 180 degrees out
of phase with respect to
the other, reducing input
ripple and minimizing capacitance needs. Internal
cycle-by-cycle current limit
provides protection against
shorted outputs while
soft-start eliminates input
current surge during start
up. The low current (<30uA
typ) shutdown provides
easy power management in
battery-powered systems.
The LT3506EDHD and
LT3506AEDHD are available
in a thermally enhanced
5mm x 4mm DFN-16 package
Photo Caption: 25V,
1.1MHz Dual 1.6A (IOUT)
Step-Down Switching Regulator