technical note Philips Magnetic Products 25 Watt DC/DC converter using integrated Planar Magnetics Philips Components 25 Watt DC/DC converter using integrated Planar Magnetics Contents Introduction 2 Converter description 3 Converter specification 4 Performance of the converter 4 Design of planar magnetics 6 PCB layout 8 Circuit diagram 11 Components list 12 1 Philips Magnetic Products 25 Watt DC/DC converter using integrated Planar Magnetics (designed in cooperation with PEI Technologies, Ireland) Introduction Planar magnetics are an attractive alternative to conventional core shapes when a low profile of magnetic devices is required. Basically this is a construction method of inductive components whose windings are fabricated using printed circuit tracks or copper stampings separated by insulating sheets, or constructed from multilayer circuit boards. These windings are placed in low profile ferrite EE-or E/PLT-core combinations. Planar devices can be constructed as stand alone components or integrated into a multilayer board with slots cut to accept the ferrite Ecore (fig.1). The aim of this demonstration board is to demonstrate the capability of Philips’ planar E cores (see Data Handbook MA01). One of these cores is used in the design of a high frequency 25 W DC/DC converter. A 6 layer PCB is used to facilitate the integration of the transformer and output inductor windings into the multilayer PCB structure. The board demonstrates the advantages over standard wire wound solutions in terms of cost, size, simplicity and reliability. It will also show that the electrical performance of the converter is excellent. 2 Philips Magnetic Products Features such as input filtering, output voltage and long term short circuit protection have been omitted from the design as the use of planar magnetics does not have an impact on these features. At 48V input, synchronous rectification will increase the efficiency by approximately 3% to 6% depending on the Rds (on) of the MOSFETS used and the switching frequency. Low Rds(on) MOSFETS increase efficiency but are more expensive. The chosen topology is the forward converter with resonant reset. A basic description of the operation of a forward converter can be found in most textbooks on switch-mode power supplies. Increased frequency will reduce the efficiency of the synchronous rectifiers due to the charging of the input capacitance once every cycle. To keep the circuit simple and low cost. the synchronous rectifiers are self driven. This means that they are driven directly with the voltage from the transformer secondary. This is not the most efficient solution particularly when the ‘dead’ time is large as at high input voltage. To counteract this, diode D1 is added in parallel to Q3. This diode will conduct during the ‘dead’ time. Converter description The schematic for the forward converter with resonant reset is shown on page 10. This converter design differs from a standard design in two ways: • It employs a resonant reset technique to reset the power transformer, T1 • It uses synchronous rectifiers Q2 and Q3, low voltage, low Rds (on) MOSFETS on the secondary side of the transformer for rectification. 1/2 planar E core In a standard forward converter a separate winding can be used to reset the transformer to ensure the flux returns to zero on each cycle. The resonant reset technique allows for the elimination of this winding which is an attractive benefit when using planar magnetics. Reset is achieved during the off time by imposing a resonant voltage on the primary winding using parasitic circuit elements. layer 1 layer 2 multilayer PCB layer 4 The frequency of this resonance is approximately equal to: fres ≈ layer 3 1 2π√ Lp • CQ1 1/2 planar E Core where Lp is the transformer primary inductance and CQ1 is the MOSFET parasitic capacitance. Fig. 1 Exploded view of a PCB transformer The advantage of this technique is that it iseasy to implement at low cost. The disadvantage is that it is a lossy solution compared to soft switching techniques. This loss is not dramatic at voltages lower than 100V, and will lead to a decrease in efficiency of approximately 1% at 48V input and 2% at 72V input voltage. The second difference in comparison with a conventional converter is the implementation of synchronous rectification. This is cost competitive with Schottky diodes at a current rating of less than 10A. 3 Philips Magnetic Products Converter specification Performance of the converter Low-profile DC/DC converter (25 W) Featuring: -planar ferrite E cores -multilayer FR4 printed circuit board(6layers) -integrated windings for transformer and output choke. 90 88 36-72V 50 mA 620 mA 5VDC ± 1% 0A 5A 50 mVpp 85 % typ ± 0.1 % ±1% 500 VDC 420 kHz 25 °C to50 °C Efficiency (%) Input voltage Max input current (no load) Max input current (full load) Output voltage Output current (min) Output current (max) Output ripple and noise Efficiency Line regulation Load regulation Isolation voltage Switching frequency Operating temperature- 86 84 82 80 35 45 50 55 60 65 70 75 Input Voltage (Volts) Fig.2 Efficiency as a function of input voltage at full load 90 Input capacitor required for operation: 10 µF , 100V. 82 Efficiency (%) All Specifications are typical at nominal line voltage(48V), full load and 25 °C unless otherwise stated. Pin J1 J2 J3 J4 40 Pin connection Vin + Vin + Output - Output 74 66 58 50 Dimensions: 60 × 57 × 6 mm 0 1 2 3 4 5 Output Current (Amps) Fig.3 Efficiency as a function of output current (Vin=48V) 4 Philips Magnetic Products vi d/ V 5 vi d / V 0 5 Oscillograms vi d/ s n 0 5 Fig.4 Primary MOSFET (Q1) gate voltage(TP6) Fig.5 Primary MOSFET (Q1) drain voltage(TP2) vi d / V 0 1 vi d/ V 5 vi d/ s n 0 5 vi d/ s n 0 5 vi d/ s n 0 5 Fig.7 Synchronous rectifier (Q3) drain voltage (TP4) vi d / V m 0 2 vi d/ V 1 Fig.6 Synchronous rectifier (Q2) drain voltage (TP3) 1 m vi d/ s vi d/ s n 0 5 Fig.8 Control IC oscillator (TP5) Fig.9 Output voltage ripple and noise (bandwidth 20 Mhz) 5 Philips Magnetic Products Design of planar magnetics Transformer losses Losses in the ferrite core and windings are estimated for a switching frequency of 400 kHz and an output current of 5 A. Transformer design (T1) In designing the power transformer the optimisation of a number of design parameters has been investigated. These are discussed here. The primary to secondary turns ratio should be approximately 4.5:1 to guarantee a secondary voltage of 5V at a minimum input voltage of 36V using a forward converter operating at a maximum duty cycle of 70%. Three turns ratios have been investigated ( 4:1, 4.5:1, 5:1) in order to determine the minimum transformer losses. The number of primary turns has been selected on the basis of a trade off between minimising core losses and copper losses. Consideration was also given to being able to accommodate the transformer windings in a 6-layer PCB construction. Hence three values of primary turns were investigated ( 5, 8 and 9 turns). Turns ratio Copper losses in the transformer have been calculated for DC only, which appears to be accurate enough for this application. Methods to predict AC losses will be treated in a.seperate application note on the winding design for planar transformers. DC resistance (mΩ) primary secondary Primary inductance (µH) 8:2 5:1 1.0 4.5 2.0 4.5 3 or 4 2 1 or 2 6 to 8 3 or 4 1 1 or 2 5 to 7 110 6 110 6 30 3 243 192 75 8:2 5:1 Primary current Primary resistance Primary loss Secondary current Secondary resistance Secondary loss Total copper loss 0.8 0.11 0.07 3.61 0.006 0.08 0.15 0.85 0.11 0.08 3.39 0.006 0.07 0.15 0.75 0.03 0.017 3.77 0.003 0.043 0.06 Core loss 0.56 0.77 2.1 Total losses (W) 0.71 0.91 2.15 table 2 The lowest overall losses are predicted for the turns tatio of 9:2, which is chosen for the design. Optimisation of switching frequency The choice of a switching frequency close to 400 kHz follows from an estimation of the total loss balance between semiconductors and magnetics. A higher frequency increases the loss in the switches, but ferrite losses are lower. A higher frequency also reduces the ripple current in the output inductor. Ferrite core: E18/4/10-3F3 + PLT18/10/2-3F3 Turns ratio 9:2 Track width (mm) primary 1.0 secondary 4.5 Number of PCB layers primary 3 or 4 econdary 2 auxiliary 1 or 2 Total 6 to 8 9:2 f (kHz) Vin (V)) Semicond. losses (W) Magnetics Total losses (W) (W) 300 36 48 72 36 48 72 36 48 72 36 48 72 36 48 72 2.11 2.38 3.19 2.13 2.52 3.58 2.33 2.67 3.98 2.61 2.84 4.39 3.05 3.01 4.81 1.34 1.27 1.19 1.20 1.13 1.05 1.16 1.09 1.01 1.22 1.15 1.07 1.22 1.15 1.07 400 500 table1 600 Note 1: 2 oz copper (70 µm) is used in all cases. 700 The primary windings can be split in such a manner that the secondary is embedded between two primary windings. This technique, known as sandwiching or interleaving, reduces leakage inductance. table 3 6 Philips Magnetic Products 3.45 3.65 4.38 3.33 3.65 4.63 3.49 3.76 4.99 3.83 3.99 5.46 4.27 4.16 5.88 Design of planar inductor (L1) The peak-to-peak ripple current in the output inductor is designed to be approximately 20% of the full load output current for the nominal input voltage of 48V. The inductance to achieve this can be calculated from the formula: L= Vsec • ton 10.66 • 1.38 µs = ∆I 1 The increased ripple current will cause an increase in ∆B which will lead to somewhat higher losses in the output inductor. Output capacitor design Output ripple voltage is calculated using the formula: 1 ∆Vo = ∫ dIL dt + ∆IL • ESR C = 14.7 µH where ∆IL is the ripple current in the output inductor and ESR is the equivalent series resistance of the output capacitors. where Vsec = Peak secondary voltage = Ns /Np . Vin = 2/9 . 48 V = 10.66 V ton = Primary MOSFET on time = 1.38 . 10-6s ∆ I = Inductor ripple current The first term is much smaller than the second due the high capacitance of the output capacitors so that the ripple voltage can be expressed as: So ideally the inductance value should be 14.7 µH. With 5 turns this means an inductance per turn of: However, a check on the flux density shows that with a peak current of 5.5 A this is too high, since: AL = L N2 = 14.7 • 10-6 25 ∆Vo = ∆IL • ESR The worst case will be at maximum input voltage. = 588 nH Vsec = 2/9 • 72V = 16V L = 10.8 µH Using the standard core E18/4-3F3-A315-P, a check on the flux density shows that with a peak current of 5.5A, the maximum value is: N • Ip • A L 5 • 5.5 • 588 • 10-9 Bmax = = = 409 mT Ae 39.5 • 10-6 Maximum ripple current follows from: ∆Imax = L = 16 • 0.92 µs 10.8 • 10-6 = 1.35 A For a ripple voltage of less than 40 mV, the equivalent ESR should be less than 30mΩ. The capacitors chosen meet this requirement. where Ip = Peak inductor current B. = Maximum flux density N = Number of turns AL = Inductance per turn Ae = Cross sectional area of core This maximum flux density of 388 mT is excessive for 3F3 material. To reduce the maximum flux density using the same core, the air-gap needs to be increased. Consequently, the maximum flux density is set to 300 mT. Using this figure and working backwards to calculate the required AL with N=5 turns and Ip=5.5 A gives: AL = Vsec • ton B • Ae 0.3 • 39.5 • 10-6 = = 431 nH 5 • 5.5 N • Ip L = AL • N2 = 431 • 10-9 • 25 = 10.8 µH 7 Philips Magnetic Products PCB layout The multilayer FR4 PCB with 70 µm of copper comprises all windings of the transformer and output inductor. These windings are divided over the separate layers in the following way: transformer primary (9turns): -5 turns in layer 1 -4 turns in layer 6 secondary (2 turns): -1 turn in layer 2 -1 turn in layer 5 sense (2 turns): -1 turn in layer 3 -1 turn in layer 4 Fig.10 Component location output inductor -1 turn in layer 1 -1 turn in layer 2 -1 turn in layer 3 -1 turn in layer 4 -1 turn in layer 5 Fig.11 Solder mask layer 1 Fig.12 Solder mask layer 6 8 Philips Magnetic Products Fig.13 PCB layer 1 Fig.14 PCB layer 2 Fig.15 PCB layer 3 Fig.16 PCB layer 4 Fig.17 PCB layer 5 Fig.18 PCB layer 6 9 Philips Magnetic Products 57 mm 60 mm The complete converter 10 Philips Magnetic Products Fig.19 Circuit diagram 11 Philips Magnetic Products Components list Reference Part No. Series Description TR1 E18/4/10-3F3 PLT18/10/2-3F3 E18/4/10-3F3 PLT18/10/2-3F3 IRF630S Si9410DY IRF7401 BCP56 BC848A MBRD320 BAV70 BZX84C12 AS3843 IL206A T1431 WCR RC-01 RC-01 WCR WCR WCR WCR WCR WCR WCR Planar E Core Plate Planar E Core Plate 200V, 0.4Ω, MOSFET 30V, 30mΩ, MOSFET 20V, 22mΩ, MOSFET 80V, 1A, NPN Trans. 30V, 100mA,NPN Trans 20V, 3A, Schottky Diode 70V, 250mA Dual Diode 12V Zener Diode PWM Controller opto-isolator Prog. Reference 100K, 0.1W 1K, 0.125W 1R, 0.25W 1K5, 0.1W 2K2, 0.1W 3K3, 0.1W 1K, 0.1W 10K, 0.1W 220R, 0.1W 15K, 0.1W 100nF,100V SMD-220 SO-8 SO-8 SOT223 SOT23 D-Pak SOT-23 SOT-23 SO-8 SO-8 SO-8 0805 1206 1206 0805 0805 0805 0805 0805 0805 0805 1812 TAJ CG,2R 100µF, 10V 100nF, 63V 220nF 22nF 22pF 15nF 10nF 500V D 1206 1206 0805 0805 0805 1206 Ll Ql Q2 Q3 Q4 Q5 Dl D3 Z1 Ul U2 U3 R1 R2 R4,R5,R18 R6 R8 R7,R9 R11,R14,R15 R10 R12 R16 C1,C21,C22, C23,C24 C3,C4,C18 C5,C11,C12 C6 C7,C10 C9 C13 C2 12 Philips Magnetic Products Package Manufacturer Philips Philips Philips Philips I.R. Siliconix I.R. Philips Motorola P.S. P.S. Astec Siemens T.I. Welwyn Philips Philips Welwyn Welwyn Welwyn Welwyn Welwyn Welwyn Welwyn Syfer AVX Philips AVX Philips Philips Kemet AVX

- Similar pages
- STMICROELECTRONICS KBP105G
- 50W Current-Mode Forward Converter Design with the
- 50W Voltage-Mode Forward Converter Design with the
- cp8030 preliminary datasheet
- ETC AS3842
- ETC HEXFETPOWERMOSFETS
- Mallory FireStorm Instructions 69000S - 69000SR
- MITSUBISHI RD16HHF1
- SANYO SFT1431
- MITSUBISHI RD01MUS2B
- Download
- AN LED Demboard
- AN 185: Thermal Management Using Heat Sinks