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technical note
Philips Magnetic Products
25 Watt DC/DC converter using
integrated Planar Magnetics
Philips
Components
25 Watt DC/DC converter using
integrated Planar Magnetics
Contents
Introduction
2
Converter description
3
Converter specification
4
Performance of the converter
4
Design of planar magnetics
6
PCB layout
8
Circuit diagram
11
Components list
12
1
Philips Magnetic Products
25 Watt DC/DC converter using integrated Planar Magnetics
(designed in cooperation with PEI Technologies, Ireland)
Introduction
Planar magnetics are an attractive alternative to
conventional core shapes when a low profile of magnetic
devices is required. Basically this is a construction method
of inductive components whose windings are fabricated
using printed circuit tracks or copper stampings separated
by insulating sheets, or constructed from multilayer circuit
boards. These windings are placed in low profile ferrite
EE-or E/PLT-core combinations. Planar devices can be
constructed as stand alone components or integrated into
a multilayer board with slots cut to accept the ferrite Ecore (fig.1).
The aim of this demonstration board is to demonstrate
the capability of Philips’ planar E cores (see Data
Handbook MA01). One of these cores is used in the
design of a high frequency 25 W DC/DC converter. A 6
layer PCB is used to facilitate the integration of the
transformer and output inductor windings into the
multilayer PCB structure.
The board demonstrates the advantages over standard wire
wound solutions in terms of cost, size, simplicity and
reliability. It will also show that the electrical performance
of the converter is excellent.
2
Philips Magnetic Products
Features such as input filtering, output voltage and long
term short circuit protection have been omitted from the
design as the use of planar magnetics does not have an
impact on these features.
At 48V input, synchronous rectification will increase the
efficiency by approximately 3% to 6% depending on the
Rds (on) of the MOSFETS used and the switching
frequency. Low Rds(on) MOSFETS increase efficiency
but are more expensive.
The chosen topology is the forward converter with
resonant reset. A basic description of the operation of a
forward converter can be found in most textbooks on
switch-mode power supplies.
Increased frequency will reduce the efficiency of the
synchronous rectifiers due to the charging of the input
capacitance once every cycle.
To keep the circuit simple and low cost. the synchronous
rectifiers are self driven. This means that they are driven
directly with the voltage from the transformer secondary.
This is not the most efficient solution particularly when
the ‘dead’ time is large as at high input voltage.
To counteract this, diode D1 is added in parallel to Q3.
This diode will conduct during the ‘dead’ time.
Converter description
The schematic for the forward converter with resonant
reset is shown on page 10. This converter design differs
from a standard design in two ways:
• It employs a resonant reset technique to reset the power
transformer, T1
• It uses synchronous rectifiers Q2 and Q3, low voltage,
low Rds (on) MOSFETS on the secondary side of the
transformer for rectification.
1/2 planar E core
In a standard forward converter a separate winding can be
used to reset the transformer to ensure the flux returns to
zero on each cycle. The resonant reset technique allows for
the elimination of this winding which is an attractive
benefit when using planar magnetics. Reset is achieved
during the off time by imposing a resonant voltage on the
primary winding using parasitic circuit elements.
layer 1
layer 2
multilayer PCB
layer 4
The frequency of this resonance is approximately equal to:
fres ≈
layer 3
1
2π√ Lp • CQ1
1/2 planar E Core
where Lp is the transformer primary inductance and CQ1
is the MOSFET parasitic capacitance.
Fig. 1 Exploded view of a PCB transformer
The advantage of this technique is that it iseasy to
implement at low cost. The disadvantage is that it is a
lossy solution compared to soft switching techniques.
This loss is not dramatic at voltages lower than 100V, and
will lead to a decrease in efficiency of approximately 1% at
48V input and 2% at 72V input voltage.
The second difference in comparison with a conventional
converter is the implementation of synchronous
rectification. This is cost competitive with Schottky diodes
at a current rating of less than 10A.
3
Philips Magnetic Products
Converter specification
Performance of the converter
Low-profile DC/DC converter (25 W)
Featuring:
-planar ferrite E cores
-multilayer FR4 printed circuit board(6layers)
-integrated windings for transformer and output choke.
90
88
36-72V
50 mA
620 mA
5VDC ± 1%
0A
5A
50 mVpp
85 % typ
± 0.1 %
±1%
500 VDC
420 kHz
25 °C to50 °C
Efficiency (%)
Input voltage
Max input current (no load)
Max input current (full load)
Output voltage
Output current (min)
Output current (max)
Output ripple and noise
Efficiency
Line regulation
Load regulation
Isolation voltage
Switching frequency
Operating temperature-
86
84
82
80
35
45
50
55
60
65
70
75
Input Voltage (Volts)
Fig.2 Efficiency as a function of input voltage at full load
90
Input capacitor required for operation: 10 µF , 100V.
82
Efficiency (%)
All Specifications are typical at nominal line voltage(48V),
full load and 25 °C unless otherwise stated.
Pin
J1
J2
J3
J4
40
Pin connection
Vin +
Vin + Output
- Output
74
66
58
50
Dimensions: 60 × 57 × 6 mm
0
1
2
3
4
5
Output Current (Amps)
Fig.3 Efficiency as a function of output current (Vin=48V)
4
Philips Magnetic Products
vi d/ V 5
vi d / V 0 5
Oscillograms
vi d/ s n 0 5
Fig.4 Primary MOSFET (Q1) gate voltage(TP6)
Fig.5 Primary MOSFET (Q1) drain voltage(TP2)
vi d / V 0 1
vi d/ V 5
vi d/ s n 0 5
vi d/ s n 0 5
vi d/ s n 0 5
Fig.7 Synchronous rectifier (Q3) drain voltage (TP4)
vi d / V m 0 2
vi d/ V 1
Fig.6 Synchronous rectifier (Q2) drain voltage (TP3)
1 m vi d/ s
vi d/ s n 0 5
Fig.8 Control IC oscillator (TP5)
Fig.9 Output voltage ripple and noise (bandwidth 20 Mhz)
5
Philips Magnetic Products
Design of planar magnetics
Transformer losses
Losses in the ferrite core and windings are estimated for a
switching frequency of 400 kHz and an output current of
5 A.
Transformer design (T1)
In designing the power transformer the optimisation of a
number of design parameters has been investigated. These
are discussed here.
The primary to secondary turns ratio should be
approximately 4.5:1 to guarantee a secondary voltage of
5V at a minimum input voltage of 36V using a forward
converter operating at a maximum duty cycle of 70%.
Three turns ratios have been investigated ( 4:1, 4.5:1, 5:1)
in order to determine the minimum transformer losses.
The number of primary turns has been selected on the
basis of a trade off between minimising core losses and
copper losses. Consideration was also given to being able
to accommodate the transformer windings in a 6-layer
PCB construction. Hence three values of primary turns
were investigated ( 5, 8 and 9 turns).
Turns ratio
Copper losses in the transformer have been calculated for
DC only, which appears to be accurate enough for this
application. Methods to predict AC losses will be treated
in a.seperate application note on the winding design for
planar transformers.
DC resistance (mΩ)
primary
secondary
Primary inductance
(µH)
8:2
5:1
1.0
4.5
2.0
4.5
3 or 4
2
1 or 2
6 to 8
3 or 4
1
1 or 2
5 to 7
110
6
110
6
30
3
243
192
75
8:2
5:1
Primary current
Primary resistance
Primary loss
Secondary current
Secondary resistance
Secondary loss
Total copper loss
0.8
0.11
0.07
3.61
0.006
0.08
0.15
0.85
0.11
0.08
3.39
0.006
0.07
0.15
0.75
0.03
0.017
3.77
0.003
0.043
0.06
Core loss
0.56
0.77
2.1
Total losses (W)
0.71
0.91
2.15
table 2
The lowest overall losses are predicted for the turns tatio
of 9:2, which is chosen for the design.
Optimisation of switching frequency
The choice of a switching frequency close to 400 kHz
follows from an estimation of the total loss balance
between semiconductors and magnetics. A higher
frequency increases the loss in the switches, but ferrite
losses are lower. A higher frequency also reduces the ripple
current in the output inductor.
Ferrite core: E18/4/10-3F3 + PLT18/10/2-3F3
Turns ratio
9:2
Track width (mm)
primary
1.0
secondary
4.5
Number of PCB layers
primary
3 or 4
econdary
2
auxiliary
1 or 2
Total
6 to 8
9:2
f
(kHz)
Vin
(V))
Semicond.
losses (W)
Magnetics Total
losses (W) (W)
300
36
48
72
36
48
72
36
48
72
36
48
72
36
48
72
2.11
2.38
3.19
2.13
2.52
3.58
2.33
2.67
3.98
2.61
2.84
4.39
3.05
3.01
4.81
1.34
1.27
1.19
1.20
1.13
1.05
1.16
1.09
1.01
1.22
1.15
1.07
1.22
1.15
1.07
400
500
table1
600
Note 1: 2 oz copper (70 µm) is used in all cases.
700
The primary windings can be split in such a manner that
the secondary is embedded between two primary
windings. This technique, known as sandwiching or
interleaving, reduces leakage inductance.
table 3
6
Philips Magnetic Products
3.45
3.65
4.38
3.33
3.65
4.63
3.49
3.76
4.99
3.83
3.99
5.46
4.27
4.16
5.88
Design of planar inductor (L1)
The peak-to-peak ripple current in the output inductor is
designed to be approximately 20% of the full load output
current for the nominal input voltage of 48V.
The inductance to achieve this can be calculated from the
formula:
L=
Vsec • ton
10.66 • 1.38 µs
=
∆I
1
The increased ripple current will cause an increase in ∆B
which will lead to somewhat higher losses in the output
inductor.
Output capacitor design
Output ripple voltage is calculated using the formula:
1
∆Vo =
∫ dIL dt + ∆IL • ESR
C
= 14.7 µH
where ∆IL is the ripple current in the output inductor and
ESR is the equivalent series resistance of the output
capacitors.
where
Vsec = Peak secondary voltage = Ns /Np . Vin
= 2/9 . 48 V = 10.66 V
ton = Primary MOSFET on time = 1.38 . 10-6s
∆ I = Inductor ripple current
The first term is much smaller than the second due the
high capacitance of the output capacitors so that the
ripple voltage can be expressed as:
So ideally the inductance value should be 14.7 µH.
With 5 turns this means an inductance per turn of:
However, a check on the flux density shows that with a
peak current of 5.5 A this is too high, since:
AL =
L
N2
=
14.7 • 10-6
25
∆Vo = ∆IL • ESR
The worst case will be at maximum input voltage.
= 588 nH
Vsec = 2/9 • 72V = 16V
L = 10.8 µH
Using the standard core E18/4-3F3-A315-P, a check on
the flux density shows that with a peak current of 5.5A,
the maximum value is:
N • Ip • A L
5 • 5.5 • 588 • 10-9
Bmax =
=
= 409 mT
Ae
39.5 • 10-6
Maximum ripple current follows from:
∆Imax =
L
=
16 • 0.92 µs
10.8 • 10-6
= 1.35 A
For a ripple voltage of less than 40 mV, the equivalent
ESR should be less than 30mΩ. The capacitors chosen
meet this requirement.
where
Ip = Peak inductor current
B. = Maximum flux density
N = Number of turns
AL = Inductance per turn
Ae = Cross sectional area of core
This maximum flux density of 388 mT is excessive for
3F3 material. To reduce the maximum flux density using
the same core, the air-gap needs to be increased.
Consequently, the maximum flux density is set to 300
mT. Using this figure and working backwards to calculate
the required AL with N=5 turns and Ip=5.5 A gives:
AL =
Vsec • ton
B • Ae
0.3 • 39.5 • 10-6
=
= 431 nH
5 • 5.5
N • Ip
L = AL • N2 = 431 • 10-9 • 25 = 10.8 µH
7
Philips Magnetic Products
PCB layout
The multilayer FR4 PCB with 70 µm of copper comprises
all windings of the transformer and output inductor.
These windings are divided over the separate layers in the
following way:
transformer
primary (9turns):
-5 turns in layer 1
-4 turns in layer 6
secondary (2 turns):
-1 turn in layer 2
-1 turn in layer 5
sense (2 turns):
-1 turn in layer 3
-1 turn in layer 4
Fig.10 Component location
output inductor
-1 turn in layer 1
-1 turn in layer 2
-1 turn in layer 3
-1 turn in layer 4
-1 turn in layer 5
Fig.11 Solder mask layer 1
Fig.12 Solder mask layer 6
8
Philips Magnetic Products
Fig.13 PCB layer 1
Fig.14 PCB layer 2
Fig.15 PCB layer 3
Fig.16 PCB layer 4
Fig.17 PCB layer 5
Fig.18 PCB layer 6
9
Philips Magnetic Products
57 mm
60 mm
The complete converter
10
Philips Magnetic Products
Fig.19 Circuit diagram
11
Philips Magnetic Products
Components list
Reference
Part No.
Series
Description
TR1
E18/4/10-3F3
PLT18/10/2-3F3
E18/4/10-3F3
PLT18/10/2-3F3
IRF630S
Si9410DY
IRF7401
BCP56
BC848A
MBRD320
BAV70
BZX84C12
AS3843
IL206A
T1431
WCR
RC-01
RC-01
WCR
WCR
WCR
WCR
WCR
WCR
WCR
Planar E Core
Plate
Planar E Core
Plate
200V, 0.4Ω, MOSFET
30V, 30mΩ, MOSFET
20V, 22mΩ, MOSFET
80V, 1A, NPN Trans.
30V, 100mA,NPN Trans
20V, 3A, Schottky Diode
70V, 250mA Dual Diode
12V Zener Diode
PWM Controller
opto-isolator
Prog. Reference
100K, 0.1W
1K, 0.125W
1R, 0.25W
1K5, 0.1W
2K2, 0.1W
3K3, 0.1W
1K, 0.1W
10K, 0.1W
220R, 0.1W
15K, 0.1W
100nF,100V
SMD-220
SO-8
SO-8
SOT223
SOT23
D-Pak
SOT-23
SOT-23
SO-8
SO-8
SO-8
0805
1206
1206
0805
0805
0805
0805
0805
0805
0805
1812
TAJ
CG,2R
100µF, 10V
100nF, 63V
220nF
22nF
22pF
15nF
10nF 500V
D
1206
1206
0805
0805
0805
1206
Ll
Ql
Q2
Q3
Q4
Q5
Dl
D3
Z1
Ul
U2
U3
R1
R2
R4,R5,R18
R6
R8
R7,R9
R11,R14,R15
R10
R12
R16
C1,C21,C22,
C23,C24
C3,C4,C18
C5,C11,C12
C6
C7,C10
C9
C13
C2
12
Philips Magnetic Products
Package
Manufacturer
Philips
Philips
Philips
Philips
I.R.
Siliconix
I.R.
Philips
Motorola
P.S.
P.S.
Astec
Siemens
T.I.
Welwyn
Philips
Philips
Welwyn
Welwyn
Welwyn
Welwyn
Welwyn
Welwyn
Welwyn
Syfer
AVX
Philips
AVX
Philips
Philips
Kemet
AVX