SC4530 - Semtech

SC4530
POWER MANAGEMENT
30V, 300mA Output Micropower
Step-Down Switching Regulator
Features
Description
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Input Voltage Range: 3V to 30V
Low Quiescent Current: Drawing 19mA from VIN when
Stepping Down from 12V to 3.3V at No Load
High Efficiency from 12V Input to 5V Output
> 80% at 650mA
> 85% at 10mA - 300mA
Up to 300mA Continuous DC Output Current
Integrated Power Switch and Schottky Diodes
Low Output Ripple
<1mA Shutdown Current
Hysteretic Current-Mode Control
Cycle-by-Cycle Current Limiting
Alternating Between Micropower Idling and Switching States at Light Loads to Conserve Power
Output Short-Circuit Protection
Solution Footprint as Small as 50mm2
Low-Profile 3mm x 2mm MLPD 8-Lead Package
Applications
Portable Equipment
 Notebook Computers
 Distributed Supplies
 Backup Power Supplies

The SC4530 is a micropower hysteretic current-mode
step-down switching regulator capable of providing up
to 300mA of output current from 3V to 30V input voltage
range. It is designed to provide very high standby efficiency
while simplifying design.
At light loads, the SC4530 switches only as needed to maintain regulation, while idling most of the time, to improve
efficiency. Typical quiescent currents from VIN and BIAS
are 7mA and 26mA respectively. The control scheme produces less than 10mV of FB voltage ripple at light loads. The
SC4530 automatically switches to continuous-conduction
mode at heavy loads.
The SC4530 has integrated power devices and on-chip
control circuitry, simplifying design and enabling a solution footprint as small as 50mm2. Only an inductor and a
few passive components are needed to complete a DC-DC
regulator. The inductor current hysteretic control of SC4530
makes it inherently short-circuit robust. The wide input
voltage range enables the device to operate from a variety
of input sources, including single- or multi-cell batteries,
system rails and wall transformers.
Typical Application Circuit
Efficiency vs Load Current
90
VIN = 12V
B IA S
EN
BST
IN
SW
V IN
7V - 30V
S C 4530
C2
4.7µF
80
C3
0.22µF ,10V
L1
OUT
33µH
5V /0.3A
C4
10pF
R1
619k
C5
33pF
R2
200k
C1
22µF
FB
GND
Efficiency (%)
OFF ON
VIN = 24V
70
60
50
VOUT = 5V
L 1 : C o ilcra ft L P S 6 2 2 5
C 1 : M u ra ta G R M 3 1 C R 6 1 A 2 2 6 K
C 2 : M u ra ta G R M 3 1 C R 7 1 H 4 7 5 K
40
0.1
1.0
10.0
100.0
Load Current (mA)
1000.0
Figure 1. 5V Output Step-Down Converter
Rev. 2.5
C 4 = 1 0 p F C 5 = 3 3 p F V e rifie d 1 /1 2 /2 0 1 2
SC4530
Pin Configuration
FB
1
B IA S
2
BST
3
SW
4
Ordering Information
9
8
EN
7
NC
6
IN
5
GND
Device
Package
SC4530WLTRT(1) (2)
MLPD-W-8 3x2
SC4530EVB
Evaluation Board
Notes:
(1) Available in tape and reel only. A reel contains 3,000 devices.
(2) Available in lead-free package only. Device is WEEE and RoHS
compliant.
θJA = 80°C/W
MLPD: 3mm x 2mm 8 Lead
Marking Information
4530
xxxx
XXXX - Lot Number
SC4530
Absolute Maximum Ratings (1)
Recommended Operating Conditions
IN………………………………………………… -0.3V to 32V
Junction Temperature Range………………… -40°C to +125°C
SW ………………………………………………… -0.6V to VIN
VIN …………………………………………………… 3V to 30V
BST ………………………………………………………
42V
Output DC Current……………………………… up to 300mA
BST Above SW……………………………………………
30V
(2)
FB ……………………………………………… -0.3V to 1.9V
Thermal Information
BIAS ……………………………………………… -0.3V to VIN
Thermal Resistance, Junction to Ambient (4) ………… 80°C/W
EN ………………………………………………… -0.3V to VIN
Maximum Junction Temperature………………………+150 °C
Storage Temperature Range …………………-65°C to +150°C
ESD Protection Level (3) …………………………………… 2kV
Peak IR Reflow Temperature (10s to 30s)……………… +260°C
Exceeding the above specifications may result in permanent damage to the device or the device may malfunction. Operation outside of the
parameters specified in the Electrical Characteristics section is not recommended.
NOTES:
(1) Unless noted otherwise, all voltage values in this section are with respect to ground.
(2) The transient negative voltage specification for the SW pin is -1V for 100ns.
(3) Tested according to JEDEC standard JS - 001- 2012.
(4) Calculated from package in still air, mounted to 3” x 4.5”, 4-layer FR4 PCB with thermal vias under the exposed pad, per JESD51 standards.
Electrical Characteristics
Unless otherwise noted, TA = 25°C for typical values, -40°C < TA = TJ < 125°C. VIN = VEN = 10V, VBST = 15V, VBIAS = 3V.
Parameter
Name
Conditions
VIN Operating Range
Min
Typ
3
VIN Quiescent Supply Current
BIAS Quiescent Supply Current
30
V
0.1
0.5
µA
Not Switching
7
11
µA
Not Switching , VBIAS = 0
34
50
µA
VEN = 0.2V
0.1
0.6
µA
Not Switching
26
40
µA
Not Switching , VBIAS = 0
0.1
1
µA
2
V
EN Pin Input Low Voltage
0.2
V
1
2.5
µA
1.232
1.245
V
VIN = 3V to 30V
0.01
0.02
%/V
VFB = 1.25V
20
60
nA
EN Pin Current
VEN = 2.5V
Feedback Voltage
VFB Falling
Feedback Voltage Line Regulation
FB Pin Bias Current
Maximum Switch Duty Cycle
Units
VEN = 0.2V
EN Pin Input High Voltage
Minimum Switch Off-time
Max
1.212
TOFF(MIN)
DMAX
90
530
ns
96
%
SC4530
Electrical Characteristics (continued)
Unless otherwise noted, TA = 25°C for typical values, -40°C < TA = TJ < 125°C. VIN = VEN = 10V, VBST = 15V, VBIAS = 3V.
Parameter
Switch Current Limit
Name
Conditions
Min
Typ
Max
Units
ILIM
VFB = 0
0.39
0.50
0.66
A
Inductor Current Hysteresis (1)
Switch Saturation Voltage
Switch Leakage Current
VFB = 0
65
ISW = -0.3A
200
VSW = 0
mA
300
mV
2
µA
Switch Minimum Bootstrap Voltage
ISW = -0.3A
1.7
2.2
V
BST Pin Current
ISW = -0.3A
7.1
12
mA
ISW = -0.3A
700
Freewheeling Diode Forward Voltage
Freewheeling Diode Reverse Leakage
VD
VSW = 10V
Bootstrap Diode Forward Voltage
IBST = 40mA
Bootstrap Diode Reverse Leakage
VSW = 10V, VBIAS = 0
mV
15
700
µA
mV
1
µA
Notes:
(1) The inductor current hysteresis is the difference between the switch current limit and the freewheeling diode valley current.
Pin Descriptions
Pin #
Pin Name
Pin Function
1
FB
Inverting input of the error amplifier. The FB pin is tied to a resistive divider between the output and ground. The
voltage divider sets the output voltage.
2
BIAS
Anode of the internal bootstrap diode. BIAS also powers the internal control circuit if VBIAS > 2.3V. Tie to the output
of the DC-DC converter if VOUT > 2.5V. Tie BIAS to IN if VOUT is set below 2.5V.
3
BST
Power transistor driver supply. Connect an external bootstrap capacitor from the SW pin to this pin to generate a
drive voltage higher than VIN to fully saturate the internal power transistor.
4
SW
The power transistor emitter and the cathode of the freewheeling diode. The SW pin is connected to an inductor
and a bootstrap capacitor.
5
GND
6
IN
Power supply to the SC4530. It must be closely bypassed to the ground pin.
7
NC
No Connection.
8
EN
The enable pin for the SC4530. Driving this pin below 0.2V completely shuts off the SC4530. Applying more than 2V
to this pin enables the SC4530. If not driven from a control circuit, tie this pin to IN. This pin cannot be floated.
9
Exposed Pad
The exposed pad at the bottom of the package serves as a thermal contact to the circuit board. It is to be soldered
to the ground plane of the PC board.
Connect this pin to the PC board power ground plane.
SC4530
Block Diagram
IN
6
R
OC
IP K
+
}
R S1
P A R A S IT IC S
VOS
B IA S
2
EN
8
D2
5 3 0 ns
M in . t O FF
BANDGAP
REFERENCE
BST
3
530 ns
1 .2 32 V
R
Q
S
Q
Q1
SW
4
IC N T L
EA
DZ
D1
+
-
-
1
+
FB
V HYS
CMP
UC
R U N / ID L E
+
IV L Y
-
R
R S2
GND
5
Figure 2. SC4530 Block Diagram
SC4530
Typical Characteristics
Efficiency vs Load Current
VOUT = 3.3V
90
Efficiency vs Load Current
VOUT = 2.5V
90
Feedback Voltage vs Temperature
1.24
VIN = 10V
70
70
VIN = 12V
60
VIN = 24V
VIN = 5V
50
40
60
50
0.1
1.0
10.0
100.0
Load Current (mA)
0.1
1000.0
VBST = 15V
550
500
25
50
75
100
12
75
100
0.60
Valley
0.45
0.40
0.35
0.30
-50
125
-25
0
25
50
75
100
125
-50
-25
o
0
25
50
75
100
Temperature ( C)
Switch Saturation Voltage
vs Switch Current
BST Pin Current vs Switch Current
Minimum Bootstrap
Voltage vs Temperature
15
125 C
200
-55oC
100
2.2
ISW = -0.39A
2.0
125oC
10
VBST - VSW (V)
Bootstrap Current (mA)
25oC
25oC
5
0.1
0.2
0.3
0.4
Switch Current (A)
0.5
0.6
1.8
1.6
1.4
1.2
0
0
125
o
-55oC
VIN = 10V
VBST = 15V
o
300
(1)
VBST = 15V
Peak
Temperature ( C)
VBST = 15V
125
VIN = 10V
Temperature ( C)
VIN = 10V
0.0
50
VFB = 0
VIN = 10V
o
400
25
Thresholds vs Temperature
0.50
13
10
400
0
0
Temperature(o C)
VBST = 15V
-25
-25
0.55
11
450
-50
-50
Current (A)
600
1.21
1.20
1000.0
14
On Time ( P s)
Off Time (ns)
1.0
10.0
100.0
Load Current (mA)
Maximum On Time vs Temperature
15
VIN = 10V
650
1.22
Peak and Valley Current DC
Minimum Off Time vs Temperature
700
1.23
Coilcraft
LPS4018-333ML
30
30
Saturation Voltage (mV)
VIN = 24V
VIN = 12V
40
Coilcraft
LPS4018-333ML
Feedback Voltage (V)
80
Efficiency (%)
Efficiency (%)
VIN = 5V
80
0.0
0.1
0.2
0.3
0.4
Switch Current (A)
0.5
0.6
-50
-25
0
25
50
75
100
125
Temperature (o C)
Notes:
(1) Circuit propagation delays and the error amplifier output voltage ripples may cause the actual inductor valley current to differ from its DC value.
SC4530
Typical Characteristics (Continued)
125oC
100
25oC
10
10
0.2
0.4
0.6
0.8
5
0
1.0
5
10
20
25
30
o
25 C
1.0
0.1
35
0.2
0.4
Reverse Voltage (V)
Voltage (V)
Bootstrap Diode
Reverse Leakage Current
VIN = 10V
125oC
Current ( P A)
25oC
IIN
30
IBIAS
-40oC
20
-40oC
1
10
0
0
IBIAS
25
40
2
1.0
30
125oC
3
0.8
Quiescent Currents vs VIN
50
4
0.6
Voltage (V)
Quiescent Currents
vs BIAS Voltage
5
Reverse Current ( P A)
15
o
-40 C
10.0
0
1
20
15
IIN
10
5
VBIAS = 3V
0
5
10
15
20
25
30
35
40
0
0.0
1.0
2.0
3.0
VBST - VBIAS (V)
BIAS Voltage (V)
VIN Quiescent Current
vs Temperature
BIAS Quiescent Current
vs Temperature
0
4.0
20
25
30
EN Pin Current vs VEN
VBIAS = 3V
10
1.5
Current ( P A)
20
-40oC
VBIAS = 3V
30
Current ( P A)
30
20
10
0
25
50
75
o
Temperature ( C)
100
125
25oC
1.0
125oC
0.5
0
0
15
VIN = 10V
VBIAS = 1V
-25
10
2.0
VIN = 10V
-50
5
VIN (V)
40
40
Current ( P A)
o
125 C
Current (mA)
-40oC
Current ( P A)
Current (mA)
125oC
100.0
15
Reverse Current ( P A)
1000
Bootstrap Diode
Forward Characteristics
Freewheeling Diode
Reverse Leakage Current
Freewheeling Diode
Forward Characteristics
0.0
-50
-25
0
25
50
75
o
Temperature ( C)
100
125
0
5
10
15
20
25
30
VEN (V)
SC4530
General Description and Operation
The SC4530 is a micropower, hysteretic current-mode
step-down switching regulator. As shown in the block
diagram in Figure 2, the converter is controlled by an error
amplifier EA and two current-sensing comparators IPK and
IVLY. IPK and IVLY monitor the switch (Q1) collector current
and the freewheeling diode (D1) current respectively. The
EA amplifies the differential voltage between the FB and
the bandgap reference, and produces a current, ICNTL,
proportional to its output voltage. ICNTL, in turn, adjusts the
switching thresholds of both the peak and valley current
comparators. The EA output voltage is high at heavy
loads, as is the peak inductor current. The Zener diode
DZ clamps the amplifier output and sets the switch peak
current limit.
When the switch Q1 is turned on, the current through Q1
ramps up until it reaches the peak threshold set by ICNTL. The
output of the IPK comparator, OC, goes high. This resets
the latch and turns off the switch. With Q1 off, the inductor
current ramps down through the freewheeling diode
D1. When D1 current ramps below the valley threshold
established by ICNTL, the output of the IVLY comparator,
UC, goes high. If Q1 has been turned off for more than
530ns, then the latch will be set and Q1 will again turn on,
starting a new cycle.
The inductor ripple current in continuous-conduction
mode is independent of ICNTL and is primarily determined
by VOS and VHYS. Continuous mode switching frequency,
therefore, depends on VIN, VOUT, the inductance L and the
propagation delay times of the current comparators.
If the regulator output is shorted to ground, then the
amplifier output will rise to DZ clamp voltage. Q1 turns
off as the inductor current reaches the peak current limit.
With the output shorted to ground, the inductor current
ramps down at a slower rate through D1. Q1 turns on again
when the inductor current crosses the valley threshold.
Therefore, short-circuiting the output merely lowers the
converter switching frequency. The inductor current
remains bounded by the peak switch current limit.
The RUN/IDLE comparator, CMP, monitors the output of
the error amplifier. If the EA output falls below the RUN/
IDLE threshold, then Q1 and all control circuits except the
reference and EA will be shut off. The output capacitor
will then supply the load, causing the output voltage
to fall. When the EA output rises above the RUN/IDLE
threshold, the control circuit wakes up and the part starts
to switch, delivering power to the output. The offset
voltage VOS at the input of the IPK comparator ensures
that any current pulse delivered to the output has some
minimum amplitude.
At very light loads, even a single minimum charge packet
delivered to the output will cause the FB voltage to rise
above the reference voltage. This causes the EA output
voltage to fall and the part to idle. The part resumes
switching when the output current discharges the FB
voltage below the reference. At light loads, the part
switches only as needed to keep the output in regulation.
By reducing the supply current drawn when idling, high
efficiency is maintained at light loads. At heavier loads,
it may take a number of consecutive minimum pulses to
bring the FB above the reference voltage. The part enters
continuous conduction mode when the amplifier output
never falls below the RUN/IDLE threshold.
Driving the base of the power transistor above the input
power supply rail minimizes the power transistor turnon voltage and maximizes efficiency. A bootstrap circuit
[formed by an internal bootstrap diode D2 (Figure 2) and
an external capacitor connected between BST and SW]
generates a voltage higher than VIN at the BST pin. The
bootstrapped voltage becomes the supply voltage of the
power transistor driver.
The internal control circuit takes its power from either the
input or from the BIAS pin if VBIAS > 2.3V. For applications
with output voltage higher than 2.5V, the BIAS pin should
be tied to the regulator output to maximize efficiency.
SC4530
Applications Information
Setting the Output Voltage
mode (CCM) primarily depends on the input and output
voltages:
D
VO U T
(3)
where VCESAT = 0.25V is the switch saturation voltage and
VD = 0.6V is the forward voltage drop of the freewheeling
diode.
SC4530
R1
VOUT VD
VIN VD VCESAT
20nA
FB
R2
Figure 3. R1 and R2 Set the Output Voltage
The SC4530 output voltage is programmed using a
resistive divider (Figure 3) with its center tap tied to the
FB pin. For a given R2, R1 can be determined:
§ V
·
R1 R 2 ˜ ¨ OUT 1¸
1
.
232
©
¹
(1)
The percentage error due to the input bias current of the
error amplifier is:
'VOUT
VOUT
20nA ˜ 100 ˜ (R1°«R 2 )
1.232V
(2)
Whenever the power switch is turned off, it is kept off for
at least 530ns. Moreover, the control circuit prevents the
power transistor from turning on for more than 13.5ms.
The inductor current pulls the SW node low as the power
switch turns off, allowing the inductor current to charge
the bootstrap capacitor. The maximum on-time ensures
that the bootstrap capacitor gets replenished after a
long switch-on interval. The minimum off-time, together
with the maximum on-time, put an upper limit on the
achievable duty cycle (≈ 0.96). From Equation (3), the
minimum VIN to avoid dropout is:
VIN(MIN)
VOUT VD
VCESAT VD
0.96
(4)
Example: Determine the output voltage error caused by
the amplifier input bias current in a 5V output converter.
If VIN falls below this minimum, then the regulator will not
be able to attain its set output voltage regardless load.
Using Equation (4), the input supply voltage must be at
least 5.5V in order to generate a 5V output.
Assuming R2 = 200kW and using Equations (1) and (2),
Inductor Selection
§ 5
·
R1 200k: ˜ ¨
1¸ | 619k:
© 1.232 ¹
'VOUT
VOUT
20nA ˜100 ˜ (200k°«619k )
0.25%
1.232V
Using large R1 and R2 helps in maintaining light-load
efficiency, since the current drawn by the feedback
resistive divider is not delivered to the converter output.
The simple calculation above shows that relatively large
R1 and R2 can be used without introducing more error
than that resulting from the tolerance of the standard 1%
resistors.
Maximum Duty Cycle Limitation
The SC4530 is a non-synchronous, step-down switching
regulator. Its duty cycle in continuous-conduction
The SC4530 uses a hysteretic current-mode control
topology. The peak-to-peak inductor ripple current, ∆IL,
is theoretically constant. However, propagation delays
of the current comparators (IPK and IVLY in Figure 2), as
well as the error amplifier (EA) output ripples, will cause
the actual inductor ripple current to vary depending on
the input voltage and the duty cycle. The inductor should
be chosen so that the valley current comparator, not the
minimum off-time, determines the switch turn-on instant.
To simplify inductance calculation, we will assume that
∆IL is constant and equal to 150mA. Furthermore, we will
use 1.5 times the typical tOFF(MIN), to allow for tolerance and
temperature variation.
LMIN
VOUT VD ˜1.5 t OFF(MIN)
'IL
(5)
SC4530
Applications Information (Continued)
For a given VIN and inductance L, the continuousconduction switching frequency is:
f
( VOUT VD )(1 D)
L ˜ 'IL
( VOUT VD )( VIN VOUT VCESAT )
(6)
L ( VIN VD VCESAT )'IL
The minimum inductance is first found using Equation
(5). Next the switching frequencies are estimated at VIN
extremes using Equation (6). The inductance is then
adjusted for achieving desired switching frequency. The
resulting switch on-time at the maximum VIN must exceed
the minimum controllable switch on-time, which can be
as high as 180ns. This prevents the inductor current from
running away when the output is shorted to ground.
Example: Select the inductor for a 3.3V output regulator,
with input voltage ranging from 10V to 26V. The desired
switching frequency is about 600kHz.
The minimum inductance is found using Equation (5):
LMIN
(3.3 0.6) u 0.53 u 1.5
| 22PH
0.15
Duty cycles and switching frequencies at the input
voltage extremes can be found using Equations (3) and (6)
respectively. The results for 22mH and 33mH are tabulated
(Table 1).
Table 1. Estimated Switching Frequencies for 3.3V Output
Input Voltage
Duty Cycle
VIN (V)
D (%)
10
26
37.7
14.8
Switching Frequency
f (kHz)
L = 22µH
L = 33µH
740
490
1000
670
The 33mH inductance will be chosen, as it gives the desired
switching frequency range.
The resulting switch on-time is checked against the
minimum controllable switch on-time. The switch ontime can be calculated using Equation (7) below:
t ON
L ˜ 'IL
VIN VCESAT VOUT
(7)
With L = 33µH, the switch on-time at 10V and 26V VIN are
770ns and 220ns respectively, above the 180ns minimum
controllable on-time.
Table 2 lists some recommended inductor values for
various output voltages.
Table 2. Recommended Inductor Values
Inductor Value (µH)
Output Voltage
VOUT (V)
VIN = 16 V
1.8
22
-
2.5
22
33
3.3
22
33
5.0
33
33
12
68
68
18
-
100
VIN = 30 V
Low-cost inductors with powder iron cores are not
suitable for high-frequency switching due to their high
core losses. Inductors with ferrite cores are recommended
for high efficiency.
It should be noted that the inductor saturation current
should be designed based on the inductor peak current
in output short circuit and startup instead of the nominal
output current. If output short circuit protection or fast
dVIN/dt (>100V/ms) at startup(EN = VIN) is required at high
VIN (up to 30V), at least a 33µH inductor with a minimum
900mA saturation current is needed. If output short
circuit does not exist and the dVIN/dt at power on (EN =
VIN) is less than 100V/ms, inductors with 600mA or more
saturation current may be used. Please check with the
inductor manufacturers for the saturation current at the
maximum inductor temperature in the real application.
Input Capacitor Selection
A step-down regulator draws pulse current from the input
power supply. A capacitor placed between the supply and
the converter filters the AC current and keeps the current
drawn from the supply to a DC constant. The input
capacitance should be high enough to filter the pulse
input current. Its equivalent series resistance (ESR) should
be low so that power dissipated in the capacitor does
not result in significant temperature rise and degrade
reliability.
10
SC4530
Applications Information (Continued)
Multi-layer ceramic capacitors, which have very low ESR
(a few mW) and can easily handle high RMS ripple current,
are the ideal choice. A single 4.7µF (X5R or X7R) ceramic
capacitor should be enough for most applications. Using
a larger capacitor (for example, 10µF) will reduce SW
node jitters if the minimum input voltage is less than 0.7V
above the output voltage. For applications with high input
voltage, a small (1µF ~ 2.2µF) ceramic capacitor can be
placed in parallel with a low ESR electrolytic capacitor to
satisfy both the ESR and bulk capacitance requirements.
Output Capacitor Selection
The output ripple voltage ∆VOUT of a step-down regulator
in continuous conduction can be expressed as:
'VOUT
§
t t ·
'IL ¨¨ ESR ON OFF ¸¸
8C OUT ¹
©
(8)
where COUT is the output capacitance.
The first term in Equation (8) results from the equivalent
series resistance (ESR) of the output capacitor while the
rest is due to the charging and discharging of COUT by the
inductor ripple current.
Substituting ∆IL = 150mA, tON + tOFF = 2µs, COUT = 22µF and
ESR = 3mΩ in Equation (8), we get:
'VOUT
0.15A ˜ (3 m: 11.4 m:)
0.45 1.7 2.2 mV
Depending on the switching period and the type of the
capacitor used, the output voltage ripple resulting from
charging/discharging of COUT may be higher than the ripple
due to the ESR. The example above also shows that the
output voltage ripple in continuous mode is very low.
The SC4530 relies on fast amplifier response to reduce the
output voltage overshoot during power-up. Neither the
error amplifier output nor the reference is ramped during
start-up. The Zener diode DZ (refer to page 5) clamps
the amplifier output, while the regulator output voltage
ramps up. As a result, the switch Q1 is turned off every
cycle at the switch current limit, ILIM (typically 0.5A). The
regulator thus delivers about 0.5A to its output until VOUT
rises to its set value. If the load is light, then the amplifier
output voltage will fall below the RUN/IDLE threshold
following regulation. This causes the regulator to idle.
However the energy previously stored in the inductor
still flows to the output, causing the output voltage to
rise above its regulation level. The minimum output
capacitance required to keep the overshoot to less than
1% of the nominal output voltage is:
C OUT !
2
50 L ILIM
VOUT VOUT VD (9)
The minimum output capacitance for various output
voltages can be estimated from Equation (9) using the
inductances given in Table 2. The results are shown in
Table 3. Smaller output capacitors may also be used if
higher output voltage overshoot is acceptable.
Table 3. Calculated Minimum Output Capacitance for 1%
VOUT Overshoot during Start-up
Minimum COUT (µF)
VOUT (V)
VIN = 16V
VIN = 30V
1.8
2.5
3.3
5.0
12
18
64
36
22
15
5.6
-
53
33
15
5.6
3.7
Ceramic capacitors are the best choice for most
applications. Sanyo TPE series polymer capacitors in Bcase, which offer large capacitors (>100µF) with slightly
higher ESR, are also good alternatives. Ripple current in
the output capacitor is not a concern because the inductor
current of a step-down converter directly feeds COUT,
resulting in very low ripple current. Avoid using Z5U or
Y5V ceramic capacitors because these types of capacitors
have high temperature and high voltage coefficients.
Bootstrapping the Power Transistor
To reduce the switch on-state voltage and maximize
efficiency, the base of the power transistor should be
driven from a power supply higher in voltage than VIN.
The required driver supply voltage (at least 2.2V higher
than the SW) is generated with a bootstrap capacitor C3
connected between the BST and the SW nodes (Figure
11
SC4530
Applications Information (Continued)
circuit reaches equilibrium when the base charge drawn
from C3 during transistor on-time is equal to the charge
replenished during the off interval.
1) and the bootstrap diode D2 (Figure 2). The D2 anode is
connected to the BIAS pin.
During startup, the power transistor in the SC4530 is first
switched on so the current flows through to the inductor.
When the transistor is switched off, the inductor current
pulls the SW voltage low, allowing C3 to be charged
through the internal bootstrap diode D2. When the power
switch is turned on again, the SW voltage goes high. This
brings the BST voltage to VSW + VC3, thus back-biasing D2.
The C3 voltage increases with each subsequent switching
cycle, as does the bootstrapped voltage at the BST pin.
After a number of switching cycles, C3 will be fully charged
to a voltage approximately equal to that applied to the
anode of D2. The minimum BST to SW voltage required
to fully saturate the power transistor is shown in the
Typical Characteristics (pages 6-7). This difference voltage
must be at least 1.72V at room temperature. This is also
specified in the Electrical Characteristics (pages 3-4) as
the Minimum Bootstrap Voltage. The minimum required
VC3 increases as temperature decreases. The bootstrap
Figure 4 summarizes various ways of bootstrapping the
SC4530. In Figure 4(a) the BIAS pin is connected to the
converter output. The bootstrap charge is obtained
from the output of the step-down converter. The inputreferred charge is reduced by the step-down ratio. This is
the most efficient configuration and it also results in the
least voltage stress at the BST pin. The maximum BST pin
voltage is about VIN + VOUT. If the output voltage is between
2.5V and 3V, then a 0.33-0.47mF bootstrap capacitor may
be needed to reduce droop. In most other cases, a 0.22mF
ceramic capacitor is adequate.
Figure 4(b) shows the SC4530 can also be bootstrapped
from the input. This way it is not as efficient as the
configuration shown in Figure 4(a). However this may
be only option if the output voltage is less than 2.5V and
there is no other supply with voltage higher than 2.5V.
M a x V B S T ˜ 2 V IN
M a x V B S T ˜ V IN + V O U T
C3
BST
V IN
BST
V O U T > 2 .5 V
BIAS
SW
IN
V IN
BIAS
IN
SC4530
C3
SW
V O U T < 2 .5 V
SC4530
GND
GND
(a)
(b)
M a x V B S T ˜ V IN + V S
V S > 2 .5 V
0 .1 µF
V IN
BIAS
BST
SW
IN
C3
VOUT
SC4530
GND
V IN
(c)
Figure 4. Methods of Bootstrapping the SC4530
12
SC4530
Applications Information (Continued)
Voltage stress at the BST pin can be somewhat higher
than 2VIN. The BST pin voltage should not exceed its
absolute maximum rating of 42V. Figure 4(c) shows how
to bootstrap the SC4530 from an independent power
supply VS with its voltage > 2.5V.
V BST
V IN
same ground level. When the power transistor is turned
on, VSW should come within a few hundred millivolts of VIN
and VBST should have at least 2.2V of headroom above VSW.
As VS is reduced to 1.9V, excessive VBST droop decreases
transistor driver headroom, as shown in Figure 5(b). The
power transistor can no longer be fully saturated (as
evidenced by the round VSW turn-off corners), resulting in
high power dissipation. When bootstrapping from a lowvoltage output or supply, checking the bootstrap voltage
is a good precaution.
V SW
All Traces
2V/div
400ns/div
(a)
V BST
V IN
Since the inductor current charges C3, the bootstrap
circuit requires some minimum load current to function.
Figures 6(a) and 6(b) show the minimum input voltage
required to saturate the power transistor and to produce
a regulated output as a function of the load current. Once
started, the bootstrap circuit is able to sustain itself down
to zero load.
Feed-Forward Compensation
All Traces
2V/div
V SW
A feed-forward capacitor C4 (connected across the upper
feedback resistor R1) is needed for stability. An initial
estimate of C4 can be found using Equation (10) below:
(b)
Figure 5. Switching Waveforms of a 10V to 5V
Regulator
(a) Sufficient Bootstrap Voltage Drives the
Power Transistor into Saturation, Minimizing
Power Loss.
(b) Excessive Droop in Bootstrap Capacitor
Voltage fails to keep the Power Transistor
Saturated near the End of its Conduction
Cycles, Causing Jitters and Low Efficiency.
To demonstrate the effect of an under-sized bootstrap
capacitor, C3 (Figure 1, page 1) is deliberately reduced to
10nF. The BIAS pin is tied to an external power supply similar
to Figure 4(c). By adjusting the external supply voltage
VS, the bootstrap voltage can be varied. Figure 5(a) shows
the switching waveforms of a correctly bootstrapped 10V
to 5V regulator with VS = 2.5V. All three traces share the
6.8 u 10 6
R1
(10)
The value of C4 can be optimized empirically by observing
the inductor current and the output voltage during load
transient. Starting with the initial estimate, C4 is tuned until
there is no excessive ringing or overshoot in the inductor
current or the output voltage during load transient.
Minimum Input Voltage
7.0
VOUT = 5V
To Start
6.5
Input Voltage (V)
400ns/div
C4
6.0
Dropout
5.5
5.0
1
10
100
1000
Load Current (mA)
(a)
13
SC4530
Applications Information (Continued)
Mode Transition and the FB Pin
Minimum Input Voltage
5.5
VOUT = 3.3V
To Start
Input Voltage (V)
5.0
4.5
4.0
Dropout
3.5
3.0
1
10
100
1000
Load Current (mA)
(b)
Figure 6. The Minimum Input Voltage Required to Start
and to Operate Before Dropout
(a) VOUT =5V
(b) VOUT = 3.3V
If the upper feedback resistor R1 (Figure 3, page 9) is large
and is about the same magnitude as R2, then fast switching
transients may couple into the FB pin, disturbing or
delaying the transition from light-load operating mode
to continuous-conduction mode (CCM). As described
previously, the output ripple voltage is very low in
continuous-conduction mode. Delayed CCM transition
extends the load range in which the converter produces
larger output voltage ripples. This disturbance becomes
more pronounced when VIN is increased above 21V and
when large feedback resistors are used. The regulator
becomes insensitive to switching disturbances after it
enters continuous-conduction mode.
VOUT
50mV/div
AC Coupled
VOUT
50mV/div
AC Coupled
IL1
200mA/div
IL1
200mA/div
VSW
20V/div
VSW
20V/div
4ms/div
(a) C4 = 10pF, C5 not Placed
S W _V
O U T _ IL=
_ 2 85V,
V to 5IV @ 1 7 4=
m A174mA
_ C 4 = 1 0 p F _ o n set C C M = 1 7 6 m A
OUT
OUT
CCM Onset IOUT = 176mA
4ms/div
(a) C4 = 22pF, C5 not Placed
S W _V
O U T _ IL=
_ 2 83.3V,
V to 3 .3 VI@ 1 4 5=
m A145mA
_ C 4 = 2 2 p F _ o n se t C C M = 1 5 0 m A
OUT
OUT
CCM Onset IOUT = 150mA
VOUT
50mV/div
AC Coupled
VOUT
50mV/div
AC Coupled
IL1
200mA/div
IL1
200mA/div
VSW
20V/div
VSW
20V/div
10ms/div
(b) C4 = 10pF, C5 = 33pF
S W _ O U TV
_ ILOUT
_ 2 8 V=to 5V,
5 V @ 1I1OUT
0 m A _=
C 4110mA
= 1 0 p F _ C 5 = 3 3 p F _ o n se t C C M = 1 2 6 m A
CCM Onset IOUT = 126mA
Figure 7. Switching Waveforms of a 28V to 5V
Converter Just Before It Enters ContinuousConduction Mode
10ms/div
(b) C4 = 22pF, C5 = 47pF
VOUT = 3.3V, IOUT = 118mA
CCM Onset IOUT = 121mA
S W _ O U T _ IL _ 2 8 V to 3 .3 V @ 1 1 8 m A _ C 4 = 2 2 p F _ C 5= 4 7 p F _ o n se t C C M = 1 2 1 m A
Figure 8. Switching Waveforms of a 28V to 3.3V
Converter Just Before It Enters ContinuousConduction Mode
14
SC4530
Applications Information (Continued)
The operating mode transition can be significantly
smoothed by filtering the FB node. A capacitor between
FB pin and ground (Capacitor C5, as shown in Figures
13(a), page 18) serves this purpose. It should be chosen
so that it improves mode transition without significantly
slowing down load transients. Switching waveforms of a
5V output regulator (Figure 13(a), page 18) immediately
before it enters continuous-conduction mode are shown
in Figure 7. The inductor current waveform appears to be
more jagged without filtering. Moreover, transition to CCM
occurs at an output current of 176mA, instead of 126mA
with FB filtering. Figure 8 compares the corresponding
switching waveforms of an output 3.3V (Figure 14(a),
page 19) regulator.
If the converter output voltage is 1.8V or less, or if R2 is
reduced to below 2kW, then C5 will not be necessary. C5 is
also optional in Figures 13(a) and 14(a) if the maximum VIN
never exceeds 21V. Bench testing shows that removing C5
from these converters still results in acceptable transitional
behavior, provided that VIN < 21V.
C5 can be estimated using the following empirical
equation:
C5
8 u 10 6
C4
R1°«R 2
(11)
FB filtering has no significant impact on the output ripple
voltage. However, it improves the converter efficiency by
0.25% to 0.5% around the mode transition point (Figure
9). Regulator efficiencies are slightly lower (< 0.25%) at
light loads when filtering the FB voltage. Positive values
in Figure 9 imply that FB filtering improves efficiency
compared to no filtering.
Reverse Input Protection
Consider a circuit board where the input power source
supplies several DC-DC converters, including an SC4530
Effect of FB Filtering on
Efficiencies vs Load Current
1.0
Efficiency Difference (%)
VOUT = 5V
0.5
0.0
-0.5
VOUT = 3.3V
VIN = 28V
-1.0
1
10
100
Load Current (mA)
1000
Figure 9. Effect of FB Filtering on Converter Efficiency (VIN = 28V)
Plotted Efficiency = the Efficiency of a FB-Filtered Converter the Efficiency of the Same Converter without FB-Filtering
15
SC4530
Applications Information (Continued)
regulator with a large output capacitor. During poweroff, the SC4530 regulator output may be held high by its
output capacitor, while VIN is discharged rapidly by other
DC-DC converters. If VIN falls to two diode voltages below
VOUT, then the parasitic junction diodes inside the SC4530
(see Figure 2, page 5) will draw current from the output
through the SW pin to the input. If the load is light and
the output capacitor is large, then high reverse current
will flow, or even damage the internal circuits.
Figure 10 shows two protection schemes. In Figure 10(a),
a Schottky diode D4 placed at the input blocks the reverse
B IA S
O FF O N
V IN
EN
BST
IN
SW
D4
O UT
S C 4530
current. This method has the disadvantage that it lowers
the converter efficiency. A PN junction diode placed from
the converter output to the input [(as shown in Figure
10(b)] shunts the reverse current away from the part, thus
protecting the part. This scheme is not suitable in a power
supply system where a backup battery is diode OR-ed
with the SC4530 regulator output and with the SC4530
input grounded.
Board Layout Considerations
In a step-down switching regulator, the input bypass
capacitor, the main power switch and the freewheeling
power diode carry pulse current with high di/dt (Figure
11). To minimize jittering, the size of the loop formed by
these components must be minimized. Since the main
power switch and the freewheeling diode are already
integrated inside the part, connecting the input bypass
capacitor close to the ground pin minimizes size of the
switched current loop.
FB
GND
(a)
D4
1N4148
O FF O N
EN
V IN
IN
VO U T
BST
S C 4530
O UT
SW
ZL
B IA S
FB
GND
(b)
Figure 11. Heavy Lines Show the Fast Switching
Current Paths in a Step-down Converter.
The Input Capacitor Should be Placed
Close to the Part for Improved Switching
Performance.
Figure 10. Reversed Input Protection Schemes
(a) D4 Blocks the Reverse Current
(b) D4 Shunts the Reverse Current
from the Part During Power-off.
16
SC4530
Applications Information (Continued)
Shortening the traces at the SW and BST nodes reduces
the parasitic trace inductance at these nodes. This not only
reduces EMI, but also decreases switching voltage spikes
at these nodes. Shielding the FB trace from the SW and
the BST nodes with ground traces is a good precaution in
mitigating switching transient disturbance.
Figure 12 shows an example of external component
placement around the SC4530. The exposed pad should
be soldered to a large power ground plane as the ground
copper acts as a heat sink for the device.
Figure 12. Suggested PCB Layout for the SC4530
17
SC4530
Typical Application Circuits
B IA S
OFF ON
EN
BST
IN
SW
V IN
7V - 30V
C3
0.22mF ,10V
L1
S C 4530
C2
4.7mF
OUT
33mH
C4
10pF
5V /0.3A
R1
619k
C1
22mF
FB
C5
33pF
GND
L 1 : C o ilcra ft L P S 6 2 2 5
R2
200k
C 1 : M u ra ta G R M 3 1 C R 6 1 A 2 2 6 K
C 2 : M u ra ta G R M 3 1 C R 7 1 H 4 7 5 K
(a)
VOUT
20mV/div
AC Coupled
C 4 = 1 0 p F C 5 = 3 3 p F V e rifie d 1 /1 2 /2 0 1 2
C 4 = 2 2 p F C 5 = 6 8 p F V e rifie d 1 /2 4 /2 0 1 1
IL1
200mA/div
VSW
10V/div
VOUT
50mV/div
AC Coupled
IL1
200mA/div
VSW
10V/div
1ms/div
4ms/div
(c)
(b)
S W _ O U T _ IL _ 1 2 V to 5 V @ 6 0 m A _ C 4 = 1 0 p F _ C 5 = 3 3 p F
S W _ O U T _ IL _ 1 2 V to 5 V @ 1 0 u A _ C 4 = 1 0 p F _ C 5 = 3 3 p F
VOUT
200mV/div
AC Coupled
VOUT
10mV/div
AC Coupled
IL1
200mA/div
IL1
200mA/div
VSW
10V/div
1ms/div
(d)
S W _ O U T _ IL _ 1 2 V to 5 V @ 3 0 0 m A _ C 4 = 1 0 p F _ C 5 = 3 3 p F
40ms/div
(e)
O U T _ IL _ 1 2 V to 5 V _ ld _ tra n _ 0 -3 0 0 m A _ C 4 = 1 0 p F _ C 5 = 3 3 p F
Figure 13. (a) 5V Output Step-Down Converter
(b) Switching Waveforms of the Figure 13(a) Circuit. VIN = 12V, IOUT = 10mA
(c) VIN = 12V, IOUT = 60mA
(d) VIN = 12V, IOUT = 300mA
(e) Load Transient. VIN = 12V, IOUT is Switched Between 0 and 300mA
18
SC4530
Typical Application Circuits (Continued)
B IA S
OFF ON
EN
BST
IN
SW
V IN
5.5V - 30V
C3
0.22µF ,6.3V
L1
S C 4530
C2
4.7µF
OUT
33µH
C4
22pF
3.3V /0.3A
R1
332k
C1
22µF
FB
GND
C5
47pF
L 1 : C o ilcra ft L P S 6 2 2 5
C 1 : M u ra ta G R M 3 1 C R 6 0 J2 2 6 K
C 2 : M u ra ta G R M 3 1 C R 7 1 H 4 7 5 K
R2
200k
(a)
VOUT
50mV/div
C 4 = 2 2 p F C 5 = 4 7 p F to re d u ce sta rt-u p o veAC
rsh oCoupled
o t 3 /1 4 /2 0 1 2
VOUT
20mV/div
AC Coupled
C 4 = 1 0 p F C 5 = 4 7 p F V e rifie d 1 /1 3 /2 0 1 2
IL1
200mA/div
C 4 = 2 2 p F C 5 = 4 7 p F V e rifie d 1 /2 6 /2 0 1 1
IL1
200mA/div
VSW
20V/div
VSW
20V/div
400ms/div
4ms/div
(b)
(c)
S W _ O U T _ IL _ 2 4 V to 3 .3 V @ 1 0 0 u A _ C 4 = 2 2 p F _ C 5= 4 7 p F
VOUT
10mV/div
AC Coupled
S W _ O U T _ IL _ 2 4 V to 3 .3 V @ 6 0 m A _ C 4 = 2 2 p F _ C 5= 4 7 p F
VOUT
200mV/div
AC Coupled
IL1
200mA/div
IL1
200mA/div
VSW
20V/div
1ms/div
40ms/div
(d)
(e)
S W _ O U T _ IL _ 2 4 V to 3 .3 V @ 3 0 0 m A _ C 4 = 2 2 p F _ C 5= 4 7 p F
O U T _ IL _ 2 4 V to 3 .3 V _ ld _ tra n _ 0 -3 0 0 m A _ C 4 = 2 2 p F _ C 5 = 4 7 p F
Figure 14. (a) 3.3V Output Step-Down Converter
(b) Switching Waveforms of the Figure 14(a) Circuit. VIN = 24V, IOUT = 100mA
(c) VIN = 24V, IOUT = 60mA
(d) VIN = 24V, IOUT = 300mA
(e) Load Transient. VIN = 24V, IOUT is Switched Between 0 and 300mA
19
SC4530
Typical Application Circuits (Continued)
B IA S
B IA S
EN
OFF ON
BST
V IN
3V - 20V
C3
0.22µF , 35V
L1
SW
IN
C2
4.7µF
BST
IN
SW
V IN
OUT
33µH
S C 4530
EN
OFF ON
1.23V /0.3A
4V - 20V
L1
C2
4.7µF
OUT
33µH
S C 4530
C1
150µF
FB
C3
0.22µF , 35V
C4
33pF
C1
100µF
R2
200k
GND
L 1 : C o ilcra ft L P S 6 2 2 5
C 1 : S a n yo P O S C A P 4 T P E 1 5 0 M A U B
C 2 : M u ra ta G R M 3 1 C R 7 1 H 4 7 5 K
Figure 15. 1.23V Output Step-Down Converter
R1
93.1k
FB
GND
L 1 : C o ilcra ft L P S 6 2 2 5
1.8V /0.3A
C 1 : M u ra ta G R M 3 1 C R 6 0 G 1 0 7 M
C 2 : M u ra ta G R M 3 1 C R 7 1 H 4 7 5 K
Figure 16. 1.8V Output Step-Down Converter
V e rifie d 2 /1 7 /2 0 1 1
B IA S
B IA S
OFF ON
EN
BST
V IN
SW
IN
4V - 30V
S C 4530
C2
10µF
LE1Nch a n g e d froBmS2T2 u H to 3 3Cu3H 1 /1 7 /2 0 1 2
OFF ON
C3
0.47µF , 6.3V
0.22mF ,16V
L1
OUT
33µH
2.5V /0.3A
C4
33pF
V IN
IN
12V - 30V
C2
4.7mF
C1
47µF
GND
C5
47pF
L 1 : C o ilcra ft L P S 6 2 2 5
C 1 : M u ra ta G R M 2 1 B R 6 0 J4 7 6 M
C 2 : M u ra ta G R M 3 2 E R 7 1 H 1 0 6 K
C 4 = 3 3 p F C 5 = 4 7 p F to re d u ce sta rt-u p o ve rsh o o t 3 /1 4 /2 0 1 2
OFF ON
EN
C 4 = 3 3 p F C 5 = 1 0 0 p F V e rifie d 1 /2 7 /2 0 1 1
R1
931k
C1
10mF
GND
C5
68pF
L 1 : C o ilcra ft L P S 6 2 2 5
C 1 : M u ra ta G R M 3 1 C R 6 1 C 1 0 6 K
C 2 : M u ra ta G R M 3 1 C R 7 1 H 4 7 5 K
R2
147k
Figure 18. 9V Output Step-Down Converter
B IA S
BST
V IN
SW
IN
15V - 30V
C4
6.8pF
9V /0.3A
FB
R2
200k
Figure 17. 2.5V Output Step-Down Converter
OUT
47mH
S C 4530
R1
205k
FB
L1
SW
S C 4530
C2
4.7µF
C3
0.22µF ,25V
L1
OUT
68µH
C4
10pF
12V /0.3A
R1
866k
C1
10µF
FB
C5
100pF
GND
L 1 : C o ilcra ft L P S 6 2 3 5
R2
100k
C 1 : M u ra ta G R M 3 1 C R 6 1 C 1 0 6 K
C 2 : M u ra ta G R M 3 1 C R 7 1 H 4 7 5 K
Figure 19.
Output
C h a n g12V
e C 4 fro
m 2 2 p F to 1 0Step-Down
p F a n d C 5 fro m 4 7 0 Converter
p F to 1 0 0 p F , 3 /1 5 /2 0 1 2
L 1 ch a n g e d fro m 4 7 u H to 6 8 u H 1 /1 6 /2 0 1 2
* L P S 4 0 1 8 6 8 u H sa tu ra tio n cu rre n t is m a rg in a lly lo w
20
SC4530
Outline Drawing – MLPD-W-8 3x2
21
SC4530
Land Pattern – MLPD-W-8 3x2
22
SC4530
© Semtech 2014
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Contact Information
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Power Management Products Division
200 Flynn Road, Camarillo, CA 93012
Phone: (805) 498-2111 Fax: (805) 498-3804
www.semtech.com
23