FAN9611 Interleaved Dual BCM PFC Controllers Features Description Sync-Lock™ Interleaving Technology for 180° Out-of-Phase Synchronization Under All Conditions Automatic Phase Disable at Light Load Minimum Restart Frequency to Avoid Audible Noise The FAN9611 interleaved dual Boundary-ConductionMode (BCM) Power-Factor-Correction (PFC) controller operates two parallel-connected boost power trains 180° out of phase. Interleaving extends the maximum practical power level of the control technique from about 300 W to greater than 800 W. Unlike the continuous conduction mode (CCM) technique often used at higher power levels, BCM offers inherent zero-current switching of the boost diodes, which permits the use of less expensive diodes without sacrificing efficiency. Furthermore, the input and output filters can be smaller due to ripple current cancellation and effective doubling of the switching frequency. Open-Feedback Protection Dead-Phase Detect Protection 2.0 A Sink, 1.0 A Source, High-Current Gate Drivers High Power Factor, Low Total Harmonic Distortion Voltage-Mode Control with (VIN)2 Feedforward Closed-Loop Soft-Start with User-Programmable Soft-Start Time for Reduced Overshoot Maximum Switching Frequency Clamp Brownout Protection with Soft Recovery Non-Latching OVP on FB Pin and Latching SecondLevel Protection on OVP Pin Power-Limit and Current Protection for Each Phase Low Startup Current of 80 µA Typical Works with DC and 50 Hz to 400 Hz AC Inputs Applications 100-1000 W AC-DC Power Supplies Large Screen LCD-TV, PDP-TV, RP-TV Power High-Efficiency Desktop and Server Power Supplies Networking and Telecom Power Supplies Solar Micro Inverters The converters operate with variable frequency, which is a function of the load and the instantaneous input / output voltages. The switching frequency is limited between 16.5 kHz and 525 kHz. The Pulse Width Modulators (PWM) implement voltage-mode control with input voltage feedforward. When configured for PFC applications, the slow voltage regulation loop results in constant on-time operation within a line cycle. This PWM method, combined with the BCM operation of the boost converters, provides automatic power factor correction. The controller offer bias UVLO of 10 V / 7.5 V, input brownout, over-current, open-feedback, output overvoltage, and redundant latching over-voltage protections. Furthermore, the converters’ output power is limited independently of the input RMS voltage. Synchronization between the power stages is maintained under all operating conditions. Related Resources Evaluation Board: FEBFAN9611_S01U300A Figure 1. Simplified Application Diagram © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com FAN9611 — Interleaved Dual BCM PFC Controllers February 2013 Part Number FAN9611MX Package 16-Lead, Small Outline Integrated Circuit (SOIC) Packing Method Packing Quantity Tape and Reel 2,500 This device passed wave soldering test by JESD22A-111. Package Outlines FAN9611 — Interleaved Dual BCM PFC Controllers Ordering Information Figure 2. SOIC-16 (Top View) Thermal Resistance Table Thermal Resistance Package 16-Lead SOIC Suffix M ΘJL(1) ΘJA(2) 35°C/W 50 – 120°C/W(3) Notes: 1. Typical ΘJL is specified from semiconductor junction to lead. 2. Typical ΘJA is dependent on the PCB design and operating conditions, such as air flow. The range of values covers a variety of operating conditions utilizing natural convection with no heatsink on the package. 3. This typical range is an estimate; actual values depend on the application. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 2 V IN L2a D2 V OUT V LINE L2b R ZCD2 C IN RIN1 L1a D1 R ZCD1 L1b AC IN EMI Filter RIN2 C 5VB 1 ZCD1 CS1 16 2 ZCD2 CS2 15 3 5VB VDD 14 4 MOT DRV1 13 5 AGND DRV2 12 6 SS PGND 11 7 COMP 8 FB R MOT R INHYST C SS R FB1 V BIAS R FB2 Q1 RG1 Q2 RG2 R OV1 R COMP VIN 10 C COMP,LF C COMP,HF RCS1 OVP 9 C VDD1 C VDD2 C INF Figure 3. Typical Application Diagram Block Diagram RCS2 R OV2 C OUT FAN9611 — Interleaved Dual BCM PFC Controllers Typical Application Diagram Figure 4. Block Diagram © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 3 FAN9611 — Interleaved Dual BCM PFC Controllers Pin Configuration Figure 5. Pin Layout (Top-View) Pin Definitions Pin # Name Description 1 ZCD1 Zero Current Detector for Phase 1 of the interleaved boost power stage. 2 ZCD2 Zero Current Detector for Phase 2 of the interleaved boost power stage. 3 5VB 5V Bias. Bypass pin for the internal supply, which powers all control circuitry on the IC. 4 MOT Maximum On-Time adjust for the individual power stages. 5 AGND Analog Ground. Reference potential for all setup signals. 6 SS 7 COMP 8 FB 9 OVP Output Voltage monitor for the independent, second-level, latched OVP protection. 10 VIN Input Voltage monitor for brownout protection and input-voltage feedforward. 11 PGND Power Ground connection. 12 DRV2 Gate Drive Output for Phase 2 of the interleaved boost power stage. 13 DRV1 Gate Drive Output for Phase 1 of the interleaved boost power stage. 14 VDD External Bias Supply for the IC. 15 CS2 Current Sense Input for Phase 2 of the interleaved boost power stage. 16 CS1 Current Sense Input for Phase 1 of the interleaved boost power stage. Soft-Start Capacitor. Connected to the non-inverting input of the error amplifier. Compensation Network connection to the output of the gM error amplifier Feedback pin to sense the converter’s output voltage; inverting input of the error amplifier. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 4 Stresses exceeding the absolute maximum ratings may damage the device. The device may not function or be operable above the recommended operating conditions and stressing the parts to these levels is not recommended. In addition, extended exposure to stresses above the recommended operating conditions may affect device reliability. The absolute maximum ratings are stress ratings only. Symbol Parameter Min. Max. Unit VDD Supply Voltage to AGND & PGND -0.3 20.0 V VBIAS 5VB Voltage to AGND & PGND -0.3 5.5 V Voltage On Input Pins to AGND (Except FB Pin) -0.3 VBIAS + 0.3 V Voltage On FB Pin (Current Limited) -0.3 VDD + 0.8 V Voltage On Output Pins to PGND (DRV1, DRV2) -0.3 VDD + 0.3 V Gate Drive Peak Output Current (Transient) 2.5 A Gate Drive Output Current (DC) 0.05 A TL Lead Soldering Temperature (10 Seconds) +260 ºC TJ Junction Temperature -40 +150 ºC TSTG Storage Temperature -65 +150 ºC IOH, IOL FAN9611 — Interleaved Dual BCM PFC Controllers Absolute Maximum Ratings Recommended Operating Conditions The Recommended Operating Conditions table defines the conditions for actual device operation. Recommended operating conditions are specified to ensure optimal performance to the datasheet specifications. Fairchild does not recommend exceeding them or designing to Absolute Maximum Ratings. Symbol Parameter Min. Typ. Max. Unit 12 18 V 5 V VDD Supply Voltage Range 9 VINS Signal Input Voltage 0 ISNK Output Current Sinking (DRV1, DRV2) 1.5 2.0 A ISRC Output Current Sourcing (DRV1, DRV2) 0.8 1.0 A (4) LMISMATCH Boost Inductor Mismatch TA ±5% Operating Ambient Temperature -40 ±10% +125 ºC Note: 4. While the recommended maximum inductor mismatch is ±10% for optimal current sharing and ripple-current cancellation, there is no absolute maximum limit. If the mismatch is greater than ±10%, current sharing is proportionately worse, requiring over-design of the power supply. However, the accurate 180° out-of-phase synchronization is still maintained, providing current cancellation, although its effectiveness is reduced. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 5 Unless otherwise noted, VDD = 12 V, TJ = -40°C to +125°C. Currents are defined as positive into the device and negative out of the device. Symbol Parameter Conditions Min. Typ. Max. Unit VDD = VON – 0.2 V 80 110 µA Output Not Switching 3.7 5.2 mA 4 6 mA Supply ISTARTUP IDD IDD_DYM Startup Supply Current Operating Current Dynamic Operating Current (5) fSW = 50 kHz; CLOAD = 2 nF VON UVLO Start Threshold VDD Increasing 9.5 10.0 10.5 V VOFF UVLO Stop Threshold Voltage VDD Decreasing 7.0 7.5 8.0 V VHYS UVLO Hysteresis VON – VOFF 2.5 TA = 25°C; ILOAD = 1 mA 5.0 V Bias Regulator (C5VB = 0.1 µF) V5VB IOUT_MAX 5VB Output Voltage Total Variation Over Line, Load, and Temperature Maximum Output Current 4.8 5.2 5.0 V mA Error Amplifier TA = 25°C 2.95 Total Variation Over Line, Load, and Temperature 2.91 3.075 Input Bias Current VFB = 1 V to 3 V; |VSS – VFB| ≤ 0.1V -0.2 0.2 µA IOUT_SRC Output Source Current VSS = 3 V; VFB = 2.9 V -13.7 -8 -4 µA IOUT_SINK Output Sink Current VSS = 3 V; VFB = 3.1 V 4 8 12 µA VOH Output High Voltage 4.5 4.7 V5VB V VOL Output Low Voltage 0.0 0.1 0.2 V gM Transconductance 50 78 115 µmho 120 195 270 mV 0 µs VEA Voltage Reference IBIAS ISINK < 100 µA 3.00 3.05 FAN9611 — Interleaved Dual BCM PFC Controllers Electrical Characteristics V PWM VRAMP,OFST PWM Ramp Offset tON,MIN Minimum On-Time TA = 25°C VFB > VSS Maximum On-Time VMOT tON,MAX Maximum On-Time Voltage R = 125 kΩ 1.16 1.25 1.30 V Maximum On-Time R = 125 kΩ; VVIN = 2.5 V; VCOMP > 4.5 V; TA = 25°C 3.4 5.0 6.6 µs VFB > VPWM_OFFSET 12.5 16.5 20.0 kHz 400 525 630 kHz Restart Timer (Each Channel) fSW,MIN Minimum Switching Frequency Frequency Clamp (Each Channel) fSW,MAX Maximum Switching Frequency(5) Continued on the following page… © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 6 Unless otherwise noted, VDD = 12 V, TJ = -40°C to +125°C. Currents are defined as positive into the device and negative out of the device. Symbol Parameter Conditions Min. Typ. Max. Unit 0.19 0.21 0.23 V 0.2 µA 85 100 ns Current Sense VCS CS Input Threshold Voltage Limit ICS CS Input Current VCSX = 0 V to 1 V CS to Output Delay CS Stepped from 0 V to 5 V tCS_DELAY -0.2 Zero Current Detection VZCD_IN Input Voltage Threshold(5) VZCD is Falling -0.1 0 0.1 V VZCD_H Input High Clamp Voltage IZCD = 0.5 mA 0.8 1.0 1.2 V VZCD_L Input Low Clamp Voltage IZCD = –0.5 mA -0.7 -0.5 -0.3 V 1 mA 10 mA IZCD_SRC IZCD_SNK tZCD_DLY (5) Source Current Capability (5) Sink Current Capability (5) Turn-On Delay ZCDx to OUTx 180 ns VOUTx = VDD/2; CLOAD = 0.1 µF 2.0 A Output ISINK ISOURCE OUTx Sink Current (5) VOUTx = VDD/2; CLOAD = 0.1 µF 1.0 tRISE Rise Time CLOAD = 1 nF, 10% to 90% 10 25 ns tFALL Fall Time CLOAD = 1 nF, 90% to 10% 5 20 ns 1 V VO_UVLO IRVS OUTx Source Current (5) Output Voltage During UVLO VDD = 5 V; IOUT = 100 µA (5) Reverse Current Withstand A 500 FAN9611 — Interleaved Dual BCM PFC Controllers Electrical Characteristics (Continued) mA Soft-Start (CSS = 0.1 µF) ISS_MAX ISS_MIN Maximum Soft-Start Current (5) Minimum Soft-Start Current VCOMP < 3.0 V -7 -5 -3 µA VCOMP > 4.5 V -0.40 -0.25 -0.10 µA 0.76 0.925 1.10 V 0.2 µA Input Brownout Protection VIN_BO Input Brownout Threshold IVINSNK VIN Sink Current VVIN > 1.1 V -0.2 VVIN < 0.8 V 1.4 2 2.5 µA V Input-Voltage Feedforward Range VFF_UL VIN Feedforward Upper Limit (5) 3.1 3.7 4.3 VFF_RATIO (5) 3.6 4.0 4.3 VFF_UL / VIN_BO Phase Management VPH,DROP Phase Dropping Threshold VCOMP Decreasing, Transition from 2 to 1 Phase, TA = 25°C 0.66 0.73 0.80 V VPH,ADD Phase Adding Threshold VCOMP Increasing, Transition from 1 to 2 Phase, TA = 25°C 0.86 0.93 1.00 V 3.15 3.25 3.35 V Over-Voltage Protection Using FB Pin – Cycle-by-Cycle (Input) VOVPNL Non-Latching OVP Threshold (+8% above VOUT_NOMINAL) VOVPNL_HYS OVP Hysteresis TA = 25°C DRV1=DRV2=0 V FB Decreasing 0.24 V Over-Voltage Protection Using OVP Pin – Latching (Input) VOVPLCH Latching OVP Threshold (+15%) DRV1=DRV2=0 V 3.36 3.50 3.65 V Note: 5. Not tested in production. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 7 There are many fundamental differences in CCM and BCM operations and the respective designs of the boost converter. 1. Boundary Conduction Mode The boost converter is the most popular topology for power factor correction in AC-to-DC power supplies. This popularity can be attributed to the continuous input current waveform provided by the boost inductor and to the fact that the boost converter’s input voltage range includes 0 V. These fundamental properties make close to unity power factor easier to achieve. The FAN9611 utilizes the boundary conduction mode control algorithm. The fundamental concept of this operating mode is that the inductor current starts from zero in each switching period, as shown in the lower waveform in Figure 7. When the power transistor of the boost converter is turned on for a fixed amount of time, the peak inductor current is proportional to the input voltage. Since the current waveform is triangular, the average value in each switching period is also proportional to the input voltage. In the case of a sinusoidal input voltage waveform, the input current of the converter follows the input voltage waveform with very high accuracy and draws a sinusoidal input current from the source. This behavior makes the boost converter in BCM operation an ideal candidate for power factor correction. This mode of control of the boost converter results in a variable switching frequency. The frequency depends primarily on the selected output voltage, the instantaneous value of the input voltage, the boost inductor value, and the output power delivered to the load. The operating frequency changes as the input voltage follows the sinusoidal input voltage waveform. The lowest frequency operation corresponds to the peak of the sine waveform at the input of the boost converter. Even larger frequency variation can be observed as the output power of the converter changes, with maximum output power resulting in the lowest operating frequency. Theoretically, under zero-load condition, the operating frequency of the boost converter would approach infinity. In practice, there are natural limits to the highest switching frequency. One such limiting factor is the resonance between the boost inductor and the parasitic capacitances of the MOSFET, the diode, and the winding of the choke, in every switching cycle. Figure 6. Basic PFC Boost Converter The boost converter can operate in continuous conduction mode (CCM) or in boundary conduction mode (BCM). These two descriptive names refer to the current flowing in the energy storage inductor of the boost power stage. FAN9611 — Interleaved Dual BCM PFC Controllers Theory of Operation Another important characteristic of the BCM boost converter is the high ripple current of the boost inductor, which goes from zero to a controlled peak value in every switching period. Accordingly, the power switch is stressed with high peak current. In addition, the high ripple current must be filtered by an EMI filter to meet high-frequency noise regulations enforced for equipment connecting to the mains. The effects usually limit the practical output power level of the converter. Figure 7. CCM vs. BCM Control As the names indicate, the current in Continuous Conduction Mode (CCM) is continuous in the inductor; while in Boundary Conduction Mode (BCM), the new switching period is initiated when the inductor current returns to zero. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 8 The FAN9611 control IC is configured to control two boost converters connected in parallel, both operated in boundary conduction mode. In this arrangement, the input and output voltages of the two parallel converters are the same and each converter is designed to process approximately half the total output power. The pulse width modulator implements voltage mode control. This control method compares an artificial ramp to the output of the error amplifier to determine the desired on-time of the converter’s power transistor to achieve output voltage regulation. Figure 9. PWM Operation Figure 8. Interleaved PFC Boost Operation In FAN9611, there are two PWM sections corresponding to the two parallel power stages. For proper interleaved operation, two independent 180degree out-of-phase ramps are needed; which necessitates the two pulse width modulators. To ensure that the two converters process the same amount of power, the artificial ramps have the same slope and use the same control signal generated by the error amplifier. Parallel power processing is penalized by the increased number of power components, but offers significant benefits to keep current and thermal stresses under control and to increase the power handling capability of the otherwise limited BCM PFC control solution. Furthermore, the switches of the two boost converters can be operated 180 degrees out of phase from each other. The control of parallel converters operating 180 degrees out of phase is called interleaving. Interleaving provides considerable ripple current reduction at the input and output terminals of the power supply, which favorably affects the input EMI filter requirements and reduces the high-frequency RMS current of the power supply output capacitor. FAN9611 — Interleaved Dual BCM PFC Controllers a sine square function. Eliminating the line frequency component from the feedback system is imperative to maintain low total harmonic distortion (THD) in the input current waveform. 2. Interleaving 4. Input-Voltage Feedforward Basic voltage-mode control, as described in the previous section, provides satisfactory regulation performance in most cases. One important characteristic of the technique is that input voltage variation to the converter requires a corrective action from the error amplifier to maintain the output at the desired voltage. When the error amplifier has adequate bandwidth, as in most DC-DC applications, it is able to maintain regulation within a tolerable output voltage range during input voltage changes. There is an obvious difficulty in interleaving two BCM boost converters. Since the converter’s operating frequency is influenced by component tolerances in the power stage and in the controller, the two converters operate at different frequencies. Therefore special attention must be paid to ensure that the two converters are locked to 180-degree out-of-phase operation. Consequently, synchronization is a critical function of an interleaved boundary conduction mode PFC controller. It is implemented in the FAN9611 using proprietary and dedicated circuitry called Sync-Lock™ interleaving technology. On the other hand, when voltage-mode control is used in power factor corrector applications; the error amplifier bandwidth, and its capability to quickly react to input voltage changes, is severely limited. In these cases, the input voltage variation can cause excessive overshoot or droop at the converter output as the input voltage goes up or down. To overcome this shortcoming of the voltage-mode PWM circuit in PFC applications, input-voltage feedforward is often employed. It can be shown mathematically that a PWM ramp proportional to the square of the input voltage rejects the effect of input voltage variations on the output voltage and eliminates the need of any correction by the error amplifier. 3. Voltage Regulation, Voltage Mode Control The power supply’s output voltage is regulated by a negative feedback loop and a pulse width modulator. The negative feedback is provided by an error amplifier that compares the feedback signal at the inverting input to a reference voltage connected to the non-inverting input of the amplifier. Similar to other PFC applications, the error amplifier is compensated with high DC gain for accurate voltage regulation, but very low bandwidth to suppress line frequency ripple present across the output capacitor of the converter. The line frequency ripple is the result of the constant output power of the converter and the fact that the input power is the product of a sinusoidal current and a sinusoidal voltage thus follows © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 9 Figure 12. PWM Cycle Start Figure 10. Input-Voltage Feedforward 6. Terminating the Conduction Interval When the PWM ramp is made proportional to the input voltage squared, the system offers other noteworthy benefits. The first is the input voltage-independent small signal gain of the closed loop power supply, which makes compensation of the voltage regulation loop much easier. The second side benefit is that the output of the error amplifier becomes directly proportional to the input power of the converter. This phenomenon is very significant and it is re-visited in Section 9 describing light-load operation. Terminating the conduction period of the boost transistor in boundary conduction mode controllers is similar to any other pulse width modulator. During normal operation, the PWM comparator turns off the power transistor when the ramp waveform exceeds the control voltage provided by the error amplifier. In the FAN9611 and in similar voltage-mode PWMs, the ramp is a linearly rising waveform at one input of the comparator circuit. 5. Starting a PWM Cycle The principle of boundary conduction mode calls for a pulse width modulator able to operate with variable frequency and initiate a switching period whenever the current in the boost inductor reaches zero. Therefore, BCM controllers cannot utilize a fixed frequency oscillator circuit to control the operating frequency. Instead, a zero current detector is used to sense the inductor current and turn on the power switch once the current in the boost inductor reaches zero. This process is facilitated by an auxiliary winding on the boost inductor. The voltage waveform of the auxiliary winding can be used for indirect detection of the zero inductor current condition of the boost inductor. Therefore it should be connected to the zero current detect input, as shown in Figure 11. FAN9611 — Interleaved Dual BCM PFC Controllers mean that the converter stops operating. To overcome these situations, a restart timer is employed to kick start the controller and provide the first turn-on command, as shown in Figure 12. Figure 13. Conduction Interval Termination In addition to the PWM comparator, the current limit circuit and a timer circuit limiting the maximum on-time of the boost transistor can also terminate the gate drive pulse of the controller. These functions provide protection for the power switch against excessive current stress. 7. Protecting the Power Components In general, power converters are designed with adequate margin for reliable operation under all operating conditions. However, it might be difficult to predict dangerous conditions under transient or certain fault situations. Therefore, the FAN9611 contains dedicated protection circuits to monitor the individual peak currents in the boost inductors and in the power transistors. Furthermore, the boost output voltage is sensed by two independent mechanisms to provide over-voltage protection for the power transistors, rectifier diodes, and the output energy storage capacitor of the converter. Figure 11. Simple Zero-Current Detection Method The auxiliary winding can also be used to generate bias for the PFC controller when an independent bias power supply is not present in the system. At startup condition and in the unlikely case of missing zero current detection, the lack of an oscillator would © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 10 The architecture and operating principle of FAN9611 also provides inherent input power limiting capability. Figure 15. Automatic Phase-Control Operation 10. Brownout Protection with Soft Recovery An additional protection function usually offered by PFC ICs is input brownout protection to prevent the converter from operating below a user-defined minimum input voltage level. For this function to work, the input voltage of the converter is monitored. When the voltage falls below the brownout protection threshold, the converter stops working. The output voltage of the boost converter falls until the load stops drawing current from the output capacitor or until the input voltage gets back to its nominal range and operation resumes. Figure 14. On-Time vs. VIN_RMS When the slope of the PWM ramp is made proportional to the square of the input RMS voltage, the maximum on-time of the boost power switch becomes inversely proportional to the square of VIN,RMS, as represented in Figure 14. In boundary-conduction mode, the peak current of the boost transistor is proportional to its on time. Therefore, controlling the maximum pulse width of the gate drive signal according to the curve shown is an effective method to implement an input-voltage independent power limit for the boost PFC. As the output falls, the voltage at the feedback pin falls proportionally, according to the feedback divider ratio. To facilitate soft recovery after a brownout condition, the soft-start capacitor – which is also the reference voltage of the error amplifier – is pulled lower by the feedback network. This effectively pre-conditions the error amplifier to provide closed-loop, soft-start-like behavior during the converter’s recovery from a brownout situation. Once the input voltage goes above the brownout protection threshold, the converter resumes normal operation. The output voltage rises back to the nominal regulation level following the slowly rising voltage across the soft-start capacitor. 9. Light-Load Operation (Phase Management) One of the parameters determining the operating frequency of a boundary conduction mode converter is the output power. As the load decreases, lower peak currents are commanded by the pulse width modulator to maintain the output voltage at the desired set point. Lower peak current means shorter on-time for the power transistor and shorter time interval to ramp the inductor current back to zero at any given input voltage. As a result, the operating frequency of the converter increases under light load condition. 11. Soft Starting the Converter As the operating frequency and corresponding switching losses increase, conduction losses diminish at the same time. Therefore, the power losses of the converter are dominated by switching losses at light load. This phenomenon is especially evident in a BCM converter. During startup, the boost converter peak charges its output capacitor to the peak value of the input voltage waveform. The final voltage level, where the output is regulated during normal operation, is reached after the converter starts switching. There are two fundamentally different approaches used in PWM controllers to control the startup characteristics of a switched-mode power supply. Both methods use some kind of soft-start mechanism to reduce the potential overshoot of the converter’s output after the desired output voltage level is reached. To improve light-load efficiency, FAN9611 disables one of the two interleaved boost converters automatically when the output power falls below approximately 13% of the maximum power limit level. By managing the number of phases used at light load, the FAN9611 can maintain high efficiency for a wider load range of the power supply. The first method is called open-loop soft-start and relies on gradually increasing the current or power limit of the converter during startup. In this case, the voltage error amplifier is typically saturated, commanding maximum current until the output voltage reaches its final value. At that time, the voltage between the error amplifier inputs changes polarity and the amplifier slowly comes out of saturation. While the error amplifier recovers and before it starts controlling the output voltage, the converter operates with full power. Thus, output voltage overshoot is unavoidable in converters utilizing the open-loop softstart scheme. Normal interleaved operation of the two boost converters resumes automatically once the output power exceeds approximately 18% of the maximum power limit level of the converter. By adjusting maximum on-time (using RMOT), the phase management thresholds can be adjusted upward, described in the “Adjusting the Phase-Management Thresholds” section of this datasheet. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 FAN9611 — Interleaved Dual BCM PFC Controllers 8. Power Limit www.fairchildsemi.com 11 FAN9611 employs closed-loop soft-start where the reference voltage of the error amplifier is slowly increased to its final value. When the current and power limits of the converter are properly taken into consideration, the output voltage of the converter follows the reference voltage. This ensures that the error amplifier stays in regulation during soft start and the output voltage overshoot can be eliminated. Functional Description be used as an indication that the chip is running. Potentially, this behavior can be utilized to control the inrush current limiting circuit. 1. Detecting Zero Inductor Current (ZCD1, ZCD2) Each ZCD pin is internally clamped close to 0 V (GND). Any capacitance on the pin is ineffective in providing any delay in ZCD triggering. The internal sense circuit is a true differentiator to catch the valley of the drain waveforms. The resistor between the auxiliary winding of the boost inductor and the ZCD pin is only used for current limiting. The maximum source current during zero current detection must be limited to 0.5 mA. If the sourcing current is larger than 0.5 mA, the internal detection circuit is saturated and the ZCD circuit can be prematurely triggered before reaching the actual ZCD valley threshold. Source and sink capability of the pin are about 1 mA and 10 mA, respectively. The stronger sinking current capability provides sufficient margin for the higher sinking current required during the conduction time of the rectifier diode. External Components Internal Circuits Figure 17. 5V Bias To arm ZCD detector PWM signal 3. Maximum On-Time Control (MOT) Maximum on-time, MOT, (of the boost MOSFET) is set by a resistor to analog ground (AGND). The FAN9611 implements input-voltage feedforward. The maximum on-time is a function of the RMS input voltage. The voltage on the MOT pin is 1.25 V during operation (constant DC voltage). The maximum on-time of the power MOSFETs can be approximated by: . 0.3/0.0V RZCD ARM (ready) 1(2) ZCD INHIBIT ZCD Valley Detector To ZCD winding ZCD signal (true differentiator) tON ,MAX = RMOT ⋅120 ⋅10−12 ⋅ Positive Negative Clamp Clamp Figure 16. Zero-Current Detect Circuit V N AUX 1 ⋅ O ⋅ 0 .5 mA 2 N BOOST 2.4 1 ⋅ 2 1.25 VINSNS , PK (2) where VINSNS,PK is the peak of the AC input voltage as measured at the VIN pin (must be divided down, see the VIN pin description). The RZCD resistor value can be approximated by: R ZCD = FAN9611 — Interleaved Dual BCM PFC Controllers This method is especially dangerous in power-factorcorrector applications because the error amplifier’s bandwidth is typically limited to a very low crossover frequency. The slow response of the amplifier can cause considerable overshoot at the output. (1) 2. 5 V Bias Rail (5VB) This is the bypass capacitor pin for the internal 5 V bias rail powering the control circuitry. The recommended capacitor value is 220 nF. At least a 100 nF, goodquality, high-frequency, ceramic capacitor should be placed in close proximity to the pin. The 5 V rail is a switched rail. It is actively held LOW when the FAN9611 is in under-voltage lockout. Once the UVLO turn-on threshold is exceeded at the VDD pin, the 5 V rail is turned on, providing a sharp edge that can © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 Figure 18. Maximum On-Time Control (MOT) www.fairchildsemi.com 12 Analog ground connection (AGND) is the GND for all control logic biased from the 5 V rail. Internally, the AGND and PGND pins are tied together by two antiparallel diodes to limit ground bounce difference due to bond wire inductances during the switching actions of the high-current gate drive circuits. It is recommended to connect AGND and PGND pins together with a short, low-impedance trace on the PCB (right under the IC). Furthermore, during brownout condition, the output voltage of the converter might fall, which is reflected at the FB pin. When FB voltage goes 0.5 V below the voltage on the SS pin, it starts discharging the soft-start capacitor. The soft-start capacitor remains 0.5 V above the FB voltage. When the brownout condition is over, the converter returns to normal operation gracefully, following the slow ramp up of the soft-start capacitor at the non-inverting input of the error amplifier. PGND is the reference potential (0 V) for the highcurrent gate-drive circuit. Two bypass capacitors should be connected between the VDD pin and the PGND pin. One is the VDD energy storage capacitor, which provides bias power during startup until the bootstrap power supply comes up. The other capacitor shall be a goodquality ceramic bypass capacitor, as close as possible to PGND and VDD pins to filter the high peak currents of the gate driver circuits. The value of the ceramic bypass capacitor is a strong function of the gate charge requirement of the power MOSFETs and its recommended value is between 1 µF and 4.7 µF to ensure proper operation. 5. Soft-Start (SS) Soft-start is programmed with a capacitor between the SS pin and AGND. This is the non-inverting input of the transconductance (gM) error amplifier. At startup, the soft-start capacitor is quickly pre-charged to a voltage approximately 0.5 V below the voltage on the feedback pin (FB) to minimize startup delay. Then a 5 µA current source takes over and charges the softstart capacitor slowly, ramping up the voltage reference of the error amplifier. By ramping up the reference slowly, the voltage regulation loop can stay closed, actively controlling the output voltage during startup. While the SS capacitor is charging, the output of the error amplifier is monitored. In case the error voltage (COMP) ever exceeds 3.5 V, indicating that the voltage loop is close to saturation, the 5 µA soft-start current is reduced. Therefore, the soft start is automatically extended to reduce the current needed to charge the output capacitor, reducing the output power during startup. This mechanism is integrated to prevent the voltage loop from saturation. The charge current of the soft-start capacitor can be reduced from the initial 5 µA to as low as 0.5 µA minimum. Figure 19. Soft-Start Programming 6. Error Amplifier Compensation (COMP) COMP pin is the output of the error amplifier. The voltage loop is compensated by a combination of RS and CS to AGND at this pin. The control range of the error amplifier is between 0.195 V and 4.3 V. When the COMP voltage is below about 0.195 V, the PWM circuit skips pulses. Above 4.3 V, the maximum on-time limit terminates the conduction of the boost switches. Due to the input-voltage feedforward, the output of the error amplifier is proportional to the input power of the converter, independent of the input voltage. In addition, also due to the input-voltage feedforward, the maximum power capability of the converter and the loop gain is independent of the input voltage. The controller’s phasemanagement circuit monitors the error amplifier output and switches to single-phase operation when the COMP voltage falls below 0.73 V and returns to two-phase operation when the error voltage exceeds 0.93 V. These thresholds correspond to about 13% and 18% of the maximum power capability of the design. In addition to modulating the soft-start current into the SS capacitor, the SS pin is clamped 0.2 V above the FB pin. This is useful in preventing the SS capacitor from running away from the FB pin and defeating the closedloop soft-start. During the zero crossing of the input source waveform, the input power is almost zero and © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 FAN9611 — Interleaved Dual BCM PFC Controllers the output voltage can not be raised. Therefore the FB voltage stays flat or even decays while the SS voltage keeps rising. This is a problem if closed-loop soft-start should be maintained. By clamping the SS voltage to the FB pin, this problem can be mitigated. 4. Analog Ground (AGND) and Power Ground (PGND) www.fairchildsemi.com 13 7. Output Voltage Feedback (FB) 8. Secondary Output Voltage Sense (OVP) The feedback pin receives the divided-down output voltage of the converter. In regulation, the FB pin should be 3 V, which is the reference used at the non-inverting input of the error amplifier. Due to the gM type error amplifier, the FB pin is always proportional to the output voltage and can be used for over-voltage protection as well. A non-latching over-voltage detection circuit monitors the FB pin and prevents the boost MOSFETs from turning on when the FB voltage exceeds 3.25 V. Operation resumes automatically when the FB voltage returns to its nominal 3 V level. A second-level latching over-voltage protection can be implemented using the OVP pin of the controller. The threshold of this circuit is set to 3.5 V. There are two ways to program the secondary OVP. Option 1, as shown in Figure 22, is to connect the OVP pin to the FB pin. In addition to the standard nonlatching OVP (set at ~8%), this configuration provides a second OVP protection (set at ~15%), which is latched. FAN9611 — Interleaved Dual BCM PFC Controllers Figure 20. Error Amplifier Compensation Circuitry In the case where redundant over-voltage protection is preferred (also called double-OVP protection), a second separate divider from the output voltage can be used, as shown by Option 2 in Figure 22. In this case, the latching OVP protection level can be independently established below or above the non-latching OVP threshold, which is based on the feedback voltage (at the FB pin). The open feedback detection circuit is also connected to the FB pin. Since the output of the boost converter is charged to the peak of the input AC voltage when power is applied to the power supply, the detection circuit monitors the presence of this voltage. If the FB pin is below 0.5 V, which would indicate a missing feedback divider (or wrong value causing dangerously-high regulation voltage), the FAN9611 does not send out gate drive signals to the boost transistors. If latching OVP protection is not desired at all, the OVP pin should be grounded (Option 3). Figure 22. Secondary Over-Voltage Protection Circuit Figure 21. Output-Voltage Feedback Circuit © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 14 The input AC voltage is sensed at the VIN pin. The input voltage is used in two functions: input under-voltage lockout (brownout protection), and input voltage feedforward in the PWM control circuit. All the functions require the RMS value of the input voltage waveform. Since the RMS value of the AC input voltage is directly proportional to its peak, it is sufficient to find the peak instead of the more complicated and slower method of integrating the input voltage over a half line cycle. The internal circuit of the VIN pin works with peak detection of the input AC waveforms. One of the important benefits of this approach is that the peak indicates the correct RMS value even at no load when the HF filter capacitor at the input side of the boost converter is not discharged around the zero crossing of the line waveform. Another notable benefit is that during line transients, when the peak exceeds the previously measured value, the input-voltage feedforward circuit can react immediately, without waiting for a valid integral value at the end of the half line period. Furthermore, lack of zero crossing detection could fool The valid range for the peak of the AC input is between approximately 0.925 V and 3.7 V. This range is optimized for universal input voltage range of operation. If the peak of the sense voltage remains below the 0.925 V threshold, input under-voltage or brownout condition is declared and the FAN9611 stops operating. When the VIN voltage exceeds 3.7 V, the FAN9611 input voltage sense circuit saturates and the feedforward circuit is not able to follow the input any higher. Consequently, the slope of the PWM ramp remains constant corresponding to the VIN = 3.7 V level amplitude for any VIN voltage above 3.7 V. The input voltage is measured by a tracking analog-todigital converter, which keeps the highest value (peak voltage) of the input voltage waveform. Once a measurement is taken, the converter tracks the input for at least 12 ms before a new value is taken. This delay ensures at least one new peak value is captured before the new value is used. FAN9611 — Interleaved Dual BCM PFC Controllers the integrator while the peak detector works properly during light-load operation. 9. Input Voltage Sensing (VIN) Figure 23. Input Voltage Sensing Circuit The measured peak value is then used in the following half-line cycle while a new measurement is executed to be used in the next half line cycle. This operation is synchronized to the zero crossing of the line waveform. Since the input voltage measurement is held steady during the line half periods, this technique does not feed any AC ripple into the control loop. If line zero crossing detection is missing, the FAN9611 measures the input voltage in every 32 ms; it can operate from a DC input © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 as well. The following figures provide detail about the input voltage sensing method of the controller. As shown in the waveforms, input voltage feedforward is instantaneous when the line voltage increases and has a half line cycle delay when the input voltage decreases. Any increase in input voltage would cause output over voltage due to the slow nature of the voltage regulation loop. This is successfully mitigated by the immediate action of the input-voltage feedforward circuit. www.fairchildsemi.com 15 The purpose of the MillerDrive™ architecture is to speed switching by providing high current during the Miller plateau region when the gate-drain capacitance of the MOSFET is being charged or discharged as part of the turn-on / turn-off process. 10. Gate Drive Outputs (DRV1; DRV2) High-current driver outputs DRV1 and DRV2 have the capability to sink a minimum of 2 A and source 1 A. Due to the low impedance of these drivers, the 1 A source current must be actively limited by an external gate resistor. The minimum external gate resistance is: FAN9611 — Interleaved Dual BCM PFC Controllers Figure 24. Input Voltage Sensing Waveforms The output pin slew rate is determined by VDD voltage and the load on the output. It is not user adjustable, but if a slower rise or fall time at the MOSFET gate is needed, a series resistor can be added. V DD (3) 1A To take advantage of the higher sink current capability of the drivers, the gate resistor can be bypassed by a small diode to facilitate faster turn-off of the power MOSFETs. Traditional fast turn-off circuit using a PNP transistor instead of a simple bypass diode can be considered as well. R GATE = It is also imperative that the inductance of the gate drive loop is minimized to avoid excessive ringing. If optimum layout is not possible or the controller is placed on a daughter card, it is recommended to use an external driver circuit located near the gate and source terminals of the boost MOSFET transistors. Small gate charge power MOSFETs can be driven by a single 1 A gate driver, such as the FAN3111C; while higher gate charge devices might require higher gate drive current capable devices, such as the single-2 A FAN3100C or the dual2 A FAN3227C family of drivers. 11. MillerDrive™ Gate Drive Technology Figure 25. Current-Sense Protection Circuits FAN9611 output stage incorporates the MillerDrive™ architecture shown in Figure 25. It is a combination of bipolar and MOS devices which are capable of providing large currents over a wide range of supply voltage and temperature variations. The bipolar devices carry the bulk of the current as OUT swings between 1/3 to 2/3 VDD and the MOS devices pull the output to the high or low rail. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 16 This is the main bias source for the FAN9611. The operating voltage range is between 8 V and 18 V. The VDD voltage is monitored by the under-voltage lockout (UVLO) circuit. At power-up, the VDD voltage must exceed 10.0 V (±0.5 V to enable operation. The FAN9611 stops operating when the VDD voltage falls below 7.5 V (±0.5 V). See PGND pin description for important bypass information. 13. Current-Sense Protection (CS1, CS2) The FAN9611 uses independent over-current protection for each of the power MOSFETs. The current-sense thresholds at the CS1 and CS2 pins are approximately 0.2 V. The current measurements are strictly for protection purposes and are not part of the control algorithm. The pins can be directly connected to the non-grounded end of the current-sense resistors because the usual R-C filters of the leading-edge current spike are integrated in the IC. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 Figure 26. Current-Sense Protection Circuits The time constant of the internal filter is approximately: τ = 27kΩ ⋅ 5 pF = 130ns or ωP = 1 = 1.2MHz 2 ⋅ π ⋅ 27 kΩ ⋅ 5 pF (4) FAN9611 — Interleaved Dual BCM PFC Controllers 12. Bias Supply (VDD) www.fairchildsemi.com 17 to the voltage on the COMP pin, minus a small offset. The voltage on the COMP pin is monitored to determine the operating power of the supply. Therefore the voltage on the SS pin can be adjusted lower to achieve the desired lower output voltage. 1. Synchronization and Timing Functions The FAN9611 employs a sophisticated synchronization sub-system. At the heart of the system is a dual-channel switching-frequency detector that measures the switching period of each channel in every switching cycle and locks their operating phase 180 degrees out of phase from each other. The slower operating frequency channel is dominant, but there is no masterslave arrangement. Moreover, as the frequency constantly changes due to the varying input voltage, either channel can be the slower dominant channel. Several possible implementations to adjust the output voltage of the boost stage at light load are described in the application note AN-8021. It includes the universal output voltage adjust implementation which is modulated by input voltage to avoid the boost converter becoming a peak rectifier at high line and light load. As opposed to the most common technique, where the phase relationship between the channels is provided by changing the on-time of one of the MOSFETs, the FAN9611 controls the phase relationship by inserting a turn-on delay before the next switching period starts for the faster running phase. As shown in the literature[1], the on-time modulation technique is not stable under all operating conditions, while the off-time modulation (or delaying the turn-on) is unconditionally stable under all operating conditions. a. Restart Timer and Dead-Phase Detect Protection Figure 27. FAN9611 Error Amplifier Configuration The restart timer is an integral part of the Sync-Lock™ synchronizing circuit. It ensures exact 180-degree outof-phase operation in restart timer operation. This is an important safety feature. In the case of a non-operating phase due to no ZCD detection, missing gate drive connection (for example no gate resistor), one of the power components failing in an open circuit, or similar errors, the other phase is locked into restart timer operation, preventing it from trying to deliver full power to the load. This is called the dead-phase detect protection. 3. Adjusting the Output Voltage with Input Voltage In some applications, the output voltage of the PFC boost converter is adjusted based on the input voltage only. This boost follower implementations increases the efficiency of the downstream DC-DC converter and therefore of the overall power supply. Implementations for both the two-level boost and the linear boost follower (or tracking boost) are described in application note AN-8021. The restart timer is set to approximately 16.5 kHz, just above the audible frequency range, to avoid any acoustic noise generation. b. 4. Adjusting the Phase-Management Thresholds In any power converter, the switching losses become dominant at light load. For an interleaved converter where there are two or more phases, light-load efficiency can be improved by shutting down one of the phases at light load (also known as phase-shedding or phase-dropping operating). Frequency Clamp Just as the restart timer, the frequency clamp is integrated into the synchronization and ensures exact 180-degree out-of-phase operation when the operating frequency is limited. This might occur at very light-load operation or near the zero crossing region of the line voltage waveform. Limiting the switching frequency at light load can improve efficiency, but has a negative effect on power factor since the converter also enters true DCM operation. The frequency clamp is set to approximately 525 kHz. The initial phase-management thresholds are fixed at approximately 13% and 18% of the maximum load power level. This means when the output power reaches 13%, the FAN9611 automatically goes from a two-phase to a single-phase operation (phase shed or phase drop). When the output power comes back up to 18%, the FAN9611 automatically goes from the singlephase to the two-phase operation (phase-add). 2. Adjusting the Output Voltage with Load Some applications, the output voltage of the PFC boost converter is decreased at low power levels to boost the light load efficiency of the power supply. The default thresholds can be adjusted upward based on the application requirement; for example, to meet the Energy STAR 5.0 or the Climate Savers Computing efficiency requirements at 20% of the load. The phase drop threshold can be adjusted upward (for example to 25%) by adjusting the maximum on time. Implementing this function with a circuit external to the FAN9611 is straightforward because the error amplifier reference (the positive input) is available on the softstart (SS) pin, as shown in Figure 27. In the FAN9611 architecture, the power of the converter is proportional © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 FAN9611 — Interleaved Dual BCM PFC Controllers Application Information www.fairchildsemi.com 18 b. Pull the FB Pin to GND. By pulling the FB pin below the open feedback protection threshold of approximately 0.5 V, the power supply can be shut down immediately. It is imperative that the FB is pulled below the threshold very quickly since the power supply keeps switching until this threshold is crossed. If the feedback is pulled LOW softly and does not cross the threshold, the power supply tries to deliver maximum power because the FB pin is forced below the reference voltage of the error amplifier on the SS pin. Eventually, as FB is pulled to GND, the SS capacitor is pulled LOW by the internal clamp between the FB and SS pins. The SS pin stays approximately 0.5 V higher than the FB pin itself. Therefore, recovery from a shut down state follows normal soft-start procedure when the FB pin is released as the voltage across the SS capacitor starts ramping from a low value. c. Pulling the COMP Pin to GND. When the COMP pin is pulled below the PWM ramp offset, approximately 0.195 V, the FAN9611 stops sending gate drive pulses to the power MOSFETs. This condition is similar to pulse skipping under no-load condition. If any load is still present at the output of the boost PFC stage, the output voltage decreases. Consequently, the FB pin decreases and the SS capacitor voltage is pulled LOW by the internal clamp between the FB and SS pins. At that point, the operation and eventual recovery to normal operation is similar to the mechanism described above. If the COMP pin is held LOW for long enough to pull the SS pin LOW, the recovery follows normal soft-start procedure when the COMP pin is released. If the SS capacitor is not pulled LOW as a result of a momentary pull-down of the COMP pin, the recovery is still soft due to the fact that a limited current source is charging the compensation capacitors at the output of the error amplifier. Nevertheless, in this case, output voltage overshoot can occur before the voltage loop enters closed-loop operation and resumes controlling the output voltage again. d. Pull the VIN Pin to GND. Since the VIN sense circuit is configured to ride through a single line cycle dropout test without shutting down the power supply, this method results in a delayed shutdown of the converter. The FAN9611 stops operation approximately 20 ms to 32 ms after the VIN pin is pulled LOW. The delay depends on the phase of the line cycle at which the pull-down occurs. This method triggers the input brownout protection (input Figure 28. Adjusting Phase Management Thresholds Since the phase management threshold is fixed at 13% and 18% of the maximum power limit level, the actual power management threshold as a percentage of nominal output power can be adjusted by the ratio between nominal power and maximum power limit level as shown in Figure 28. The second plot shows an example where the maximum power limit level is 1.4 times of nominal output power. By adjusting the maximum on-time (using RMOT), the phase management thresholds can be adjusted upward. Phase management is implemented such that the output of the error amplifier (VCOMP) does not have to change when the system toggles between single-phase and two-phase operations, as shown in Figure 29. The output of the error amplifier is proportional to the output power of the converter independently, whether one or both phases are operating in the power supply. Furthermore, because the maximum on-time limit is applied independently to each pulse-width modulator, the power handling capability of the converter with only one phase running is approximately half of the total output power that can be delivered when both phases are utilized. Additional details on adjusting phase management are provided in the application note AN-6086. Figure 29. VCOMP vs. tONMAX 5. Disabling the FAN9611 There are four ways to disable the FAN9611. It is important to understand how the part reacts for the various shutdown procedures. a. Pull the SS Pin to GND. This method uses the error amplifier to stop the operation of the power supply. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 FAN9611 — Interleaved Dual BCM PFC Controllers By pulling the SS pin to GND, the error amplifier’s non-inverting input is pulled to GND. The amplifier senses that the inverting input (FB pin) is higher than the reference voltage and tries to adjust its output (COMP pin) to make the FB pin equal to the reference at the SS pin. Due to the slow speed of the voltage loop in PFC applications, this might take several line cycles. Thus, it is important to consider that by pulling the SS pin to GND, the power supply is not shut down immediately. Recovery from a shut down follows normal soft-start procedure when the SS pin is released. www.fairchildsemi.com 19 Power Ground and Analog Ground Power ground (PGND) and analog ground (AGND) should meet at one point only. The return path for the gate drive current and VDD capacitor should be connected to the PGND pin. Minimize the ground loops between the driver outputs (DRV1, DRV2), MOSFETs, and PGND. Adding the by-pass capacitor for noise on the VDD pin is recommended. It should be connected as close to the pin as possible. Gate Drive The gate drive pattern should be wide enough to handle 1 A peak current. For high-power applications, two or more PCB layers are recommended to effectively use the ground pattern to minimize the switching noise interference. The FAN9611 incorporates fast-reacting input circuits, short propagation delays, and strong output stages capable of delivering current peaks over 1.5 A to facilitate fast voltage transition times. Many high-speed power circuits can be susceptible to noise injected from their own output or external sources, possibly causing output re-triggering. These effects can be especially obvious if the circuit is tested in breadboard or nonoptimal circuit layouts with long input or output leads. The following guidelines are recommended for all layout designs, but especially strongly for the single-layer PCB designs. (For example of a 1-layer PCB design, see the Application Note AN-6086.) Keep the controller as close to the MOSFETs as possible. This minimizes the length and the loop area (series inductance) of the high-current gate drive traces. The gate drive pattern should be as short as possible to minimize interference. Current Sensing Current sensing should be as short as possible. To minimize switching noise, current sensing should not make a loop. Input Voltage Sensing (VIN) Since the impedance of voltage divider is large and FAN9611 detects the peak of the line voltage, the VIN pin can be sensitive to the switching noise. The trace connected to this pin should not cross traces with high di/dt to minimize the interference. General Keep high-current output and power ground paths separate from analog input signals and signal ground paths. For best results, make connections to all pins as short and direct as possible. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 All the control components should be connected to AGND without sharing the trace with PGND. 6. Layout and Connection Guidelines FAN9611 — Interleaved Dual BCM PFC Controllers under-voltage lockout), which gradually discharges the compensation capacitor. As the output voltage decreases, the FB pin falls, pulling LOW the SS capacitor voltage. Similarly to the shutdown, once the VIN pin is released, operation resumes after several milliseconds of delay needed to determine that the input voltage is above the turn-on threshold. At least one line cycle peak must be detected above the turn-on threshold before operation can resume at the following line voltage zero-crossing. The converter starts following normal soft-start procedure. The noise bypass capacitor for VIN should be connected as close to the pin as possible. www.fairchildsemi.com 20 The FAN9611 can be configured following the next steps outlined in this section. This Quick Setup Guide refers to the schematic diagram and component references of Figure 32. It uses the equations derived and explained in Application Note AN-6086. Input Voltage Sense Divider In preparation to calculate the setup component values, the power supply specification must be known. Furthermore, a few power stage components must be pre-calculated before the controller design begins as their values determine the component selections. An Excel design tool is also available to ease calculations. Bypass Capacitor for VDD_HF Description Name VLINE.ON Minimum AC RMS Input (Turn-Off) VLINE.OFF Minimum Line Frequency fLINE,MIN Nominal DC Output VOUT,RIPPLE Latching Output OVP VOUT,LATCH Nominal Output Power (to Load) POUT Desired Hold-Up Time tHOLD Minimum DC Output (End of tHOLD) Gate Drive Resistor RG1, RG2 Gate Drive Speed-Up Diode DG1, DG2 Current Sense Resistor CVDD1 2.2μ CVDD2 47μ RCS1, RCS2 Step 1: Input Voltage Range Value FAN9611 utilizes a single pin (VIN) for input voltage sensing. The VIN pin must be above 0.925 V (VIN_BO) to enable operation. The converter turns on at a higher VIN voltage (VIN_ON) set independently by the designer. The input voltage information is used for feedforward in the control algorithm as well. The input-voltage feedforward operates over a four-to-one range from 0.925 V (VIN_BO) to 3.7 V (VFF_UL), as measured at the VIN pin. VOUT Output Voltage Ripple (2 · fLINE) RINHYST Startup Energy Storage for VDD From Power Supply Specification: Minimum AC RMS Input (Turn-On) RIN2 Brownout Hysteresis Set FAN9611 — Interleaved Dual BCM PFC Controllers Quick Setup Guide VOUT,MIN Minimum Switching Frequency fSW,MIN Maximum DC Bias VDDMAX Pre-Calculated Power Stage Parameters: Estimated Conversion Efficiency Maximum Output Power per Channel Output Capacitance Boost Inductance per Channel Maximum On-Time per Channel Turns Ratio (NBOOST / NAUX) η 0.95 PMAX,CH Figure 30. VIN Turn-on and Turn-off Thresholds COUT At VIN voltages above the 3.7 V upper limit (VFF_UL), input voltage feedforward is not possible. The input voltage-sense circuitry saturates at this point and the PWM ramp is modulated any longer. Above VFF_UL, the converter’s output power becomes a function of the input voltage as shown in Figure 31. It can also be expressed analytically as: L tON,MAX N 10 Other Variables Used During the Calculations: Peak Inductor Current Maximum DC Output Current (to Load) IL,PK VIN POUT_NO_FF = PMAX • 3.7 IO,MAX Calculated Component Values: Zero Current Detect Resistor RZCD1, RZCD2 Bypass Capacitor for 5V Bias C5VB Maximum On-Time Set RMOT Soft-Start Capacitor CSS Compensation Capacitor Compensation Resistor Compensation Capacitor Feedback Divider (5) where VIN is the voltage at the VIN pin and PMAX is the desired maximum output power of the converter which will be maintained constant while the input voltage feedforward circuit is operational. 0.15μ CCOMP,LF RCOMP CCOMP,HF RFB1 Feedback Divider RFB2 Over Voltage Sense Divider ROV1 Over Voltage Sense Divider ROV2 Input Voltage Sense Divider RIN1 © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 2 Figure 31. VIN Feedforward Range As can be seen, the converter’s output power capability will follow a square function above VFF_UL. www.fairchildsemi.com 21 FAN9611 — Interleaved Dual BCM PFC Controllers Figure 32. Interleaved BCM PFC Schematic Using FAN9611 Step 2: Estimated Conversion Efficiency Step 3: Maximum Output Power per Channel Use the estimated full-load power conversion efficiency. Typical value for an interleaved BCP PFC converter is in the 0.92 to 0.98 range. The efficiency is in the lower half of the range for low-power applications. Using state-ofthe-art semiconductors, good quality ferrite inductors and selecting lower limit for minimum switching frequency positively impacts the efficiency of the system. In general, the value of 0.95 can be used unless a more accurate power budget is available. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 PMAX, CH = 1.2 ⋅ POUT 2 (6) A margin of 20% has been added to the nominal output power to cover reference inaccuracy, internal component tolerances, inductance mismatch, and current-sense resistor variation to the per-channel power rating. www.fairchildsemi.com 22 COUT(RIPPLE) COUT(HOLD) = Step 9: Zero Current Detect Resistors POUT = 4 ⋅ fLINE,MIN ⋅ VOUT ⋅ VOUT,RIPPLE R ZCD1 = R ZCD2 = (7) 2 ⋅ POUT ⋅ t HOLD 2 V 2 VOUT − OUT,RIPPLE − VOUT,MIN 2 (8) Step 10: Maximum On-Time Setting Resistor RMOT = 4340 ⋅ 10 6 ⋅ t ON,MAX Step 11: Output Voltage Setting Resistors (Feedback) R FB2 = L LINE,MAX = ( ) (9) ( ) (10) 2 η ⋅ VLINE,MAX ⋅ VOUT − 2 ⋅ VLINE,MAX 2 ⋅ fSW,MIN ⋅ VOUT ⋅ PMAX,CH The minimum switching frequency can occur either at the lowest or at the highest input line voltage. Accordingly, two boost inductor values are calculated and the lower of the two inductances must be selected. This L value keeps the minimum operating frequency above fSW,MIN under all operating conditions. t ON,MAX = 2 η ⋅ VLINE, OFF RFB2 = 2 ⋅ VLINE,OFF L ⋅ t ON,MAX (11) 2 ⋅ PMAX, CH VOUT © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 0.12mA ⋅ VOUT V RFB1 = OUT − 1 ⋅ RFB2 3V (12) Step 8: Maximum DC Output Current IOUT,MAX = [ ] 3V ⋅ 2 ⋅ VLINE,ON − (12.5V + 3 ⋅ 0.7V ) (17) where 3 V is the reference voltage of the error amplifier at its non-inverting input; 12.5 V is the controller’s UVLO turn-on threshold; 0.12 mA is the worst-case startup current required to start operation; and 3·0.7 V accounts for the forward voltage drop of three diodes in series of the startup current. Once the value of RFB2 is determined, RFB1 is given by the following formula: Step 7: Peak Inductor Current per Channel IL,PK = (16) If the feedback divider is used to provide startup power for the FAN9611 (see AN-6086 for implementation details), the following equation is used to calculate RFB2: Step 6: Maximum On-Time per Channel 2 ⋅ L ⋅ PMAX,CH 3V ⋅ VOUT 3V = PFB IFB where 3 V is the reference voltage of the error amplifier at its non-inverting input and PFB or IFB are selected by the designer. If the power loss associated to the feedback divider is critical to meet stand-by power consumption regulations, it might be beneficial to start the calculation by choosing PFB. Otherwise, the current of feedback divider, IFB should be set to approximately 0.4 mA at the desired output voltage set point. This value ensures that parasitic circuit board and pin capacitances do not introduce unwanted filtering effect in the feedback path. Step 5: Boost Inductance per Channel 2 ⋅ fSW, MIN ⋅ VOUT ⋅ PMAX,CH (15) where RMOT should be between 40 kΩ and 130 kΩ. The second expression yields the minimum output capacitance based on the required hold-up time based on the power supply specification. Ultimately, the larger of the two values satisfies both design requirements and has to be selected for COUT. 2 η ⋅ VLINE, OFF ⋅ VOUT − 2 ⋅ VLINE, OFF (14) where 0.5·VOUT is the maximum amplitude of the resonant waveform across the boost inductor during zero current detection; N is the turns ratio of the boost inductor and the auxiliary winding utilized for the zero current detection; and 0.5 mA is the maximum current of the ZCD pin during the zero current detection period. The output capacitance must be calculated by two different methods. The first equation determines the capacitor value based on the allowable ripple voltage at the minimum line frequency. It is important to remember that the scaled version of this ripple is present at the FB pin. The feedback voltage is continuously monitored by the non-latching over voltage protection circuit. Its threshold is about 8% higher the nominal output voltage. To avoid triggering the OVP protection during normal operation, VOUT,RIPPLE should be limited to less 12% of the nominal output voltage, VOUT. L LINE,OFF = 0.5 ⋅ VOUT N ⋅ 0.5mA FAN9611 — Interleaved Dual BCM PFC Controllers Step 4: Output Capacitance (18) RFB1 can be implemented as a series combination of two or three resistors; depending on safety regulations, maximum voltage, and or power rating of the selected resistor type. (13) www.fairchildsemi.com 23 C SS = 5 μA ⋅ C OUT ⋅ (R FB1 + R FB2 ) 0.3 ⋅ IOUT, MAX ⋅ R FB2 Step 14: Over-Voltage Protection Setting (OVP) (19) where 5 μA is the charge current of the soft-start capacitor and 0.3·IOUT,MAX is the maximum output current charging the output capacitor of the converter during the soft-start process. It is imperative to limit the charge current of the output capacitor to be able to maintain closed-loop soft-start of the converter. The 0.3 factor used in the CSS equation can prevent output over voltage at the end of the soft-start period and provides sufficient margin to supply current to the load while the output capacitor is charging. R OV2 = gM ⋅ IOUT,MAX 4.1V ⋅ COUT ⋅ (2 ⋅ π ⋅ f0 ) 2 ⋅ RFB2 RFB1 + RFB2 VOUT, LATCH R OV1 = − 1 ⋅ R OV2 3.5V Step 15: Input Line Voltage Sense Resistors R IN2 = To guarantee closed-loop soft-start operation under all conditions, it is recommended that: 1 2 ⋅ π ⋅ f0 ⋅ CCOMP,LF CCOMP,HF = 1 2 ⋅ π ⋅ fHFP ⋅ RCOMP (21) 2 LINE,MAX (26) 2 • VLINE,MIN • PINSNS 2 ⋅ VLINE,MIN RIN1 = − 1 ⋅ RIN2 0.925V (27) RIN1 can be implemented as a series combination of two or three resistors; depending on safety regulations, maximum voltage, and or power rating of the selected resistor type. (22) (23) RINHYST 2 ⋅ VLINE,ON ⋅ RIN2 − 0.925V RIN1 + RIN2 = 2μA (28) where 0.925 V is the threshold voltage of the line under-voltage lockout comparator and 2 μA is the sink current provided at the VIN pin during line undervoltage (brownout) condition. The sink current, together with the terminating impedance of the VIN pin determines the hysteresis between the turn-on and turn-off thresholds. where fHFP is the frequency of a pole implemented in the error amplifier compensation network against highfrequency noise in the feedback loop. The pole should be placed at least a decade higher than f0 to ensure that it does not interfere with the phase margin of the voltage regulation loop at its crossover frequency. It should also be sufficiently lower then the switching frequency of the converter so noise can be effectively attenuated. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 0.925 V • V where 0.925 V is the brownout protection threshold at the VIN pin. VLINE,MIN is the minimum input RMS operating voltage. Its divided down level at the VIN pin corresponds to the 0.925 V brownout protection threshold. VLINE,MAX is the maximum input RMS voltage anticipated in the design and PINSNS is the total power dissipation of the RIN1 - RIN2 divider when the input voltage equals VLINE,MAX. Typical PINSNS power loss is in the 50 mW to 100 mW range. This relationship is determined by the ratio between the maximum output current of the gM error amplifier to the maximum charge current of the soft-start capacitor. Observing this correlation between the two capacitor values ensures that the compensation capacitor voltage can be adjusted faster than any voltage change taking place across the soft-start capacitor. Therefore, during startup the voltage regulation loop’s response to the increasing soft-start voltage is not limited by the finite current capability of the error amplifier. RCOMP = (25) ROV1 can be implemented as a series combination of two or three resistors; depending on safety regulations, maximum voltage, and or power rating of the selected resistor type. (20) where 4.1 V is the control range of the error amplifier and f0 is the desired voltage loop crossover frequency. It is important to consider that the lowest output ripple frequency limits the voltage loop crossover frequency. In PFC applications, that frequency is two times the AC line frequency. Therefore, the voltage loop bandwidth (f0), is typically in the 5 Hz to 15 Hz range. C COMP,HF < 4 ⋅ C SS (24) POVP where 3.5 V is the threshold voltage of the OVP comparator and POVP is the total dissipation of the resistive divider network. Typical POVP power loss is in the 50 mW to 100 mW range. Step 13: Compensation Components CCOMP,LF = 3.5V ⋅ VOUT, LATCH FAN9611 — Interleaved Dual BCM PFC Controllers The recommended fHFP frequency is around 120 Hz in PFC applications. Step 12: Soft-Start Capacitor www.fairchildsemi.com 24 It is recommended to place at least a 15 Ω resistor between each of the gate drive outputs (DRV1, DRV2) and their corresponding power devices. The gate drive resistors have a beneficial effect to limit the current drawn from the VDD bypass capacitor during the turn-on of the power MOSFETs and to attenuate any potential oscillation in the gate drive circuits. R G1 = R G2 = VDD MAX 1.0 A In addition to the high-speed turn-off, another advantage of this circuit is that the FAN9611 does not have to sink the high peak discharge current from the MOSFET, reducing the internal power dissipation in the gate drive circuitry by a factor of two. Instead, the current is discharged locally in a tighter, more controlled loop, minimizing parasitic trace inductance while protecting the FAN9611 from injected disturbances associated with ground bounce and ringing due to high-speed turn-off. (29) where 1.0 A is the recommended peak value of the gate drive current. Step 17: Current-Sense Resistors R CS1 = R CS2 = 0.18V IL,PK (30) FAN9611 — Interleaved Dual BCM PFC Controllers The FAN9611 sources high peak current to the MOSFET gate through RG and DON, where RG is used to control the turn-on transition time. When the MOSFET is commanded to turn off, QOFF conducts, shorting the gate to the source, where the turn-off speed can be controlled by the value of ROFF. Where maximum turnoff time is desired, the value of ROFF can be 0 Ω. DON serves the dual purpose of protecting the QOFF baseemitter junction and blocking the MOSFET discharge current from sinking back through the FAN9611. Step 16: Gate Resistors Figure 33. Recommended Gate Drive Schematic where 0.18 V is the worst-case threshold of the current limit comparator. The size and type of current sense resistors depends on their power dissipation and manufacturing considerations. A speed-up discharge diode that feeds switching current back into the IC is not recommended. 1 4⋅ 2 ⋅V LINE,OFF 2 ⋅ R CS1 ⋅ − PRCS1 = 1.5 ⋅ IL,PK 6 9 ⋅ π ⋅ VOUT RG DG RCS (31) where the 1.5 factor is used for the worst-case effect of the current-limit threshold variation. When the currentsense resistor is determined, the minimum currentsense threshold must be used to avoid activating overcurrent protection too early as the power supply approaches full-load condition. The worst-case power dissipation of the current sense resistor occurs when the current-sense threshold is at its maximum value defined in the datasheet. The ratio between the minimum and maximum thresholds squared (since the square of the current determines power dissipation) yields exactly the 1.5 factor used in the calculation. Figure 34. Discharge Diode is Not Recommended In cases where it is desirable to control the MOSFET turn-on and turn-off transition times independently, the circuit of Figure 35 can be used. Figure 35. Gate Drive Schematic with Independent Turn-On and Turn-Off © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 25 Typical characteristics are provided at TA = 25°C and VDD = 12 V unless otherwise noted. Figure 36. ISTARTUP vs. Temperature Figure 37. Operating Current vs. Temperature Figure 38. UVLO Thresholds vs. Temperature Figure 39. UVLO Hysteresis vs. Temperature © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 26 FAN9611 — Interleaved Dual BCM PFC Controllers Typical Performance Characteristics — Supply Typical characteristics are provided at TA = 25°C and VDD = 12 V unless otherwise noted. Figure 40. Transfer Function (Maximum On Time vs. VIN) Figure 41. Maximum On Time vs. Temperature Figure 42. EA Transconductance (gM) vs. Temperature Figure 43. EA Reference vs. Temperature Figure 44. 5 V Reference vs. Temperature Figure 45. Soft-Start Current vs. Temperature © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 FAN9611 — Interleaved Dual BCM PFC Controllers Typical Performance Characteristics — Control www.fairchildsemi.com 27 FAN9611 — Interleaved Dual BCM PFC Controllers Typical Performance Characteristics — Control Typical characteristics are provided at TA = 25°C and VDD = 12 V unless otherwise noted. Figure 46. Phase-Control Thresholds vs. Temperature Gate Drive 1 Gate Drive 1 Gate Drive 2 Gate Drive 2 Inductor Current 1 Inductor Current 1 Inductor Current 2 Inductor Current 2 Figure 47. Phase-Dropping Operation © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 Figure 48. Phase-Adding Operation www.fairchildsemi.com 28 Typical characteristics are provided at TA = 25°C and VDD = 12 V unless otherwise noted. Figure 49. CS Threshold vs. Temperature Figure 50. CS to OUT Delay vs. Temperature Figure 51. Restart Timer Frequency vs. Temperature Figure 52. Maximum Frequency Clamp vs. Temperature FAN9611 — Interleaved Dual BCM PFC Controllers Typical Performance Characteristics — Protection Figure 53. Brownout Threshold vs. Temperature © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 29 Typical characteristics are provided at TA = 25°C and VDD = 12 V unless otherwise noted. Figure 54. Non-Latching OVP vs. Temperature Figure 55. Latching OVP vs. Temperature FAN9611 — Interleaved Dual BCM PFC Controllers Typical Performance Characteristics — Protection Figure 56. OVP Hysteresis vs. Temperature © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 30 Typical characteristics are provided at TA = 25°C and VDD = 12 V unless otherwise noted. IL1 IL1 IL2 IL2 IL1 + IL2 IL1 + IL2 Figure 57. Ripple-Current Cancellation (110 VAC) Figure 58. Ripple-Current Cancellation (110 VAC) VGATE VGATE VOUT VOUT FAN9611 — Interleaved Dual BCM PFC Controllers Typical Performance Characteristics — Operation Line Current Line Current Figure 59. No-Load Startup at 115 VAC Figure 60. Full-Load Startup at 115 VAC Line Vol VOUT 110VAC 220VAC COMP Line Current Figure 61. Input Voltage Feedforward Note: 6. For full performance operational characteristics at both low line (110 VAC) and high line (220 VAC), as well as at no-load and full-load, refer to FEB388 Evaluation Board User Guide: 400 W Evaluation Board. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 31 FEB388: 400-W Evaluation Board Using FAN9611 FEB388 is one of the evaluation boards for an interleaved dual boundary-conduction-mode PFC converter. It is rated at 400 W (400 V/1 A) power. With phase management, efficiency is maintained above 96% even down at 10% of the rated output power. The efficiencies for full-load condition exceed 96% as shown below. Figure 62 and Figure 63 show the phase management with the default minimum threshold values of the IC. They can be adjusted upwards to achieve a different efficiency profile (Figure 64 and Figure 65) where phase management thresholds are adjusted to 30% / 44% of the full load. For full specification, design schematic, bill of materials and test results; see FEB388 — FAN9611/12 400-W Evaluation Board User Guide (AN-9717). Input Voltage Rated Output Power Output Voltage (Rated Current) VIN Nominal: 85 V~264 VAC VDD Supply: 13 V~18 VDC 400 W 400 V (1 A) Figure 63. Measured Efficiency at 230VAC (Default Thresholds) FAN9611 Efficiency vs. Load FAN9611 Efficiency vs. Load (115 VAC Input, 400 VDC Output, 400 W) (230 VAC Input, 400 VDC Output, 400 W) 100 100 95 95 Efficiency (%) Efficiency (%) Figure 62. Measured Efficiency at 115VAC (Default Thresholds) FAN9611 — Interleaved Dual BCM PFC Controllers Evaluation Board With Phase Management 90 Without Phase Management With Phase Management 90 Without Phase Management 85 85 80 80 10 20 30 40 50 60 70 80 90 10 100 30 40 50 60 70 80 90 100 Figure 65. Measured Efficiency at 230VAC (Adjusted Thresholds) Figure 64. Measured Efficiency at 115VAC (Adjusted Thresholds) © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 20 Output Power (%) Output Power (%) www.fairchildsemi.com 32 Related Products Part Number FAN6961 Description Green Mode PFC FAN7527B Boundary Mode PFC Control IC PFC Control Number of Pins Comments Industry Standard Pin-Out with Green Mode Functions Single BCM (CRM) 8 Single BCM (CRM) 8 Industry Standard Pin-Out FAN7528 Dual Output Critical Conduction Mode PFC Controller Single BCM (CRM) 8 Low THD for Boost-Follower Implementation FAN7529 Critical Conduction Mode PFC Controller Single BCM (CRM) 8 Low THD FAN7530 Critical Conduction Mode PFC Controller Single BCM (CRM) 8 Low THD, Alternate Pin-Out of FAN7529 (Pins 2 and 3 Reversed) FAN7930 Critical Conduction Mode PFC Controller Single BCM (CRM) 8 PFC Ready pin, Frequency Limit, AC-Line-Absent Detection, SoftStart to Minimize Overshoot, Integrated THD Optimizer, TSD FAN9611 Interleaved Dual BCM PFC Controller Dual BCM (CRM) 16 Dual BCM (CRM), 180° Out-of-Phase, 10.0V UVLO FAN9612 Interleaved Dual BCM PFC Controller Dual BCM (CRM) 16 Dual BCM (CRM), 180° Out-of-Phase, 12.5V UVLO FAN9611 — Interleaved Dual BCM PFC Controllers Table 1. Related Resources AN-6086: Design Consideration for Interleaved Boundary Conduction Mode (BCM) PFC Using FAN9611/12 AN-9717: Fairchild Evaluation Board User Guide FEB388: 400W Evaluation Board using FAN9611/12 AN-8021: Building Variable Output Voltage Boost PFC Converters Using FAN9611/12 Fairchild 300-W Low Profile Evaluation Board: FEBFAN9611_S01U300A References 1. 2. L. Huber, B. Irving, C. Adragna and M. Jovanovich, “Implementation of Open-Loop Control for Interleaved DCM/BCM Boundary Boost PFC Converters”, Proceedings of APEC ’08, pp. 1010-1016. C. Bridge and L. Balogh, “Understanding Interleaved Boundary Conduction Mode PFC Converters”, Fairchild Power Seminars, 2008-2009. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 33 10.00 9.80 A 8.89 16 9 B 4.00 3.80 6.00 PIN ONE INDICATOR 1.75 1 5.6 8 0.51 0.35 1.27 (0.30) 0.25 M 1.27 C B A 0.65 LAND PATTERN RECOMMENDATION 1.75 MAX 1.50 1.25 SEE DETAIL A 0.25 0.10 C FAN9611 — Interleaved Dual BCM PFC Controllers Physical Dimensions 0.25 0.19 0.10 C 0.50 0.25 X 45° (R0.10) NOTES: UNLESS OTHERWISE SPECIFIED GAGE PLANE A) THIS PACKAGE CONFORMS TO JEDEC MS-012, VARIATION AC, ISSUE C. B) ALL DIMENSIONS ARE IN MILLIMETERS. C) DIMENSIONS ARE EXCLUSIVE OF BURRS, MOLD FLASH AND TIE BAR PROTRUSIONS D) CONFORMS TO ASME Y14.5M-1994 E) LANDPATTERN STANDARD: SOIC127P600X175-16AM F) DRAWING FILE NAME: M16AREV12. (R0.10) 0.36 8° 0° SEATING PLANE 0.90 0.50 (1.04) DETAIL A SCALE: 2:1 Figure 66. 16-Lead SOIC Package Package drawings are provided as a service to customers considering Fairchild components. Drawings may change in any manner without notice. Please note the revision and/or date on the drawing and contact a Fairchild Semiconductor representative to verify or obtain the most recent revision. Package specifications do not expand the terms of Fairchild’s worldwide terms and conditions, specifically the warranty therein, which covers Fairchild products. Always visit Fairchild Semiconductor’s online packaging area for the most recent package drawings: http://www.fairchildsemi.com/packaging/. © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 34 FAN9611 — Interleaved Dual BCM PFC Controllers © 2008 Fairchild Semiconductor Corporation FAN9611 • Rev. 1.1.7 www.fairchildsemi.com 35