www.fairchildsemi.com Application Note AN4149 Design Guidelines for Quasi-Resonant Converters Using KA5Q-series Fairchild Power Switch (FPSTM) Abstract In general, a Quasi-Resonant Converter (QRC) shows lower EMI and higher power conversion efficiency compared to the conventional hard switched converter with a fixed switching frequency. Therefore, it is well suited for color TV applications that are noise sensitive. This application note presents practical design considerations of Quasi-Resonant Converters for color TV applications employing KA5Qseries FPSTM (Fairchild Power Switch). It includes designing the transformer, output filter and sync network, selecting the components and closing the feedback loop. The step-by-step design procedure described in this application note will help engineers design quasi-resonant converter easily. DR(n) LP(n) VO(n) NS(n) CO(n) Np DR1 CP(n) LP1 VO1 (B+) NS1 AC IN CO1 KA5Q-series Sync CP1 Drain PWM LP2 DR2 Cr VO2 (Sound) GND NS2 VFB CB Vcc Ra Ca Da DSY CP2 CO2 Linear regulator Rd RSY1 Rstr CSY MCU Na H11A817A RSY2 KA431 R3 Rbias R1 RF CF R2 R1 Q Picture ON Figure 1. Basic Quasi Resonant Converter Using KA5Q-series (Color TV Application) 1. Introduction system reliability and productivity. The KA5Q-series FPSTM (Fairchild Power Switch) is an integrated Pulse Width Modulation (PWM) controller and a Sense FET specifically designed for quasi-resonant off-line Switch Mode Power Supplies (SMPS) with minimal external components. Compared with a discrete MOSFET and PWM controller solution, it can reduce total cost, component count, size and weight while simultaneously increasing efficiency, Figure 1 shows the basic schematic of a quasi-resonant converter using KA5Q-series for the color TV application, which also serves as the reference circuit for the design process described in this paper. Vo1 is the output voltage that powers horizontal deflection circuit while Vo2 is the output voltage that supplies power to the Micro Controller Unit (MCU) through a linear regulator. Rev. 1.0.0 ©2005 Fairchild Semiconductor Corporation AN4149 APPLICATION NOTE 2. Step-by-step Design Procedure 1. Define the system specifications (Vlinemin, Vlinemax, fL , Po , Eff ) In this section, a design procedure is presented using the schematic of Figure 1 as a reference. Figure 2 illustrates the design flow chart. The detailed design procedures are as follows: [STEP-1] Define the system specifications 2. Determine DC link capacitor (CDC) and DC link voltage range - Line voltage range (Vlinemin and Vlinemax). - Line frequency (fL). 3. Determine the reflected output voltage (VRO) - Maximum output power (Po). - Estimated efficiency (Eff) : The power conversion efficiency must be estimated to calculate the maximum input power. If no reference data is available, set Eff = 0.7~0.75 for low voltage output applications and Eff = 0.8~0.85 for high voltage output applications. In the case of Color TV applications, the typical efficiency is 80~83%. 4. Determine the transformer primary side inductance (Lm) 5. Choose proper FPS considering input power and Idspeak 6. Determine the proper core and the minimum primary turns (Npmin) With the estimated efficiency, the maximum input power is given by P P in = ------oE ff 7. Determine the number of turns for each output For multiple output SMPS, the load occupying factor for each output is defined as 8. Determine the startup resistor Po ( n ) K L ( n ) = -----------Po 9. Determine the wire diameter for each winding Is the winding window area (Aw) enough ? (1) Y (2) where Po(n) is the maximum output power for the n-th output. For single output SMPS, KL(1)=1. It is assumed that Vo1 is the reference output that is regulated by the feedback control in normal operation. N Y Is it possible to change the core ? [STEP-2] Determine DC link capacitor (CDC) and the DC link voltage range. N 10. Choose the secondary side rectifier diodes It is typical to select the DC link capacitor as 2-3uF per watt of input power for universal input range (85-265Vrms) and 1uF per watt of input power for European input range (195V265Vrms). With the DC link capacitor chosen, the minimum DC link voltage is obtained as 11. Determine the output capacitors V DC min = 2 ⋅ ( V line 12. Design the synchronization network 13. Design the voltage drop circuit for burst operation P in ⋅ ( 1 – D ch ) ) – -----------------------------------C DC ⋅ f L min 2 (3) where CDC is the DC link capacitor and Dch is the duty cycle ratio for CDC to be charged as defined in Figure 3, which is typically about 0.2. Pin, Vlinemin and fL are specified in STEP-1. 14. Design the feedback control circuit The maximum DC link voltage is given as Design finished Figure 2. Flow Chart of Design Procedure 2 V DC max = 2V line max (4) where Vlinemax is specified in STEP-1. ©2005 Fairchild Semiconductor Corporation APPLICATION NOTE AN4149 [STEP-4] Determine the transformer primary side inductance (Lm) Figure 5 shows the typical waveforms of MOSFET drain current, secondary diode current and the MOSFET drain voltage of a Quasi Resonant Converter. During TOFF, the current flows through the secondary side rectifier diode and the MOSFET drain voltage is clamped at (VDC+VRO). When the secondary side current reduces to zero, the drain voltage begins to drop by the resonance between the effective output capacitor of the MOSFET and the primary side inductance (Lm). In order to minimize the switching loss, the KA5Qseries is designed to turn on the MOSFET when the drain voltage reaches its minimum voltage (VDC -VRO). Minimum DC link voltage DC link voltage T1 Dch = T1 / T2 = 0.2 T2 Figure 3. DC Link Voltage Waveform [STEP-3] Determine the reflected output voltage (VRO) Figure 4 shows the typical waveforms of the drain voltage of quasi-resonant flyback converter. When the MOSFET is turned off, the DC link voltage (VDC) together with the output voltage reflected to the primary (VRO) are imposed on the MOSFET. The maximum nominal voltage across the MOSFET (Vdsnom) is V ds nom = V DC max + V RO Ids ID (5) where VDCmax is as specified in equation (4). By increasing VRO, the capacitive switching loss and conduction loss of the MOSFET are reduced. However, this increases the voltage stress on the MOSFET as shown in Figure 4. Therefore, determine VRO by a trade-off between the voltage margin of the MOSFET and the efficiency. Typically, VRO is set as 120~180V so that Vdsnorm is 490~550V (75~85% of MOSFET rated voltage). V ds V DC +V RO V RO V RO V DC V DC -V RO T ON - + + VDC Lm - FPS Drain GND + Cr To determine the primary side inductance (Lm), the following variables should be determined beforehand : + Vds - VRO VRO Vdsnom VDC max VRO Vdsnom VRO 0V Figure 4. The Typical Waveform of MOSFET Drain Voltage for Quasi Resonant Converter ©2005 Fairchild Semiconductor Corporation TF Figure 5. Typical Waveforms of Quasi-Resonant Converter VO VRO - T OFF TS • The minimum switching frequency (fsmin) : The minimum switching frequency occurs at the minimum input voltage and full load condition and should be higher than the minimum switching frequency of FPS (20kHz). By increasing fsmin, the transformer size can be reduced. However, this results in increased switching losses. Therefore, determine fsmin by a trade-off between switching losses and transformer size. It is typical to set fsmin to be around 25kHz. • The falling time of the MOSFET drain voltage (TF) : As shown in Figure 5, the MOSFET drain voltage fall time is half of the resonant period of the MOSFET’s effective output capacitance and primary side inductance. By increasing TF, EMI can be reduced. However, this forces an increase of the resonant capacitor (Cr) resulting in increased switching losses. The typical value for TF is 2-2.5us. 3 AN4149 APPLICATION NOTE After determining fsmin and TF, the maximum duty cycle is calculated as V RO min D max = ------------------------------------⋅ ( 1 – fs × TF ) min V RO + V DC (6) where VDCmin is specified in equation (3) and VRO is determined in STEP-3. Then, the primary side inductance is obtained as min Lm 2 ( V DC ⋅ D max ) = --------------------------------------------min 2 ⋅ fs ⋅ P in (7) where Pin, VDCmin and Dmax are specified in equations (1), (3), and (6), respectively and fsmin is the minimum switching frequency. Once Lm is determined, the maximum peak current and RMS current of the MOSFET in normal operation are obtained as min I ds peak I ds rms V DC D max = ----------------------------------min Lm fs = (8) D max peak -------------- ⋅ I ds 3 (9) where VDCmin, Dmax and Lm are specified in equations (3), (6) and (7), respectively and fsmin is the minimum switching frequency. [STEP-5] Choose the proper FPS considering input power and peak drain current. With the resulting maximum peak drain current of the MOSFET (Idspeak) from equation (8), choose the proper FPS whose pulse-by-pulse current limit level (ILIM) is higher than Idspeak. Since FPS has ± 12% tolerance of ILIM, there should be some margin for ILIM when choosing the proper FPS device. Table 1 shows the lineups of KA5Q-series with rated output power and pulse-by-pulse current limit. Maximum Output Power PRODUCT 230Vac ±15% 85~ 265Vac KA5Q0740RT 90 W (85~170Vac) KA5Q0565RT ILIM Min Typ Max 4.4A 5A 5.6A 75 W 60 W 3.08A 3.5A 3.92A KA5Q0765RT 100 W 85 W 4.4A 5A 5.6A KA5Q1265RT 150 W 120 W 5.28A 6A 6.72A KA5Q1265RF 210 W 170 W 7.04A 8A 8.96A KA5Q1565RF 250 W 210 W 10.12A 11.5A 12.88A [STEP-6] Determine the proper core and the minimum primary turns. Table 2 shows the commonly used cores for C-TV application for different output powers. When designing the transformer, consider the maximum flux density swing in normal operation (∆B) as well as the maximum flux density in transient (Bmax). The the maximum flux density swing in normal operation is related to the hysteresis loss in the core while the maximum flux density in transient is related to the core saturation. With the chosen core, the minimum number of turns for the transformer primary side to avoid the over temperature in the core is given by peak NP min L m I ds 6 = -------------------------- × 10 ∆ BA e (10) where Lm is specified in equation (7), Idspeak is the peak drain current specified in equation (8), Ae is the crosssectional area of the transformer core in mm2 as shown in Figure 6 and ∆B is the maximum flux density swing in tesla. If there is no reference data, use ∆B =0.25~0.30 T. Since the MOSFET drain current exceeds Idspeak and reaches ILIM in a transient or fault condition, the transformer should be designed not to be saturated when the MOSFET drain current reaches ILIM . Therefore, the maximum flux density (Bmax) when drain current reaches ILIM should be also considered as NP min L m I LIM 6 = -------------------- × 10 B max A e (11) where Lm is specified in equation (7), ILM is the pulse-bypulse current limit, Ae is the cross-sectional area of the core in mm2 as shown in Figure 6 and Bmax is the maximum flux density in tesla. Figure 7 shows the typical characteristics of ferrite core from TDK (PC40). Since the core is saturated at low flux density as the temperature goes high, consider the high temperature characteristics. If there is no reference data, use Bmax =0.35~0.4 T. The primary turns should be determined as less than Npmin values obtained from equation (10) and (11). Aw (mm2) Table 1. FPS Lineups with Rated Output Power Ae (mm2) Figure 6. Window Area and Cross Sectional Area 4 ©2005 Fairchild Semiconductor Corporation APPLICATION NOTE AN4149 where n is obtained in equation (12) and Np and Ns1 are the number of turns for the primary side and the reference output, respectively. M agnetization Curves (typical) M aterial :PC40 25 ℃ The number of turns for the other output (n-th output) is determined as 500 60 ℃ Flux density B (mT) Vo ( n ) + VF ( n ) N s ( n ) = --------------------------------⋅ N s1 V o1 + V F1 100 ℃ 400 ( 14 ) where Vo(n) is the output voltage and VF(n) is the diode (DR(n)) forward voltage drop of the n-th output. 300 200 NS(n) + VF(n) DR(n) + VO(n) 100 - 0 0 800 M agnetic field H (A/m) 1600 Np NS2 - Figure 7. Typical B-H Characteristics of Ferrite Core (TDK/PC40) + VF2 DR2 VO2 + VRO - + Output Power Core 70-100W EER35 100-150W EER40 EER42 150-200W EER49 Ra Vcc + - VFa + N a Linear Regulator NS1 + VF1 - Da DR1 + VO1 - Table 2. Commonly Used Cores for C-TV Applications Figure 8. Simplified Diagram of the Transformer [STEP-7] Determine the number of turns for each output Figure 8 shows the simplified diagram of the transformer. It is assumed that Vo1 is the reference output which is regulated by the feedback control in normal operation. It is also assumed that the linear regulator is connected to Vo2 to supply a stable voltage for MCU. First, calculate the turns ratio (n) between the primary winding and reference output (Vo1) winding as a reference V R0 n = ------------------------V o1 + V F1 - Vcc winding design : KA5Q-series drops all the outputs including the Vcc voltage in standby mode in order to minimize the power consumption. Once KA5Q-series enters into standby mode, Vcc voltage is hysteresis controlled between 11V and 12V as shown in Figure 9. The sync threshold voltage is also reduced from 2.6V to 1.3V in burst mode. Therefore, design the Vcc voltage to be around 24V in normal operation for proper quasi-resonant switching in standby mode as can be observed by (12) where VRO is determined in STEP-3 and Vo1 is the reference output voltage and VF1 is the forward voltage drop of diode (DR1). ( 11 + 12 ) ⁄ 2- 1.3 ------------------------------≅ -------24 2.6 (15) Then, determine the proper integer for Ns1 so that the resulting Np is larger than Npmin as N p = n ⋅ N s1 > N p ©2005 Fairchild Semiconductor Corporation min (13) 5 AN4149 APPLICATION NOTE Vcc C DC 12V 11V Normal mode Standby mode AC line Vsync I sup R str Ra Vcc Da KA5Q-series 4.6V 2.6V Ca 3.6V 1.3V Figure 9. Burst operation in standby mode In general, switched mode power supply employs an error amplifier and an opto-coupler to regulate the output voltage. However, Primary Side Regulation (PSR) can be used for a low cost design if output regulation requirements are not very tight. PSR scheme regulates the output voltage indirectly by controlling the Vcc voltage without an optocoupler. KA5Q-series has an internal error amplifier with a fixed reference voltage of 32.5V for PSR applications. If PSR is used, set Vcc to 32.5V. After determining Vcc voltage in normal operation, the number of turns for the Vcc auxiliary winding (Na) is obtained as V cc + V Fa - ⋅ N s1 N a = ------------------------V o1 + V F1 ( turns ) ( 16 ) where VFa is the forward voltage drop of Da as defined in Figure 8. [STEP-8] Determine the startup resistor Figure 10 shows the typical startup circuit for KA5Q-series. Because some protections are implemented as latch mode, AC startup is typically used to provide a fast reset. Initially, FPS consumes only startup current (max 200uA) before it begins switching. Therefore, the current supplied through the startup resistor (Rstr) can charge the capacitors Ca1 and Ca2 while supplying startup current to FPS. When Vcc reaches a start voltage of 15V (VSTART), FPS begins switching, and the current consumed by FPS increases. Then, the current required by FPS is supplied from the transformer’s auxiliary winding. 6 Figure. 10 Startup Resistor and Vcc Auxiliary Circuit - Startup resistor (Rstr) : The average of the minimum current supplied through the startup resistor is given by I sup avg min ⎛ 2⋅V ⎞ V line start 1 ⎜ = ------------------------------------- – -----------------⎟ ⋅ -----------⎜ π 2 ⎟ R str ⎝ ⎠ ( 17 ) where Vlinemin is the minimum input voltage, Vstart is the start voltage (15V) of FPS and Rstr is the startup resistor. The startup resistor should be chosen so that Isupavg is larger than the maximum startup current (200uA). If not, Vcc can not be charged up to the start voltage and FPS will fail to start up. The maximum startup time is determined as T str max V start = C a ⋅ -------------------------------------------------avg max ( I sup – I start ) ( 18 ) Where Ca is the Vcc capacitor and Istartmax is the maximum startup current (200uA) of FPS. Once the startup resistor (Rstr) is determined, the maximum approximate power dissipation in Rstr is obtained as max⎞ 2 2 max⎞ ⎛ ⎛V +V ⋅V 2 2⋅V ⎜ ⎝ line ⎟ ⎠ start 1 start line P = ------------ ⋅ ⎜ ---------------------------------------------------------------- – -----------------------------------------------------------------⎟ str R str ⎜ π 2 ⎟ ⎝ ⎠ ( 19 ) where Vlinemax is the maximum input voltage, which is specified in STEP-1. The startup resistor should have a proper dissipation rating based on the value of Pstr. ©2005 Fairchild Semiconductor Corporation APPLICATION NOTE AN4149 [STEP-9] Determine the wire diameter for each winding based on the RMS current of each output. voltage and current margins for the rectifier diode are as follows V RRM > 1.3 ⋅ V D ( n ) The RMS current of the n-th secondary winding is obtained as I sec ( n ) rms = I ds V RO ⋅ K L ( n ) 1 – D max ----------------------- ⋅ -------------------------------------( Vo ( n ) + VF ( n ) ) D max rms ( 20 ) where Dmax and Idsrms are specified in equations (6) and (9), Vo(n) is the output voltage of the n-th output, VF(n) is the diode (DR(n)) forward voltage drop, VRO is specified in STEP-3 and KL(n) is the load occupying factor for n-th output defined in equation (2). The current density is typically 5A/mm2 when the wire is long (>1m). When the wire is short with a small number of turns, a current density of 6-10 A/mm2 is also acceptable. Do not use wire with a diameter larger than 1 mm to avoid severe eddy current losses as well as to make winding easier. For high current output, it is recommended using parallel windings with multiple strands of thinner wire to minimize skin effect. Check if the winding window area of the core, Aw (refer to Figure 6) is enough to accommodate the wires. The required winding window area (Awr) is given by A wr = A c ⁄ K F (21) where Ac is the actual conductor area and KF is the fill factor. Typically the fill factor is 0.2~0.25 for single output applications and 0.15~0.2 for multiple output applications. If the required window (Awr) is larger than the actual window area (Aw), go back to the STEP-6 and change the core to a bigger one. Sometimes it is impossible to change the core due to cost or size constraints. In that case, reduce VRO in STEP-3 or increase fsmin, which reduces the primary side inductance (Lm) and the minimum number of turns for the primary (Npmin) as can be seen in equation (7) and (10). [STEP-10] Choose the proper rectifier diodes in the secondary side based on the voltage and current ratings. The maximum reverse voltage and the rms current of the rectifier diode (DR(n)) of the n-th output are obtained as max V DC ⋅ ( Vo ( n ) + VF ( n ) ) V D ( n ) = V o ( n ) + --------------------------------------------------------------V RO ( 22 ) V RO K L ( n ) 1 – D max ----------------------- ⋅ -------------------------------------( Vo ( n ) + VF ( n ) ) D max ( 23 ) ID ( n ) rms = I ds rms I F > 1.5 ⋅ I D ( n ) (24) rms (25) where VRRM is the maximum reverse voltage and IF is the average forward current of the diode. A quick selection guide for the Fairchild Semiconductor rectifier diodes is given in Table 3. In this table, trr is the maximum reverse recovery time. Ultra Fast Recovery Diode Products VRRM IF trr Package EGP10B 100 V 1A 50 ns DO-41 UF4002 100 V 1A 50 ns DO-41 EGP20B 100 V 2A 50 ns DO-15 EGP30B 100 V 3A 50 ns DO-210AD FES16BT 100 V 16 A 35 ns TO-220AC EGP10C 150 V 1A 50 ns DO-41 EGP20C 150 V 2A 50 ns DO-15 EGP30C 150 V 3A 50 ns DO-210AD FES16CT 150 V 16 A 35 ns TO-220AC EGP10D 200 V 1A 50 ns DO-41 UF4003 200 V 1A 50 ns DO-41 EGP20D 200 V 2A 50 ns DO-15 EGP30D 200 V 3A 50 ns DO-210AD FES16DT 200 V 16 A 35 ns TO-220AC EGP10F 300 V 1A 50 ns DO-41 EGP20F 300 V 2A 50 ns DO-15 EGP30F 300 V 3A 50 ns DO-210AD EGP10G 400 V 1A 50 ns DO-41 UF4004 400 V 1A 50 ns DO-41 EGP20G 400 V 2A 50 ns DO-15 EGP30G 400 V 3A 50 ns DO-210AD UF4005 600 V 1A 75 ns DO-41 EGP10J 600 V 1A 75 ns DO-41 EGP20J 600 V 2A 75 ns DO-15 EGP30J 600 V 3A 75 ns DO-210AD UF4006 800 V 1A 75 ns TO-41 UF4007 1000 V 1A 75 ns TO-41 Table 3. Fairchild Diode Quick Selection Table max, rms where KL(n), VDC Dmax and Ids are specified in equations (2), (4), (6) and (9), respectively, VRO is specified in STEP-3, Vo(n) is the output voltage of the n-th output and VF(n) is the diode (DR(n)) forward voltage drop. The typical ©2005 Fairchild Semiconductor Corporation 7 AN4149 APPLICATION NOTE [STEP-11] Determine the output capacitors considering the voltage and current ripple. The ripple current of the n-th output capacitor (Co(n)) is obtained as Np rms ( ID ( n ) = + rms 2 ) – Io ( n ) 2 Drain (26) where Io(n) is the load current of the n-th output and ID(n)rms is specified in equation (23). The ripple current should be smaller than the maximum ripple current specification of the capacitor. The voltage ripple on the n-th output is given by CO Cr Ids 4.6/2.6V Sync Vcc D I peak V R K ( Vo ( n ) + VF ( n ) ) Na Co (n ) fs min D SY V sync where Co(n) is the capacitance, Rc(n) is the effective series resistance (ESR) of the n-th output capacitor, KL(n), Dmax and Idspeak are specified in equations (2), (6) and (8) respectively, VRO is specified in STEP-3, Io(n) and Vo(n) are the load current and output voltage of the n-th output, respectively and VF(n) is the diode (DR(n)) forward voltage drop. Sometimes it is impossible to meet the ripple specification with a single output capacitor due to the high ESR of the electrolytic capacitor. In those cases, additional L-C filter stages (post filter) can be used to reduce the ripple on the output. Da Ca o ( n ) max ds RO C ( n ) L ( n ) ∆ V o ( n ) = -------------------------+ ---------------------------------------------------------- (27) + V ds - GND R cc I V o1 Lm Sync comparator I cap ( n ) N s1 KA5Q-series C SY R SY1 R SY2 Figure. 11 Synchronization Circuit The peak value of the sync signal is determined by the voltage divider network RSY1 and RSY2 as V sync pk R SY2 = ---------------------------------- ⋅ V cc R SY1 + R SY2 ( 28 ) [STEP-12] Design the synchronization network. KA5Q-series employs a quasi resonant switching technique to minimize the switching noise as well as switching loss. In this technique, a capacitor (Cr) is added between the MOSFET drain and source as shown in Figure 11. The basic waveforms of a quasi-resonant converter are shown in Figure 12. The external capacitor lowers the rising slope of drain voltage, which reduces the EMI caused by the MOSFET turn-off. To minimize the MOSFET switching loss, the MOSFET should be turned on when the drain voltage reaches its minimum value as shown in Figure 12. The optimum MOSFET turn-on time is indirectly detected by monitoring the Vcc winding voltage as shown in Figure 11 and 12. The output of the sync detect comparator (CO) becomes high when the sync voltage (Vsync) exceeds 4.6V and low when the Vsync reduces below 2.6V. The MOSFET is turned on at the falling edge of the sync detect comparator output (CO). Choose the voltage divider RSY1 and RSY2 so that the peak value of sync voltage (Vsyncpk) is lower than the OVP threshold voltage (12V) in order to avoid triggering OVP in normal operation. Typically, Vsyncpk is set to 8~10V. To synchronize the Vsync with the MOSFET drain voltage, choose the sync capacitor (CSY) so that TF is same as TQ as shown in Figure 12. TF and TQ are given, respectively, as T F = π ⋅ L m ⋅ C eo R SY2 V cc T Q = R SY2 ⋅ C SY ⋅ ln ⎛ --------- ⋅ ----------------------------------⎞ ⎝ 2.6 R SY1 + R SY2⎠ (29) (30) where Lm is the primary side inductance of the transformer, Ns and Na are the number of turns for the output winding and Vcc winding, respectively and Ceo is the effective MOSFET output capacitance (Coss+Cr). ©2003 Fairchild Semiconductor Corporation 8 APPLICATION NOTE AN4149 Assuming that both Vo1 and Vo2 drop to half of their normal values, the maximum value of R3 for proper burst operation is given by Vds VRO ( V 02 ⁄ 2 – 0.7 – 2.5 ) ⋅ R 1 ⋅ R 2 R 3 = -----------------------------------------------------------------------------2.5 ⋅ ( R 1 + R 2 ) – ( R 2 ⋅ V 01 ⁄ 2 ) VRO VDC (32) TF Vsync VFB Vovp (12V) 1V Vsyncpk Ids 4.6V 2.6V Ibpk TQ CO Vcc MOSFET Gate 12V ON ON 11V Figure. 12 Synchronization Waveforms Normal Mode Figure 13. Burst Operation Waveforms [STEP-13] Design voltage drop circuit for the burst operation. To minimize the power consumption in the standby mode, KA5Q-series employs burst operation. Once FPS enters into burst mode, all the output voltages as well as effective switching frequencies are reduced as shown in Figure 13. Figure 14 shows the typical output voltage drop circuit for C-TV applications. Under normal operation, the picture on signal is applied and the transistor Q1 is turned on, which decouples R3 and D1 from the feedback network. Therefore, only Vo1 is regulated by the feedback circuit in normal operation and is determined as V o1 R1 + R2 = 2.5 ⋅ ⎛ --------------------⎞ ⎝ R2 ⎠ VO2 (31) Linear Regulator VO1 (B+) M icom RD Rbias R1 CF C KA431 In standby mode, the picture on signal is disabled and the transistor Q1 is turned off, which couples R3 and D1 to the reference pin of KA431. If R3 is small enough to make the reference pin voltage of KA431 higher than 2.5V, the current through the opto LED pulls down the feedback voltage (VFB) of FPS and forces FPS to stop switching. Once FPS stops switching, Vcc decreases, and when Vcc reaches 11V, it resumes switching with a predetermined peak drain current until Vcc reaches 12V. When Vcc reaches 12V, the switching operation is terminated again until Vcc reduces to 11V. In this way, Vcc is hysteresis controlled between 11V and 12V in the burst mode operation. ©2005 Fairchild Semiconductor Corporation Standby Mode A RF D1 R3 Q1 Picture ON R R2 Figure 14. Typical Feedback Circuit to Drop Output Voltage in Standby Mode [STEP-14] Design the feedback control circuit. Since the KA5Q-series employs current mode control as shown in Figure 15, the feedback loop can be easily implemented with a one-pole and one-zero compensation circuit. The current control factor of FPS, K is defined as 9 AN4149 APPLICATION NOTE I pk I LIM K = --------- = ----------------V FB V FBsat (33) where Ipk is the peak drain current and VFB is the feedback voltage for a given operating condition, ILIM is the current limit of the FPS and VFBsat is the internal feedback saturation voltage, which is typically 2.5V. In order to express the small signal AC transfer functions, the small signal variations of feedback voltage (vFB) and controlled output voltage (vo1) are introduced as vˆFB and vˆo1. vo1 vbias FPS vFB RB CB RD ibias Rbias iD CTR :1 CF RF R1 When the converter has more than one output, the low frequency control-to-output transfer function is proportional to the parallel combination of all load resistance, adjusted by the square of the turns ratio. Therefore, the effective load resistance is used in equation (34) instead of the actual load resistance of Vo1. Notice that there is a right half plane (RHP) zero (wrz) in the control-to-output transfer function of equation (34). Because the RHP zero reduces the phase by 90 degrees, the crossover frequency should be placed below the RHP zero. The Figure 16 shows the variation of a quasi-resonant flyback converter’s control-to-output transfer function for different input voltages. This figure shows the system poles and zeros together with the DC gain change for different input voltages. The gain is highest at the high input voltage condition and the RHP zero is lowest at the low input voltage condition. Figure 17 shows the variation of a quasi-resonant flyback converter’s control-to-output transfer function for different loads. This figure shows that the gain between fp and fz does not change for different loads and the RHP zero is lowest at the full load condition. The feedback compensation network transfer function of Figure 15 is obtained as KA431 R2 ˆ w i 1 + s ⁄ w zc v FB ----- --------------------------------ˆ - = - s ⋅ 1 + s ⁄ w pc v o1 Ipk ( 35 ) R B ⋅ CTR 1 1 where w i = ------------------------ , w zc = --------------- , w pc = --------------R1 RD CF RF CF RB CB MOSFET current Figure 15. Control Block Diagram For quasi-resonant flyback converters, the control-to-output transfer function using current mode control is given by vˆ o1 G vc = -------vˆ and RB is the internal feedback bias resistor of FPS, which is typically 2.8kΩ, CTR is the current transfer ratio of opto coupler and R1, RD, RF, CF and CB are shown in Figure 15. 40 dB FB K ⋅ R L V DC ( N p ⁄ N s1 ) ( 1 + s ⁄ w z ) ( 1 – s ⁄ w rz ) = ----------------------------------------------------- ⋅ ---------------------------------------------------------1 + s ⁄ wp 2 ( 2V RO + v DC ) fp ( 34 ) where VDC is the DC input voltage, RL is the effective total load resistance of the controlled output, which is defined as Vo12/Po. Additionally, Np and Ns1 are specified in STEP-7, VRO is specified in STEP-3, Vo1 is the reference output voltage, Po is specified in STEP-1 and K is specified in equation (33). The pole and zeros of equation (34) are defined as 20 dB fp Low input voltage 2 fz -20 dB frz fz frz -40 dB 1Hz RL ( 1 – D ) 1 (1 + D) w z = -------------------- , w rz = ---------------------------------------- and w p = ------------------2 R c1 C o1 R L C o1 DL m ( N s1 ⁄ N p ) High input voltage 0dB 10Hz 100Hz 1kHz 10kHz 100kHz Figure 16. QR Flyback Converter Control-to Output Transfer Function Variation for Different Input Voltages where Lm is specified in equation (7), D is the duty cycle of the FPS, Co1 is the output capacitor of Vo1 and RC1 is the ESR of Co1. 10 ©2005 Fairchild Semiconductor Corporation APPLICATION NOTE AN4149 When determining the feedback circuit component, there are some restrictions as described below: 40 dB fp (a) Design the voltage divider network of R1 and R2 to provide 2.5V to the reference pin of the KA431. The relationship between R1 and R2 is given as Light load 20 dB fp 2.5 ⋅ R 1 R 2 = -----------------------V o1 – 2.5 0dB Heavy load -20 dB frz frz fz -40 dB 1Hz 10Hz 100Hz 1kHz 10kHz 100kHz where Vo1 is the reference output voltage. (b) The capacitor connected to feedback pin (CB) is related to the shutdown delay time in an overload condition by Figure 17. QR Flyback Converter Control-to Output Transfer Function Variation for Different Loads When the input voltage and the load current vary over a wide range, determining the worst case for the feedback loop design is difficult. The gain together with zeros and poles varies according to the operating conditions. One simple and practical solution to this problem is designing the feedback loop for low input voltage and full load condition with enough phase and gain margin. The RHP zero is lowest at low input voltage and full load condition. The gain increases only about 6dB as the operating condition is changed from the lowest input voltage to the highest input voltage condition under universal input condition. The procedure to design the feedback loop is as follows (a) Set the crossover frequency (fc) below 1/3 of RHP zero to minimize the effect of the RHP zero. Set the crossover frequency below half of the minimum switching frequency (fsmin). (36) T delay = ( V SD – 2.5 ) ⋅ C B ⁄ I delay (37) where VSD is the shutdown feedback voltage and Idelay is the shutdown delay current. Typical values for VSD and Idelay are 7.5V and 5uA, respectively. In general, a delay of 20 ~ 50 ms is typical for most applications. Because CB also determines the high frequency pole (wpc) of the compensator transfer function as shown in equation (35), too large a CB can limit the control bandwidth by placing wpc at too low a frequency. Typical value for CB is 10-50nF. Application circuit to extend the shutdown time without limiting the control bandwidth is shown in Figure 19. By setting the zener breakdown voltage (Vz) slightly higher than 2.7V, the additional delay capacitor (Cz) is de-coupled from the feedback circuit in normal operation. When the feedback voltage exceeds the zener breakdown voltage (Vz), Cz and CB determine the shutdown time. (b) Determine the DC gain of the compensator (wi/wzc) to cancel the control-to-output gain at fc. FPS (c) Place a compensator zero (fzc) around fc/3. IFB (d) Place a compensator pole (fpc) around 3fc. Idelay vFB CB Loop gain T Cz 40 dB Vz fzc 20 dB Compensator fpc fp 0 dB Control to output fc V SD frz -20 dB VZ 2.7V fz -40 dB 1Hz 10Hz 100Hz 1kHz 10kHz Figure 18. Compensator Design 100kHz T delay Figure 19. Delayed Shutdown ©2005 Fairchild Semiconductor Corporation 11 AN4149 APPLICATION NOTE (c) The resistors Rbias and RD used together with the optocoupler H11A817A and the shunt regulator KA431 should be designed to provide proper operating current for the KA431 and to guarantee the full swing of the feedback voltage for the FPS device chosen. In general, the minimum values of cathode voltage and current for the KA431 are 2.5V and 1mA, respectively. Therefore, Rbias and RD should be designed to satisfy the following conditions: V bias – V OP – 2.5 --------------------------------------------- > I FB RD V OP ------------- > 1mA R bias ( 38 ) (39) where Vbias is the KA431 bias voltage as shown in Figure 16 and VOP is opto-diode forward voltage drop, which is typically 1V. IFB is the feedback current of FPS, which is typically 1mA. 12 ©2005 Fairchild Semiconductor Corporation APPLICATION NOTE AN4149 Design Example I (KA5Q0765RT) Application Device Input Voltage Output Power Output Voltage (Rated Current) Color TV KA5Q0765RT 85-265Vac 82W 125V (0.4A) (60Hz) 20V (0.5A) 16V (1.0A) 12V (0.5A) Schematic D201 EGP20D T1 EER3540 RT101 5D-9 C102 220uF 400V R103 68kΩ 0.5W BD101 10 1 3 11 4 13 L101 BEAD R104 68kΩ 0.5W D105 1N4937 1 Drain 3 Vcc IC101 SYNC 5 KA5Q0765RT GND 2 C104 47uF 50V FB 4 C109 47nF 50V C107 1nF 1kV ZD101 4.7V 0.5W C103 100nF 50V 12 6 D106 R106 D103 R107 1N4148 680Ω 1N4937 5.1Ω 0.25W 0.25W R105 470Ω 0.25W 14 15 16 C105 3.9nF 50V C206 470pF 1kV 17 7 PC301 817A ©2005 Fairchild Semiconductor Corporation 12V, 0.5A L203 C208 BEAD 1000uF 35V L202 C214 BEAD 100uF 160V 125V, 0.4A C215 47uF 160V D202 EGP20D 18 F101 FUSE 250V 3.0A C207 470pF 1kV 20V, 0.5A L204 C210 BEAD 1000uF 35V D203 EGP20J LF101 C101 330nF 275VAC C212 470pF 1kV D205 EGP20D C205 470pF 1kV VR201 30kΩ R201 1kΩ 0.25W R203 1kΩ R204 0.25W 39kΩ 0.25W C108 2.2nF 16V, 1A L201 C202 BEAD 1000uF 35V C203 22nF 50V R206 220kΩ 0.25W VR202 30kΩ D201 Q201 KA431 R205 4.7kΩ 0.25W Q202 KSC945 SW 201 R207 5.1kΩ 0.25W R208 5.1kΩ 0.25W 13 AN4149 APPLICATION NOTE Transformer Specifications EER 3540 N p1 1 18 2 17 3 16 Na N p2 N20V N125V/2 15 N 1 2 5V/2 4 Na N 16 V Np2 5 14 6 13 7 12 N125V/2 8 11 Np1 9 10 N 12 5 V /2 N12V N 12 V N16V N 20 V Transformer Schematic Diagram Winding Specifications No Pin (s→f) Wire φ 0.6 × 1 Turns Winding Method 35 Center Winding Np1 1-3 N125V/2 16 - 15 0.6 × 1 28 Center Winding N16V 18 - 17 0.4φ ×2 8 Center Winding N12V 12 - 13 0.5 × 1 6 Center Winding Np2 3-4 0.6 × 1 35 Center Winding N125V/2 15 - 14 0.5φ ×1 28 Center Winding N20V 11 - 10 0.5 × 1 10 Center Winding Na 7-6 11 Center Winding φ φ φ φ φ 0.3 × 1 Electrical Characteristics Inductance Leakage Inductance Pin Specification 1-4 565uH ± 5% 1-4 10uH Max Remarks 1kHz, 1V 2 nd all short Core & Bobbin Core : EER 3540 Bobbin : EER3540 Ae : 109 mm2 14 ©2005 Fairchild Semiconductor Corporation APPLICATION NOTE AN4149 Design Example II (KA5Q1265RF) Application Device Input Voltage Output Power Output Voltage (Rated Current) Color TV KA5Q1265RF 85-265Vac 154W 125V (0.8A) (60Hz) 20V (0.5A) 16V (2.0A) 12V (1.0A) Schematic D201 EGP20D T1 EER4242 RT101 10D-9 10 1 3 C102 470uF 400V R103 68kΩ 0.5W BD101 11 L101 BEAD R104 68kΩ 0.5W 4 3 Vcc IC101 SYNC 5 KA5Q1265RF GND 2 C104 47uF 50V FB 4 C109 47nF 50V ZD101 4.7V 0.5W C103 100nF 50V 13 C107 1.5nF 1kV D105 1N4937 1 Drain 12 6 D106 R106 D103 R107 1N4148 680Ω 1N4937 5.1Ω 0.25W 0.25W R105 470Ω 0.25W 14 15 16 C105 2.7nF 50V 17 7 PC301 817A ©2005 Fairchild Semiconductor Corporation C206 470pF 1kV L202 C214 BEAD 220uF 200V 125V, 0.8A C215 100uF 200V D202 EGP30D 18 F101 FUSE 250V 5.0A C207 470pF 1kV 12V, 1A L203 C208 BEAD 2200uF 35V D203 EGP30J LF101 C101 330nF 275VAC C212 470pF 1kV D205 EGP20D 20V, 0.5A L204 C210 BEAD 1000uF 35V C205 470pF 1kV VR201 30kΩ R201 1kΩ 0.25W R203 1kΩ R204 0.25W 39kΩ 0.25W C108 2.2nF 16V, 2A L201 C202 BEAD 2200uF 35V C203 22nF 50V R206 220kΩ 0.25W VR202 30kΩ D201 Q201 KA431 R205 4.7kΩ 0.25W Q202 KSC945 SW201 R207 5.1kΩ 0.25W R208 5.1kΩ 0.25W 15 AN4149 APPLICATION NOTE Transformer Specifications EER4242 18 N p1 1 Na 17 N 2 N p2 Na N 20V 16V 3 16 4 15 N 125V/2 N p2 5 14 N 12V 6 13 7 12 8 11 9 10 N 125V/2 N 125V /2 N 16V N 12 V N 125V/2 N p1 N 20 V Transformer Schematic Diagram Winding Specifications No Pin (s→f) Wire φ 0.5 × 2 Turns Winding Method 22 Center Winding Np1 1-3 N125V/2 16 - 15 0.5 × 2 18 Center Winding N16V 18 - 17 0.5φ ×2 5 Center Winding N12V 12 - 13 0.4 × 2 4 Center Winding Np2 3-4 0.5 × 2 22 Center Winding N125V/2 15 - 14 0.5φ ×2 18 Center Winding N20V 11 - 10 0.5 × 1 6 Center Winding Na 7-6 7 Center Winding φ φ φ φ φ 0.3 × 1 Electrical Characteristics Inductance Leakage Inductance Pin Specification 1-4 385uH ± 5% 1-4 10uH Max Remarks 1kHz, 1V 2 nd all short Core & Bobbin Core : EER 4242 Bobbin : EER4242 Ae : 234 mm2 16 ©2005 Fairchild Semiconductor Corporation APPLICATION NOTE AN4149 Design Example III (KA5Q1565RF) Application Device Input Voltage Output Power Output Voltage (Rated Current) Color TV KA5Q1565RF 85-265Vac 217W 125V (1.0A) (60Hz) 20V (1.0A) 16V (3.0A) 12V (2.0A) Schematic D201 EGP20D T1 EER5345 RT101 10D-9 10 1 3 C102 470uF 400V R103 68kΩ 0.5W BD101 11 4 Drain 3 Vcc IC101 SYNC 5 KA5Q1565RF GND 2 C104 47uF 50V FB 4 C109 47nF 50V ZD101 4.7V 0.5W C103 100nF 50V 13 C107 2nF 1kV D105 1N4937 1 12 6 D106 R106 D103 R107 1N4148 680Ω 1N4937 5.1Ω 0.25W 0.25W R105 470Ω 0.25W 14 15 16 C105 2.7nF 50V 17 7 PC301 817A ©2005 Fairchild Semiconductor Corporation C206 470pF 1kV L202 C214 BEAD 330uF 200V 125V, 1A C215 220uF 200V D202 FFPF05U20S 18 F101 FUSE 250V 5.0A C207 470pF 1kV 12V, 2A L203 C208 BEAD 2200uF 35V D203 FFPF05U60S LF101 C101 330nF 275VAC 20V, 1A L204 C210 BEAD 1000uF 35V D205 EGP30D L101 BEAD R104 68kΩ 0.5W C212 470pF 1kV C205 470pF 1kV VR201 30kΩ R201 1kΩ 0.25W R203 1kΩ R204 0.25W 39kΩ 0.25W C108 2.2nF 16V, 3A L201 C202 BEAD 2200uF 35V C203 22nF 50V R206 220kΩ 0.25W VR202 30kΩ D201 Q201 KA431 R205 4.7kΩ 0.25W Q202 KSC945 SW 201 R207 5.1kΩ 0.25W R208 5.1kΩ 0.25W 17 AN4149 APPLICATION NOTE Transformer Specifications EER5345 Np1 1 18 2 17 3 16 4 15 5 14 6 13 7 12 N p2 Na N20V N16V N125V/2 Np2 N125V/2 N12V N125V /2 N16V Na N12 V N125V/2 Np1 8 11 9 10 N20 V Transformer Schematic Diagram Winding Specifications No Pin (s→f) Wire φ 0.6 × 2 Turns Winding Method 21 Center Winding Np1 1-3 N125V/2 16 - 15 0.6 × 2 17 Center Winding N16V 18 - 17 0.6φ ×3 5 Center Winding N12V 12 - 13 0.6 × 2 4 Center Winding Np2 3-4 0.6 × 2 21 Center Winding N125V/2 15 - 14 0.6φ ×2 17 Center Winding N20V 11 - 10 0.5 × 1 6 Center Winding Na 7-6 7 Center Winding φ φ φ φ φ 0.3 × 1 Electrical Characteristics Inductance Leakage Inductance Pin Specification 1-4 325uH ± 5% 1-4 10uH Max Remarks 1kHz, 1V 2 nd all short Core & Bobbin Core : EER 5345 Bobbin : EER5345 Ae : 318 mm2 18 ©2005 Fairchild Semiconductor Corporation AN4149 APPLICATION NOTE Hangseok Choi, Ph.D Power Conversion Team / Fairchild Semiconductor Phone : +82-32-680-1383 Facsimile : +82-32-680-1317 E-mail : hangseok.choi@fairchildsemi.com DISCLAIMER FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS. LIFE SUPPORT POLICY FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR CORPROATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, or (c) whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in significant injury to the user. 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. www.fairchildsemi.com 9/20/05 0.0m 002 © 2005 Fairchild Semiconductor Corporation

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