® RT6220 6A, 23V, 500kHz, ACOTTM Synchronous Buck Converter General Description Features The RT6220 is a synchronous Buck converter with Advanced Constant On-Time (ACOTTM) mode control, 4.5V to 23V Input Voltage Range which provides a very fast transient response with no external compensators. The RT6220 operates from 4.5V to 23V input voltage, provides complete protection functions including Over Current Protection (OCP), Under Voltage Protection (UVP) and Over Voltage Protection (OVP). This IC also provides a 1.5ms internal soft-start function and an open-drain power good indicator. Adjustable from 0.6V to 5V Output Range Up to 98% Duty for 2S Battery Application 500kHz Switching Frequency ACOTTM Mode Performs Fast Transient Response Integrated MOSFETs 31mΩ Ω of High-Side MOSFET 20mΩ Ω of Low-Side MOSFET Supports MLCC Output Capacitors Internal Soft-Start (1.5ms typ) Built-in OVP/UVP/OCP Power Good Indicator Thermal Shutdown Ordering Information RT6220 Package Type QUF : UQFN-16L 3x3 (FC) (U-Type) Applications Lead Plating System G : Green (Halogen Free and Pb Free) PWM Operation / VOUT Protection A : with DEM/Latch AH : with DEM/Hiccup BL : without DEM/Latch BH : without DEM/Hiccup Note : Richtek products are : Laptop Computers Tablet PCs Networking Systems Servers Personal Video Recorders Flat Panel Television and Monitors Distributed Power Systems RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020. Suitable for use in SnPb or Pb-free soldering processes. Simplified Application Circuit VIN BOOT VIN CIN CB RT6220 L Chip Enable EN CEN VSYS VBYP PGND AGND Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 SW VOUT R1 VOUT FB PGOOD VCC COUT R2 RPGOOD VCC CVCC is a registered trademark of Richtek Technology Corporation. www.richtek.com 1 RT6220 Marking Information RT6220AGQUF RT6220BLGQUF 6J= : Product Code 6J=YM DNN 7M= : Product Code YMDNN : Date Code 7M=YM DNN RT6220AHGQUF RT6220BHGQUF 7P= : Product Code 7P=YM DNN YMDNN : Date Code 7N= : Product Code YMDNN : Date Code 7N=YM DNN Pin Configurations YMDNN : Date Code AGND EN FB VCC BOOT (TOP VIEW) 14 13 12 11 10 VIN 1 PGND 2 15 9 SW 8 SW SW 16 SW 4 5 6 7 VBYP PGOOD AGND AGND VOUT 3 UQFN-16L 3x3 (FC) Functional Pin Description Pin No. Pin Name Pin Function 1 VIN Power Input Connect to High-Side MOSFET Drain. 2 PGND Power Ground. 3 VBYP Switch Over Input Supply Voltage for VCC. Do not connect to VCC pin. 4 PGOOD Open-Drain Power Good Indicator Output. 5, 6, 14 AGND Analog Ground. 7 VOUT Output Voltage Sense Input. An internal discharging circuit is connected to this pin. SW Switch Node. 10 BOOT Bootstrap Supply for High-Side Gate Driver. A capacitor is needed to drive the power switch's gate above the supply voltage. It is connected between the SW and BOOT pins to form a floating supply across the power switch driver. 11 VCC 5V Linear Regulator Output for Internal Control Circuit. Bypass VCC to PGND with a 1F capacitor. VCC can only supply internal circuits. Do not connect to external loads. 12 FB Feedback Voltage Input. 13 EN Enable Control Input. Do not leave this pin floating. The slew rate of EN is recommended to be slower than 4.8V/s. 8, 9, 15, 16 Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 2 is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 Function Block Diagram VBYP VCC POR & Reference Soft-Start FB VREF VFB VCC VCC Switch-Over + + - On-Time One shot VIN Gate Control Logic Min off Time BOOT SW VCC EN PGND VOUT SW VOC 125% x VREF - + - OVP AGND Fault Logic + 91.5% x VREF PGOOD OCP + POK 58% x VREF + UVP VCC VIN VCC Regulator - Operation Overall OCP The RT6220 is a synchronous step-down converter with advanced constant on-time control mode. Using the ACOTTM control mode can reduce the output capacitance and provide fast transient response. It can minimize the component size without additional external compensation network. The inductor valley current is monitored cycle-by-cycle via the internal swiches, preventing an on-time until the current drops below the current limit. Power Good After soft-start is finished, the power good output goes high. The PGOOD pin is an open-drain output. Internal VCC Regulator The regulator provides 5V power to supply the internal control circuit. Connecting a 1μF ceramic capacitor for decoupling and stability is required. VCC Switch-Over Soft-Start Power Off In order to prevent the converter output voltage from overshooting during the startup period, the soft-start function is necessary. The soft-start time is internal setting and the duration is around 1.5ms. There is an internal discharging circuit to discharge the residual charge of output capacitor when converter is power off. Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 The internal regulator output will switch over to VBYP if VBYP level is higher than 4.6V. is a registered trademark of Richtek Technology Corporation. www.richtek.com 3 RT6220 Absolute Maximum Ratings (Note 1) VIN to PGND -------------------------------------------------------------------------------------------------- −0.3V to 27V SW to PGND -------------------------------------------------------------------------------------------------- −0.3V to (V IN + 0.3V) <30ns ----------------------------------------------------------------------------------------------------------- −5V to 28V BOOT to PGND ----------------------------------------------------------------------------------------------- (VSW − 0.3V) to (VSW + 6V) EN, FB to AGND --------------------------------------------------------------------------------------------- −0.3V to 27V VBYP to AGND ----------------------------------------------------------------------------------------------- −0.3V to 5.3V VCC, PGOOD, VOUT to AGND -------------------------------------------------------------------------- −0.3V to 6V PGND to AGND ----------------------------------------------------------------------------------------------- −0.3V to 0.3V Power Dissipation, PD @ TA = 25°C UQFN-16L 3x3 (FC) ------------------------------------------------------------------------------------------ 1.4W Package Thermal Resistance (Note 2) UQFN-16L 3x3 (FC), θJA ------------------------------------------------------------------------------------ 70°C/W UQFN-16L 3x3 (FC), θJC ------------------------------------------------------------------------------------ 15°C/W Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------ 260°C Junction Temperature ---------------------------------------------------------------------------------------- 150°C Storage Temperature Range ------------------------------------------------------------------------------- −65°C to 150°C ESD Susceptibility (Note 3) HBM (Human Body Model) --------------------------------------------------------------------------------- 2kV Recommended Operating Conditions (Note 4) Supply Input Voltage, VIN ---------------------------------------------------------------------------------- 4.5V to 23V Junction Temperature Range ------------------------------------------------------------------------------- −40°C to 125°C Ambient Temperature Range ------------------------------------------------------------------------------- −40°C to 85°C Electrical Characteristics (VIN = 12V, TA = 25°C, unless otherwise specified) Parameter Symbol Test Conditions Min Typ Max Unit -- 2.5 5 A (RT6220A/AH) -- 100 130 (RT6220BL/BH) -- 110 150 -- -- 2.5 RDS(ON)_H VBOOT – VSW = 5V -- 31 -- RDS(ON)_L -- 20 -- 7.6 -- 11.4 A 450 500 550 kHz -- 200 -- ns Supply Current Shutdown Current VEN = 0V Quiescent Current VEN = 2V, No Switching BOOT to SW Leakage Current BOOT to SW Leakage Current Switch On-Resistance VBYP = 5V, VEN = 0V Switch On-Resistance A A m Current Limit Current Limit IOC Valley current of low-side switch Switching Frequency and Minimum Off Timer Switching Frequency f SW Minimum Off-Time TOFF Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 4 is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 Parameter Symbol Test Conditions Min Typ Max Unit 120 125 130 % -- 5 -- s 53 58 63 % -- 5 -- s 0.594 0.600 0.606 V 1 1.5 2 ms 1.25 1.35 1.45 V 50 200 250 mV VEN = 2V -- 1 -- VEN = 0V -- 0 -- Protections OVP Trip Threshold VOVP OVP Propagation Delay TOVPDLY UVP Trip Threshold VUVP UVP Propagation Delay TUVPDLY With respect to output voltage With respect to output voltage Reference and Soft-Start Feedback Reference Voltage VREF Soft-Start Time TSS From EN high to PGOOD high Enable and UVLO EN Input High Voltage VENH EN Hysteresis VENHYS EN Input Current IEN VCC UVLO Rising VCCUVLO 3.8 4.2 4.45 V VCC UVLO Hysteresis VCCHYS 75 400 650 mV 4.805 5 5.295 V 4.4 4.6 4.8 V 150 200 400 mV -- 3 5 A VCC Regulator VCC Regulator VVCC VCC Switch Over Threshold to VBYP VCC Switch Over Hysteresis VBYP Rising Edge Switch Over On-Resistance Power Good Indicator PGOOD Threshold From Lower VOUT Rising 86.5 91.5 96.5 % PGOOD Low Hysteresis VOUT Falling -- 10 -- % -- 0.5 -- ms PGOOD Low to High Delay TPGDLY PGOOD Sink Current Capability VPGSINK Sink 4mA -- -- 0.4 V PGOOD Leakage Current IPGLEAK VPGOOD = 5V -- -- 100 nA TSD TJ Rising 135 150 -- °C -- 25 -- °C Thermal Shutdown Thermal Shutdown Threshold Thermal Shutdown Hysteresis Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect device reliability. Note 2. θJA is measured at TA = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. Note 3. Devices are ESD sensitive. Handling precaution is recommended. Note 4. The device is not guaranteed to function outside its operating conditions. Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 5 RT6220 Typical Application Circuit VIN 5.2V to 23V 1 CIN 10µF x 2 RB 2.2 (Optional) BOOT 10 VIN RT6220 13 VSYS 5V 3 2 5, 6, 14 EN SW 8, 9, 15, 16 CB 0.1µF L 2.2µH VOUT 7 FB 12 VBYP PGND PGOOD 4 VCC 11 AGND CFF 22pF (Optional) RPGOOD 100k COUT 22µF x 3 R1 22k VOUT 5V/6A R2 3k VCC CVCC 1µF Figure 1. Typical Application Circuit for VOUT = 5V VIN 4.5V to 23V 1 CIN 10µF x 2 RB 2.2 (Optional) BOOT 10 VIN RT6220 13 EN SW 3 VBYP VSYS 5V 2 5, 6, 14 2.2µH VOUT 7 FB 12 PGND AGND 8, 9, 15, 16 CB 0.1µF L PGOOD 4 VCC 11 CFF 4.7pF (Optional) RPGOOD 100k R1 162k COUT 22µF x 3 VOUT 3.3V/6A R2 36k VCC CVCC 1µF Figure 2. Typical Application Circuit for VOUT = 3.3V VIN 4.5V to 23V 1 CIN 10µF x 2 CEN 0.1µF VSYS 5V VIN RT6220 13 Chip Enable RB 2.2 (Optional) BOOT 10 3 2 5, 6, 14 SW EN AGND 1.2µH VOUT 7 FB 12 VBYP PGND 8, 9, 15, 16 CB 0.1µF L PGOOD 4 VCC 11 CFF 22pF (Optional) RPGOOD 100k R1 20k COUT 22µF x 4 VOUT 1V/6A R2 30k VCC CVCC 1µF Figure 3. Typical Application Circuit for VOUT = 1V Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 6 is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 Typical Operating Characteristics Performace waveforms are tested on the evaluation board of the Typical Application Circuit, VIN = 12V, VOUT = 1V, L = 1μH, TJ = 25°C, unless otherwise noted. Efficiency vs. Load Current Efficiency vs. Load Current 100 100 95 95 Efficiency (%) Efficiency (%) 90 85 VIN = 7.4V VIN = 12V VIN = 19V 80 75 90 VIN = 7.4V VIN = 12V VIN = 19V 85 80 70 65 60 0.001 75 VOUT = 1V, fSW = 500kHz, L = 1μH, DCR = 3.3mΩ, EN = 2V 0.01 0.1 1 70 0.001 10 VOUT = 3.3V, fSW = 500kHz, L = 2.2μH, DCR = 7mΩ, EN = 2V 0.01 Load Current (A) Efficiency vs. Load Current 10 Switching Frequency (kHz)1 600 95 Efficiency (%) 1 Switching Frequency vs. Load Current 100 VIN = 7.4V VIN = 12V VIN = 19V 90 85 80 75 VBYP = VOUT = 5V, fSW = 500kHz, L = 2.2μH, DCR = 7mΩ, EN = 2V 70 0.001 0.01 0.1 1 500 400 300 200 100 VIN = 7.4V, VOUT = 1V, EN = 2V 0 0.001 10 0.01 Load Current (A) Switching Frequency vs. Load Current 1 10 Switching Frequency vs. Load Current Switching Frequency (kHz)1 600 500 400 300 200 100 0 0.001 0.1 Load Current (A) 600 Switching Frequency (kHz)1 0.1 Load Current (A) VIN = 12V, VOUT = 1V, EN = 2V 0.01 0.1 1 Load Current (A) Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 10 500 400 300 200 100 0 0.001 VIN = 19V, VOUT = 1V, EN = 2V 0.01 0.1 1 10 Load Current (A) is a registered trademark of Richtek Technology Corporation. www.richtek.com 7 RT6220 Quiescent Current vs. Input Voltage Shutdown Current vs. Input Voltage 5 105 Shutdown Current (μA)1 Quiescent Current (μA) 110 100 95 90 85 5 7 9 11 13 15 17 19 21 3 2 1 EN = 0V EN = 2V, No Switching 80 4 0 23 5 7 9 Input Voltage (V) 11 13 15 17 19 21 23 Input Voltage (V) Output Voltage vs. Load Current Output Voltage vs. Load Current 1.05 3.50 1.04 3.45 Output Voltage (V) Output Voltage (V) 1.03 1.02 1.01 1.00 0.99 0.98 3.40 3.35 3.30 3.25 0.97 3.20 0.96 VIN = 12V, VOUT = 1V, EN = 2V 0.95 0.001 0.01 0.1 1 10 3.15 0.001 VIN = 12V, VOUT = 3.3V, EN = 2V 0.01 0.1 1 Load Current (A) Load Current (A) Output Voltage vs. Load Current Power On Through EN 10 5.25 5.20 VOUT (1V/Div) Output Voltage (V) 5.15 5.10 5.05 V CC (5V/Div) 5.00 4.95 4.90 4.85 4.80 4.75 0.001 VIN = 12V, VOUT = 5V, EN = 2V 0.01 0.1 1 10 PGOOD (5V/Div) EN (5V/Div) No Load, VIN = 12V Time (500μs/Div) Load Current (A) Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 8 is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 Power Off Through EN Load Transient Response VIN = 12V, EN = High, VOUT = 1V, COUT = 4 x 22μF, L = 1.2μH, IOUT = 0.6A to 6A @ 2.5A/μs VOUT (1V/Div) VOUT (20mV/Div) V CC (5V/Div) SW (20V/Div) PGOOD (5V/Div) EN (5V/Div) No Load, VIN = 12V IL (5A/Div) Time (500μs/Div) Time (50μs/Div) Load Transient Response Load Transient Response VIN = 12V, EN = High, VOUT = 3.3V, COUT = 3 x 22μF, L = 2.2μH, IOUT = 0.6A to 6A @ 2.5A/μs VIN = 12V, EN = High, VOUT = 5V, COUT = 3 x 22μF, L = 2.2μH, IOUT = 0.6A to 6A @ 2.5A/μs VOUT (100mV/Div) VOUT (200mV/Div) SW (20V/Div) SW (20V/Div) IL (5A/Div) IL (5A/Div) Time (50μs/Div) Time (50μs/Div) UVP OVP VOUT (1V/Div) VOUT (1V/Div) PGOOD (5V/Div) PGOOD (5V/Div) SW (10V/Div) VIN (10V/Div) IL (5A/Div) VIN = 12V, EN = High Time (50μs/Div) Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 VIN = 12V, VOUT = 1.25V, EN = High Time (50μs/Div) is a registered trademark of Richtek Technology Corporation. www.richtek.com 9 RT6220 Application Information The RT6220 is high-performance 500kHz 6A step-down regulators with internal power switches and synchronous rectifiers. It features an Advanced Constant On-Time (ACOT TM) control architecture that provides stable operation for ceramic output capacitors without complicated external compensation, among other benefits. The input voltage range is from 4.5V to 23V, and the output voltage is adjustable from 0.6V to 5V. The proprietary ACOT TM control scheme improves conventional constant on-time architectures, achieving nearly constant switching frequency over line, load, and output voltage ranges. Since there is no internal clock, response to transients is nearly instantaneous and inductor current can ramp quickly to maintain output regulation without large bulk output capacitance. ACOTTM Control Architecture In order to achieve good stability with low-ESR ceramic capacitors, ACOTTM uses a virtual inductor current ramp generated inside the IC. This internal ramp signal replaces the ESR ramp normally provided by the output capacitor's ESR. The ramp signal and other internal compensations are optimized for low-ESR ceramic output capacitors. Making the on-time proportional to VOUT and inversely proportional to VIN is not sufficient to achieve good constant-frequency behavior for several reasons. First, voltage drops across the MOSFET switches and inductor cause the effective input voltage to be less than the measured input voltage and the effective output voltage to be greater than the measured output voltage as sensing input and output voltage. When the load changes, the switch voltage drops change causing a switching frequency variation with load current. Also, at light loads if the inductor current goes negative, the switch deadtime between the synchronous rectifier turn-off and the high-side switch turn-on allows the switching node to rise to the input voltage. This increases the effective on-time and causes the switching frequency to drop noticeably. One way to reduce these effects is to measure the actual switching frequency and compare it to the desired range. This has the added benefit eliminating the need to sense Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 10 the actual output voltage, potentially saving one pin connection. The ACOTTM uses this method, measuring the actual switching frequency and modifying the on-time with a feedback loop to keep the average switching frequency in the desired range. ACOTTM One-shot Operation The RT6220 control algorithm is simple to understand. The feedback voltage, with the virtual inductor current ramp added, is compared to the reference voltage. When the combined signal is less than the reference, the on-time one-shot is triggered, as long as the minimum off-time one-shot is clear and the measured inductor current (through the synchronous rectifier) is below the current limit. The on-time one-shot turns on the high-side switch and the inductor current ramps up linearly. After the ontime, the high-side switch is turned off and the synchronous rectifier is turned on and the inductor current ramps down linearly. At the same time, the minimum off-time one-shot is triggered to prevent another immediate on-time during the noisy switching time and allow the feedback voltage and current sense signals to settle. The minimum off-time is kept short (200ns typical) so that rapidly-repeated ontimes can raise the inductor current quickly when needed. Diode Emulation Mode (DEM) In diode emulation mode, the RT6220 automatically reduces switching frequency at light load conditions to maintain high efficiency. This reduction of frequency is achieved smoothly. As the output current decreases from heavy load conditions, the inductor current is also reduced, and eventually comes to the point that its current valley touches zero, which is the boundary between continuous conduction and discontinuous conduction modes. To emulate the behavior of diodes, the low-side MOSFET allows only partial negative current to flow when the inductor free wheeling current becomes negative. As the load current is further decreased, it takes longer and longer time to discharge the output capacitor to the level that requires the next “ON” cycle. In reverse, when the output current increases from light load to heavy load, the switching frequency increases to the preset value as the is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 inductor current reaches the continuous conduction. The transition load point to the light load operation is shown in Figure 4 and can be calculated as follows : IL Slope = (VIN - VOUT) / L Current Limit IPEAK ILOAD = IPEAK / 2 tON t Figure 4. Boundary Condition of CCM/DEM ILOAD (VIN VOUT ) tON 2L where tON is the on-time. The switching waveforms may appear noisy and asynchronous when light load causes diode emulation operation. This is normal and results in high efficiency. Trade offs in DEM noise vs. light load efficiency is made by varying the inductor value. Generally, low inductor values produce a broader efficiency vs. load curve, while higher values result in higher full load efficiency (assuming that the coil resistance remains fixed) and less output voltage ripple. Penalties for using higher inductor values include larger physical size and degraded load transient response (especially at low input voltage levels). During discontinuous switching, the on-time is immediately increased to add “hysteresis” to discourage the IC from switching back to continuous switching unless the load increases substantially. The IC returns to continuous switching as soon as an on-time is generated before the inductor current reaches zero. The on-time is reduced back to the length needed for 500kHz switching and encouraging the circuit to remain in continuous conduction, preventing repetitive mode transitions between continuous switching and discontinuous switching. Linear Regulators (VCC) The RT6220 includes a 5V linear regulator (VCC). The VCC regulator steps down input voltage to supply both internal circuitry and gate drivers. Do not connect the VCC Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 pin to external loads. When PGOOD is pulled high and BYP pin voltage is above 4.6V, an internal 3Ω P-MOSFET switch connects VCC to the BYP pin while the VCC linear regulator is simultaneously turned off. The RT6220 current limit is a cycle-by-cycle “valley” type, measuring the inductor current through the synchronous rectifier during the off-time while the inductor current ramps down. The current is determined by measuring the voltage between Source and Drain of the synchronous rectifier, adding temperature compensation for greater accuracy. If the current exceeds the current limit, the on-time oneshot is inhibited until the inductor current ramps down below the current limit. If the output current exceeds the available inductor current (controlled by the current limit mechanism), the output voltage will drop. If it drops below the output under-voltage protection level (see next section), the IC will stop switching to avoid excessive heat. Output Over-Voltage Protection and Under-Voltage Protection The RT6220 features an output Over-Voltage Protection (OVP). For RT6220A and RT6220BL, if the output voltage rises above the regulation level, the IC stops switching and is latched off. On the other hand, for RT6220AH and RT6220BH, the IC will stop switching and restart automatically after a short period which is the so-called Hiccup mode. The RT6220 also features an output UnderVoltage Protection (UVP). If the output voltage drops below the UVP trip threshold for longer than 5μs (typical), the UVP is triggered, and the IC will shutdown. Likewise, for RT6220A and RT6220BL, the IC stops switching and is latched off. On the other hand, for RT6220AH and RT6220BH, the IC will stop switching and enter the Hiccup mode. To restart operation from latch off, toggle EN or power the IC off and then turn on again. Input Under-Voltage Lockout In addition to the enable function, the RT6220 features an Under-Voltage Lockout (UVLO) function that monitors the input voltage. To prevent operation without fully-enhanced internal MOSFET switches, this function inhibits switching when input voltage drops below the UVLO-falling threshold. is a registered trademark of Richtek Technology Corporation. www.richtek.com 11 RT6220 The IC resumes switching when input voltage exceeds the UVLO-rising threshold. Over-Temperature Protection The RT6220 features an Over-Temperature Protection (OTP) circuitry to prevent overheating due to excessive power dissipation. The OTP shuts down switching operation when the junction temperature exceeds 150°C. Once the junction temperature cools down by approximately 25°C the IC resumes normal operation with a complete soft-start. For continuous operation, provide adequate cooling so that the junction temperature does not exceed 150°C. Note that the VCC regulator remains on as the OTP is triggered. Enable and Disable The enable input (EN) has a logic-low level of 1.15V. When VEN is below this level, the IC enters shutdown mode and supply current drops to less than 5μA (typical). When VEN exceeds its logic-high level (1.35V typical), the IC is fully operational. Soft-Start The RT6220 provides an internal soft-start function to prevent large inrush current and output voltage overshoot when the converter starts up. The soft-start (SS) automatically begins once the chip is enabled. During softstart, it clamps the ramp of internal reference voltage which is compared with FB signal. And it will correct the output voltage more accurately after soft-start. The typical softstart duration is 1.5ms. Power Off When VEN is pulled to GND or lower than the logic-low level of 1.15V, there is an internal discharging resistor to discharge the residual charge inside the output capacitors. Besides, the value of discharging resistor is about twenty ohms. Power Good Output (PGOOD) The power good output is an open-drain output that requires a pull-up resistor. When the output voltage is 20% (typical) below its set voltage, PGOOD will be pulled low. It is held low until the output voltage returns to 90% of its set voltage Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 12 once more. During soft-start, PGOOD is actively held low and only allowed to be pulled high after soft-start is over and the output reaches 90% of its set voltage and the PGOOD low to high delay(500μs typical) has passed. There is a 2μs PGOOD high to low delay built into PGOOD circuitry to prevent false triggering. External Bootstrap Capacitor (CBOOT) Connect a 0.22μF low ESR ceramic capacitor between the BOOT and SW pins. This bootstrap capacitor provides the gate driver supply voltage for the high-side N-MOSFET switch. The internal power MOSFET switch gate driver is optimized to turn the switch on fast enough for low power loss and good efficiency, and slow enough to reduce EMI. Switch turn-on is when most EMI occurs since VSW rises rapidly. During switch turn-off, SW is discharged relatively slowly by the inductor current during the dead-time between high-side and low-side switch on-times. In some cases it is desirable to reduce EMI further, at the expense of some additional power dissipation. The switch turn-on can be slowed by placing a small (<10Ω) resistance between BOOT and the external bootstrap capacitor. This will slow the high-side switch turn-on and VSW's rise. Setting the Output Voltage The output voltage of RT6220 is adjustable and with valley control. There is an easy way to determine the output voltage only by two resistors, R1 and R2. As the feedback circuit shown in Figure 5. the relation of VOUT and VREF can be derived as VOUT = (1+R1/R2) x VREF readily. Generally, the stability is a serious issue for converter. In order to achieve better performance on stability and transient, a feed-forward capacitor, CFF, is added to increase the noise margin and transient response of loop control. However, there is a tradeoff of adding a feed-forward capacitor. An additional dc offset will be generated on output voltage due to the amplified feedback ripple by feedforward compensator. This is not always the case that every CFF makes the same value of dc offset, it is based on different pole and zero placement generated by R1, R2 and C FF. For simplicity, a symbol named Vdc,offset is supposed to be the value of dc offset. This value may influence the performance (e.g. regulation or peak value is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 of VOUT) of converter slightly, the suggested CFFis to select a pair of pole and zero to provide the maximum phase lead at switching frequency. VOUT,valley = 1+ R1 VREF +Vdc,offset R2 VOUT,valley is the valley of output voltage, and Vdc,offset is used for describing the additional dc offset on VOUT, the value is related to the output voltage ripple and CFF. VOUT R1 CFF (opt.) FB RT6220 R2 GND Figure 5. The Equivalent Circuit of Feedback Loop Inductor Selection Selecting an inductor involves specifying its inductance and also its required peak current. The exact inductor value is generally flexible and is ultimately chosen to obtain the best mix of cost, physical size, and circuit efficiency. Lower inductor values benefit from reduced size and cost and they can improve the circuit's transient response. However, they increase the inductor ripple current and output voltage ripple and reduce the efficiency due to the resulting higher peak currents. Conversely, higher inductor values increase efficiency, but the inductor will either be physically larger or have higher resistance since more turns of wire are required and transient response will be slower since more time is required to change current (up or down) in the inductor. A good compromise between size, efficiency, and transient response is to use a ripple current (ΔIL) about 20-50% of the desired full output load current. Calculate the approximate inductor value by selecting the input and output voltages, the switching frequency (fSW), the maximum output current (IOUT(MAX)) and estimating a ΔIL as some percentage of that current. Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 V (VIN VOUT ) L OUT VIN fSW IL Once an inductor value is chosen, the ripple current (ΔIL) is calculated to determine the required peak inductor current. V (VIN VOUT ) I IL OUT and IL(PEAK) IOUT(MAX) L VIN fSW L 2 To guarantee the required output current, the inductor needs a saturation current rating and a thermal rating that exceeds IL(PEAK). These are minimum requirements. To maintain control of inductor current in overload and shortcircuit conditions, some applications may desire current ratings up to the current limit value. However, the IC's output under-voltage shutdown feature make this unnecessary for most applications. For best efficiency, choose an inductor with a low DC resistance that meets the cost and size requirements. For low inductor core losses some type of ferrite core is usually best and a shielded core type, although possibly larger or more expensive, will probably give fewer EMI and other noise problems. Input Capacitor Selection High quality ceramic input decoupling capacitor, such as X5R or X7R, with values greater than 20μF are recommended for the input capacitor. The X5R and X7R ceramic capacitors are usually selected for power regulator capacitors because the dielectric material has less capacitance variation and more temperature stability. Voltage rating and current rating are the key parameters when selecting an input capacitor. Generally, selecting an input capacitor with voltage rating 1.5 times greater than the maximum input voltage is a conservatively safe design. The input capacitor is used to supply the input RMS current, which can be calculated using the following equation : IRMS VOUT V I 2 (1 OUT ) IOUT 2 L VIN VIN 12 The next step is to select a proper capacitor for RMS current rating. One good design uses more than one capacitor with low Equivalent Series Resistance (ESR) in parallel to form a capacitor bank. The input capacitance is a registered trademark of Richtek Technology Corporation. www.richtek.com 13 RT6220 value determines the input ripple voltage of the regulator. The input voltage ripple can be approximately calculated using the following equation : VIN IOUT VIN V (1 OUT ) CIN fSW VOUT VIN The typical operating circuit is recommended to use two 10μF low ESR ceramic capacitors on the input. Output Capacitor Selection The RT6220 is optimized for ceramic output capacitors and best performance will be obtained by using them. The total output capacitance value is usually determined by the desired output voltage ripple level and transient response requirements for sag (undershoot on positive load steps) and soar (overshoot on negative load steps). Output ripple at the switching frequency is caused by the inductor current ripple and its effect on the output capacitor's ESR and stored charge. These two ripple components are called ESR ripple and capacitive ripple. Since ceramic capacitors have extremely low ESR and relatively little capacitance, both components are similar in amplitude and both should be considered if ripple is critical. VRIPPLE VRIPPLE(ESR) VRIPPLE(C) VRIPPLE(ESR) IL RESR VRIPPLE(C) IL 8 COUT fSW In addition to voltage ripple at the switching frequency, the output capacitor and its ESR also affect the voltage sag (undershoot) and soar (overshoot) when the load steps up and down abruptly. The ACOT transient response is very quick and output transients are usually small. However, the combination of small ceramic output capacitors (with little capacitance), low output voltages (with little stored charge in the output capacitors), and low duty cycle applications (which require high inductance to get reasonable ripple currents with high input voltages) increases the size of voltage variations in response to very quick load changes. Typically, load changes occur slowly with respect to the IC's 500kHz switching frequency. However, some modern digital loads can exhibit nearly instantaneous load changes and the following section shows how to calculate the worst-case voltage swings in response to very fast load steps. Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 14 The amplitude of the ESR step up or down is a function of the load step and the ESR of the output capacitor : VESR_STEP IOUT RESR The amplitude of the capacitive sag is a function of the load step, the output capacitor value, the inductor value, the input-to-output voltage differential, and the maximum duty cycle. The maximum duty cycle during a fast transient is a function of the on-time and the minimum off-time since the ACOTTM control scheme will ramp the current using on-times spaced apart with minimum off-times, which is as fast as allowed. Calculate the approximate on-time (neglecting parasitics) and maximum duty cycle for a given input and output voltage as : VOUT tON t ON and DMAX VIN fSW tON + tOFF(MIN) The actual on-time will be slightly longer as the IC compensates for voltage drops in the circuit, but we can neglect both of these since the on-time increases compensations for the voltage losses. Calculate the output voltage sag as : VSAG L (IOUT )2 2 COUT ( VIN(MIN) DMAX VOUT ) The amplitude of the capacitive soar is a function of the load step, the output capacitor value, the inductor value and the output voltage : VSOAR L ( IOUT )2 2 COUT VOUT Most applications never experience instantaneous full load steps and the RT6220's high switching frequency and fast transient response can easily control voltage regulation at all times. Therefore, sag and soar are seldom an issue except in very low-voltage CPU core or DDR memory supply applications, particularly for devices with high clock frequencies and quick changes into and out of sleep modes. In such applications, simply increasing the amount of ceramic output capacitor (sag and soar are directly proportional to capacitance) or adding extra bulk capacitance can easily eliminate any excessive voltage transients. In any application with large quick transients, it should calculate soar and sag to make sure that over-voltage protection and under-voltage protection will not be triggered. is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 Thermal Considerations Layout Considerations For continuous operation, do not exceed absolute maximum junction temperature. The maximum power dissipation depends on the thermal resistance of the IC package, PCB layout, rate of surrounding airflow, and difference between junction and ambient temperature. The maximum power dissipation can be calculated by the following formula : Layout is very important in high frequency switching converter design. The PCB can radiate excessive noise and contribute to converter instability with improper layout. Certain points must be considered before starting a layout using the RT6220. PD(MAX) = (TJ(MAX) − TA) / θJA where TJ(MAX) is the maximum junction temperature, TA is the ambient temperature, and θJA is the junction to ambient thermal resistance. For recommended operating condition specifications, the maximum junction temperature is 125°C. The junction to ambient thermal resistance, θJA, is layout dependent. For UQFN-16L 3x3 (FC) package, the thermal resistance, θJA, is 70°C/W on a standard JEDEC 51-7 four-layer thermal test board. The maximum power dissipation at TA = 25°C can be calculated by the following formula : P D(MAX) = (125°C − 25°C) / (70°C/W) = 1.4W for UQFN-16L 3x3 (FC) package Maximum Power Dissipation (W)1 The maximum power dissipation depends on the operating ambient temperature for fixed T J(MAX) and thermal resistance, θJA. The derating curve in Figure 6 allows the designer to see the effect of rising ambient temperature on the maximum power dissipation. 2.0 Make traces of the high current paths as short and wide as possible. Put the input capacitor as close as possible to the device pins (VIN and PGND). The SW node encounters high frequency voltage swings so it should be kept in a small area. Keep sensitive components away from the SW node to prevent noise coupling. The PGND pin should be connected to a strong ground plane for heat sinking and noise protection. Avoid using vias in the power path connections that have switched currents (from CIN to PGND and CIN to VIN) and the switching node (SW). The ground of VCC is recommended to connect to GND layer through via. An example of PCB layout guide is shown in Figure 7 for reference. Four-Layer PCB 1.6 1.2 0.8 0.4 0.0 0 25 50 75 100 125 Ambient Temperature (°C) Figure 6. Derating Curve of Maximum Power Dissipation Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 15 RT6220 CFF (opt.) GND The optional compensating components must be connected as close to the IC as possible R1 R2 VIN EN F B VCC BOOT The input capacitor must be placed as close to the IC as possible. AGND VIN 14 13 12 11 10 The output capacitor must be placed near the IC 1 15 9 SW 8 SW L VOUT SW 2 16 3 4 5 6 7 VBYP PGOOD AGND AGND VOUT SW GND GND CBOOT CIN PGND CVCC VOUT COUT SW should be connected to inductor by wide and short trace. Keep sensitive components away from this trace. Impedance between PGND and AGND should be as small as possible for unified ground voltage. Figure 7. PCB Layout Guide Copyright © 2014 Richtek Technology Corporation. All rights reserved. www.richtek.com 16 is a registered trademark of Richtek Technology Corporation. DS6220-03 April 2015 RT6220 Outline Dimension Symbol Dimensions In Millimeters Dimensions In Inches Min. Max. Min. Max. A 0.500 0.600 0.020 0.024 A1 0.000 0.050 0.000 0.002 A3 0.100 0.175 0.004 0.007 D 2.900 3.100 0.114 0.122 E 2.900 3.100 0.114 0.122 b 0.150 0.250 0.006 0.010 b1 0.100 0.200 0.004 0.008 L 0.350 0.450 0.014 0.018 L1 0.750 0.850 0.030 0.033 L2 0.550 0.650 0.022 0.026 e 0.400 0.016 K 0.975 0.038 K1 1.335 0.053 K2 1.675 0.066 K3 1.935 0.076 K4 0.975 0.038 K5 1.675 0.066 U-Type 16L QFN 3x3 (FC) Package Copyright © 2014 Richtek Technology Corporation. All rights reserved. DS6220-03 April 2015 is a registered trademark of Richtek Technology Corporation. www.richtek.com 17 RT6220 Richtek Technology Corporation 14F, No. 8, Tai Yuen 1st Street, Chupei City Hsinchu, Taiwan, R.O.C. Tel: (8863)5526789 Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries. www.richtek.com 18 DS6220-03 April 2015