Data Sheet

PRM® Regulator
Remote Sense Only
PRM48Bx480y600A00
High Efficiency Remote Sense PRM Converter FEATURES
DESCRIPTION
The VI Chip PRM® Regulator is a high efficiency
converter, operating from a 38 to 55 Vdc input to
generate a regulated 5 to 55 Vdc output. The ZVS Buck
– Boost topology enables high switching frequency (~1
MHz) operation with high conversion efficiency. High
switching frequency reduces the size of reactive
components enabling power density up to 2,072 W/in3.
 45V (38 to 55 VIN), non-isolated ZVS buck-boost
regulator
 5 to 55 V adjustable output range
 Building block for high efficiency DC-DC systems
 600W Output Power in 1.11 in2 footprint
 97.6% typical efficiency, at full load
 2,072 W/in3 (124 W/cm3) Power Density
 Enables a 48 V to 1.5 V, 345 A isolated, regulated
solution with total footprint of 3.3in2 (21cm2)
 Flexible “Remote Sense” architecture optimizes
regulation / feedback loop design to fit application
requirements
 Current Feedback signal allows dynamic adjustment
of current limit setpoint
 6.36 MHrs MTBF (MIL-HDBK-217Plus Parts Count)
The full VI Chip package is compatible with standard
pick-and-place and surface mount assembly processes
with a planar thermal interface area and superior thermal
conductivity.
In a Factorized Power Architecture™ system, the
PRM48BF480T600A00
and
downstream
VTM™
transformer minimize distribution and conversion losses
in a high power solution.
TYPICAL APPLICATIONS






An external control loop and current sensor maintain
regulation and enable flexibility both in the design of
voltage and current compensation loops to control of
output voltages and currents.
High Efficiency Server Processor and Memory Power
High Density ATE system DC-DC power
Telecom NPU and ASIC core power
LED drivers
High Density Power Supply DC-DC rail outputs
Non-isolated power converters
48 V to 1.5 V, 230A Voltage Regulator
Voltage
Control
Feedback
Enable/
Disable
Voltage
Reference
PC
PR
+IN
38 to 55
Vdc Input
TM
+OUT
PC
VIZ0051
PRM
-IN
IF RE
+OUT1
+OUT2
+IN
VIV0007TFJ
-IN
-OUT
SG VC
VC
Current
Sense
-OUT1
-OUT2
PC
+OUT1
+OUT2
+IN
VIV0007TFJ
Constant
Vc
-IN
VC
-OUT1
-OUT2
PRM® Regulator Rev 1.2 vicorpower.com Page 1 of 22 07/2015 800 927.9474 Load
PRM48Bx480y600A00
1.0 ABSOLUTE MAXIMUM RATINGS
The ABSOLUTE MAXIMUM ratings below are stress ratings only. Operation at or beyond these maximum ratings can
cause permanent damage to device. Electrical specifications do not apply when operating beyond rated operating
conditions. All voltages are specified relative to SG unless otherwise noted. Positive pin current represents current flowing
out of the pin.
2.0 ELECTRICAL CHARACTERISTICS
Specifications apply over all line and load conditions, TJ = 25 ºC and output voltage from 20V to 55V, unless otherwise
noted. Boldface specifications apply over the temperature range of -40 ºC < TJ < 125 ºC (T-grade).
PRM® Regulator Rev 1.2 vicorpower.com Page 2 of 22 07/2015 800 927.9474 PRM48Bx480y600A00
3.0 SIGNAL CHARACTERISTICS
Specifications apply over all line and load conditions, TJ = 25 ºC and Output Voltage from 20V to 55V, unless otherwise
noted. Boldface specifications apply over the temperature range of -40 ºC < TJ < 125 ºC (T-grade).
Voltage Source
VS
• Intended to power feedback components and/or auxiliary circuits.
• 9 V, 5mA regulated voltage source
• With > 5% output load, VS ripple typically 100mV
Signal Type
State
Attribute
VS Voltage
Regular
VS Available Current
Operation
Analog Output
VS Voltage Ripple
Transition
VS Capacitance (External)
VS Fault Response Time
Reference Enable
RE
• RE signals succesfull startup and powertrain ready to operate
• Intended to power and enable the feedback circuit reference
• 3.3V, 8mA regulated voltage source
Signal Type
State
Attribute
RE Voltage
Regular
RE Available Current
Operation
RE Regulation
Analog Output
Transition
RE Voltage Ripple
PC to RE Delay
RE Capacitance (External)
VS to RE Delay
Control Node
PR
• Modulator control node input
• 0.5mA constant current sink when externally driven
• 0.79V, up to 2mA voltage source when externally pulled low
Signal Type
State
Attribute
PR Voltage Active Range
PR Source Current
Regular
Analog Input
Operation
PR Sink Current
PR Resistance to SG
Symbol
VVS
IVS
VVS_PP
CVS_EXT
T FR_VS
Conditions / Notes
Iout = 0A, Cvs_ext=0. Maximum specification
includes powertrain operation in burst mode.
Symbol
VPR
IPR
IPR_Low
RPR
Conditions / Notes
Typ
9.00
100
From fault recognition to VS = 1.5 V
Symbol
VRE
IRE
%RE
VRE_PP
T PC_RE
CRE_EXT
T VS_RE
Min
8.55
5
Max
9.45
Unit
V
mA
400
mV
0.04
F
s
Max
3.6
Unit
V
30
Min
3.0
Typ
3.3
8.0
2.5
100
100
across load and temperature
in burst mode
Fault detected
0.1
VS = 8.1 V to RE high, VIN > VIN_UVLO-
Conditions / Notes
1
Min
0.79
Typ
Max
7.40
2
Unit
V
mA
250
500
93.3
750
A
kΩ
VPR  0.79V
VPR  0.79V
Current Feedback
IF
• A voltage proportional to the PRM output current must be supplied externally to the IF pin in order for the device to properly protect overcurrent events and to enable output current limit (clamp)
• Overcurrent protection trip will cause instantaneous powertrain disable, latched after TBLNK
Signal Type
State
Conditions / Notes
Min
Typ
Max
Attribute
Symbol
VIN = 45 V; TJ = 25 °C
VIF_IL
Current Limit (clamp) Threshold
1.90
2.00
2.10
Not Production Tested; Guaranted by Design;
VIF_OC
Regular
2.58
2.69
2.80
Overcurrent Protection Threshold
Analog Input
TJ = 25 °C
Operation
RIF
IF Input Impedance
2.13
2.11
2.15
BWIL
Current Limit Bandwidth
2
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%
mV
s
F
ms
Unit
V
kΩ
kHz
PRM48Bx480y600A00
Temperature Monitor
TM
• The TM pin monitors the internal temperature of the PRM analog control IC to within ±5 °C.
• Room temperature setpoint is ~ 3 V, approximate gain is 10 mV/°C.
• "Power Good" flag to verify that the PRM is operating
Signal Type
State
Attribute
Symbol
VTM
TM Voltage
VTM_AMB
TM Voltage reference
Regular
Analog Output
VVS_PP
TM Voltage Ripple
Operation
ITM
TM Available Current
ATM
TM Gain
Digital Output [Fault Flag]
Fault or Standby TM Disabled Current
Signal Ground
SG
• All control signals must be referenced to this pin, with the exception of VC
• SG is internally connected to -IN and -OUT
Signal Type
State
Attribute
Analog Input / Output
Any
Maximum Allowable Current
VTM Control
VC
• Used to synchronize start up of downstream VTM converter.
• 14V nominal, 10ms voltage pulse
• If not used, must be resistively terminated to -OUT
Signal Type
State
Attribute
VC Voltage
VC Current Limit
Startup
Analog Output
VC duration
VC Slew Rate
ITM_DIS
Conditions / Notes
Full temperature range
TJ = 27 °C
Min
2.12
2.94
powertrain in burst mode
Typ
3.00
200
Max
4.04
3.06
100
DC state with TM Voltage +/- 0.5V. This is a high
impedance state.
Unit
V
V
mV
A
10
mV/°C
0.0
mA
Symbol
ISG
Conditions / Notes
Min
-100
Typ
Max
100
Unit
mA
Symbol
VVC
Conditions / Notes
RVC = 68
VC = 14 V, VIN > 20 V
Min
13
200
7
Typ
Max
500
10
20
16
Unit
V
mA
ms
V/s
IVC
TVC
dVC/dt
RVC = 1k
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4.0 FUNCTIONAL BLOCK DIAGRAM
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5.0 HIGH LEVEL FUNCTIONAL STATE DIAGRAM
Conditions that cause state transitions are shown along arrows. Sub-sequence activities listed inside the state bubbles.
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6.0 TIMING DIAGRAMS
Module Inputs are shown in blue; Module Outputs are shown in brown; Timing diagrams assumes the following:
 Single PRM (no array)
 VS powers error amplifier
 RE powers voltage reference and output current transducer
 IOUT is sensed, scaled, and fed back to IF pin such that IF = 2.00 V at full load
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7.0 APPLICATIONS CHARACTERISTICS
The following figures present typical performance at TC = 25ºC, unless otherwise noted. See associated figures for
general trend data.
No Load Power Dissipation vs. Line
Module Enabled, Nominal VOUT
Power Dissipation vs. Line
Module Disabled, PC=Low
0.4
Power Dissipation [W]
3
2
1
0.3
0.2
0.1
0
0
38
40
42
44
46
48
50
52
38
54
40
42
44
-40 ºC
T CASE:
25 ºC
100 ºC
48
50
52
54
25 ºC
100 ºC
Figure 2 - No load power dissipation vs. VIN, module
disabled
Efficiency & Power Dissipation
VOUT = 20 V
TCASE = -40 ºC
Efficiency & Power Dissipation
VOUT = 48 V
TCASE = -40 ºC
32
98
32
96
28
97
28
94
24
92
20
90
16
88
12
86
8
84
4
82
0
0
1
2
3
4
5
6
7
8
9
10 11 12
Efficiency [%]
98
Power Dissipation [W]
Efficiency [%]
-40 ºC
T CASE:
Figure 1 - No load power dissipation vs. VIN, module
enabled
24
96
20
95
16
94
12
93
8
92
4
91
13
0
0
1
2
Load Current [A]
38
VIN:
45
55
38
3
4
5
6
7
8
9
10 11 12
13
Load Current [A]
45
55
Figure 3 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=20V, TCASE=-40ºC
46
Input Voltage [V]
Input Voltage [V]
Power Dissipation [W]
Power Dissipation [W]
4
VIN:
38
45
55
38
45
55
Figure 4 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=48V, TCASE=-40ºC
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32
98
28
96
28
94
24
24
95
94
20
93
16
92
12
91
8
90
89
4
88
0
1
2
3
4
5
6
7
8
9
10
11
92
20
90
16
88
12
86
84
8
82
4
80
12
0
0
1
2
3
4
Load Current [A]
45
55
45
55
Efficiency [%]
97
96
94
93
6
7
8
9
10 11 12
55
20
94
16
93
12
92
8
91
4
0
0
13
1
2
3
4
45
55
94
24
92
20
90
16
88
12
86
8
84
4
82
0
2
3
4
5
6
7
8
45
55
9
10 11 12
8
9
10
11
12
38
45
55
38
45
55
32
28
97
24
96
20
16
95
12
94
8
93
4
92
13
0
0
1
2
3
4
5
6
7
8
9
10 11 12
13
Load Current [A]
45
55
Figure 9 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=20V, TCASE=100ºC
7
Efficiency & Power Dissipation
VOUT = 48 V
T CASE = 100 ºC
Load Current [A]
38
VIN:
6
Figure 8 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=55V, TCASE=25ºC
98
Power Dissipation [W]
Efficiency [%]
28
38
VIN:
32
96
1
5
Load Current [A]
38
Efficiency & Power Dissipation
VOUT = 20 V
T CASE = 100 ºC
0
32
90
Figure 7 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=48V, TCASE=25ºC
98
55
24
Efficiency [%]
45
45
Efficiency & Power Dissipation
VOUT = 55 V
T CASE = 25 ºC
Load Current [A]
38
VIN:
38
Figure 6 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=20V, TCASE=25ºC
0
5
55
95
4
92
45
28
8
4
13
96
12
3
10 11 12
97
16
95
2
9
98
20
1
8
28
24
0
7
32
Power Dissipation [W]
Efficiency & Power Dissipation
VOUT = 48 V
T CASE = 25 ºC
38
VIN:
Figure 5 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=55V, TCASE=-40ºC
98
6
Load Current [A]
38
Efficiency [%]
38
VIN:
5
Power Dissipation [W]
0
32
Power Dissipation [W]
Efficiency [%]
96
Efficiency & Power Dissipation
VOUT = 20 V
T CASE = 25 ºC
Power Dissipation [W]
97
Efficiency [%]
98
Power Dissipation [W]
Efficiency & Power Dissipation
VOUT = 55 V
TCASE = -40 ºC
VIN:
38
45
55
38
45
55
Figure 10 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=48V, TCASE=100ºC
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6.5
32
28
96
24
95
20
94
16
93
12
92
8
91
4
90
6
1
2
3
4
5
6
7
8
9
10
11
6.09
4.5
4.52
4.49
4
-40
12
-20
0
Load Current [A]
38
VIN:
45
55
38
45
20
40
Temperature [ºC]
60
4.21
80
6.25
IOUT:
55
Figure 11 – Total efficiency and power dissipation vs. VIN
and IOUT, VOUT=55V, TCASE=100ºC
5.82
5
0
0
6.13
5.5
VPR [V]
97
Power Dissipation [W]
98
Efficiency [%]
VPR vs. Case Temperature
VIN = 45 V; VOUT = 48 V
Efficiency & Power Dissipation
VOUT = 55 V
T CASE = 100 ºC
100
12.5
Figure 12 – Typical control node voltage vs. TCASE, IOUT;
VIN=45V, VOUT=48V
Powertrain switching frequency and periodic
output charge vs. input voltage, Full load
fSW [kHz]
36
fsw
1000
32
975
28
950
24
925
20
900
16
C
875
12
850
8
825
4
800
Total output charge
per
switching cycle [ C]
1025
0
38
40
42
44
46
48
50
52
54
56
Input Voltage [V]
Figure 13 – Typical output voltage ripple waveform,
TCASE= 30ºC, VIN=45V, VOUT=48V, IOUT=12.5A, no external
capacitance.
55
VOUT
24
925
20
900
16
875
12
C
8
825
4
800
0
38
40
42
44
46
48
50
52
54
55
20
48
700
12
600
10
500
8
400
6
300
4
200
2
100
0
0
5
56
20
48
Figure 15 – Powertrain switching frequency and periodic
input charge vs. VIN, VOUT; IOUT=12.5A
10
15
20
25
30
35
40
45
50
Output Voltage [V]
55
48
14
Input Voltage [V]
VOUT:
20
Current
Power
Figure 16 – DC Output Safe Operating Area
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60
Output Power [W]
950
Output Current [A]
28
Total input charge per
switching cycle [ C]
fSW [kHz]
32
975
850
55
DC Safe Operating Area
36
f sw
1000
48
Figure 14 – Powertrain switching frequency and periodic
output charge vs. VIN, VOUT; IOUT=12.5A
Powertrain switching frequency and periodic
input charge vs. input voltage, Full load
1025
20
PRM48Bx480y600A00
14
12
gpr
280
18
24
240
16
20
200
160
6
120
4
80
req_out
2
0
0
1
2
3
4
5
6
7
8
9
10
11
gpr
14
gPR [dBS]
8
req_out [ ]
gPR [dBS]
10
DC modulator gain and powertrain equivalent
output resistance vs. output current, VOUT = 20V
12
12
10
40
8
0
6
8
req_out
45
55
1
2
3
4
45
55
12
90
60
6
45
4
30
req_out
req_out [ ]
gPR [dBS]
75
gpr
15
0
2
3
4
5
6
7
8
9
45
38
45
55
38
45
55
8
6
4
2
0
5
10
15
20
25
30
35
40
45
50
55
Voltage [V]
38
45
55
Figure 20 – Effective internal input
capacitance vs. voltage – ceramic type
and
output
Powertrain equivalent input resistance
vs. output current, VOUT = 55V
12
Typical min
10
Nominal
Typical max
8
req_in [ ]
Output Power [W]
55
10
Output Power vs. VPR
VIN = 45V, VOUT = 48V, T C=25ºC
6
4
2
0
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6.0
6.5
0
7.0
1
2
3
4
5
6
7
8
9
10
11
Output Current [A]
PR Voltage [V]
VIN:
Figure 21 – Output Power vs. VPR; VIN=45V, VOUT=48V,
TCASE=25ºC
10 11 12 13
12
10 11 12 13
Figure 19 – Powertrain characteristics vs. IOUT;
Resistive load, VOUT=48V, various VIN
600
560
520
480
440
400
360
320
280
240
200
160
120
80
40
0
9
14
Output Current [A]
VIN:
8
0
0
1
7
Effective internal capacitance vs. applied
voltage, Input (CIN_INT) and Output (COUT_INT)
Effective capacitance [ F]
105
0
6
Figure 18 – Powertrain characteristics vs. IOUT;
Resistive load, VOUT=20V, various VIN
14
2
38
VIN:
DC modulator gain and powertrain equivalent
output resistance vs. output current, VOUT = 48V
8
5
Output Current [A]
38
Figure 17 – Powertrain characteristics vs. IOUT;
Resistive load, VOUT=55V, various VIN
10
4
0
0
12
Output Current [A]
38
VIN:
16
req_out [ ]
DC modulator gain and powertrain equivalent
output resistance vs. output current, VOUT = 55V
38
45
55
Figure 22 – Magnitude of powertrain dynamic input
impedance vs. VIN, IOUT; VOUT=55V
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Powertrain equivalent input resistance
vs. output current, VOUT = 20V
Powertrain equivalent input resistance
vs. output current, VOUT = 48V
60
14
50
12
10
req_in [ ]
req_in [ ]
40
30
20
8
6
4
10
2
0
0
0
1
2
3
4
5
6
7
8
9
10
11
12
13
0
1
2
3
Output Current [A]
VIN:
38
45
4
5
6
7
8
9
10
11
12
13
Output Current [A]
55
VIN:
Figure 23 – Magnitude of powertrain dynamic input
impedance vs. VIN, IOUT; VOUT=20V
38
45
55
Figure 24 – Magnitude of powertrain dynamic input
impedance vs. VIN, IOUT; VOUT=48V
8.0 GENERAL CHARACTERISTICS
GENERAL CHARACTERISTICS
Specifications apply over all line and load conditions, and trim from 20V to 55V, unless otherwise noted;
Boldface specifications apply over the temperature range of -40 ºC < TJ < 125 ºC (T-Grade); All Other specifications are at TJ = 25 ºC unless otherwise noted.
Conditions / Notes
Attribute
Symbol
Min
Typ
Max
Unit
MECHANICAL
Length
L
32.3 / [1.27] 32.5 / [1.28] 32.8 / [1.29] mm / [in]
Width
W
21.8 / [0.86] 22.0 / [0.87] 22.3 / [0.88] mm / [in]
Height
H
Volume
Vol
Weight
W
6.60 / [0.26] 6.73 / [0.26] 6.86 / [0.27] mm / [in]
4.81 / [0.29]
cm 3 / [in3]
No Heatsink
13.6
Nickel
Palladium
Gold
Lead Finish
THERMAL
Operating and Storage Junction
Temperature
Operating Case Temperature
Thermal Capacity
TJ
TC
Storage Temperature
2.03
0.15
0.051
-40
-40
125
100
10
ASSEMBLY
Peak Compressive Force Applied to
Case (Z-axis)
ESD Rating
g
0.51
0.02
0.003
Supported by J-Lead only
TST
ESDHBM
Human Body Model, "JEDEC JESD 22-A114C.01"
-40
1000
ESDCDM
Charged Device Model, "JEDEC JESD 22-C101D"
400
m
ºC
ºC
Ws/ºC
6
5.41
lbs / in 2
lbs
125
ºC
V
SOLDERING
Peak Temperature During Reflow
MSL 4 (Datecode 1528 and later)
Maximum Time Above [217] ºC
Peak Heating Rate During Reflow
Peak Cooling Rate Post Reflow
1.5
2.5
245
ºC
150
2
3
s
ºC / s
ºC / s
SAFETY
MTBF
Agency Approvals / Standards
Telcordia Issue 2 - Method I Case 1; Ground Benign, Controlled
MIL-HDBK-217Plus Parts Count - 25C Ground Benign, Stationary, Indoors / Computer Profile
C TUVUS
CE Mark
ROHS 6 of 6
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6.36
MHrs
MHrs
PRM48Bx480y600A00
9.0 PRODUCT OUTLINE DRAWING AND RECOMMENDED PCB FOOTPRINT
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10.0
PRODUCT DETAILS AND DESIGN
GUIDELINES
minimum load. Figure 25 shows a reference AC smallsignal model.
10.1
Current feedback (IF) is the input for the module output
overcurrent protection and current limit features (see
functional block diagram in section 4.0). A voltage
proportional to the powertrain output current must be
applied to IF in order for overcurrent protection to
operate properly.
If the IF voltage exceeds the IF pin’s overcurrent
protection threshold, the powertrain will stop switching. If
the IF voltage falls below the overcurrent protection
threshold within TBLANK time, then the powertrain will
immediately resumes switching. Otherwise a fault is
latched.
The current limit threshold for the IF pin is set lower than
the protection threshold. When the IF pin average
voltage exceeds the current limit threshold, an internal
integrator will activate a clamp amplifier which overrides
the modulator input maximum level. This causes the
powertrain to maintain a constant output current.
The bandwidth of this current limit integrator is
significantly slower than that of the PR control node
input. Therefore this current limit can not be used in lieu
of properly compensating the (external) PR control loop
to avoid exceeding maximum current or power ratings
for the device.
If the IF pin is not driven, it must be resistively
terminated to SG.
A 1k resistor to SG is
recommended in this case.
Control pins description and characteristics
Control node (PR) is the input to the control node which
determines the powertrain timing and ultimately the
module output power (Figure 21). An internal 0.5mA
current sink is always active. The bi-directional buffer
between PR and the control node has two states. In
normal operation, PR will be above the 0.79V switching
threshold, and will drive the control node through the
buffer. An internal 7.4V clamp determines the maximum
output power that can be requested of the modulator.
When PR falls below 0.79 V, the converter will stop
switching. An internal circuit clamps the modulator input
control node to 0.79 V, and a buffer will source up to 2.5
mA out of the pin at that clamp level. For this reason, the
output impedance of the amplifier driving PR must be
taken into account. A rail-to-rail operational amplifier
with low output impedance is always recommended.
The powertrain small signal (plant) response consists of
a single pole determined by the load resistance, the
powertrain equivalent output resistance, and the total
output capacitance (internal and external to the module).
Both the modulator gain and the equivalent output
resistance vary as a function of line, load and output
voltage, as shown in Figures 17, 18 and 19. As the load
increases, the powertrain pole moves to higher
frequency. As a result, the closed loop crossover
frequency will be the highest at full load and lowest at
Figure 25 – PRM48BF480T600A00 AC small signal model
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VTM Control (VC) pin supplies an initial VCC voltage to
downstream VTMs, enabling them and synchronizing
their startup with the PRM. The VCC voltage is a pulse,
typically 10ms duration at 14V.
If VC is not loaded by a VTM, it must be terminated with
a 1kresistor to –VOut.
Primary Control (PC) is both an input and an output. It
can provide the following features:
• Delayed start: upon application of voltage (>UVLO) to
the module power input and after TOFF, the PC pin will
source a constant 90μA current.
• Output disable: PC may be pulled down externally in
order to disable the module. Pull down resistance should
be less than 300 Ω to SG.
• Fault detection flag: The PC 5 V voltage source is
internally turned off when a fault condition is latched.
Note that aside from the Short Circuit fault condition, PC
does not have significant current sinking capability.
Therefore in the case of an array of PRMs with
interconnected PC pins, PC does not in general reflect
the fault state of all PRMs. The common PC line will not
disable neighboring modules when a fault is detected
except for a latched Output Short Circuit fault.
Conversely any unit in the array latching a Short Circuit
fault will disable the array for TSCR.
Temperature Monitor (TM) pin outputs a voltage
proportional to the absolute temperature of the converter
analog control IC. It can be used to accomplish the
following functions:
• Monitor the control IC temperature: The gain and
setpoint of TM are such that the temperature, in Kelvin,
of the PRM controller IC is equal to the voltage on the
TM pin scaled by 100. (i.e. 3.0 V = 300 K = 27ºC).
• Closed loop thermal management at the system level
(e.g. variable speed fans or coolant flow)
• Fault detection flag: The TM voltage source is turned
off as soon as a fault is detected. For system monitoring
purposes (microcontroller interface) faults are detected
on falling edges of TM.
Reference Enable (RE) pin outputs a regulated 3.3V,
8mA voltage source. It is enabled only after successful
startup of the PRM powertrain (see chapters 5.0 and
6.0.) RE is intended to power the output current
transducer and also the voltage reference for the control
loop. Powering the reference generator with RE helps
provide a controlled startup, since the output voltage of
the system is able to track the reference level as it
comes up.
Voltage Source (VS) pin outputs a gated (e.g. mirrors
PC status), non-isolated, regulated 9V, 5mA voltage
source. It can be used to power external control circuitry;
it always leads RE.
Signal Ground (SG) pin provides a Kelvin connection to
the PRM’s internal signal ground. It should be used as
the reference for PR, TM, IF, and should return all PC,
VS and RE pin currents. In array configurations with
common ground control circuits, a series resistor (~1)
is recommended in order to decouple power and signal
current returns.
10.2
Control
procedure
circuit
requirements
and
design
The PRM48BF480T600A00 is an intelligent powertrain
module designed to fully exploit external output voltage
feedback and current sensing sub-circuits. These two
external circuits are illustrated in Figure 26, which shows
an example of the PRM in a standalone application with
local voltage feedback and high side current sensing.
In general, these circuits include a precision voltage
reference, an operational amplifier which provides
closed loop feedback compensation, and a high side
current sense circuit which includes a shunt and current
sense IC.
The following design procedures refer to the circuit
shown in Figure 26.
10.2.1 Setting the output voltage level
The output voltage setpoint is a function of the voltage
reference and the output voltage sense ratio. With
reference to Fig. 26, R1 and R2 form the output voltage
sensing divider which provides the scaled output voltage
to the negative input of the error amplifier; a dedicated
reference IC provides the reference voltage to the
positive input of the error amplifier. Under normal
operation, the error amplifier will keep the voltages at the
inverting and non-inverting inputs equal, and therefore
the output voltage is defined by:
VOUT  Vref 
R1  R 2
R2
Note that the component R1 will also factor into the
compensation as described in a later section.
It is important to apply proper slew rate to the reference
voltage rise when the control loop is initially enabled.
The recommended range for reference rise time is 1 ms
to 9 ms. The lower rise time limit will ensure optimized
modulator timing performance during startup, and to
allow the current limit feature (through IF pin) to fully
protect the device during power-up. The upper rise time
limit is needed to guarantee a sufficient factorized bus
voltage is provided to any downstream VTM input before
the end of the VC pulse.
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10.2.2 Setting the output current limit and overcurrent
protection level
The current limit and overcurrent protection set points
are linked, and scale together against the current sense
shunt, and the gain of the current sense amplifier. The
output of the current sense IC provides the IF voltage
which has VIF_IL and VIF_OC thresholds for the two
functions respectively. The set points are therefore
defined by:
I IL 
VIF _ IL

RS  GCS
and
I OC 
Powertrain equivalent resistance rEQ: See
Figures 17, 18, 19
 Internal output capacitance: see Figure 20
 External output capacitance value
In the case of ceramic capacitors, the ESR can be
considered low enough to push the associated zero well
above the frequency of interest. Applications with high
ESR capacitor may require a different type of
compensation, or cascade control.
The system poles and zeros of the closed loop can then
be defined as follows:

Powertrain pole, assuming the external
capacitor ESR can be neglected:
RCOUT _ EXT 
VIF _ OC
RS  GCS

rEQ _ OUT  RLOAD
rEQ _ OUT  RLOAD
Main pole frequency:
where GCS is the gain of the current sense amplifier.
10.2.3 Control loop compensation requirements
In order to properly compensate the control loop, all
components which contribute to the closed loop
frequency response should be identified and understood.
Figure 25 shows the AC small signal model for the
module. Modulator DC gain gPR and powertrain
equivalent resistance rEQ_OUT are shown.
These
modeling parameters will support a design cut-off
frequency up to 50kHz.
Standard Bode analysis should be used for calculating
the error amplifier compensation and analyzing the
closed loop stability. The recommended stability criteria
are as follows:
1) Phase Margin > 45º : for the closed loop response,
the phase should be greater than 45º where the gain
crosses 0dB.
2) Gain Margin > 10dB : The closed loop gain should be
lower than -10dB where the phase crosses 0º.
3) Gain Slope = -20dB/decade : The closed loop gain
should have a slope of -20dB/decade at the crossover
frequency.
The compensation characteristics must be selected to
meet these stability criteria. Refer to Figure 27 for a
local sense, voltage-mode control example based on the
configuration in Figure 26. In this example, it is assumed
that the maximum crossover frequency (FCMAX) has been
selected to occur between B and C. Type-2
compensation (Curve IJKL) is sufficient in this case.
The following data must be gathered in order to proceed:
 Modulator Gain gPR: See Figures 17, 18, 19
FP 

1
2 π
rEQ _ OUT  RLOAD
rEQ _ OUT  RLOAD
Compensation Mid-Band Gain:
G MB  20 log

R3
R1
[1]
Compensation Zero:
FZ1 

 COUT _ INT  COUT _ EXT 
1
2 π R 3  C1
[2]
Compensation Pole:
FP 2 
1
R C C
2 π 3 1 2
C1  C2
and for FP2>>FZ1 (C1 + C2 ≈ C1):
FP 2 
1
2  R3  C2
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PRM48Bx480y600A00
10.2.4 Midband Gain Design (R1,R3):
10.2.5 Compensation Zero Design (C1):
With reference to Figure 27: curve ABC is the:
 minimum output voltage in the application
 maximum input voltage expected in the
application
 maximum load
PRM open loop response, and is where the maximum
crossover frequency occurs. In order for the maximum
crossover frequency to occur at the design choice FCMAX,
the compensation gain must be equal and opposite of
the powertrain gain at this frequency. For stability
purposes, the compensation should be in the Mid-band
(J-K) at the crossover. Using Equation [1], the mid-band
gain can be selected appropriately.
With reference to Figure 27: curve EFG is the:
 maximum output voltage in the application
 minimum input voltage expected in the
application
 minimum load in the application
PRM open loop response, and is where the minimum
crossover frequency FCMIN occurs. Based on stability
criteria, the compensation must be in the mid-band at
the minimum crossover frequency, therefore FCMIN will
occur where EFG is equal and opposite of GMB. C1 can
be selected using Equation [2] so that FZ1 occurs prior to
FCMIN.
C2
C1
R3
+
Vref
R2
R1
F1
+IN
VS
PR
RS
+OUT
PRM
VIZ0051
CIN_EXT
CIN_INT
-IN
Vref
IF RE
COUT_EXT
COUT_INT
SG
-OUT
I sense
IC
Vref IC
Figure 26 – Control circuit example
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Open Loop Gain vs. Frequency
80
60
Gain (dB)
40
20
10MHz GBW
I
Compensation Gain
F
E
PRM Open Loop Min Load
B
A
PRM Open Loop Max Load
J
K
L
FCMIN
0
FCMAX
-20
C
G
-40
Frequency (Hz)
Figure 27 – reference asymptotic Bode plot for the considered system
10.2.6
High Frequency Pole Design (C2):
based on the ratio of the “kick” to “droop” (as defined in
Fig. 28).
Using Equation [3], C2 should be selected so that FP2 is
at least one decade above FCMAX and prior to the gain
bandwidth product of the operational amplifier (10MHz
for this example). For applications with a higher desired
crossover frequency the use of a high gain bandwidth
product amplifier may be necessary to ensure that the
real pole can be set at least one decade above the
maximum crossover frequency.
10.2.7 Verifying Stability:
The preferred method for verifying stability is to use a
network analyzer, measuring the closed loop response
across various lines and load conditions.
In the absence of a network analyzer, a load step
transient response can be used in order to estimate
stability.
Figure 28 illustrates an example of a load step response.
Equation [4] can be used to predict the phase margin
Figure 28 – load step response example and “droop”
vs. “kick” definition
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2
 k
 ln 
 d
 m  100
2
 k
2
 ln   
 d
[4]
Figure 20 provides the effective internal capacitance of
the module. A conservative estimate of input and output
peak-peak voltage ripple at nominal line and trim is
provided by equation [5]:
QTOT 
V 
10.3
Burst Mode Operation:
At light loads, the PRM will operate in a burst mode due
to minimum timing constraints.
An example burst
operation waveform is illustrated in Figure 29.
For very light loads, and also for higher input voltages,
the minimum time power switching cycle from the
powertrain will exceed the power required by the load. In
this case the external error amplifier will periodically
drive PR below the switching threshold in order to
maintain regulation. Switching will cease momentarily
until the error amplifier once again drives PR voltage
above the threshold.
CINT
I FL  0.4
f SW
 C EXT
[5]
QTOT is the total input (Fig. 15) or output (Fig. 14) charge
per switching cycle at full load, while CINT is the module
internal effective capacitance at the considered voltage
(Fig. 20) and CEXT is the external effective capacitance at
the considered voltage.
10.5
Input filter stability
The PRM can provide very high dynamic transients. It is
therefore very important to verify that the voltage supply
source as well as the interconnecting line are stable and
do not oscillate. For this purpose, the converter dynamic
input impedance magnitude
rEQ _ IN
is provided in
Figures 22, 23, 24. It is recommended to provide
adequate design margin with respect to the stability
conditions illustrated in 10.5.1 and 10.5.2 .
10.5.1 Inductive source and local, external input
decoupling capacitance with negligible ESR (i.e.:
ceramic type)
Figure 29 – light load burst mode of operation
Note that during the bursts of switching, the powertrain
frequency is constant, but the number of pulses as well
as the time between bursts is variable. The variability
depends on many factors including input voltage, output
voltages, load impedance, and external error amplifier
output impedance.
In burst mode, the gain of the PR input to the plant which
is modeled in the previous sections is time varying.
Therefore the small signal analysis can not be directly
applied to burst mode operation.
10.4
Input and Output filter design
Figures 14 and 15 provide the total input and output
charge per cycle, as well as switching frequency, of the
PRM at full load under various input and output voltages
conditions.
The voltage source impedance can be modeled as a
series RlineLline circuit. The high performance ceramic
decoupling capacitors will not significantly damp the
network because of their low ESR; therefore in order to
guarantee stability the following conditions must be
verified:
Rline 
(C IN _ INT
Rline  rEQ _ IN
Lline
 C IN _ EXT )  rEQ _ IN
[6]
[7]
It is critical that the line source impedance be at least an
octave lower than the converter’s dynamic input
resistance, [7]. However, Rline cannot be made arbitrarily
low otherwise equation [6] is violated and the system will
show instability, due to under-damped RLC input
network.
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10.5.2 Inductive source and local, external input
decoupling capacitance with significant RCIN_EXT ESR
(i.e.: electrolytic type)
In order to simplify the analysis in this case, the voltage
source impedance can be modeled as a simple inductor
Lline. Notice that, the high performance ceramic
capacitors CIN_INT within the PRM should be included in
the external electrolytic capacitance value for this
purpose. The stability criteria will be

array. Imbalances in sharing are not only due to
current sharing accuracy specifications, but also
temperature differences among PRMs, Vin
variations, and error terms in the buffering of the
error amplifier output to the PR pins.
Control loop compensation procedures above
will hold for an array, in general, although many
parameters must be scaled against the number
of PRMs in the system.
Please contact Vicor Applications for assistance.
rEQ _ IN  RC IN _ EXT
[8]
Lline
 rEQ _ IN
C IN _ EXT  RC IN _ EXT
[9]
Equation [9] shows that if the aggregate ESR is too
small – for example by using very high quality input
capacitors (CIN_EXT) – the system will be under-damped
and may even become destabilized. Again, an octave of
design margin in satisfying [8] should be considered the
minimum.
10.6
Arrays
Up to ten PRMs of the same type may be placed in
parallel to expand the power capacity of the system. The
following high-level guidelines must be followed in order
for the resultant system to start up and operate properly,
and to avoid overstress or exceeding any absolute
maximum ratings.
 –IN pins of all PRMs must be connected
together. Both inductance and resistance from
the common power source to each PRM should
be minimized, and matched.
 Input voltage to all PRMs must be the same.
Independent fuses for each PRM are
recommended.
 PC pins must be connected together for
synchronization and proper fault response.
 Reference supply to the control loop voltage
reference and current sense circuitry must be
enabled when all modules’ RE pins have
reached their operational voltage levels.
 There must be one single external voltage
control loop. The control loop must drive each
PR pin relative to each modules’ SG pin, and the
local PR voltage must be the same across all
modules.
 Each PRM must have its own local current shunt
and current sense circuitry to drive it’s IF pin.
 The number of PRMs required to achieve a
given array capacity must consider all sources of
mismatch to avoid overstress of any PRM in the
10.7
Input Fuse Recommendations
A fuse should be incorporated at the input to each PRM,
in series with the +IN pin. A 15A or smaller input fuse
(Littelfuse® NANO2® 451/453 Series, or equivalent) is
required to safety agency conditions of acceptability.
Always ascertain and observe the safety, regulatory, or
other agency specifications that apply to your specific
application.
10.8
Layout considerations
Application Note AN:005 details board layout using
VI Chip components. Additional consideration must be
given to the external control circuit components.
The current sense shunt signal voltage is highly
sensitive to noise. As such, current sensing circuitry
should be located close to the shunt to minimize the
length of the sense signals. A Kelvined connection at the
shunt is recommended for best results.
The control signal from a remote voltage sense circuit to
the PRM should be shielded. Avoid routing this, or other
control signals directly underneath the PRM, if possible.
Components that tie directly to the PRM should be
located close to their respective pins. It is also critical
that all control components be referenced to SG, and
that SG not be tied to any other ground in the system,
including –IN or –OUT of the PRM.
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Warranty
Vicor products are guaranteed for two years from date of shipment against defects in material or workmanship when
in normal use and service. This warranty does not extend to products subjected to misuse, accident, or improper
application or maintenance. Vicor shall not be liable for collateral or consequential damage. This warranty is extended
to the original purchaser only.
EXCEPT FOR THE FOREGOING EXPRESS WARRANTY, VICOR MAKES NO WARRANTY, EXPRESS OR
IMPLIED, INCLUDING, BUT NOT LIMITED TO, THE WARRANTY OF MERCHANTABILITY OR FITNESS FOR A
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warranty, the buyer must contact Vicor to obtain a Return Material Authorization (RMA) number and shipping
instructions. Products returned without prior authorization will be returned to the buyer. The buyer will pay all charges
incurred in returning the product to the factory. Vicor will pay all reshipment charges if the product was defective
within the terms of this warranty.
Information published by Vicor has been carefully checked and is believed to be accurate; however, no responsibility
is assumed for inaccuracies. Vicor reserves the right to make changes to any products without further notice to
improve reliability, function, or design. Vicor does not assume any liability arising out of the application or use of any
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policy does not recommend the use of its components in life support applications wherein a failure or malfunction may
directly threaten life or injury. Per Vicor Terms and Conditions of Sale, the user of Vicor components in life support
applications assumes all risks of such use and indemnifies Vicor against all damages.
Vicor’s comprehensive line of power solutions includes high density AC-DC and DC-DC
modules and accessory components, fully configurable AC-DC and DC-DC power
supplies, and complete custom power systems.
Information furnished by Vicor is believed to be accurate and reliable. However, no responsibility is assumed by Vicor
for its use. Vicor components are not designed to be used in applications, such as life support systems, wherein a
failure or malfunction could result in injury or death. All sales are subject to Vicor’s Terms and Conditions of Sale,
which are available upon request.
Specifications are subject to change without notice.
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Vicor and its subsidiaries own Intellectual Property (including issued U.S. and Foreign Patents and pending patent
applications) relating to the products described in this data sheet. Interested parties should contact Vicor's Intellectual
Property Department.
The products described on this data sheet are protected by the following U.S. Patents Numbers:
5,945,130; 6,403,009; 6,710,257; 6,911,848; 6,930,893; 6,934,166; 6,940,013; 6,969,909; 7,038,917;
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Tel: 800-735-6200
Fax: 978-475-6715
email
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