Drive Piezoelectric Actuators With Fast, High

Sam Robinson
••• [email protected]
Drive Piezoelectric Actuators
With Fast, High-Power Op Amps
Think you can’t drive your actuator at 80 kHz with
300-V p-p signals using MOSFET IC op amps? Then
try out this bridge configuration.
In the last 15 years, high-speed piezoelectric actuators have become less
expensive to manufacture. Consequently, they find themselves as a
favorite design choice for a growing
number of applications.
Piezoelectric actuators made their initial entry in medical devices, including
surgical tools and ultrasonic testing in
the late 1980s. It was for good reason.
Piezoelectric actuators are the fastestresponding positioning element available with microsecond time constants.
They also can produce motions in subnanometer increments. So, it’s no surprise to see a dramatic rise in the number of companies designing products
that employ these devices.
Piezoelectric actuators require highvoltage drivers that can deliver hun-
+175 V
dreds of volts, peak-to-peak. In addition, because a typical actuator looks
virtually like a pure capacitance to the
driving amplifier, nearly all power dissipation becomes the burden of the driving amplifier.
A quick look through the abundance of
high-speed, small-signal op amps will disclose many amplifiers with bandwidths in
the hundreds of megahertz. But to combine that speed with more than 12 V, as
you must if you’re planning to drive a highspeed piezoelectric device, the choices of
suitable amplifiers shrink pretty fast.
Moreover, choosing a design based upon
a MOSFET monolithic amplifier has compelling benefits.
The piezoelectric actuator circuit in
the example presented here requires a
300-V p-p source at 80 kHz. It will drive
2.2 pF 8.2k
V1 + 15 V p-p
80 kHz
1 nF
+175 V
–5 V
+175 V
2.2 pF 8.2k
–5 V
–5 V
1. This circuit shows a bridge-connected configuration. Two PA78s drive the
piezoelectric actuator. Asymmetric power supplies at +175 V and –5 V power
the PA78s.
our actuator, which can be represented
by a 1-nF capacitance in series with a 1 resistance (Fig. 1).
A LOOK AT THE OPTIONS • We evaluated several alternatives for our application
before deciding on the best approach to
drive a piezoelectric actuator:
• Single amplifier: Cost is the issue with
this option. Because a piezoelectric actuator will require a swing between +150 V
and –150 V, the only device on the
market that meets this requirement at
a frequency of 80 kHz is going to be a
hybrid, which will price out at over
$100 each.
• Small-signal amplifier with level
shifting: A design that begins with a
small-signal amplifier, followed by level shifting, means configuring a
design from scratch with a large number of discretes. In this case, the nonrecurring-engineering effort will be
large, and the design time will be long
and costly.
• High-voltage, high-speed, low-current
MOSFET op-amp IC: The monolithic
PA78, which we chose for this design,
uses a class A/B driver stage to drive
output MOSFETs and an innovative input
stage to achieve very high slew rates—
without the high quiescent currents of
traditional op-amp designs. Also, this
design will require two PA78s. And
because this amplifier is an IC design, it
will cost approximately $15 each. That’s
a considerable savings vis-à-vis the
hybrid at over $100.
As depicted in Figure 1, we’ve configured two PA78s in a bridge circuit.1 In
this configuration, the amplifiers supply
an output-voltage swing twice that of a
2. These output waveforms show the left module output (a), the right module
output (b), and the waveform appearing across the piezoelectric actuator (c).
single op amp. This configuration doubles the slew rate. Any nonlinearities
become symmetrical, reducing secondharmonic distortion compared to a singleamplifier circuit.
The sinusoidal source applies 15 V
p-p at 80 kHz to drive the amplifier
pair, which, in turn, drives the piezoelectric actuator. In our case, we shall
assume that the piezoelectric actuator
will exhibit an equivalent impedance of
a 1-nF capacitor in series with a 1-
resistor, as shown.
A FLOATING LOAD • In our application,
the load is floating. In other words, it’s not
ground-connected at all. When the left
output VOUTA swings from 10 to 160 V (Fig.
2a) and the right output VOUTB descends
from 160 V to 10 V (Fig. 2b), a voltage
swing of 300 V (–150 V to +150 V) develops across the load (Fig. 2c).
The outputs of the two amplifiers are
now out of phase. The overall gain of the
bridge-configured PA78s is +20, so that
300 V p-p is delivered to the piezoelectric actuator, as required. The feedback
circuit comprising resistors R3 and R4
center both of the PA78 modules’ outputs around 85 V. As shown in Figure 1,
a dual-source, asymmetric power supply
delivers +175 V and –5
V to the two amplifier
But it’s the amplifier’s common-mode
input range (CMR) positive and negative
values (specified in the PA78 data sheet)
that play a significant role in governing
the values of +VS and –VS in this asymmetrical sourcing arrangement.
In the case of the PA78, the specified
value of the CMR negative is –VS + 3 V.
This means the input should approach
the negative rail no closer than 3 V.
Therefore, by choosing –VS equal to –5 V,
both VOUTA and VOUTB (having negative
excursions to 10 V) will approach the
negative rail no closer than 15 V. With
the CMR positive, the value is +VS – 2 V.
This means that the most positive-going
excursion of both VOUTA and VOUTB must
stay at least 2 V below +VS.
A second issue with regard to the +VS
rail is the voltage drop at the output
when the modules are delivering peak
current. In this application, the peak current is approximately 75 mA. A graph in
the PA78 data sheet called “Output Voltage Swing” says that if you’re driving
this much current, you will lose approximately 8 V. The sum of the two, 2 V and
8 V, is 10 V, which says the +VS must be
at least 10 V above our maximum voltage swing of 150 V. Choosing a +VS of
HEADROOM • The values of +VS and –VS
must be carefully chosen to ensure sufficient
headroom during the
positive and negative
excursions of both VOUTA
and VOUTB. The output
VOUTA – VOUTB will swing 3. Maximum power dissipation is computed using the
from +150 V to –150 V. equivalent circuit shown here.
175 V means an additional headroom
margin of 15 V.
With any piezoelectric actuator circuit, it’s essential to prevent signals
from inadvertently feeding back to the
amplifier. A piezo transducer can convert mechanical into electrical energy
just as easily as going from electrical to
mechanical. So if something were to
bump the transducer, it may create lots
of energy that would travel backwards
into the output of the amplifier. Of
course, that could be rather destructive.
Yet simply connecting several ultra-fast
MUR160 diodes (CR1 to CR4) from the
output of each amplifier to its corresponding power-supply rails will protect
each amplifier.
impedance of the piezoelectric cartridge
is given by the expression:
= 1+
j2 π 80 × 10 3 1 × 10 −9
= 1 − j1989 ≈ − j1989Ω
assuming that R = 1 , C = 1 nF, and =
80 kHz.
To compute the maximum power per
module, we can devise the equivalent
circuit shown in Figure 3. To do this, the
Figure 1 circuit is split into two parts,
with each part comprising a 2-nF capacitor and a 0.5- resistor, while assuming
a virtual ground denoted by the dotted
line and symbol. Because the real part of
the impedance (1 ) is negligible compared with the total capacitive reactance
of 1989 , we shall neglect it.
In our equivalent circuit, the applied
voltage will be one half the total potential applied to each module:
0.5 ( + VS ) − ( − VS ) =
0.5[(175 V) − ( −5 V)] = 90 V
The circuit for each half will drive half
the capacitive reactance, which is 1989
divided by 2 each, or 994.5 . Determining power dissipation begins with knowing the phase difference between V and I
in the load. This is a simple case
because we’ve modeled our load as a
pure capacitor, so the phase angle is
90°. The formula for determining the
maximum power dissipated when there’s
a reactive load for a phase angle greater
than 40° is given by:3
PD (max) =
2 VS 2
πZ L
where VS is the magnitude of each supply and ZL is the magnitude of the load
PD ( max) =
2(90) 2
= 5.18 W
Because the load is totally reactive, the
5.18 W are dissipated by each PA78 amplifier IC and none by the load. We can then
go on to select a heatsink and confirm that
the maximum allowable junction temperature of each PA78 won’t be exceeded.
heatsink has been selected for mounting
each PA78 IC. The thermal resistance of
each is 5.3°C/W and, as we have determined, the dissipation of each amplifier
will be 5.18 W.
We must confirm that the junction
temperatures of the MOSFET devices
in the PA78 won’t exceed a safe value. The familiar thermal resistance
equation is:
Pθ JA = TJ − TA
P(θ JC + θ CA ) = TJ − TA
We can modify the above equation by
substituting the thermal resistance of
the heatsink HS for CA:
P(θ JC + θ HS ) = TJ − TA
We want to solve this for TJ to confirm
that we won’t exceed the maximum junction temperature. Rearranging the terms,
we have:
TJ = P(θ JC + θ HS ) + TA
In our case the power per device is
5.18 W and the JC, according to the
PA78 data sheet, is 5.5°C/W. The uHS
for the heatsink is 7.8°C/W, and the rise
in temperature above the ambient is
48.2°C. (For the graphs that show the
heatsink’s thermal resistance
as a function of power, and
the temperature rise at the
interface, go to www.elecdesign.
com and see Drill Deeper 11366.)
Thus, the maximum junction temperature will be:
TJ = P(θ JC + θ HS ) + TA
= 5.18(5.5 + 7.8) + 25°C
= 68.9°C + 25°C
= 93.9°C
Therefore, the actual TJ will never rise
above 93.9°C. This is far below the maximum permissible value of 150°C specified in the PA78 data sheet.
It’s essential when applying high power
to a highly reactive load, such as a piezoelectric actuator, to check the dissipation
and the safe operating area. The former is
discussed in the Application Note “General Operating Conditions,”3 and the latter
is covered in the PA78 data sheet.
In the past, industrial-grade power
amplifiers have traded off bandwidth to
ensure unity-gain stability. Bipolar designs
haven’t always met the linearity requirements of demanding applications4, such
as the piezoelectric actuator design in
this article. But with the availability of a
MOSFET-based architecture in devices,
the possibilities have changed. Now new
standards for bandwidth and linearity can
be created for IC power amplifiers.
ED Online 11302
Sam Robinson is an applications engineer specializing in power analog electronics at Apex Microtechnology Corp.
(, Tucson, Ariz.
He holds a BSEE from the University of
Alabama in Huntsville.
1. Application Note 20, “Bridge Mode
Operation of Power Operational Amplifiers,” Apex Microtechnology Corp.,
2. Application Note 21, Section 3.1,
“Single Supply Operation of Power Operational Amplifiers,” ibid.
3. Application Note 1, Section 7.2, “General Operating Considerations,” ibid.
4. Application Note 17, “Wide Band
Low Distortion Techniques,” ibid.