An Improved 2-Switch Forward Converter Application Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions Generalities About the 1-Switch Forward Converter PROs It is a transformer-isolated buck-derived topology It requires a single transistor, ground referenced Non-pulsating output current reduces rms content in the caps CONs Smaller power capability than a full or half-bridge topology Limited in duty-cycle (duty ratio) excursion because of core reset The drain voltage swings to twice the input voltage or more Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions Transformer Core Reset: Why? Without transformer core reset: X1 Lmag D1 L D2 C R Vin Q1 Q1 0 t ILmag t The current builds up at each switching cycle It brings the core into saturation Transformer Core Reset: Why? With transformer core reset: D1 L X1 Lmag D2 C R Vin Q1 D3 Q1 t ILmag 0 t The current does not build up at each switching cycle Volt-seconds average to zero during each cycle The voltage reverses over Lmag and resets it Core Reset Techniques: How ? Energy is stored in the magnetizing inductor This energy does not participate to the power transfer It needs to be released to avoid flux walk away 3 common standard techniques for the core reset: Tertiary winding RCD clamp 2-switch forward Core Reset Techniques: Tertiary Winding • Reset with the 3rd winding ☺ Duty ratio can be > 50% But Q1 peak voltage can be > 2 • Vin 3rd winding for the transformer D1 L X1 Lmag D2 C Vin Q1 D3 0 3rd winding R Core Reset Techniques: RCD Clamp • Reset with RCD clamp ☺ Duty ratio can be > 50% But Writing equation and simulation are required for checking the correct reset Lower cost than 3rd winding technique D1 L Lmag Cclamp Vin Rclamp X2 D2 Dclamp XFMR1 Q1 RCD clamp 0 C R Core Reset Techniques: 2-switch Forward • Reset with a 2-switch forward ☺ Easy to implement ☺ Q1 peak voltage is equal to Vin But Additional power MOSFET (Q2) + high side driver 2 High voltage, low power diodes (D3 & D4) Q2 X1 D1 L Vin D4 2-switch forward reset D3 Lmag Q1 D2 Note : Q1 & Q2 have same drive command 0 C R 2-Switch Forward: How Does It Works? IL Q2 X1 L D1 t Vin D4 D3 Lmag D2 C R ILmag t Q1 Step 1 Step 2 Step 3 0 Note : Primary controller status • “on time” : Step1 • “off time”: Step 2 + Step 3 Q1 & Q2 D1 D2 D3 & D4 Step 1 ON ON OFF OFF Step 2 OFF OFF ON ON Step 3 OFF OFF ON OFF Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions NCP1252 – Fixed Frequency Controller Featuring Skip Cycle and Latch OCP Value Proposition The NCP1252 offers everything needed to build a cost-effective and reliable ac-dc switching power supply. Unique Features Adjustable switching freq. Delayed operation upon startup • Latched Short circuit protection timer based. • skip cycle mode Benefits Design flexibility independent of the aux. winding Allow temporary over load and latch permanent fault Achieve real no load operation Others Features Adjustable soft start duration Internal ramp compensation Auto-recovery brown-out detection Vcc up to 28 V with auto-recovery UVLO Frequency jittering ±5% of the switching frequency Duty cycle 50% with A Version, 80% with B version Main differences with the UC384X series NCP1252 UC3843/5 Startup current < 100 µA 500 µA Leading Edge Blanking (LEB) Yes No Internal Ramp Compensation Adj. No Frequency jittering 300 Hz, ±5% No Skip Cycle (light load behavior) Yes No Brown-Out with shutdown feature Yes No Pre-short protection Latch-off, 15 ms delay No Delay on startup 120 ms No Soft start Adj. No 5 V voltage reference No Yes Market & Applications ATX Power supply AC adapters 14 CCPG – Jun-09 Ordering & Package Information NCP1252ADR2G: 50% Duty Cycle SOIC8 NCP1252BDR2G: 80% Duty Cycle SOIC8 UC3843/5 Application Exemple UC3843/5 Delay upon startup SS Pre-short protection BO UC384X does not include brown-out, soft-start and overload detection the external implementation cost of these functions is $0.07 NCP1252 includes them all, reducing cost and improving reliability Spec Review: NCP1252’s Demo Board • • • • • • • • Input voltage range: 340-410 V dc Output voltage: 12 V dc, ± 5% Nominal output power: 96 W (8 A) Maximal output power: 120 W (5 seconds per minute) Minimal output power: real no load (no dummy load!) Output ripple : 50 mV peak to peak Maximum transient load step: 50% of the max load Maximum output drop voltage: 250 mV (from Iout = 50% to Full load (5 A 10 A) in 5 µs) Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions Power Components Calculation: Transformer (1/3) • Step 1: Turns ratio calculation in CCM: Vout = η ⋅ Vbulk min ⋅ DCmax ⋅ N ⇔ Where: • Vout is the output voltage • η is the targeted efficiency • Vbulkmin is the min. input voltage • DCmax is the max duty cycle of the NCP1252 • N is the transformer turn ratio Vout N= η ⋅ Vbulk min ⋅ DCmax 12 N= 0.9 × 350 × 0.45 N = 0.085 Power Components Calculation: Transformer (2/3) • Step 2: Verification: Maximum duty cycle at high input line DCmin (Based on the previous equation) Vout = η ⋅ Vbulk max ⋅ DCmin ⋅ N ⇔ Where: • Vout is the output voltage • η is the targeted efficiency • Vbulkmax is the max. input voltage • N is the transformer turn ratio DCmin = DCmin DCmin Vout η ⋅ Vbulk max ⋅ N 12 = 0.9 × 410 × 0.085 = 38.2% Power Components Calculation: Transformer (3/3) • Step 3: Magnetizing inductor value. – For resetting properly the core, a minimal magnetizing current is needed to reverse the voltage across the winding. • (Enough energy must be stored so to charge the capacitance) – Rule of thumb: Magnetizing current = 10% primary peak current ( Lmag ILmag_pk = 10% Ip_pk) Vbulk _ min 350 = = = 13.4 mH 10%I p _ pk 0.1× 0.94 0.45 TON 125k Ip t ILmag DCminTsw t Power Components Calculation: LC Output Filter (1/4) • Step 1: Crossover frequency (fc) selection – arbitrarily selected to 10 kHz. – fc > 10 kHz requires noiseless layout due to switching noise (difficult). Crossover at higher frequency is not recommended • Step 2: Cout & RESR estimation – If we consider a ΔVout = 250 mV dictated by fc, Cout & ΔIout, we can write the following equation: Cout ≥ ΔI out 5 ≥ ⇒ Cout ≥ 318µF 2π f c ΔVout 2π × 10k × 0.25 R ESR ≤ 1 1 ≤ ⇒ RESR ≤ 50 mΩ 2π f c Cout 2π × 10k × 318µ Where: • fc crossover frequency • ΔIout is the max. step load current • ΔVout is the max. drop voltage @ ΔIout Power Components Calculation: LC Output Filter (2/4) • Step 3: Capacitor selection dictated by ESR rather than capacitor value: – Selection of 2x1000 µF, FM capacitor type @ 16 V from Panasonic. – Extracted from the capacitor spec: • Ic,rms = 5.36 A (2*2.38 A) @ TA = +105 °C • RESR,low = 8.5 mΩ (19 mΩ/2) @ TA = +20 °C • RESR,high = 28.5 mΩ (57 mΩ/2) @ TA = -10 °C – ΔVout calculation @ ΔIout = 5 A • ΔVout = ΔI out RESR ,max = 5 × 28.5m = 142 mV Tips: Rule of thumb: RESR ,high ESR( step 2 ) 2 Is acceptable given a specification at 250 mV Power Components Calculation: LC Output Filter (3/4) • Step 4: Maximum peak to peak output current ΔI L ≤ Vripple RESR ,max ≤ 50m ≤ 2.27 A 22m RESR,max = 22 mΩ @ 0 °C Δ IL • Step 5: Inductor value calculation Vout ΔI L ≥ (1 − DCmin ) Tsw L V 12 1 ⇔ L ≥ out (1 − DCmin ) Tsw = (1 − 0.38) 2.27 125k ΔI L L ≥ 26 µH – Let select a standardized value of 27 µH IL DCminTsw (1-DCmin)Tsw t Power Components Calculation: LC Output Filter (4/4) • Step 6: rms current in the output capacitor I Cout ,rms = I out where 1 − DCmin τL = 12τ L Lout Vout 1 I out Fsw = 10 × = ICout,rms (1.06 A) < IC,rms (5.36 A) 1 − 0.38 12 × 2.813 27 µ = 2.813 12 1 10 125k = 1.06 A Note: τL is the normalized inductor time constant No need to adjust or change the output capacitors Power Components Calculation: Transformer Current • RMS current on primary and secondary side – secondary currents: ΔI L 2.27 = 10 + = 11.13 A 2 2 = I L _ pk − ΔI L = 11.13 − 2.27 = 8.86 A IL_pk Δ IL IL_valley IL I L _ pk = I out + I L _ valley – Primary current can calculated by multiplying the secondary current with the turns ratio: I p _ pk = I L _ pk N = 11.13 × 0.085 = 0.95 A t Ip Ip_pk DCTsw (1-DC)Tsw Ip_valley t I p _ valley = I L _ valley N = 8.86 × 0.085 = 0.75 A ⇒ I p ,rms 2 ⎛ ΔI L N ) 2 ( = DCmax ⎜ ( I p _ pk + 10% ) − ( I p _ pk + 10% ) ΔI L N + ⎜ 3 ⎝ ⎞ ⎟ = 0.63 A ⎟ ⎠ Note: Ip,rms has been calculated by taking into account the magnetizing current (10% of Ip_pk). Power Components Calculation: MOSFET (1/3) • With a 2-switch forward converter max voltage on power MOSFET is limited to the input voltage • Usually a derating factor is applied on drain to source breakdown voltage (BVDSS) equal to 15%. • If we select a 500-V power MOSFET type, the derated max voltage should be 425 V (500 V x 0.85). • FDP16N50 has been selected: – – – – – Package TO220 BVDSS = 500 V RDS(on) = 0.434 Ω @ Tj = 110 °C Total Gate charge: QG = 45 nC Gate drain charge: QGD = 14 nC Power Components Calculation: MOSFET (2/3) • Losses calculation: – Conduction losses: Pcond = I p ,rms ,10% 2 RDS ( on ) @T j = 110°C = 0.6322 × 0.434 = 173 mW – Switch ON losses: Δt PSW ,on = Fsw I D ( t )VDS ( t ) dt ∫ Δt 0 Vbulk 2 VDS(t) Vbulk Δt I p _ valleyVbulk Δt 2 Fsw = Fsw = 6 12 0.75 × 410 × 46.7 n = × 125k = 149 mW 12 I p _ valley PSW ,on t Overlap (Δt) is extracted from Δt = QGD I DRV _ pk ID(t) Ip_valley = 14n = 46.7 ns 0 .3 PSW,on losses Power Components Calculation: MOSFET (3/3) – Switch OFF losses: based on the same equation of switch ON PSW ,off = I p _ valleyVbulk ,max Δt 6 Fsw = 1.04 × 410 × 40n × 125k = 355 mW 6 Δt Overlap (Δt ) is extracted from Δt = QGD I DRV _ pk = VDS(t) Ip_pk Vbulk 14n = 40 ns 0.35 ID(t) t – Total losses: Plosses = Pcond + PSW ,on + PSW ,off = 173 + 149 + 355 = 677 mW PSW,off losses Power Components Calculation: Diode (1/2) • Secondary diodes: D1 and D2 sustain same Peak Inverse Voltage (PIV): – Where kD is derating factor of the diodes (40%) NVbulk max 0.085 × 410 = = 58 V PIV = 1 − kD 0.6 Q2 X1 D1 L Vin D4 D3 Lmag Q1 0 D2 C PIV < 100 V Schottky diode can be selected: R MBRB30H60CT (30 A, 60 V in TO-220) Power Components Calculation: Diode (2/2) • Diode selection: MBRB30H60CT (30 A, 60 V in TO-220) • Losses calculation: – During ON time : Worst case @ low line (DCmax) Pcond , forward = I outV f DCmax = 10 × 0.5 × 0.45 = 2.25 W – During OFF time : Worst case @ High line (DCmin) Pcond , freewheel = I outV f (1 − DCmin ) = 10 × 0.5 × (1 − 0.39 ) = 3.05 W 0.5V @ 125°C Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop Feedback: simulations and compensation 7. Demo board schematics & Picture. 8. Board performance review 9. Conclusions NCP1252 Components Calculation: Rt • Switching frequency selection: a simple resistor allows to select the switching frequency from 50 to 500 kHz: Rt = 1.95 × 109 VRt Fsw If we assume Fsw = 125 kHz Rt = 1.95 × 10 × 2.2 = 34.3 k Ω 125k 9 Where: • VRt is the internal voltage reference (2.2 V) present on Rt pin ≈ 33 kΩ NCP1252 Components Calculation: Sense Resistor • • NCP1252 features a max peak current sensing voltage to 1 V. The sense resistor is computed with 20% margin of the primary peak current (Ip,rms,20%): 10% for the magnetizing current + 10% for overall tolerances. FCS 1 = = 884 mΩ Rsense = I p _ pk + 20% 0.946 × 1.2 PRsense = Rsense I p ,rms + 20% 2 = 0.884 × 0.6952 = 427 mW If we select 1206 SMD type of resistor, we need to place 2 resistors in parallel to sustain the power: 2 x 1.5 Ω. Where: • Ip_pk is the primary peak current • Ip,rms,20% is the primary rms current with a 20% margin on the peak current NCP1252 Components Calculation: Ramp Compensation (1/5) • Ramp compensation prevents sub-harmonic oscillation at half of the switching frequency, when the converter works in CCM and duty ratio close or above 50%. • With a forward it is important to take into account the natural compensation due to magnetizing inductor. • Based on the requested ramp compensation (usually 50% to 100%), only the difference between the ramp compensation and the natural ramp could be added externally – Otherwise the system will be over compensated and the current mode of operation can be lost, the converter will work more like a voltage mode than current mode of operation. NCP1252 Components Calculation: Ramp Compensation (2/5) • How to build it? Where: • Vramp = 3.5 V, Internal ramp level. • Rramp = 26.5 kΩ, Internal pull-up resistance NCP1252 Components Calculation: Ramp Compensation (3/5) • Calculation: Targeted ramp compensation level: 100% – Internal Ramp: Sint = Vramp DCmax Fsw = 3.5 125k = 875 mV/µs 0.50 – Natural primary ramp S natural = Vbulk 350 0.75 = 20.19 mV/µs Rsense = −3 Lmag 13 ⋅10 – Secondary down slope S sense = (Vout + V f ) N s (12 + 0.5) Rsense = 0.087 × 0.75 = 30.21 mV/µs −6 Lout Np 27 ⋅10 – Natural ramp compensation δ natural _ comp = S natural 20.19 = = 66.8% 30.21 S sense Where: • Vout = 12 V • Lout = 27 µH • Vf = 0.5 V (Diode drop) • Rsense : 0.75 Ω • Fsw : 125 kHz • Vbulk,min = 350 V • DCmax = 50% • Lmag = 13 mH • N = 0.087 NCP1252 Components Calculation: Ramp Compensation (4/5) • As the natural ramp comp. (67%) is lower than the targeted 100% ramp compensation, we need to calculate a compensation of 33% (100-67). Ratio = ( S sense δ comp − δ natural _ comp Sint Rcomp = Rramp Rsense1 1.5R Rsense2 1.5R CS pin 0 0 875 Ratio 0.0114 = 26.5 ⋅103 = 305 Ω 1 − Ratio 1 − 0.0114 Rcomp 330R CCS 680pF ) = 30.21(1.00 − 0.67 ) = 0.0114 • RcompCCS network filtering need time constant around 220 ns: CCS = τ RC 220n = = 666 pF RComp 330 NCP1252 Components Calculation: Ramp Compensation (5/5) • Illustration of correct filtering on CS pin switching noise is filtered CS pin current information is not distorted NCP1252 Components Calculation: Brown-Out • Dedicated pin for monitoring the bulk voltage to protects the converter against low input voltage. IBO current source is connected when BO pin voltage is below VBO reference: its creates the BO hysteresis NCP1252 Components Calculation: Brown-Out • From the previous schematic, we can extract the brown-out resistors ⎛ Vbulkon − VBO ⎞ 1 ⎛ 370 − 1 ⎞ − 1⎟ = − 1 = 5731 Ω ⎜ ⎜ Vbulkoff − VBO ⎟ 10µ ⎜⎝ 350 − 1 ⎟⎠ ⎝ ⎠ = 5.1 kΩ + 680 Ω RBOlo = RBOlo VBO I BO RBOup = Vbulkon − Vbulkoff I BO = 370 − 350 = 2.0 MΩ 10 µ RBOup = 2 × 1 MΩ Where : • Vbulkon = 370 V, starting point level • Vbulkoff = 350 V, stopping point level • VBO = 1 V (fixed internal voltage reference) • IBO = 10 µA (fixed internal current source) Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions Small Signal Analysis: Model NCP1252’s small signal model is available for running and validating the closed loop regulation, as well as the step load response of the power supply with very fast simulation time. DC 2 V1 {Vin} IN FB DC 3 U5 NCP1252_AC 4 OUT GND 1 5 • D3 FS = 125K L = {L1/(2*N1**2)} RI = {RSENSE} SE = {SP} 0 2 U2 1 L1 {L1} R1 V12V 13.3m R2 {Rdelay} MBRB30H60CT 1 3 XFMR1 0 2 C1 2000u RATIO = {N1} V12V 0 R6 {Rled} Cpole = {Cpopto} CTR = {CTR} FB U3 opto R7 1k R4 {Rupper} C2 U4 TL431 C4 1n {Czero} R5 4.7k 0 Example of schematic for studying closed loop regulation R3 {CTR_a} Small Signal Analysis: Power Stage 1 40 180d 32 144d 24 16 ⎪G(s)⎪ -23 dB -66° @ FC = 6 kHz @ FC = 6 kHz 108d 72d 8 36d 0 0d Arg(G(s))-8 -36d -16 -72d -24 -108d -32 -144d >> -40 100Hz 1 -180d DB(V(V12V)) 2 1.0KHz P(V(V12V)) 10KHz 100KHz Frequency If we want a crossover @ Fc = 6 kHz, we need to measure: ⎪G(6 kHz)⎪ = -23 dB Arg(G(6 kHz)) = -66° 2 Small Signal Analysis: Open Loop After applying the K factor method @ Fc = 6 kHz and phase margin = 70°, with the help of an automated Orcad simulation, we obtain: PARAMETERS: Vout = 12V L1 = 27u L2 = {L1*(N2/N1)**2} N1 = 0.0870 N2 = 0.0498 Rsense = 0.75 Rupper = {(Vout-2.5)/532u} 1 80 180d 64 144d Simulated with the help of Orcad 48 Fc = 6k PM = 70 108d 32 72d 16 36d 0 0d GFc = -25 PFc = -66 G = {10**(-GFc/20)} boost = {PM-PFc-90} K = {tan((boost/2+45)*pi/180)} -16 -36d -32 -72d C2 = {1/(2*pi*Fc*G*K*Rupper)} C1 = {C2*(PWR(K,2)-1)} R2 = {K/(2*pi*Fc*C1)} Fzero = {Fc/K} -48 Measured on a bench -108d Fpole = {K*Fc} Rpullup = 4k -64 RLED = {CTR*Rpullup/G} >> -80 100Hz 1 Czero = {1/(2*pi*Fzero*Rupper)} Cpole = {1/(2*pi*Fpole*Rpullup)} CTR = 0.7 Lmag = 12.3mH Sp = {(Vin/Lmag)*Rsense} Vin = 390V Cfb = {Cpole-Cpopto} Cpopto = 3nF -144d DB(V(FB)) 2 1.0KHz P(V(FB)) 10KHz Frequency -180d 100KHz 2 Step Load Stability Validation of the closed loop stability with a step load test 165 mV < 250 mV targeted Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions NCP1252 Demo Board Schematic (1/2) Vbulk FDP16N50 M1 DRV_HI (Drive and Vcc circuits are shown on the next slide) D2 MURA160 J2 IN_GND R1 105k R2 22R 2W 0 R3 47k D3 1SMA5931 Vin J1 C1 47uF 450V C3 10pF 450V D4 MUR160 1 DRV_HI_ref C2 2.2nF 100V 1 T1 D6 MUR160 5 0 FDP16N50 M2 R10 47k D8 1SMA5931 D7 MURA160 C8 10pF 450V CS R17 200k 1% C14 1nF R18 100 1% 0 R6 2.2nF 10R R12 1R5 R8 1.5k FB U2 SFH615A_4 R13 1R5 R9a 9k R11 1k C11 1nF 0 U4 NCP1252 1 3 SS BO Vcc CS 4 R19 1k FB RT C15 220pF R20 39k 0 0 DRV GND 8 C10 33nF 0 0 7 0 U3 TL431 R9b 9k C9 10nF VCC 6 DRV 5 C13 100nF 0 R15 4.7k 0 NCP1252 controller 12 Vout J3 C4 1000uF/FM 16V C5 1000uF/FM 16V Out_GND J4 FB 2 CS R21 6200 1% R7 105k 0 R14 1M 1% R16 1M 1% C6 2.2nF 100V 6 XFMR1 C7 DRV_LO Vbulk 2 2306-H-RC R4 22R 2W D5 MBRB30H60 10 L1 27uH 2-Switch forward converter NCP1252 Demo Board Schematic (2/2) D301 MMSD4148 XFMR2 6 MMBT489LT1G Q301 4 C302 220nF 1 2 3 1 2 3 4 U104 NCP1010P60 VCC GND NC GND GND FB DRAIN 5 C301 10n R305 47R 1 8 7 5 2 U301 U102 SFH615A_4 C101 1n 1 D303 MMSD4148 DRV_HI DRV_HI_ref DRV_LO DRV_LO_ref R306 1k Q304 MMBT589LT1G Vbulk + C102 47uF/25V Q303 MMBT589LT1G 3 MMBT589LT1G Q302 J203 HEADER 3 D302 MMSD4148 R304 1k R302 47 Vcc DRV GND R301 47R L101 2 D102 MUR160 Vcc : Auxiliary power supply D101 BZX84C13/ZTX R102 1k Vcc 2.2mH + C103 47uF/25V 0 J302 HEADER 5 High side and low side driver R101 1k 0 0 1 2 3 4 5 NCP1252 Demo Board: Pictures Top view Bottom view Link to demoboard web page: http://www.onsemi.com/PowerSolutions/evalBoard.do?id=NCP1252TSFWDGEVB Or from the page of the NCP1252: http://www.onsemi.com/PowerSolutions/product.do?id=NCP1252 Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions NCP1252 Demo Board: Efficiency 40% of max load Efficiency > 90% NCP1252 Demo Board: No Load Operation Time (400 µs/div) • Thanks to the skip cycle feature implemented on the NCP1252, it is possible to achieve a real no load regulation without triggering any overvoltage protection. The demonstration board does not have any dummy load and ensure a correct no load regulation. This regulation is achieved by skipping some driving cycles and by forcing the NCP1252 in burst mode of operation. NCP1252 Demo Board: Soft Start One dedicated pin allows to adjust the soft start duration and control the peak current during the startup NCP1252 Demo Board: Performance Improvements • Synchronous rectification on the secondary side of the converter will save few percent of the efficiency from middle to high load. • Stand-by power: The NCP1252 can be shut down by grounding the BO pin less than 100 µA is sunk on Vcc rail when NCP1252 is shutdown. Agenda 1. Generalities on forward converters 2. Core reset: tertiary winding, RCD clamp, 2-switch forward 3. Specs review of the NCP1252’s demo board 4. Power components calculation 5. NCP1252 components calculation 6. Closed-loop feedback: simulations and compensation 7. Demo board schematics & picture. 8. Board performance review 9. Conclusions Conclusion • NCP1252 features high-end characteristics in a small 8-pin package • Added or improved functions make it powerful & easy to use • Low part-count • Ideal candidate for forward applications, particularly adapters, ATX power supplies and any others applications where a low standby power is requested. References • Datasheet: NCP1252/D “Current Mode PWM Controller for Forward and Flyback Applications” • Application note: AND8373/D “2 Switch-Forward Current Mode Converter” Detailed all the calculations presented in this document. • C. Basso, Director application engineer at ON Semiconductor. “Switch Mode Power Supplies: SPICE Simulations and Practical Designs”, McGraw-Hill, 2008. • Note : Datasheet and application note are available on www.onsemi.com. For More Information • View the extensive portfolio of power management products from ON Semiconductor at www.onsemi.com • View reference designs, design notes, and other material supporting the design of highly efficient power supplies at www.onsemi.com/powersupplies

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