APPLICATION NOTE APN1013: A Differential VCO Design for GSM Handset Applications Introduction The VCO Model The differential pair of bipolar transistors is the common building block in modern RF integrated circuits. An advantage of this architecture is its high loop gain making it popular for differential Voltage Controlled Oscillator (VCO) designs in RFICs. Designers of discrete (or hybrid) VCO circuits usually use the more traditional Colpitts design, avoiding the added complexity of the differential VCO configuration. In the circuit in Figure 1 the transistor pair X1 and X2 form a differential configuration with X3 as a current source. All of the transistors are NEC NE68519 with 12 GHz fT. We chose this transistor for this demonstration because it is commonly used in many VCO applications and its SPICE parameters are available. The RF signal from collector of X1 is fed through the coupling capacitor SRLC3 to the resonator. The resonator is formed by the parallel connection of capacitive branches SRLC4 and SRLC5/X4, and the inductive branch formed by microstrip line TL1. The output signal from the resonator is fed through the coupling capacitor SRLC2 to the base of transistor X2, which is in differential feedback with X1. There it closes the feedback path. Ideally, the phase shift between the base of X2 and the collector of X1 in the differential stage is 0°, and the oscillation should occur at the exact parallel resonance of the tank circuit. In reality, however, the phase shift in the differential pair is not 0°, because of unavoidable transit times and parasitic reactances. Because of its primary usage in proprietary RFICs, only limited differential VCO design and modeling information has been made available. In this application note, we will attempt to build a bridge toward better understanding of the differential VCO by describing a design covering 910–980 MHz which is appropriate for use in GSM handsets. This design will also demonstrate how the SMV1493-079 varactor diode, an essential element in this VCO, may be used advantageously in designs aimed at other wireless applications including AMPS, CDMA, and PCS. For example, the phase shift in this loop changes from -50° to -100° in the frequency range of 0.8–1.5 GHz. To compensate this (capacitive) phase shift, the resonator should be inductive and provide an opposite phase shift. It is interesting to note that compared to the Colpitts VCO configuration, the resonant frequency may be set at the exact oscillation frequency. The Colpitts design cannot be used at the natural resonance frequency of the resonator due to losses in the feedback path. Therefore, the differential design has the advantage of both high phase slope at resonance and low loss in the feedback path. Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200323 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 1 APPLICATION NOTE • APN1013 Spice Model for SMV1493-079 Figure 1. VCO Model Schematic Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 2 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200323 Rev. A APPLICATION NOTE • APN1013 Capacitors SRLC2 and SRLC3 are used to obtain the required phase shift at the center of the tank circuit resonant frequency. By changing the values of these capacitors, optimum phase noise may be achieved. Resistor R5 suppresses higher frequency oscillation modes caused by the parasitic resonance between SRLC4 and varactor branch SRLC5/X4. A resistance of 20–50 Ω is normally enough to perform this function without serious degradation of loop power and phase noise. In the VCO model test bench in Figure 2 we define open loop gain, Ku = VOUT/VIN, as the ratio of voltage phasors at the input and output ports of an OSCTEST component. Defining the oscillation point is a technique to balance input (loop) power to provide zero gain for a zero loop phase shift. Once the oscillation point is defined, the frequency and output power may be “measured.” We do not recommend the use of the OSCTEST2 component for closed loop analysis, since it may not converge and does not allow clear insight to VCO behavior. The differential stage bias is set by voltage dividers R6/PRC1 and R8/R7. The base currents are limited by R3 and R5 for the differential pair and R2 for the current source. In this case, the 3 V, VCC current is set to about 7 mA. Figure 2. VCO Model Test Bench Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200323 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 3 APPLICATION NOTE • APN1013 . Figure 3. VCO Model Default Bench Figure 3 shows the default bench where some circuit variables were defined in the “Variables and Equations” component for the convenience of “tuning” during performance analysis and optimization SMV1493-079 SPICE Model SMV1493-079 is a low series resistance, hyperabrupt junction varactor diode. It features the industry’s smallest plastic package SC-79 with a body size of 47 x 31 x 24 mm (the total length with leads is 62 mm). The SPICE model for the SMV1493-079 varactor diode, defined for the Libra IV environment, is shown in Figure 1 with a description of the parameters employed. Table 1 describes the model parameters. It shows default values appropriate for silicon varactor diodes that may be used by the Libra IV simulator. Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 4 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200323 Rev. A APPLICATION NOTE • APN1013 Parameter Unit Default IS Saturation current (with N, determine the DC characteristics of the diode) Description A 1e-14 RS Series resistance Ω 0 N Emission coefficient (with IS, determines the DC characteristics of the diode) - 1 TT Transit time S 0 CJO Zero-bias junction capacitance (with VJ and M define nonlinear junction capacitance of the diode) F 0 VJ Junction potential (with VJ and M define nonlinear junction capacitance of the diode) V 1 M Grading coefficient (with VJ and M define nonlinear junction capacitance of the diode) - 0.5 EG Energy gap (with XTI, helps define the dependence of IS on temperature) eV 1.11 XTI Saturation current temperature exponent (with EG, helps define the dependence of IS on temperature) - 3 KF Flicker noise coefficient - 0 AF Flicker noise exponent - 1 FC Forward bias depletion capacitance coefficient - 0.5 BV Reverse breakdown voltage V Infinity IBV Current at reverse breakdown voltage A 1e-3 ISR Recombination current parameter A 0 NR Emission coefficient for ISR - 2 IKF High injection knee current A Infinity NBV Reverse breakdown ideality factor - 1 IBVL Low-level reverse breakdown knee current A 0 NBVL Low-level reverse breakdown ideality factor - 1 TNOM Nominal ambient temperature at which these model parameters were derived °C 27 FFE Flicker noise frequency exponent - 1 Table 1. Silicon Diode Default Values in Libra IV According to the SPICE model, the varactor capacitance, CV, is a function of the applied reverse DC voltage, VR, and may be expressed as follows: CV = CJO M ( 1 + VVAR ) Part CJO Number M (pF) VJ CP (V) RS (pF) LS Ω) (Ω (nH) SMV1493-079 29 0.47 0.63 0 0.25 1.7 Table 2. SPICE Parameters for SMV1493-079 + CP VJ 30 Note, that in the Libra model above, CP is given in picofarads, while CGO is given in farads to comply with the default unit system used in Libra. 25 Capacitance (pF) This equation is a mathematical expression of the capacitance characteristic. The model is accurate for abrupt junction varactors (like the SMV1408). For hyperabrupt junction varactors, the model is less accurate because the coefficients are dependent on the applied voltage. To make the equation work better for the hyperabrupt varactors, the coefficients were optimized for the best capacitance vs. voltage fit as shown in Table 2 and Figure 4. Measurement Data Model 20 15 10 5 0 1 2 3 4 5 Varactor Voltage (V) Figure 4. SMV1493-079 CV Curve Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200323 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 5 APPLICATION NOTE • APN1013 VCO Design, Materials, Layout and Performance The differential VCO schematic diagram appropriate for an RFIC in many wireless or cordless applications is shown in Figure 5. This circuit is supplied from the 3 V voltage source. The ICC current is established at approximately 8 mA, and the RF output signal is fed from the VCO through capacitor C4 (1 pF). The PCB layout is shown in Figure 6. The board is made of standard 30 mil thick FR4 material. Table 3 lists the bill of materials. VCC +3 V L1 33 n R4 3k R2 3k R7 3k R6 3k R10 51 C6 100 p R1 47 C3 2p C2 3.6 p C4 1p RF Out V2 NE68519 V1 NE68519 C8 6p R9 3.9 k R5 3k R3 9.1 k C1 100 p V3 NE68519 C7 2p M1 4 x 0.5 mm R8 20 M2 6.5 x 0.2 mm D1 SMV1493-079 VVAR C5 100 p Figure 5. VCO Schematic Diagram Figure 6. Differential VCO PC Board Layout Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 6 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200323 Rev. A APPLICATION NOTE • APN1013 Designator Value Part Number Footprint Manufacturer C1 100 p 0603AU101JAT9 0603 AVX C2 3.6 p 0603AU3R6JAT9 0603 AVX C3 2p 0603AU2R0JAT9 0603 AVX C4 1p 0603AU1R0JAT9 0603 AVX C5 100 p 0603AU101JAT9 0603 AVX C6 100 p 0603AU101JAT9 0603 AVX C7 2p 0603AU2R0JAT9 0603 AVX C8 6p 0603AU6R0JAT9 0603 AVX D1 SMV1493-079 SMV1493-079 SOD-323 Skyworks Solutions Coilcraft L1 33 n 0603CS-33NX_BC 0603 M1 4 x 0.5 mm MSL 4 x 0.5 mm M2 6.5 x 0.2 mm MSL 6.5 x 0.2 mm R1 47 CR10-470J-T 0603 AVX R10 51 CR10-510J-T 0603 AVX R2 3k CR10-302J-T 0603 AVX R3 9.1 k CR10-912J-T 0603 AVX R4 3k CR10-302J-T 0603 AVX R5 3k CR10-302J-T 0603 AVX R6 3k CR10-302J-T 0603 AVX R7 3k CR10-302J-T 0603 AVX R8 20 CR10-200J-T 0603 AVX R9 3.9 k CR10-392J-T 0603 AVX V1 NE68519 NE68519 SOT-416 NEC V2 NE68519 NE68519 SOT-416 NEC V3 NE68519 NE68519 SOT-416 NEC Table 3. VCOs Bill of Materials Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200323 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 7 APPLICATION NOTE • APN1013 The measured performance of this circuit and the simulated results obtained with the model are shown in Figure 7. Fairly good compliance of frequency response vs. varactor tuning voltage confirms the validity of the varactor model for predicting the frequency behavior of the VCO. However, the power response simulation indicates more variation of power in the tuning range than the measured response. This may be due to the transistor modeling accuracy or the accuracy of the overall VCO model. For example, simulations show that the power may strongly depend on high order harmonics which may not be accurately described by the VCO model. 1.00 -5 Frequency Simulated 0.98 -6 0.96 -7 0.94 -8 -9 0.92 Output Power (dBm) Frequency (GHz) example, after exchanging the varactor with a lower loss discrete ceramic capacitor, the phase noise was improved by 5–6 dB. In another example, increasing the solder thickness on the microstrip line improved phase noise as much as 3 dB. In comparison, typically there is less phase noise sensitivity to circuit components in a Colpitts VCO design, where the oscillation frequency is always below the resonance frequency of the resonator. Power Simulated -10 0.90 0 1 2 3 Figure 8. The VCO Phase Noise vs. Carrier Offset Frequency Tuning Voltage (V) Frequency Meas. Power Meas. Figure 7. Measured and Simulated Frequency and Power Responses of the Differential VCO Figure 8 shows the phase noise of the VCO measured in the range of 100 Hz to 8 MHz offset from the carrier. This measurement was made using the Aeroflex PN9000 Phase Noise Test Set with a 40 ns delay line. The best phase noise occurred with circuit components very close to the originally established values in the preliminary simulation. Only small changes were introduced such as: C2 from its original 4 pF to 3.6 pF, C3 from 1.5 pF to 2 pF and C4 from 1.5 pF to 1 pF. Phase noise was approximately -93 dBc/Hz at a 10 kHz offset, which is about 5–10 dB poorer than a typical Colpitts low noise VCO having similar bias condition. The reason for the poorer phase noise may be that the resonator in a differential VCO works at its resonance frequency (or very near to it). This results in high reactive currents in the resonator, causing high power losses, causing much higher phase noise sensitivity to resonator component losses. This was confirmed by replacing the resonator circuit components with similar ones having less loss. For List of Available Documents The Differential RF VCO Simulation Project Files for Libra IV. The Differential RF VCO Circuit Schematic and PCB Layout for Protel EDA Client, 1998 Version. The Differential RF VCO PCB Gerber Photo-plot Files. VCO Related Application Notes APN1004, “Varactor SPICE Models for RF VCO Applications.” APN1006, “A Colpitts VCO for Wideband (0.95– 2.15 GHz) Set-Top TV Tuner Applications.” APN1005, “A Balanced Wideband VCO for Set-Top TV Tuner Applications.” APN1007, “Switchable Dual-Band 170/420 MHz VCO for Handset Cellular Applications.” “An RF VCO Design for Wireless and Broadband Applications.” Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 8 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200323 Rev. A APPLICATION NOTE • APN1013 Copyright © 2002, 2003, 2004, 2005, Skyworks Solutions, Inc. All Rights Reserved. Information in this document is provided in connection with Skyworks Solutions, Inc. (“Skyworks”) products or services. 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Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200323 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 9