APPLICATION NOTE APN1015: A Dual-Band Switchable IF VCO for GSM/PCS Handsets Introduction Many of today’s handset cellular telephones are multifunctional, multiband units. They are complex RF systems with frequency plans requiring multiple RF sources. To accomplish this, the number of VCOs can be increased, however this is expensive and requires more PCB area. This approach strongly contradicts current market trends. A straightforward solution proposed in this application note is using switchable (multiband) VCOs. Skyworks application note APN1007, Switchable Dual-Band 170/470 MHz VCO for Handset Cellular Applications, discusses a switchable Colpitts VCO design switching between the 170 MHz and 420 MHz range. Described here is a solution for higher frequency range switching, 450/640 MHz, using a DC Cascode Colpitts configuration for the VCO. This design is optimized for the lowest phase noise meeting GSM/PCS handset requirements. The Colpitts VCO Fundamentals Fundamental Colpitts VCO operation is illustrated in Figures 1a and 1b. Figure 1a shows the Colpitts VCO circuit as it is usually implemented. In Figure 1b, the same circuit is shown as a common emitter amplifier with parallel feedback. The transistor junction and package capacitors CEB, CCB and CCE are separated from the rest of the transistor parasitic components to demonstrate their direct effect on the VCO tank circuit. In a real low noise VCO circuit, the capacitor, CVAR, may have a more complicated structure including series and parallel connected discrete capacitors used to set required oscillation frequency and tuning sensitivity. The parallel connection of resonator inductor, LRES, and varactor capacitive branch, CVAR , constitutes a parallel resonator (or simply resonator). A fundamental property of the parallel resonator in a Colpitts VCO implementation is that it always shows inductive impedance at the oscillation frequency. This means that its parallel resonant frequency is always above the oscillation frequency. Loss in the resonator increases as the frequency approaches resonance in the feedback loop, acting as a stop-band filter at resonance. Thus, the nearer the oscillation frequency to parallel resonance, the more loss incurred in the feedback path. However, since more reactive energy is stored in the resonator nearer to the resonance frequency, higher loaded Q (QL) is achieved. Obviously, low loss resonators, such as crystals or dielectric resonators, allow oscillation buildup closer to parallel resonance with much lower loss compared to microstrip or discrete component-based resonators. The proximity of the parallel resonant frequency to the oscillation frequency is established by the value of capacitor, CSER. If the capacitance of CSER were reduced, then the parallel resonator would be more inductive to compensate for the increased capacitive reactance. This means that the oscillation frequency should move closer to the parallel resonance and would result in higher QL and higher feedback losses. VCC CCB CCE CSER CVCC CCB CDIV1 LRES CVAR POUT CEB CSER LRES CVCC CDIV2 RL CEB CVAR CDIV1 CDIV2 RL CCE Figure 1a. Basic Colpitts VCO Configuration Figure 1b. Common Emitter View of the Colpitts VCO Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200325 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 1 APPLICATION NOTE • APN1015 is that the control component (PIN diode) is not placed in the resonator current path. That way it exercises control over a small reactance portion of the overall tank circuit. This has significant impact on VCO performance, especially if more than 10% of the switching gap (the difference between designated frequency bands) is required. The Leeson equation, establishing connection between tank circuit QL and its losses, states: ξ ( ƒm) = FkT 2P 1+ ƒ2 4 Q L2 ƒm2 Where F is the large signal noise figure of the amplifier shown in Figure 1b, P is the loop or feedback power (the one which measured at the input of the transistor), and QL is loaded Q. These three parameters have significant consequences for phase noise in a low noise RF VCO. In designing a low noise VCO, we need to define the conditions for minimum F and maximum P and QL. The above discussion shows that the loop power and QL are contradictory parameters. That is, an increase of QL leads to more losses in the feedback path resulting in lower loop power. The optimum conditions for noise also contradict maximum loop power, and largely depend on transistor choice. Usually the best noise is achieved with high gain transistors with maximum gain coinciding with minimum noise at the large signal condition. Because no such specifications are currently available for industry-standard transistors, we can base our transistor choice only on experience. A Switchable Resonator VCO Circuit Switchable resonator VCO designs are shown in Figures 2a and 2b. In the switchable resonator concept, two or more separate resonators, tuned and optimized according frequency bands, are activated (switched) by low resistance PIN-diodes (D3 and D4). The advantage of switching the entire resonator rather than switching an element within the resonator (capacitor or inductor) To understand the impact of switching on the resonator losses, consider the following example. Assume switching between 0.47 and 0.62 GHz bands using these two concepts. In the intraresonator-switching scheme in Figure 3a, the capacitance changes from 10 pF to 5.8 pF. In this case, 4.2 pF was added to jump to the lower frequency band. Alternately, in the interresonator switching scheme, 2 pF was added to the switching path. Simple analysis shows that the current flowing through the switching component in the intraresonator scheme may be more than double. This results in more than 6 dB additional loss in the lower band compared to the interresonator concept, which may be enough to prevent any oscillation. Even if oscillation could be sustained due to the excess of gain in the oscillator’s active portion, there is still the problem of balancing loaded Q and the feedback loop power to optimize the phase noise performance. Other problems with the intraresonator switching scheme include the lack of flexibility in providing optimum tuning in both frequency bands and extra noise modulation. The PIN diode control current comes from the same source that feeds the rest of the handset circuitry. This current may be carrying lowfrequency noise that may not be filterable. Even though these noise fluctuations of DC current are small and relatively fast, the PIN diode is still a semiconductor device with inevitable nonlinear and/or parametric consequences that may result in excess modulation noise. VSW1: Band 1 VCTL: 0.5–2.5 V VSW1: Band 1 VCC: 3 V C5 C1 C9 R1 R6 VCC: 3 V C1 L3 C5 C11 C9 R1 R6 L1 L1 D1 D2 L2 V1 D3 D4 R2 R3 C2 C3 C12 D1 C6 C4 R4 C10 C8 D4 C14 POUT L4 R5 V1 C6 D2 C7 D3 L2 C2 R3 R4 C7 C10 R2 POUT C8 C13 R5 C4 VCTL: 0.5–2.5 V C3 VSW2: Band 2 Figure 2a. Switchable Resonator VCO with Simplified Resonator Design VSW2: Band 2 Figure 2b. Switchable Resonator VCO with High-Performance Resonator Design Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 2 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200325 Rev. A APPLICATION NOTE • APN1015 RS 2 CPAR 3.8 p LRES 12 nH Relative Tuning Sensitivity Variation (%) CSER 2p ISW1 C 4.2 p a) Intraresonator Switching CSER 2p CSER 2p RS 2 LRES 12 nH CPAR 50 45 40 35 6 pF 30 4.5 pF 25 LRES 20 LRES 12 nH CPAR 3.8 p CSER2 CPAR 15 3 pF 10 SMV1763 2 pF 5 3 RS 2 CPAR 8p 55 4 5 6 7 8 9 10 Series Capacitance (CSER2) (pF) Figure 4. Relative Tuning Sensitivity (Kf) Variation in the Range of 0.5–2.5V for SMV1763-079 as a Function of CSER2 and CPAR ISW2 <0.5ISW1 b) Interresonator Switching Figure 3. VCO Band Switching Concepts The two circuits shown in Figure 2 differ only in resonator design. Figure 2a shows a simplified resonator design, minimizing the component count, thus minimizing cost and space. Resonator inductors L1 and L2 are also used as varactor biasing chokes. However, this circuit has some drawbacks. One is a lack of design flexibility caused by discrete choices of available varactors (D1 and D2) and inductors (L1 and L2) values. Capacitors may be added in parallel to the varactors to improve design flexibility, but this may degrade the tuning sensitivity. Tuning linearity may also suffer in such a circuit because there is no capacitor in series with the varactor. Alternately, the resonator in Figure 2b has more components, which allows more design flexibility and better tuning linearity, but the component count is higher. The effect of capacitor, CSER2, in series with the varactor is demonstrated in Figure 4. The relative tuning sensitivity is defined as: K ƒ = 1 ∂ƒ ; ƒ ∂V V C S2 K ƒ = C JO K DV = K DV C 2VAR (C S + C PAR ) + C VAR C S ( 2 C PAR + C S ) + C 2V J V 1 + VAR M? VJ K ƒVAR = K ƒ 0.5 V K K M +1 ƒ 2.5 V ƒ 1.5 V % 2 S C PAR ; Where VVAR is varactor DC bias in the middle of the tuning range,CSER2 and CPAR are capacitors in series with the varactor (C11 and C13 in Figure 2b) and resonator parallel capacitors (C12 and C14 in Figure 2b), and CJO, VJ, M are parameters describing varactor capacitance [1]: See SMV1763-079 SPICE model section. According to this equation, relative tuning sensitivity variation is defined as the percent variation of tuning sensitivity per volt in the tuning range from 0.5–2.5 V — typical for most battery handset applications. The graph shows that higher values of CSER2 cause larger tuning sensitivity variations. Consequently, high variation of tuning sensitivity would occur across the tuning range without CSER2 in Figure 2a. VCO Model Description In the circuit in Figure 5 transistors, X3 and X9 are connected in DC Cascode sharing the base biasing network consisting of R1 (RDIV2), R4 (RDIV1), and R7 (RDIV3). The bias resistor values were selected to evenly distribute DC voltages between X3 and X9. Resistor R5 (RL) was chosen as low as 100 Ω to minimize the DC voltage drop for the specified 8 mA DC current. At RF frequencies, X9 works as a common emitter amplifier with the emitter grounded through capacitor SRLC1. The oscillator stage output is fed to the buffer transistor through coupling capacitor C17 (CCPL). The output circuit of the buffer stage consists of discrete inductor, L4, modeled with parallel capacitor, C5 (0.38 pF), and output capacitor C1 (COUT). For flatter power response over the specified 450/640 MHz range, capacitor SLC2 (for values less than 2 pF), in parallel with inductor L4 (in Figure 5), may be used for fine trimming. Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200325 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 3 APPLICATION NOTE • APN1015 The dual-band switchable resonator circuit in Figure 6 consists of two identical parts (low band on the left, and high band on the right). The PIN diodes were modeled as parallel RC networks PRC1 and PRC2, with switching resistors, RSW_L and RSW_H, in the low band and high band branches respectively. Figure 5. Open Loop Analysis VCO Schematic Bench Figure 6. Dual-Band Resonator Schematic Bench Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 4 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200325 Rev. A APPLICATION NOTE • APN1015 The appropriate biasing resistors, R1, R2, and R3, are shown as shunt elements to ground. The truth table showing the values of RSW_L and RSW_H for the appropriate low/high switching is shown below: RSW_L RSW_H State 3 Ω 3000 Ω Low band 3000 Ω 3 Ω High band Resonator inductors, L1 and L4, are modeled as inductance losses with parallel capacitors, C4 and C5 respectfully. The resonator’s parallel capacitors are presented as series equivalent models, SRLC1 and SRLC2, each with 1.5 nH inductance and 0.2 Ω series resistance. Other discrete capacitors are modeled with similar series SRLC networks. The different inductor values were selected based on the individual RF path layout. Varactor SMV1763-079, used in this design, is described by the PN-junction diode SPICE model described in the next section. The variable values used in the circuit are given in the Variable Equation module of the Default Bench shown in Figure 7. The Test Bench is shown in Figure 8. For open loop analysis we use the OSCTEST component supplied in the Libra IV library. This component allows us to observe the open loop VCO feedback gain as a function of frequency and power (in the loop) preserving feedback loading integrity. This is the way the VCO would see it when the feedback loop is closed. (Refer to Libra IV manual for further details of OSCTEST operation). Figure 7. VCO Default Bench Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200325 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 5 APPLICATION NOTE • APN1015 Figure 8. VCO Test Bench SMV1763-079 SPICE Model SMV1763-079 is a low series resistance, hyperabrupt varactor diode. It has the industry’s smallest plastic package SC-79 with a plastic body size of 47 x 31 x 24 mils (the total length with leads is 62 mils). The SPICE model for the SMV1763-079 varactor diode, defined for the Libra IV environment, is shown in Figure 9 with a description of the parameters employed. Figure 9. SMV1763-079 SPICE Model for Libra IV Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 6 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200325 Rev. A APPLICATION NOTE • APN1015 Table 1 describes the model parameters. It shows the appropriate default values for silicon varactor diodes that may be used by the Libra IV simulator. This equation is a mathematical expression of the capacitor characteristic. The model is most accurate for abrupt junction varactors (like the SMV1408). However, for hyperabrupt junction varactors the model is less accurate because the coefficients are dependent on the applied voltage. To make the above equation work better for hyperabrupt junction varactors, the coefficients were optimized for the best capacitance vs. voltage fit. These simulated coefficients may not have physical meaning. According to the SPICE model the varactor capacitance, CV, is a function of the applied reverse DC voltage, VR, and may be expressed as follows: C JO CV = V 1+ R VJ M +CP Note that in the Libra model, CP, is given in picoFarads, while CGO is given in Farads to comply with the default unit system used in Libra. Parameter Unit Default IS Saturation current (with N, determine the DC characteristics of the diode) Description A 1e-14 RS Series resistance Ω 0 N Emission coefficient (with IS, determines the DC characteristics of the diode) - 1 TT Transit time S 0 CJO Zero-bias junction capacitance (with VJ and M define nonlinear junction capacitance of the diode) F 0 VJ Junction potential (with VJ and M define nonlinear junction capacitance of the diode) V 1 M Grading coefficient (with VJ and M define nonlinear junction capacitance of the diode) - 0.5 EG Energy gap (with XTI, helps define the dependence of IS on temperature) EV 1.11 XTI Saturation current temperature exponent (with EG, helps define the dependence of IS on temperature) - 3 KF Flicker noise coefficient - 0 AF Flicker noise exponent - 1 FC Forward-bias depletion capacitance coefficient - 0.5 BV Reverse breakdown voltage V Infinity IBV Current at reverse breakdown voltage A 1e-3 ISR Recombination current parameter A 0 NR Emission coefficient for ISR - 2 IKF High injection knee current A Infinity NBV Reverse breakdown ideality factor - 1 0 IBVL Low-level reverse breakdown knee current A NBVL Low-level reverse breakdown ideality factor - 1 TNOM Nominal ambient temperature at which these model parameters were derived °C 27 FFE Flicker noise frequency exponent - 1 Table 1. Silicon Diode Default Values in Libra IV Part Number CJO (pF) M VJ (V) CP (pF) Ω) RS (Ω LS (nH) SMV1763-079 8.87 2.7 4.3 0.1 0.5 0.9 Table 2. SPICE Parameters for SMV1763-079 Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200325 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 7 APPLICATION NOTE • APN1015 VCO Design, Materials and Layout The VCO schematic diagram is shown in Figure 10. The circuit is powered from a 3 V voltage source. The ICC current was set at approximately 8 mA. The RF output signal is coupled from the VCO through capacitor C18 (5 pF). Band selection is accomplished D1, D2 SMV1763-079 by forward biasing either of the PIN diodes D3 or D4. Bias current is set by resistors R2, R3 and R4. In the “ON” state, the PIN diodes (SMP1320-079) have about 2 Ω RF resistance with 1 mA control current. D3, D4 SMV1320-079 VCTL: 0.5–2.5 V L1 100 nH R1 50 VCC: 3 V 8 mA VSW_460 MHz C1 100 pF C3 3p C2 8p C4 2p D1 R2 1.0 k C12 100 pF C9 100 pF R6 3.9 k L3 15 nH D2 L4 10 nH L2 100 nH C6 6p C8 3p POUT: 0 dBm C17 2p D4 C5 1.5 p L5 15 nH V2 NE68019 V1 NE68119 C11 100 pF D3 C16 100 p R7 270 R3 1.0 k R4 1.5 k R5 6.8 k C13 10 p C14 6p C10 100 pF C18 5p C15 1p R8 100 C7 100 pF VSW _600 MHz Figure 10. The Switchable VCO Circuit Diagram Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 8 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200325 Rev. A APPLICATION NOTE • APN1015 The PCB layout is shown in Figure 11. The board was made of standard, 30 mil thick, FR4 material. Figure 11. PCB Layout Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200325 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 9 APPLICATION NOTE • APN1015 The bill of materials used is given in the Table 3. Designator Value Part Number Footprint C1 100 p 0402AU101KAT 0402 AVX Manufacturer C2 8p 0402AU8R0KAT 0402 AVX C3 3p 0402AU3R0KAT 0402 AVX C4 2p 0402AU2R0KAT 0402 AVX C5 1.5 p 0402AU1R5KAT 0402 AVX C6 6p 0402AU6R0KAT 0402 AVX C7 100 p 0402AU101KAT 0402 AVX C8 3p 0402AU3R0KAT 0402 AVX C9 100 p 0402AU101KAT 0402 AVX C10 100 p 0402AU101KAT 0402 AVX C11 100 p 0402AU101KAT 0402 AVX C12 100 p 0402AU101KAT 0402 AVX C13 10 p 0402AU100KAT 0402 AVX C14 6p 0402AU6R0KAT 0402 AVX C15 1p 0402AU1R0KAT 0402 AVX C16 100 p 0402AU101KAT 0402 AVX C17 2p 0402AU2R0KAT 0402 AVX C18 5p 0402AU5R0KAT 0402 AVX D1 SMV1763-079 SMV1763-079 SC-79 Skyworks Solutions D2 SMV1763-079 SMV1763-079 SC-79 Skyworks Solutions D3 SMP1320-079 SMP1320-079 SC-79 Skyworks Solutions D4 SMP1320-079 SMP1320-079 SC-79 Skyworks Solutions L1 100 nH LL1608-FHR10 0603 TOKO L2 100 nH LL1608-FHR10 0603 TOKO L3 15 nH 0402CS-15NXJB 0402 COILCRAFT L4 10 nH 0402CS-10NXJB 0402 COILCRAFT L5 15 nH LL1005-F15NS 0402 TOKO R1 50 CR10-500J-T 0402 AVX/KYOCERA R2 1k CR10-102J-T 0402 AVX/KYOCERA R3 1k CR10-102J-T 0402 AVX/KYOCERA R4 1.5 k CR10-152J-T 0402 AVX/KYOCERA R5 6.8 k CR10-682J-T 0402 AVX/KYOCERA R6 3.9 k CR10-392J-T 0402 AVX/KYOCERA R7 270 CR10-271J-T 0402 AVX/KYOCERA R8 100 CR10-101J-T 0402 V1 NE68119 NE68119 SOT-416 NEC/CEL V2 NE68019 NE68019 SOT-416 NEC/CEL AVX/KYOCERA Table 3. The VCO’s Bill of Materials Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 10 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200325 Rev. A APPLICATION NOTE • APN1015 3.0 The measured performance of this circuit and the simulated results obtained with the model above are shown in Figures 12 and 13. The simulated results agree with the measurements confirming the validity of the VCO model. A difference of about 1 dB in simulation of the output power at low band may be attributed to the effects of higher harmonics. A more precise VCO simulation would require more accurate modeling of miscellaneous parasitic components such as pad capacitances to the ground, transmission lines and discontinuities. Note that in the simulations, the low band resonator inductor (L3 = 15 nH in Figure 10) was replaced with a 16.5 nH measured for that type of inductor in our lab. Another reason for the divergence of simulated power response from the measurements may be in the precision of the transistor models used. These models are usually derived for small-signal or relatively weak-signal amplifier applications, and may not reflect the highly nonlinear operation of a VCO. 510 2.5 500 2.0 490 1.5 480 1.0 470 0.5 460 0 450 -0.5 440 -1.0 430 -1.5 -2.0 420 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 Varactor Voltage (V) F (meas) F (simu) P (meas) POUT (simu) Frequency Deviation (MHz) Figure 12. Low Band VCO Tuning Performance 660 0 650 -0.2 640 -0.4 630 -0.6 620 -0.8 610 -1.0 600 -1.2 590 -1.4 580 -1.6 570 -1.8 560 Output Power (dBm) Both measured and simulated VCO output power variations in the tuning ranges are less than 1 dB. The measured frequency tuning sensitivity in Figures 12 and 13 are relatively linear at 27 MHz/V (low band) and 28 MHz/V (high band) in the 0.5–2.5 V range for battery applications. The simulated frequency tuning response is very similar to the measured one, although a slightly higher tuning voltage was observed in simulations for high band over the 2 V varactor bias voltage. Frequency Deviation (MHz) 520 Output Power (dBm) Measurement and Simulation Results -2.0 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 Varactor Voltage (V) F (meas) F (simu) P (meas) POUT (simu) Figure 13. High Band VCO Tuning Performance Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200325 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 11 APPLICATION NOTE • APN1015 Phase noise measurements vs. frequency offset for both low band and high band are shown in Figure 14. It shows -100 dBc/Hz in high band and -105 dBc/Hz in low-band at 10 kHz offset and -120 dBc/Hz and -126 dBc/Hz at 100 kHz offset. The 20 dB/Decade slope is fairly constant to 5 MHz. The measurements were done using a Comstron-Aeroflex PN9000 Phase Noise Test Set in the range of less than 7 MHz offset because of the 100 ns delay-line setup used. Figure 14. Measured VCO Phase Noise for Low and High Bands at VCC = 3 V, VVAR = 1.5 V List of Available Documents VCO Related Application Notes The GSM/PCS Switchable IF VCO Simulation Project Files for Libra IV. APN1004: Varactor SPICE Models for RF VCO Applications The GSM/PCS Switchable IF VCO Circuit Schematic and PCB Layout for Protel EDA Client 1998 version. The GSM/PCS Switchable IF VCO PCB Gerber Photo-plot files. APN1006: A Colpitts VCO for Wide Band (0.95 GHz 2.15 GHz) Set-Top TV Tuner Applications. APN1005: A Balanced Wide Band VCO for Set-Top TV Tuner Applications. APN1007: Switchable Dual-Band 170/420 MHz VCO For Handset Cellular Applications. An RF VCO Design for Wireless and Broadband A Differential VCO for GSM Handset Applications Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 12 July 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200325 Rev. A APPLICATION NOTE • APN1015 Copyright © 2002, 2003, 2004, 2005, Skyworks Solutions, Inc. All Rights Reserved. Information in this document is provided in connection with Skyworks Solutions, Inc. (“Skyworks”) products or services. These materials, including the information contained herein, are provided by Skyworks as a service to its customers and may be used for informational purposes only by the customer. Skyworks assumes no responsibility for errors or omissions in these materials or the information contained herein. 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Skyworks assumes no liability for applications assistance, customer product design, or damage to any equipment resulting from the use of Skyworks products outside of stated published specifications or parameters. Skyworks, the Skyworks symbol, and “Breakthrough Simplicity” are trademarks or registered trademarks of Skyworks Solutions, Inc., in the United States and other countries. Third-party brands and names are for identification purposes only, and are the property of their respective owners. Additional information, including relevant terms and conditions, posted at www.skyworksinc.com, are incorporated by reference. Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com 200325 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 13