DS6212AB 02

RT6212A/B
2A, 18V, 500kHz, ACOTTM Step-Down Converter
General Description
Features
The RT6212A/B is a high-efficiency, monolithic
synchronous step-down DC/DC converter that can
deliver up to 2A output current from a 4.5V to 18V input
supply. The RT6212A/B adopts ACOT architecture to
allow the transient response to be improved and keep
in constant frequency. Cycle-by-cycle current limit
provides protection against shorted outputs and
soft-start eliminates input current surge during start-up.
Fault conditions also include output under voltage

protection, output over current protection, and thermal
shutdown.









Package Type
J6F : TSOT-23-6 (FC)


LCD TVs
Lead Plating System
G : Green (Halogen Free and Pb Free)
Marking Information
UVP Option
H : Hiccup
RT6212AHGJ6F
0B= : Product Code
DNN : Date Code
0B=DNN
PSM/PWM
A : PSM Mode
B : PWM Mode
RT6212BHGJ6F
Note :
0A= : Product Code
DNN : Date Code
0A=DNN
Richtek products are :

Set-Top Boxes
Portable TVs
Access Point Routers
DSL Modems

RT6212A/B
0.8V  1.5% Voltage Reference
Internal Start-Up into Pre-biased Outputs
Compact Package : TSOT-23-6 Pin
Input Under-Voltage Lockout
Over-Current Protection and Hiccup
Applications

Ordering Information
Integrated 163m/86m MOSFETs
4.5V to 18V Supply Voltage Range
500kHz Switching Frequency
ACOT Control
RoHS compliant and compatible with the current
requirements of IPC/JEDEC J-STD-020.

Suitable for use in SnPb or Pb-free soldering processes.
Simplified Application Circuit
RT6212A/B
BOOT
VIN
VIN
CIN
CBOOT
L
VOUT
LX
Enable
EN
R1
GND
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DS6212A/B-02
May 2016
CFF
COUT
FB
R2
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1
RT6212A/B
Pin Configurations
(TOP VIEW)
BOOT EN
6
FB
5
4
2
3
GND LX VIN
TSOT-23-6 (FC)
Functional Pin Description
Pin No.
Pin Name
Pin Function
1
GND
System Ground. Provides the ground return path for the control circuitry and
low-side power MOSFET.
2
LX
Switch Node. LX is the switching node that supplies power to the output and
connect the output LC filter from LX to the output load.
3
VIN
Power Input. Supplies the power switches of the device.
4
FB
Feedback Voltage Input. This pin is used to set the desired output voltage via
an external resistive divider. The feedback voltage is 0.8V typically.
5
EN
Enable Control Input. Floating this pin or connecting this pin to GND can
disable the device and connecting this pin to logic high can enable the device.
6
BOOT
Bootstrap Supply for High-Side Gate Driver. Connect a 100nF or greater
capacitor from LX to BOOT to power the high-side switch.
Function Block Diagram
BOOT
VIN
VIN
PVCC
Reg
Minoff
PVCC
VIBIAS
VREF
UGATE
OC
Control
LX
Driver
LGATE
UV &OV
GND
GND LX
PVCC
Ripple
Gen.
EN
+
--
LX
Comparator
EN
VIN
On-Time
LX
FB
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DS6212A/B-02
May 2016
RT6212A/B
Operation
The RT6212A/B is a synchronous step-down converter
with advanced constant on-time control mode. Using
the ACOTTM control mode can reduce the output
capacitance and provide fast transient response. It can
minimize the component size without additional
external compensation network.
UVLO Protection
To protect the chip from operating at insufficient supply
voltage, the UVLO is needed. When the input voltage
of VIN is lower than the UVLO falling threshold voltage,
the device will be lockout.
Current Protection
Thermal Shutdown
The inductor current is monitored via the internal
switches cycle-by-cycle. Once the output voltage drops
under UV threshold, the RT6212A/B will enter hiccup
mode.
When the junction temperature exceeds the OTP
threshold value, the IC will shut down the switching
operation. Once the junction temperature cools down
and is lower than the OTP lower threshold, the
converter will autocratically resume switching.
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RT6212A/B
Absolute Maximum Ratings
(Note 1)

Supply Input Voltage --------------------------------------------------------------------------------- 0.3V to 20V

Switch Node Voltage, LX ---------------------------------------------------------------------------- 0.3V to (VIN + 0.3V)

BOOT Pin Voltage ------------------------------------------------------------------------------------ (VLX – 0.3V) to (VIN + 6.3V)

Other Pins ----------------------------------------------------------------------------------------------- 0.3V to 6V

Power Dissipation, PD @ TA = 25C
TSOT-23-6 (FC) --------------------------------------------------------------------------------------- 1.667W

Package Thermal Resistance
(Note 2)
TSOT-23-6 (FC), JA --------------------------------------------------------------------------------- 60C/W
TSOT-23-6 (FC), JC --------------------------------------------------------------------------------- 8C/W

Lead Temperature (Soldering, 10 sec.) ---------------------------------------------------------- 260C

Junction Temperature -------------------------------------------------------------------------------- 150C

Storage Temperature Range ----------------------------------------------------------------------- 65C to 150C

ESD Susceptibility
(Note 3)
HBM (Human Body Model) ------------------------------------------------------------------------- 2kV
Recommended Operating Conditions
(Note 4)

Supply Input Voltage --------------------------------------------------------------------------------- 4.5V to 18V

Ambient Temperature Range----------------------------------------------------------------------- 40C to 85C

Junction Temperature Range ---------------------------------------------------------------------- 40C to 125C
Electrical Characteristics
(VIN = 12V, TA = 25C, unless otherwise specified)
Parameter
Symbol
Test Conditions
Min
Typ
Max
4.5
--
18
Unit
Supply Voltage
VIN Supply Input Operating
Voltage
VIN
Under-Voltage Lockout
Threshold
VUVLO
Under-Voltage Lockout
Threshold Hysteresis
VUVLO
Shutdown Current
ISHDN
Quiescent Current
IQ
V
3.6
3.9
4.2
--
340
--
mV
VEN = 0V
--
--
5
µA
VEN = 2V, VFB = 0.85V
--
0.5
--
mA
--
1000
--
µS
VEN Rising
1.4
1.5
1.6
VEN Falling
1.18
1.28
1.38
Soft-Start
Soft-Start Time
Enable Voltage
Enable Voltage Threshold
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V
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DS6212A/B-02
May 2016
RT6212A/B
Parameter
Symbol
Test Conditions
Min
Typ
Max
Unit
0.788
0.8
0.812
V
--
163
--
--
86
--
High-Side Switch Current Limit ILIM_H
--
5.8
--
Low-Side Switch Valley Current
ILIM_L
Limit
2.6
3.3
--
fOSC
400
500
640
Minimum On Time
tON(MIN)
35
60
85
Minimum Off Time
tOFF(MIN)
185
240
315
--
125
--
%
--
10
--
µS
UVP Detect
45
50
55
Hysteresis
--
10
--
--
5
--
Feedback Voltage
VFB_TH
4.5V ≤ VIN ≤ 18V
High-Side On-Resistance
RDS(ON)_H
VBOOT − VLX = 4.8V
Low-Side On-Resistance
RDS(ON)_L
Feedback Voltage Threshold
Internal MOSFET
mΩ
Current Limit
A
Switching Frequency
Switching Frequency
kHz
On-Time Timer Control
nS
Output Under Voltage and Over Voltage Protections
OVP Trip Threshold
OVP Detect
OVP Propagation Delay
UVP Trip Threshold
UVP Propagation Delay
%
µS
Thermal Shutdown
Thermal Shutdown Threshold
TSD
--
150
--
Thermal Shutdown Hysteresis
TSD
--
20
--
°C
Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the
operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may affect
device reliability.
Note 2. JA is measured at TA = 25C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. The first
layer of copper area is filled JC is measured at the exposed pad of the package.
Note 3. Devices are ESD sensitive. Handling precaution recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.
Copyright © 2016 Richtek Technology Corporation. All rights reserved.
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RT6212A/B
Typical Application Circuit
RT6212A/B
3
VIN
CIN
22μF
Enable
VIN
BOOT
LX
5
6
2
FB
VOUT
L
2.2μH
EN
GND
1
CBOOT
0.1μF
R1
12k
CFF
Open
COUT
22μF
4
R2
24k
Table 1. Recommended Components Selection
VOUT (V)
R1 (k)
R2 (k)
CFF (pF)
L (H)
COUT (F)
5.0
126
24
10 to 22
4.7
22
3.3
75
24
10 to 22
3.6
22
2.5
51
24
10 to 22
3.6
22
1.8
30
24
10 to 22
2.2
22
1.5
21
24
--
2.2
22
1.2
12
24
--
2.2
22
1.0
6
24
--
2.2
22
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RT6212A/B
Typical Operating Characteristics
Efficiency vs. Output Current
100
90
90
80
80
70
VIN = 4.5V
60
VIN = 12V
50
VIN = 18V
Efficiency (%)
Efficiency (%)
Efficiency vs. Output Current
100
40
30
20
70
VIN = 4.5V
60
VIN = 12V
50
VIN = 18V
40
30
20
10
10
RT6212A, VOUT = 1.2V
0
0
0.5
1
1.5
RT6212B, VOUT = 1.2V
0
2
0
0.5
Output Current (A)
Efficiency vs. Output Current
100
90
90
2
80
70
VOUT = 5V
60
VOUT = 3.3V
Efficiency (%)
Efficiency (%)
1.5
Efficiency vs. Output Current
100
80
VOUT = 1.2V
50
40
30
20
70
VOUT = 5V
60
VOUT = 3.3V
VOUT = 1.2V
50
40
30
20
10
10
RT6212A, VIN = 12V
0
RT6212B, VIN = 12V
0
0
0.5
1
1.5
2
0
0.5
Output Current (A)
1.5
2
Output Voltage vs. Input Voltage
1.23
1.23
1.22
1.22
1.21
Output Voltage (V)
1.24
RT6212A
1.21
1
Output Current (A)
Output Voltage vs. Output Current
Output Voltage (V)
1
Output Current (A)
1.20
RT6212B
1.19
1.18
IOUT = 1A
1.20
1.19
IOUT = 2A
1.18
1.17
1.16
1.17
VIN = 12V, VOUT = 1.2V
1.15
1.16
0
0.5
1
1.5
Output Current (A)
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2
4
6
8
10
12
14
16
18
Input Voltage (V)
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RT6212A/B
Switching Frequency vs. Temperature
700
650
650
Switching Frequency (kHz)1
Switching Frequency (kHz)1
Switching Frequency vs. Output Current
700
600
550
500
450
400
350
600
550
500
450
400
350
IOUT = 1A
RT6212B
300
300
0
0.5
1
1.5
2
-50
-25
0
Output Current (A)
50
75
100
125
Feedback Voltage vs. Temperature
0.84
0.84
0.83
0.83
Feedback Voltage (V)
Feedback Voltage (V)
Feedback Voltage vs. Input Voltage
0.82
0.81
0.80
0.79
0.78
0.82
0.81
0.80
0.79
0.78
0.77
0.77
0.76
0.76
4
6
8
10
12
14
16
-50
18
-25
0
25
50
75
100
125
Temperature (°C)
Input Voltage (V)
4.5
4.0
3.5
3.0
2.5
2.0
Low-Side Switch
1.5
Inductor Valley Current Limit vs. Temperature
Inductor Valley Current Limit (A)1
Inductor Valley Current Limit vs. Input Voltage
Inductor Valley Current Limit (A)1
25
Temperature (°C)
4.5
4.0
3.5
3.0
2.5
2.0
Low-Side Switch
1.5
4
6
8
10
12
14
16
Input Voltage (V)
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18
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT6212A/B
Shutdown Current vs. Temperature
18
3.5
15
Shutdown Current (μA)1
Shutdown Current (μA)1
Shutdown Current vs. Input Voltage
4.0
3.0
2.5
2.0
1.5
VEN = 0V
1.0
4
6
8
10
12
14
16
12
9
6
3
VEN = 0V
0
18
-50
-25
0
Quiescent Current vs. Input Voltage
50
1200
1100
1100
1000
100
125
900
800
700
1000
900
800
700
600
600
VEN = 2V, VFB = 0.85V
500
VEN = 2V, VFB = 0.85V
500
4
6
8
10
12
14
16
18
-50
-25
0
Input Voltage (V)
4.0
1.6
Enable Threshold (V)
1.7
Rising
3.8
3.7
Falling
3.6
50
75
100
125
Enable Threshold vs. Temperature
4.1
3.9
25
Temperature (°C)
Input UVLO vs. Temperature
Input UVLO (V)
75
Quiescent Current vs. Temperature
1200
Quiescent Current (μA)
Quiescent Current (μA)
25
Temperature (°C)
Input Voltage (V)
3.5
1.5
Rising
1.4
1.3
Falling
1.2
1.1
3.4
1.0
-50
-25
0
25
50
75
100
Temperature (°C)
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May 2016
125
-50
-25
0
25
50
75
100
125
Temperature (°C)
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RT6212A/B
Load Transient Response
Load Transient Response
RT6212B
RT6212A
VOUT
VOUT
(100mV/Div)
(100mV/Div)
IOUT
(1A/Div)
IOUT
(1A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 0 to 2A
Time (100s/Div)
Time (100s/Div)
Voltage Ripple
Voltage Ripple
VLX
VLX
(10V/Div)
(10V/Div)
VOUT
VOUT
(50mV/Div)
(50mV/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 1A
VIN = 12V, VOUT = 1.2V, IOUT = 2A
Time (1s/Div)
Time (1s/Div)
Power On from Input Voltage
Power Off from Input Voltage
VIN
VIN
(20V/Div)
(20V/Div)
VLX
(10V/Div)
VLX
(10V/Div)
VOUT
(1V/Div)
VOUT
(1V/Div)
IOUT
(2A/Div)
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2A
Time (2.5ms/Div)
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VIN = 12V, VOUT = 1.2V, IOUT = 0 to 2A
VIN = 12V, VOUT = 1.2V, IOUT = 2A
Time (25ms/Div)
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RT6212A/B
Power On from Enable
Power Off from Enable
VEN
(5V/Div)
VEN
(5V/Div)
VLX
(10V/Div)
VLX
(10V/Div)
VOUT
(1V/Div)
VOUT
(1V/Div)
IOUT
(2A/Div)
IOUT
(2A/Div)
VIN = 12V, VOUT = 1.2V, IOUT = 2A
Time (2.5ms/Div)
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May 2016
VIN = 12V, VOUT = 1.2V, IOUT = 2A
Time (2.5ms/Div)
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RT6212A/B
Application Information
Inductor Selection
Selecting an inductor involves specifying its inductance
and also its required peak current. The exact inductor
value is generally flexible and is ultimately chosen to
obtain the best mix of cost, physical size, and circuit
efficiency. Lower inductor values benefit from reduced
size and cost and they can improve the circuit's
transient response, but they increase the inductor
ripple current and output voltage ripple and reduce the
efficiency due to the resulting higher peak currents.
Conversely, higher inductor values increase efficiency,
but the inductor will either be physically larger or have
higher resistance since more turns of wire are required
and transient response will be slower since more time
is required to change current (up or down) in the
inductor. A good compromise between size, efficiency,
and transient response is to use a ripple current (IL)
about 20% to 50% of the desired full output load
current. Calculate the approximate inductor value by
selecting the input and output voltages, the switching
frequency (f SW ), the maximum output current
(IOUT(MAX)) and estimating a IL as some percentage of
that current.
L=
VOUT   VIN  VOUT 
VIN  fSW  IL
Once an inductor value is chosen, the ripple current
(IL) is calculated to determine the required peak
inductor current.
IL =
VOUT   VIN  VOUT 
I
and IL(PEAK) = IOUT(MAX)  L
VIN  fSW  L
2
To guarantee the required output current, the inductor
needs a saturation current rating and a thermal rating
that exceeds IL(PEAK). These are minimum requirements.
To maintain control of inductor current in overload and
short circuit conditions, some applications may desire
current ratings up to the current limit value. However,
the IC's output under-voltage shutdown feature make
this unnecessary for most applications.
IL(PEAK) should not exceed the minimum value of IC's
upper current limit level or the IC may not be able to
meet the desired output current. If needed, reduce the
inductor ripple current (IL) to increase the average
inductor current (and the output current) while ensuring
that IL(PEAK) does not exceed the upper current limit
level.
For best efficiency, choose an inductor with a low DC
resistance that meets the cost and size requirements.
For low inductor core losses some type of ferrite core is
usually best and a shielded core type, although
possibly larger or more expensive, will probably give
fewer EMI and other noise problems.
Considering the Typical Operating Circuit for 1.2V
output at 2A and an input voltage of 12V, using an
inductor ripple of 0.4A (20%), the calculated inductance
value is :
L
1.2  12  1.2 
 5.4μH
12  500kHz  0.4A
The ripple current was selected at 0.4A and, as long as
we use the calculated 5.4H inductance, that should be
the actual ripple current amount. The ripple current and
required peak current as below :
IL =
1.2  12  1.2 
= 0.4A
12  500kHz  5.4μH
and IL(PEAK) = 2A  0.4A = 2.2A
2
For the 5.4H value, the inductor's saturation and
thermal rating should exceed at least 2.2A. For more
conservative, the rating for inductor saturation current
must be equal to or greater than switch current limit of
the device rather than the inductor peak current.
Input Capacitor Selection
The input filter capacitors are needed to smooth out the
switched current drawn from the input power source
and to reduce voltage ripple on the input. The actual
capacitance value is less important than the RMS
current rating (and voltage rating, of course). The RMS
input ripple current (IRMS) is a function of the input
voltage, output voltage, and load current :
IRMS = IOUT(MAX) 
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VOUT
VIN
VIN
1
VOUT
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May 2016
RT6212A/B
Ceramic capacitors are most often used because of
their low cost, small size, high RMS current ratings, and
robust surge current capabilities. However, take care
For the Typical Operating Circuit for 1.2V output and an
inductor ripple of 0.4A, with 1 x 22F output
capacitance each with about 5m ESR including PCB
when these capacitors are used at the input of circuits
supplied by a wall adapter or other supply connected
through long, thin wires. Current surges through the
inductive wires can induce ringing at the RT6212A/B
input which could potentially cause large, damaging
voltage spikes at VIN. If this phenomenon is observed,
some bulk input capacitance may be required. Ceramic
capacitors (to meet the RMS current requirement) can
be placed in parallel with other types such as tantalum,
electrolytic, or polymer (to reduce ringing and
overshoot).
trace resistance, the output voltage ripple components
are :
Choose capacitors rated at higher temperatures than
required. Several ceramic capacitors may be paralleled
to meet the RMS current, size, and height requirements
of the application. The typical operating circuit uses two
10F and one 0.1F low ESR ceramic capacitors on
VRIPPLE(ESR) = 0.4A  5m = 2mV
0.4A
 4.545mV
8  22μF  500kHz
VRIPPLE = 2mV  4.545mV  6.545mV
VRIPPLE(C) =
Output Transient Undershoot and Overshoot
In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load
steps up and down abruptly. The ACOT transient
response is very quick and output transients are
usually small.
However, the combination of small ceramic output
the input.
capacitors (with little capacitance), low output voltages
Output Capacitor Selection
(with little stored charge in the output capacitors), and
The RT6212A/B are optimized for ceramic output
capacitors and best performance will be obtained using
them. The total output capacitance value is usually
determined by the desired output voltage ripple level
and transient response requirements for sag
(undershoot on positive load steps) and soar
(overshoot on negative load steps).
Output Ripple
Output ripple at the switching frequency is caused by
the inductor current ripple and its effect on the output
capacitor's ESR and stored charge. These two ripple
components are called ESR ripple and capacitive ripple.
Since ceramic capacitors have extremely low ESR and
relatively little capacitance, both components are
similar in amplitude and both should be considered if
ripple is critical.
VRIPPLE = VRIPPLE(ESR)  VRIPPLE(C)
VRIPPLE(ESR) = IL  RESR
VRIPPLE(C) =
IL
8  COUT  fSW
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DS6212A/B-02
May 2016
low duty cycle applications (which require high
inductance to get reasonable ripple currents with high
input voltages) increases the size of voltage variations
in response to very quick load changes. Typically, load
changes occur slowly with respect to the IC's 650kHz
switching frequency.
But some modern digital loads can exhibit nearly
instantaneous load changes and the following section
shows how to calculate the worst-case voltage swings
in response to very fast load steps.
The output voltage transient undershoot and overshoot
each have two components : the voltage steps caused
by the output capacitor's ESR, and the voltage sag and
soar due to the finite output capacitance and the
inductor current slew rate. Use the following formulas
to check if the ESR is low enough (typically not a
problem with ceramic capacitors) and the output
capacitance is large enough to prevent excessive sag
and soar on very fast load step edges, with the chosen
inductor value.
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RT6212A/B
The amplitude of the ESR step up or down is a function
of the load step and the ESR of the output capacitor :
VESR _STEP = IOUT x RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,
the input-to-output voltage differential, and the
maximum duty cycle. The maximum duty cycle during a
fast transient is a function of the on-time and the
minimum off-time since the ACOTTM control scheme
will ramp the current using on-times spaced apart with
minimum off-times, which is as fast as allowed.
Calculate the approximate on-time (neglecting
parasites) and maximum duty cycle for a given input
and output voltage as :
tON =
VOUT
tON
and DMAX =
VIN  fSW
tON  tOFF(MIN)
The actual on-time will be slightly longer as the IC
compensates for voltage drops in the circuit, but we
can neglect both of these since the on-time increase
compensates for the voltage losses. Calculate the
output voltage sag as :
VSAG =
L  (IOUT )2
2  COUT   VIN(MIN)  DMAX  VOUT 
The amplitude of the capacitive soar is a function of the
load step, the output capacitor value, the inductor value
and the output voltage :
VSOAR =
L  (IOUT )2
2  COUT  VOUT
For the Typical Operating Circuit for 1.2V output, the
circuit has an inductor 5.4H and 1 x 22F output
capacitance with 5m ESR each. The ESR step is 2A
x 5m = 10mV which is small, as expected. The output
voltage sag and soar in response to full 0A-2A-0A
instantaneous transients are :
1.2V
= 200ns
12V  500kHz
200ns
and DMAX =
= 0.455
200ns  240ns
tON =
where 240ns is the minimum off time.
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VSAG 
5.4μH  (2A)2
 115mV
2  22μF  12V  0.455  1.2V 
VSOAR 
5.4μH  (2A)2
 409mV
2  22μF  1.2V
The sag is about 9.58% of the output voltage and the
soar is a full 34% of the output voltage. The ESR step
is negligible here but it does partially add to the soar,
so keep that in mind whenever using higher-ESR
output capacitors.
The soar is typically much worse than the sag in high
input, low-output step-down converters because the
high input voltage demands a large inductor value
which stores lots of energy that is all transferred into
the output if the load stops drawing current. Also, for a
given inductor, the soar for a low output voltage is a
greater voltage change and an even greater
percentage of the output voltage.
Any sag is always short-lived, since the circuit quickly
sources current to regain regulation in only a few
switching cycles. With the RT6212B, any overshoot
transient is typically also short-lived since the converter
will sink current, reversing the inductor current sharply
until the output reaches regulation again. The
RT6212A discontinuous operation at light loads
prevents sinking current so, for that IC, the output
voltage will soar until load current or leakage brings the
voltage down to normal.
Most applications never experience instantaneous full
load steps and the RT6212A/B high switching
frequency and fast transient response can easily
control voltage regulation at all times. Also, since the
sag and soar both are proportional to the square of the
load change, if load steps were reduced to 1A (from the
3A examples preceding) the voltage changes would be
reduced by a factor of almost ten. For these reasons
sag and soar are seldom an issue except in very
low-voltage CPU core or DDR memory supply
applications, particularly for devices with high clock
frequencies and quick changes into and out of sleep
modes. In such applications, simply increasing the
amount of ceramic output capacitor (sag and soar are
directly proportional to capacitance) or adding extra
bulk capacitance can easily eliminate any excessive
voltage transients.
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May 2016
RT6212A/B
In any application with large quick transients, always
calculate soar to make sure that over-voltage
protection will not be triggered. Under-voltage is not
likely since the threshold is very low (50%), that
function has a long delay (5s), and the IC will quickly
return the output to regulation. Over-voltage protection
has a minimum threshold of 125% and short delay of
10s and can actually be triggered by incorrect
component choices, particularly for the RT6212A which
does not sink current.
Feed-forward Capacitor (Cff)
The RT6212A/B are optimized for ceramic output
capacitors and for low duty cycle applications. However
for high-output voltages, with high feedback attenuation,
the circuit's response becomes over-damped and
transient response can be slowed. In high-output
voltage circuits (VOUT > 3.3V) transient response is
improved by adding a small “feed-forward” capacitor
(Cff) across the upper FB divider resistor (Figure 1), to
increase the circuit's Q and reduce damping to speed
up the transient response without affecting the
steady-state stability of the circuit. Choose a suitable
capacitor value that following below step.

Get the BW the quickest method to do transient
response form no load to full load. Confirm the
damping frequency. The damping frequency is BW.

Cff can be calculated base on below equation :
Cff 
1
2  3.1412  R1 BW  0.8
Enable Operation (EN)
For automatic start-up the low-voltage EN pin can be
connected to VIN through a 100k resistor. Its large
hysteresis band makes EN useful for simple delay and
timing circuits. EN can be externally pulled to VIN by
adding a resistor-capacitor delay (REN and CEN in
Figure 2). Calculate the delay time using EN's internal
threshold where switching operation begins (1.5V,
typical).
An external MOSFET can be added to implement
digital control of EN when no system voltage above 2V
is available (Figure 3). In this case, a 100k pull-up
resistor, REN, is connected between VIN and the EN
pin. MOSFET Q1 will be under logic control to pull
down the EN pin. To prevent enabling circuit when VIN
is smaller than the VOUT target value or some other
desired voltage level, a resistive voltage divider can be
placed between the input voltage and ground and
connected to EN to create an additional input under
voltage lockout threshold (Figure 4).
EN
VIN
REN
EN
RT6212A/B
CEN
GND
Figure 2. External Timing Control
BW
VIN
REN
100k
EN
Q1
Enable
RT6212A/B
GND
VOUT
Figure 3. Digital Enable Control Circuit
R1
Cff
VIN
FB
RT6212A/B
R2
GND
Figure 1. Cff Capacitor Setting
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May 2016
REN1
EN
REN2
RT6212A/B
GND
Figure 4. Resistor Divider for Lockout Threshold
Setting
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RT6212A/B
Output Voltage Setting
Set the desired output voltage using a resistive divider
from the output to ground with the midpoint connected
to FB. The output voltage is set according to the
following equation :
VOUT = 0.8V x (1 + R1 / R2)
VOUT
placing a small (<47) resistance between BOOT and
the external bootstrap capacitor. This will slow the
high-side switch turn-on and VLX's rise. To remove the
resistor from the capacitor charging path (avoiding poor
enhancement due to undercharging the BOOT
capacitor), use the external diode shown in Figure 6 to
charge the BOOT capacitor and place the resistance
between BOOT and the capacitor/diode connection.
5V
R1
FB
RT6212A/B
R2
GND
BOOT
RT6212A/B
0.1μF
LX
Figure 5. Output Voltage Setting
Place the FB resistors within 5mm of the FB pin.
Choose R2 between 10k and 100k to minimize
power consumption without excessive noise pick-up
and calculate R1 as follows :
R1 
R2  (VOUT  VREF )
VREF
For output voltage accuracy, use divider resistors with
1% or better tolerance.
External BOOT Bootstrap Diode
When the input voltage is lower than 5.5V it is
recommended to add an external bootstrap diode
between VIN (or VINR) and the BOOT pin to improve
enhancement of the internal MOSFET switch and
improve efficiency. The bootstrap diode can be a low
cost one such as 1N4148 or BAT54.
External BOOT Capacitor Series Resistance
The internal power MOSFET switch gate driver is
optimized to turn the switch on fast enough for low
power loss and good efficiency, but also slow enough
to reduce EMI. Switch turn-on is when most EMI occurs
since VLX rises rapidly. During switch turn-off, LX is
discharged relatively slowly by the inductor current
during the dead time between high-side and low-side
switch on-times. In some cases it is desirable to reduce
EMI further, at the expense of some additional power
dissipation. The switch turn-on can be slowed by
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Figure 6. External Bootstrap Diode
Thermal Considerations
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature.
The maximum power dissipation can be calculated by
the following formula :
PD(MAX) = (TJ(MAX)  TA) / JA
where TJ(MAX) is the maximum junction temperature,
TA is the ambient temperature, and JA is the junction to
ambient thermal resistance.
For recommended operating condition specifications,
the maximum junction temperature is 125C. The
junction to ambient thermal resistance, JA, is layout
dependent. For TSOT-23-6 (FC) package, the thermal
resistance, JA, is 60C/W on a standard four-layer
thermal test board. The maximum power dissipation at
TA = 25C can be calculated by the following formula :
PD(MAX) = (125C  25C) / (60C/W) = 1.667W for
TSOT-23-6 (FC) package
The maximum power dissipation depends on the
operating ambient temperature for fixed TJ(MAX) and
thermal resistance, JA. The derating curve in Figure 7
allows the designer to see the effect of rising ambient
temperature on the maximum power dissipation.
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DS6212A/B-02
May 2016
RT6212A/B
Layout Considerations
Maximum Power Dissipation (W)1
2.0
Four-Layer PCB
Follow the PCB layout guidelines for optimal

performance of the device.
1.5

Keep the traces of the main current paths as short
and wide as possible.

Put the input capacitor as close as possible to VIN
1.0
and VIN pins.
0.5

LX node is with high frequency voltage swing and
should be kept at small area. Keep analog
components away from the LX node to prevent stray
capacitive noise pickup.

Connect feedback network behind the output
0.0
0
25
50
75
100
125
Ambient Temperature (°C)
capacitors. Keep the loop area small. Place the
feedback components near the device.
Figure 7. Derating Curve of Maximum Power
Dissipation

Connect all analog grounds to common node and
then connect the common node to the power ground
behind the output capacitors.

For better thermal performance, to design a wide
and thick plane for GND pin or to add a lot of vias to
GND plane.

An example of PCB layout guide is shown in Figure
8 for reference.
Place the feedback as close
to the IC as possible.
VOUT
R1
BOOT EN
The only GND pin
must be connected
to a wide and thick
plane for thermal
consideration.
6
R2
FB
5
4
2
3
GND LX VIN
GND
CBOOT
CIN
LX should be connected
L
to inductor by wide and
short trace, and keep
VOUT
sensitive components
COUT
away from this trace.
Place the input and output capacitors
as close to the IC as possible.
Figure 8. PCB Layout Guide
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DS6212A/B-02
May 2016
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RT6212A/B
Outline Dimension
Dimensions In Millimeters
Symbol
Dimensions In Inches
Min.
Max.
Min.
Max.
A
0.700
1.000
0.028
0.039
A1
0.000
0.100
0.000
0.004
B
1.397
1.803
0.055
0.071
b
0.300
0.559
0.012
0.022
C
2.591
3.000
0.102
0.118
D
2.692
3.099
0.106
0.122
e
0.950
0.037
H
0.080
0.254
0.003
0.010
L
0.300
0.610
0.012
0.024
TSOT-23-6 (FC) Surface Mount Package
Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot assume
responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be accurate and
reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may
result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.
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is a registered trademark of Richtek Technology Corporation.
DS6212A/B-02
May 2016