Thermocouple Measurement

Application Note 28
February 1988
Thermocouple Measurement
Jim Williams
Introduction
Thermocouples in Perspective
In 1822, Thomas Seebeck, an Estonian physician, accidentally joined semicircular pieces of bismuth and copper
(Figure 1) while studying thermal effects on galvanic arrangements. A nearby compass indicated a magnetic disturbance. Seebeck experimented repeatedly with different
metal combinations at various temperatures, noting relative
magnetic field strengths. Curiously, he did not believe that
electric current was flowing, and preferred to describe the
effect as “thermo-magnetism.” He published his results in
a paper, “Magnetische Polarisation der Metalle und Erze
durch Temperatur-Differenz” (see references).
Temperature is easily the most commonly measured
physical parameter. A number of transducers serve temperature measuring needs and each has advantages and
considerations. Before discussing thermocouple-based
measurement it is worthwhile putting these sensors in
perspective. Figure 2’s chart shows some common contact
temperature sensors and lists characteristics. Study reveals
thermocouple strengths and weaknesses compared to
other sensors. In general, thermocouples are inexpensive,
wide range sensors. Their small size makes them fast and
their low output impedance is a benefit. The inherent voltage output eliminates the need for excitation.
Subsequent investigation has shown the “Seebeck Effect”
to be fundamentally electrical in nature, repeatable, and
quite useful. Thermocouples, by far the most common
transducer, are Seebeck’s descendants.
JUNCTION
•
COPPER
•
N
W
BISMUTH
JUNCTION
E
S
COMPASS
AN28 F01
Figure 1. The Arrangement for Dr. Seebeck’s Accidental
Discovery of “Thermo-Magnetism”
an28f
AN28-1
RANGE OF
OPERATION
–270°C to
1800°C
–100°C to
450°C
–250°C to
900°C
–270°C to
175°C
–85°C to
125°C
Typical
TYPE
Thermocouples
(All Types)
AN28-2
Thermistors
and Thermistor
Composites
Platinum
Resistance Wire
Diodes and
Transistors
Integrated Circuit
COST
Typically
Several
Seconds
1/8 to 1/4 In.
Typical. Smaller
Sizes Available
Over –55°C to
125°C
Within 1° (0.2°
from 0°C to
70°C) Typical
Several
Seconds
TO-18 Transistor
Package Size.
Also MiniDIP
±2°C to ±5°C Over Within 2° Over 1 to 10 Sec. is Standard Diode
–55°C to 125°C Operating Range Standard. Small and Transistor
Diode Packages Case Sizes. Glass
Permit Speeds Passivated Chips
in ms Range Permit Extremely
Small Sizes
±0.1°C Readily
Nearly Linear
Available. ±0.01°C
Over Large
in Precision
Spans; Typically
Standards—Lab Within 1° Over
Units
200°C Ranges
$1 to $10
Below 50¢.
Cryogenic
Units More
Expensive
Glass, Metal
Metal, Plastic
$25 to $1000
Depending On
Specs; Most
Industrial
Types Below
$100
Glass, Epoxy,
Ceramic,
Teflon, Metal,
Etc.
Figure 2. Characteristics of Some Contact Temperature Sensors (Chart Adapted from Reference 2)
0.4%/°C Typical
–2.2mV/°C
(Approx.
0.33%/°C)
Approximately
0.5%/°C
Metallic
Bead, Variety
of Probes
Available
PACKAGE
COMMENTS
Current and Voltage Outputs
Available
Require Individual
Calibration. Must be Driven
from Current Source for
Optimum Performance.
Extremely Inexpensive.
Calibrated Cryogenic Types
Available
Sets Standard for Stability
Over Long Term. Has Wider
Temperature Range Than
Thermistor, but Lower
Sensitivity
Highest Temperature
Sensitivity of Any Common
Sensor. Special Units
Required for Long-Term
Stability Above 100°C
SIZE
0.02 In. Bead
Typical. 0.0005
In. Units are
Available
Glass,
$2 to $10
≈5%/°C for
±0.1°C Standard
±0.2°C for
1 to 10 Sec. is Beads Can be as
Thermistors.
from –40°C to
Linearized
Standard; 3ms Small as 0.005 Epoxy, Teflon for Standard
Units. $10
≈0.5%/°C for
100°C; ±0.01°C Composite Units to 100ms Types In., But 0.04 to Encapsulated,
Metal
to $350 for
Linearized Units from 0°C to 60°C
Over 100°C
are Available 0.1 In. is Typical.
Available
Ranges
“Flake” Types are Housing, Etc. High Precision
Types and
Only 0.001 In.
Specials
Thick
SPEED IN
STIRRED OIL
Poor Over Wide Typically 1 Sec.
Range, Better Some Types are
Over ≈100°C
Faster
LINEARITY
Requires Reference. Low
Level Output Requires
Stable Signal Conditioning
Components
±0.5°C with
Reference
ACCURACY
$1 to $50
Depending
On Type,
Specifications
and Package
Typically Less
Than 50μV/°C
SENSITIVITY AT
25°C
Application Note 28
an28f
Application Note 28
JUNCTION MATERIALS
Copper—Constantan
Iron—Constantan
Chromel—Alumel
Chromel—Constantan
Platinum 10%—Rhodium/Platinum
Platinum 13%—Rhodium/Platinum
APPROXIMATE
SENSITIVITY IN
μV/°C AT 25°C
40.6
51.70
40.6
60.9
6.0
6.0
USEFUL TEMPERATURE RANGE (°C)
APPROXIMATE
VOLTAGE SWING OVER
RANGE
LETTER DESIGNATION
–270 to 600
–270 to 1000
–270 to 1300
–270 to 1000
0 to 1550
0 to 1600
25.0mV
60.0mV
55.0mV
75.0mV
16.0mV
19.0mV
T
J
K
E
S
R
Figure 3. Temperature vs Output for Some Thermocouple Types
Signal Conditioning Issues
Potential problems with thermocouples include low level
outputs, poor sensitivity and nonlinearity (see Figures 3
and 4). The low level output requires stable signal conditioning components and makes system accuracy difficult
to achieve. Connections (see Appendix A) in thermocouple
systems must be made with great care to get good accuracy.
Unintended thermocouple effects (e.g., solder and copper
create a 3μV/°C thermocouple) in system connections
make “end-to-end” system accuracies better than 0.5°C
difficult to achieve.
KmSCALE
2.5
1
5.0
2
JmSCALE
7.5
3
10.0
4
12.5
5
SCALEkE
15.0
6
17.5
7
SCALEkT
20.0
0
50
*A practical example of this technique appears in LTC Application Note
AN-25, “Switching Regulators for Poets.”
+
MEASUREMENT
THERMOCOUPLE
VOUTPUT =
VMEASUREMENT –
VCOLDJUNCTION
–
0
ERROR FOR TYPE J AND K (°C)
ERROR FOR TYPE E AND T (°C)
0
0°C in an ice bath. Ice baths, while inherently accurate, are
impractical in most applications. Another approach servo
controls a Peltier cooler, usually at 0°C, to electronically
simulate the ice bath (Figure 6). This approach* eliminates
ice bath maintenance, but is too complex and bulky for
most applications.
–
+
“COLD
JUNCTION”
THERMOCOUPLE
ICE BATH
(0°C)
AN28 F05
Figure 5. Ice Bath Based Cold Junction Compensator
+V
8
+
100 150 200 250 300 350 400
TEMPERATURE (°C)
SERVO
AMPLIFIER
AN28 F04
Figure 4. Thermocouple Nonlinearity for Types J, K, E and T Over
0°C to 400°C. Error Increases Over Wider Temperature Ranges
–
TEMPERATURE
SENSOR MATED TO
PELTIER COOLER
–
Cold Junction Compensation
The unintended, unwanted and unavoidable parasitic thermocouples require some form of temperature reference
for absolute accuracy. (See Appendix A for a discussion
on minimizing these effects). In a typical system, a “cold
junction” is used to provide a temperature reference
(Figure 5). The term “cold junction” derives from the
historical practice of maintaining the reference junction at
+V
+
POWER
STAGE
AN28 F06
PELTIER
COOLER
–
MEASUREMENT
THERMOCOUPLE
+
VOUTPUT =
VMEASUREMENT –
VCOLDJUNCTION
Figure 6. A 0°C Reference Based on Feedback Control of a
Peltier Cooler (Sensor is Typically a Platinum RTD)
an28f
AN28-3
Application Note 28
AMBIENT
TEMPERATURE
SENSOR
COLD
JUNCTION
THERMOCOUPLE
COLD
JUNCTION
COMPENSATION
CIRCUITRY
TEMPERATURE
TO BE
MEASURED
COPPER
THERMOCOUPLE
WIRES
WIRES
(e.g., IRON-CONSTANTAN, ETC.)
COMPENSATED
OUTPUT
AN28 F07
Figure 7. Typical Cold Junction Compensation Arrangement.
Cold Junction and Compensation Circuitry Must be Isothermal
Figure 7 conveniently deals with the cold junction requirement. Here, the cold junction compensator circuitry does
not maintain a stable temperature but tracks the cold
junction. This temperature tracking, subtractive term
has the same effect as maintaining the cold junction at
constant temperature, but is simpler to implement. It is
designed to produce 0V output at 0°C and have a slope
equal to the thermocouple output (Seebeck coefficient)
over the expected range of cold junction temperatures.
For proper operation, the compensator must be at the
same temperature as the cold junction.
Figure 8 shows a monolithic cold junction compensator
IC, the LT®1025. This device measures ambient (e.g., cold
junction) temperature and puts out a voltage scaled for
use with the desired thermocouple. The low supply current minimizes self-heating, ensuring isothermal operation
with the cold junction. It also permits battery or low power
operation. The 0.5°C accuracy is compatible with overall
achievable thermocouple system performance. Various
compensated outputs allow one part to be used with many
thermocouple types. Figure 9 uses an LT1025 and an
amplifier to provide a scaled, cold junction compensated
output. The amplifier provides gain for the difference
between the LT1025 output and the type J thermocouple.
C1 and C2 provide filtering, and R5 trims gain. R6 is a
typical value, and may require selection to accommodate
R5’s trim range. Alternately, R6 may be re-scaled, and R5
enlarged, at some penalty in trim resolution. Figure 10 is
similar, except that the type K thermocouple subtracts from
the LT1025 in series-opposed fashion, with the residue
fed to the amplifier. The optional pull-down resistor allows
readings below 0°C.
R1
10k
1%
E
60.9μV/°C
VIN
+
BUFFER
10mV/°C
OUTPUT
10mV/°C
TEMPERATURE
SENSOR
–
+
R3
1M
1%
–
C1
0.01μF
J
51.7μV/°C
K, T
40.6μV/°C
V+
R6
8.4k
V+
–
R7
6.8k
LT1025
J
GND
GND
0.5°C ACCURACY
RESISTOR
VO
10mV/°C COMMON
4V TO 36V OPERATION
80μA SUPPLY CURRENT
COMPATIBLE WITH TYPE E, J, K, R, S AND T THERMOCOUPLES
AUXILIARY 10mV/°C OUTPUT
R5 FULL-SCALE
2k ADJUST
TYPE J
VIN
R, S
6μV/°C
R4
10k
R–
C2
0.01
+
LT1001
VOUT
10mV/°C
AN28 F09
V–
AN28 F08
Figure 8. LT1025 Thermocouple Cold Junction Compensator
Figure 9. LT1025 Cold Junction Compensates a Type J
Thermocouple. The Op Amp Provides the Amplified Difference
Between the Thermocouple and the LT1025 Cold Junction Output
an28f
AN28-4
Application Note 28
R2
100Ω
FULL-SCALE
TRIM
R1
1k
1%
current exceeds 500μA. These leads can generate both DC
and AC offset terms in the presence of thermal gradients
in the package and/or external air motion.
R3
255k
1%
C2
0.1μF
V+
V+
–
VIN
+
K
–
+
LT1025
VO
GND
R–
TYPE K
R4*
C1
0.1μF
LTKA0x
VOUT
10mV/°C
AN28 F10
V–
V–
. R4 IS NOT REQUIRED (OPEN)
30μA
FOR LT1025 TEMPERATURES ≥ 0°C
*R4 ≤
V–
Figure 10. LT1025 Compensates a Type K Thermocouple. The
Amplifier Provides Gain for the LT1025-Thermocouple Difference
Amplifier Selection
The operation of these circuits is fairly straightforward,
although amplifier selection requires care.
Thermocouple amplifiers need very low offset voltage and
drift, and fairly low bias current if an input filter is used.
The best precision bipolar amplifiers should be used for
type J, K, E and T thermocouples which have Seebeck
coefficients to 40μV/°C to 60μV/°C. In particularly critical
applications, or for R and S thermocouples (6μV/°C to
15μV/°C), a chopper-stabilized amplifier is required. Linear
Technology offers two amplifiers specifically tailored for
thermocouple applications. The LTKA0x is a bipolar design
with extremely low offset (30μV), low drift (1.5μV/°C), very
low bias current (1nA), and almost negligible warm-up
drift (supply current is 400μA).
For the most demanding applications, the LTC ®1052
CMOS chopper-stabilized amplifier offers 5μV offset and
0.05μV/°C drift. Input bias current is 30pA, and gain is typically 30 million. This amplifier should be used for R and S
thermocouples, especially if no offset adjustments can be
tolerated, or where a large ambient temperature swing is
expected. Alternatively, the LTC1050, which has similar drift
and slightly higher noise can be used. If board space is at
a premium, the LTC1050 has the capacitors internally.
Regardless of amplifier type, for best possible performance
dual-in-line (DIP) packages should be used to avoid
thermocouple effects in the kovar leads of TO-5 metal
can packages. This is particularly true if amplifier supply
In many situations, thermocouples are used in high noise
environments, and some sort of input filter is required.
To reject 60Hz pick-up with reasonable capacitor values,
input resistors in the 10k to 100k range are needed. Under
these conditions, bias current for the amplifier needs to
be less than 1nA to avoid offset and drift effects.
To avoid gain error, high open-loop gain is necessary
for single-stage thermocouple amplifiers with 10mV/°C
or higher outputs. A type K amplifier, for instance, with
100mV/°C output, needs a closed-loop gain of 2,500. An
ordinary op amp with a minimum loop of 50,000 would
have an initial gain error of (2,500)/(50,000) = 5%! Although
closed-loop gain is commonly trimmed, temperature drift
of open-loop gain will have a deleterious effect on output
accuracy. Minimum suggested loop gain for type E, J, K
and T thermocouples is 250,000. This gain is adequate for
type R and S if output scaling is 10mV/°C or less.
Additional Circuit Considerations
Other circuit considerations involve protection and common mode voltage and noise. Thermocouple lines are
often exposed to static and accidental high voltages,
necessitating circuit protection. Figure 11 shows two
suggested approaches. These examples are designed
to prevent excessive overloads from damaging circuitry.
The added series resistance can serve as part of a filter.
Effects of the added components on overall accuracy
should be evaluated. Diode clamping to supply lines is
effective, but leakage should be noted, particularly when
large current limiting resistors are used. Similarly, IC bias
currents combined with high value protection resistors
can generate apparent measurement errors. Usually, a
favorable compromise is possible, but sometimes the
circuit configuration will be dictated by protection or noise
rejection requirements.
Differential Thermocouple Amplifiers
Figure 12a shows a way to combine filtering and full differential sensing. This circuit features 120dB DC common
mode rejection if all signals remain within the LTC1043
supply voltage range. The LTC1043, a switched-capacitor building block, transfers charge between the input
an28f
AN28-5
Application Note 28
+VS
RLIMIT
CIRCUITRY
OPTIONAL
FILTER
CIRCUITRY
AN28 F11
ACCEPTABLE WHERE
GROUND INTEGRITY IS
ASSURED OR FOR
BATTERY OPERATION
OPTIONAL
FILTER
OPTIONAL
FILTER
–VS
USEFUL WHERE GROUND INTEGRITY IS UNCERTAIN.
INCLUDING OPEN THERMOCOUPLE LINE
Figure 11. Input Protection Schemes
255k*
100Ω
+V
1k*
0.1μF
9.1k
VIN
OUTPUT
10mV/°C
VO
5V
–
GND
470k
R–
–15V
LTKA0x†
1/2 LTC1043
6
LT1025
+
5
–5V
2
TYPE K
1μF
1μF
* = METAL FILM
† = LTC1050 CAN BE USED
3
18
15
AN28 F12a
1M
16
0.01μF
Figure 12a. Full Differential Input Thermocouple Amplifiers
“flying” capacitor and the output capacitor. The LTC1043’s
commutating frequency, which is settable, controls rate
of charge transfer, and hence overall bandwidth. The differential inputs reject noise and common mode voltages
inside the LTC1043’s supply rails. Excursions outside
these limits require protection networks, as previously
discussed. As in Figure 9, an optional resistor pull-down
permits negative readings. The 1M resistor provides a
bias path for the LTC1043’s floating inputs. Figure 12b,
for use with grounded thermocouples, subtracts sensor
output from the LT1025.
Isolated Thermocouple Amplifiers
In many cases, protection networks and differential
operation are inadequate. Some applications require
continuous operation at high common mode voltages
with severe noise problems. This is particularly true in
industrial environments, where ground potential differences of 100V are common. Under these conditions the
thermocouple and signal conditioning circuitry must
be completely galvanically isolated from ground. This
requires a fully isolated power source and an isolated
an28f
AN28-6
Application Note 28
R2
100Ω
FULL-SCALE TRIM
R1
1k
R3
255k
0.1μF
5V
–
LTKA0x*
1/2 LTC1043
6
+
5
TYPE J
VOUT
10mV/°C
–5V
2
V+
*LTC1050 CAN BE USED
1μF
VIN
1μF
3
LT1025
J
GND
18
15
R–
AN28 F12b
16
0.01μF
Figure 12b
signal transmission path to the ground referred output.
Thermocouple work allows bandwidth to be traded for DC
accuracy. With careful design, a single path can transfer
floating power and isolated signals. The output may be
either analog or digital, depending on requirements.
Figure 13 shows an isolated thermocouple signal conditioner which provides 0.25% accuracy at 175V common
mode. A single transformer transmits isolated power and
data. 74C14 inverter I1 forms a clock (Trace A, Figure 14).
I2, I3 and associated components deliver a stretched pulse
to the 2.2k resistor (Trace B). The amplitude of this pulse
is stabilized because A1’s fixed output supplies 74C14
power. The resultant current through the 2.2k resistor drives
L1’s primary (Trace E). A pulse appears at L1’s secondary
(Trace F, Q2’s emitter). A2 compares this amplitude with
A5’s signal conditioned thermocouple voltage. To close
its loop, A2’s output (Trace G) drives Q2’s base to force
L1’s secondary (Pins 3 to 6) to clamp at A5’s output value.
Q2 operates in inverted mode, permitting clamping action
even for very low A5 outputs. When L1’s secondary (Trace
F) clamps, its primary (Trace E) also clamps. After A2
settles, the clamp value is stable. This stable clamp value
represents A5’s thermocouple related information. Inverter
I4 generates a clock delayed pulse (Trace C) which is fed
to A3, a sample-hold amplifier. A3 samples L1’s primary
winding clamp value. A4 provides gain scaling and the
LT1004 and associated components adjust offset. When
the clock pulse (Trace A) goes low, sampling ceases. When
Trace B’s stretched clock pulse goes low, the I5-I6 inverter
chain output (Trace D) is forced low by the 470k-75pF
differentiator’s action. This turns on Q1, forcing substantial
energy into L1’s primary (Trace E). L1’s secondary (Trace
F) sees large magnetic flux. A2’s output (Trace G) moves
as it attempts to maintain its loop. The energy is far too
great, however, and A2 rails. The excess energy is dumped
into the Pin 1-Pin 4 winding, placing a large current pulse
(Trace H) into the 22μF capacitor. This current pulse occurs
with each clock pulse, and the capacitor charges to a DC
voltage, furnishing the circuit’s isolated supply. When the
470k-75pF differentiator times out, the I5-I6 output goes
high, shutting off Q1. At the next clock pulse the entire
cycle repeats.
an28f
AN28-7
OUTPUT
0VDC TO 5VDC
20k
F. S. TRIM
4.99k
+
–
7
8
A3
LT398A
I4
0.02
15V
430k*
I3
–15V
75pF
I5
20k
10k
–15V
I6
430k*
50mV
TRIM
5
2
Q1
2N3906
+VREG
100k*
6.2k
2.2k*
•
•
•
4
1
6
3
Q2
2N3904
+
= FLOATING COMMON
+VISOL
(≈10V)
A2
1/2 LT1013
+VISOL
150pF
* = 1% FILM RESISTOR
= 74C14
= 1N4148
L1 = PC-SSO-32 (UTC)
22
2k
39pF
Figure 13. 0.25% Thermocouple Isolation Amplifier
LT1004
2.5V
100k*
470k
+VREG
A1
1/2 LT1013
20k
30k
15V
4.7k
470k
150pF
I2
330k
–15V
150pF
+VREG
≈10.7V
A4
1/2 LT1013
0.01
I1
+
+VREG
–
390k
A5
1/2 LT1013
+VISOL
0.1
1.2M*
4.7k (TYPICAL
SEE TEXT)
K
1k*
0.1
–
+
R–
AN28 F13
GND
LT1025
VIN
VISOL
+
+
AN28-8
–
–
15k
TYPE K
Application Note 28
an28f
Application Note 28
A = 50V/DIV
B = 50V/DIV
C = 50V/DIV
D = 50V/DIV
E = 10V/DIV
F = 10V/DIV
G = 10V/DIV
H = 50mA/DIV
HORIZ = 50μs/DIV
AN28 F14
Figure 14. Waveforms for Figure 13’s Thermocouple
Isolation Amplifier
Proper operation of this circuit relies on several considerations. Achievable accuracy is primarily limited by
transformer characteristics. Current during the clamp
interval is kept extremely low relative to transformer
core capacity. Additionally, the clamp period must also
be short relative to core capacity. The clamping scheme
relies on avoiding core saturation. This is why the power
refresh pulse occurs immediately after data transfer, and
not before. The transformer must completely reset before
the next data transfer. A low clock frequency (350Hz)
ensures adequate transformer reset time. This low clock
frequency limits bandwidth, but the thermocouple data
does not require any speed.
Gain slope is trimmed at A5, and will vary depending upon
the desired maximum temperature and thermocouple type.
The “50mV” trim should be adjusted with A5’s output at
50mV. The circuit cannot read A5 outputs below 20mV
(0.5% of scale) due to Q2’s saturation limitations.
Drift is primarily due to the temperature dependence of
L1’s primary winding copper. This effect is swamped by
the 2.2k series value with the 60ppm/°C residue partially
compensated by I3’s saturation resistance tempco. Overall tempco, including the LT1004, is about 100ppm/°C.
Increased isolation voltages are possible with higher
transformer breakdown ratings.
Figure 15’s thermocouple isolation amplifier is somewhat
more complex, but offers 0.01% accuracy and typical
drift of 10ppm/°C. This level of performance is useful
in servo systems or high resolution applications. As in
Figure 13, a single transformer provides isolated data and
power transfer. In this case the thermocouple information is width modulated across the transformer and then
demodulated back to DC. I1 generates a clock pulse
(Trace A, Figure 16). This pulse sets the 74C74 flip-flop
(Trace B) after a small delay generated by I2, I3 and associated components. Simultaneously, I4, I5 and Q1 drive L1’s
primary (Trace C). This energy, received by L1’s secondary
(Trace H), is stored in the 47μF capacitor and serves as
the circuit’s isolated supply. L1’s secondary pulse also
clocks a closed-loop pulse width modulator composed
of C1, C2, A3 and A4. A4’s positive input receives A5’s
LT1025-based thermocouple signal. A4 servo-biases C2
to produce a pulse width each time C1 allows the 0.003μF
capacitor (Trace E) to receive charge via the 430k resistor.
C2’s output width is inverted by I6 (Trace F), integrated to
DC by the 47k-0.68μF filter and fed back to A4’s negative
input. The 0.68μF capacitor compensates A4’s feedback
loop. A4 servo controls C2 to produce a pulse width that
is a function of A5’s thermocouple related output. I6’s
low loss MOS switching characteristics combined with
A3’s supply stabilization ensure precise control of pulse
width by A4. Operating frequency, set by the I1 oscillator
on L1’s primary side, is normally a stability concern, but
ratios out because it is common to the demodulation
scheme, as will be shown.
I6’s output width’s (Trace F) negative-going edge is differentiated and fed to I7. I7’s output (Trace G) drives Q3.
Q3 puts a fast spike into L1’s secondary (Trace H). “Sing
around” behavior by C1 is gated out by the diode at C2’s
positive input. Q3’s spike is received at L1’s primary, Pins 7
and 3. Q2 serves as a clocked synchronous demodulator,
pulling its collector low (Trace D) only when its base is
high and its emitter is low (e.g., when L1 is transferring
data, not power). Q2’s collector spike resets the 74C74
flip-flop. The MOS flip-flop is driven from a stable source
(A1) and it is also clocked at the same frequency as the
pulse-width modulator. Because of this, the DC average
of its Q output depends on A5’s output. Variations with
supply, temperature and I1 oscillator frequency have no
effect. A2 and its associated components extract the DC
average by simple filtering. The 100k potentiometer permits desired gain scaling. Because this scheme depends
on edge timing at the flip-flop, the delay in resetting the
0.003μF capacitor causes a small offset error. This term
is eliminated by matching this delay in the 74C74 “set”
line with the previously mentioned I2-I3 delay network.
This delay is set so that the rising edge of the flip-flop
output (Trace B) corresponds to I6’s rising edge. No such
an28f
AN28-9
AN28-10
OUT
–
+
A2
1/2 LT1013
15V
I1
+VREG
270k
0.01
12k
200k
+V R
74C74
Q
C
D
S
+VREG
I3
10k
510pF
I2
510pF
15k
LT1034
2.5V
0.68
15V
10k
100k
+
Q2
3k
+VREG
I5
I4
330
A1
1/2 LT1013
–
•
3•
6
7
2
5
•1
1k
I7
33k
100pF
0.68
47k
HP-5082-2810
10k
+
47k
+VISOL
= 1N4148
10k
† = A5 CAN ALSO BE LTC1050
= FLOATING COMMON
= 74C14
–
+
A4
1/2 LT1013
* = 1% METAL FILM TRW MAR-6
= 2N3904
0.68
+VISOL
0.003
430k
L1 = PC-SSO-32 (UTC)
+
100k
LT1034
2.5V
C1
1/2 LT1018
–
+
–
C2
1/2 LT1018
+VISOL
47k
47μF
I6
10k
+
–
A3
1/2 LT1013
Figure 15. 0.01% Thermocouple Isolation Amplifier
0.015
L1
Q3
Q1
+VISO
15V
1k*
3.8k*
+VREG
K
–
+
R–
AN28 F15
GND
LT1025
VIN
+VISOL
0.1
A5
LT1006†
+VISOL
0.1
1.2M*
(TYPICAL)
+
68k*
–
100k*
1k*
TYPE K
Application Note 28
an28f
Application Note 28
compensation is required for falling edge data because
circuit elements in this path (I7, Q3, L1 and Q2) are
wideband. With drift matched LT1034s and the specified
resistors, overall drift is typically 10ppm/°C with 0.01%
linearity.
A = 20V/DIV
B = 20V/DIV
C = 10V/DIV
D = 20V/DIV
E = 2V/DIV
Digital Output Thermocouple Isolator
F = 10V/DIV
G = 10V/DIV
H = 20V/DIV
Figure 17 shows another isolated thermocouple signal
conditioner. This circuit has 0.25% accuracy and features
a digital (pulse width) output. I1 produces a clock pulse
(Trace A, Figure 18). I2-I5 buffers this pulse and biases Q1
to drive L1. Concurrently, the 680pF-10k values provide a
differentiated spike (Trace B), setting the 74C74 flip-flop
(Trace C). L1’s primary drive is received at the secondary.
AN28 F16
HORIZ = 50μs/DIV
Figure 16. Pulse-Width-Modulation Based Thermocouple
Isolation Amplifier Waveforms
TYPE K
+VISOL
+
A1
LT1006
0.1
K
–
+VISO
I2
VIN
I3
LT1025
0.1
GND
1k*
15V
R–
330
7.5k
I4
1.5M
I5
I1
Q1
1.2M*
–
LT1004
2.5V
Q3
301k*
I10
680pF
I11 0.01
S
•1
5
+
Q
WIDTH
OUTPUT
7
+VISOL
3
74C906
0.05
POLYSTYRENE
I9
C1
1/2 LT1017
100k*
0.01
10k
+VISOL
+VISOL
Q2
15V
I8
I7
I6
33k
8•
74C74
15V
4
100pF
Q4
+
(ALL SECTIONS
PARALLELED)
+VISOL
10μF
2k
C
R
1k
D
10k
L1 = PC-SSO-19 (UTC)
100k
= 1N4148
* = 1% METAL FILM
PNP = 2N3809 DUAL
NPN = 2N3904
Q5
150pF
15V
AN28 F17
= 74C14
Figure 17. Digital Output Thermocouple Isolator
an28f
AN28-11
Application Note 28
is low. This condition occurs during data transfer, but not
during power transfer. The demodulated output (Trace H)
contains a single negative spike synchronous with C1’s
(e.g., I11’s) output transition. This spike resets the flip-flop,
providing the circuit output. The 74C74’s width output
thus varies with thermocouple temperature.
A = 20V/DIV
B = 20V/DIV
C = 20V/DIV
D = 0.05V/DIV
E = 20V/DIV
F = 20V/DIV
Linearization Techniques
G = 50V/DIV
H = 20V/DIV
It is often desirable to linearize a thermocouple-based
signal. Thermocouples’ significant nonlinear response
requires design effort to get good accuracy. Four techniques are useful. They include offset addition, breakpoints,
analog computation, and digital correction. Offset addition schemes rely on biasing the nonlinear “bow” with a
constant term. This results in the output being high at
low scale and low at high scale with decreased errors
between these extremes (Figure 19). This compromise
reduces overall error. Typically, this approach is limited
to slightly nonlinear behavior over wide ranges or larger
nonlinearity over narrow ranges.
AN28 F18
HORZ = 50μs/DIV
Figure 18. Waveforms for Digital-Output Thermocouple Isolator
The 10μF capacitor charges to DC, supplying isolated
power. The pulse received at L1’s secondary also resets
the 0.05μF capacitor (Trace D) via the inverters (I6, I7,
I8) and the 74C906 open-drain buffer. When the received
pulse ends, the 0.05μF capacitor charges from the Q2-Q3
current source. When the resultant ramp crosses C1’s
threshold (A1’s thermocouple related output voltage)
C1 switches high, tripping the I9-I11 inverter chain. I11
(Trace E) drives L1’s secondary via the 0.01μF capacitor
(Trace F). The 33k-100pF filter prevents regenerative
“sing around”. The resultant negative-going spike at L1’s
primary biases Q4, causing its collector (Trace G) to go
low. Q4 and Q5 form a clocked synchronous demodulator
which can pull the 74C74 reset pin low only when the clock
Figure 20 shows a circuit utilizing offset linearization for
a type S thermocouple. The LT1025 provides cold junction compensation and the LTC1052 chopper-stabilized
amplifier is used for low drift. The type S thermocouple
output slope varies greatly with temperature. At 25°C it
R2
100Ω
FULL-SCALE
TRIM
R1
1k
1%
R3
909k
1%
R4
2.7k
0.1μF
ERROR BEFORE OFFSETTING
15V
OUTPUT (V)
VH
V+
ERROR AFTER OFFSETTING
2
OFFSET AMPLIFIER
VIN
SIMPLE
AMPLIFIER
THERMOCOUPLE
VL
AN28 F19
Figure 19. Offset Curve Fitting
R,S
–
+
LT1025
R4
1μF 1.37M
1%
TYPE S
GND
TL T1/6
TM
T5/6 TH
TEMPERATURE (°C)
0
–
7
VOUT
V+
6
10mV/°C
LTC1052*
800°C TO 1200°C
3 +
8
V–
4
1
0.1μF
0.1μF
R5
AN28 F20
10k
OFFSET
*LTC1050 CAN BE USED
TRIM
5
VO
R–
R7
750k
–15V
R6
12k
LT1009
2.5V
Figure 20. Offset-Based Linearization
an28f
AN28-12
Application Note 28
is 6μV/°C, with an 11μV/°C slope at 1000°C. This circuit
gives 3°C accuracy over the indicated output range. The
circuit, similar to Figure 10, is not particularly unusual
except for the offset term derived from the LT1009 and
applied through R4. To calibrate, trim R5 for VOUT = 1.669
at VIN = 0.000mV. Then, trim R2 for VOUT = 9.998V at
T = 1000°C or for VIN (+ input) = 9.585mV.
Figure 21, an adaption of a configuration shown by Sheingold (reference 3), uses breakpoints to change circuit
gain as input varies. This method relies on scaling of
the input and feedback resistors associated with A2-A6
and A7’s reference output. Current summation at A8 is
linear with the thermocouple’s temperature. A3-A6 are
the breakpoints, with the diodes providing switching
when the respective summing point requires positive
bias. As shown, typical accuracy of 1°C is possible over a
0°C to 650°C sensed range.
Figure 22, derived from Villanucci (reference 8), yields
similar performance but uses continuous function analog
computing to replace breakpoints, minimizing amplifiers
and resistors. The AD538 combines with appropriate scaling to linearize response. The causality of this circuit is
similar to Figure 22; the curve fit mechanism (breakpoint
vs continuous function) is the primary difference.
Digital techniques for thermocouple linearization have
become quite popular. Figure 23, developed by Guy M.
Hoover and William C. Rempfer, uses a microprocessor fed from a digitized thermocouple output to achieve
linearization. The great advantage of digital techniques is
elimination of trimming. In this scheme a large number
of breakpoints are implemented in software.
The 10-bit LTC1091A A/D gives 0.5°C resolution over a
0°C to 500°C range. The LTC1052 amplifies and filters the
thermocouple signal, the LT1025A provides cold junction
compensation and the LT1019A provides an accurate reference. The J type thermocouple characteristic is linearized
digitally inside the processor. Linear interpolation between
known temperature points spaced 30°C apart introduces
less than 0.1°C error. The 1024 steps provided by the
LTC1091 (24 more that the required 1000) ensure 0.5°C
resolution even with the thermocouple curvature.
Offset error is dominated by the LT1025 cold junction
compensator which introduces 0.5°C maximum. Gain
error is 0.75°C max because of the 0.1% gain resistors
and, to a lesser extent, the output voltage tolerance of the
LT1019A and the gain error of the LTC1091A. It may be
reduced by trimming the LT1019A or gain resistors. The
LTC1091A keeps linearity better than 0.15°C. The LTC1052’s
5μV offset contributes negligible error (0.1°C or less).
Combined errors are typically inside 0.5°C. These errors
don’t include the thermocouple itself. In practice, connection and wire errors of 0.5°C to 1°C are not uncommon.
With care, these errors can be kept below 0.5°C.
The 20k-10k divider on CH1 of the LTC1091 provides low
supply voltage detection (the LT1019A reference requires
a minimum supply of 6.5V to maintain accuracy). Remote
location is possible with data transferred from the MCU
to the LTC1091 via the 3-wire serial port.
Figure 24 is a complete software listing* of the code
required for the 68HC05 processor. Preparing the circuit
involves loading the software and applying power. No
trimming is required.
*Including of a software-based circuit was not without attendant conscience
searching and pain on the author’s part. Hopefully, the Analog Faithful will
tolerate this transgression ...I’m sorry everybody, it just works too well!
References
1. Seebeck, Thomas Dr., “Magnetische Polarisation der
Metalle und Erze durch Temperatur-Differenz”, Abhaandlungen der Preussischen Akademic der Wissenschaften
(1822-1823), pg. 265-373.
2. Williams, J., “Designer’s Guide to Temperature Sensors”,
EDN, May 5, 1977.
3. Sheingold, D.H., “Nonlinear Circuits Handbook”, Analog
Devices, Inc., pg. 92-97.
4. “Omega Temperature Measurement Handbook”, Omega
Engineering, Stamford Connecticut.
5. “Practical Temperature Measurements”, Hewlett-Packard Applications Note #290, Hewlett-Packard.
6. Thermocouple Reference Tables, NBS Monograph 125,
National Bureau of Standards.
7. Manual on the Use of Thermocouples in Temperature
Measurement, ASTM Special Publication 470A.
8. Villanucci, Robert S., “Calculator and IC Simplify
Linearization”, EDN, January 21, 1991.
an28f
AN28-13
Application Note 28
199k*
1k*
R
R
–
0.47μF
1.0464R
+
–
15V
R
VIN
LT1025
GND
0.65R
A2
+
E
R
2.174R
+
TYPE E
+
–
R
3.726R
12.74R
R
8.826R
R
11.111R
–
+
A5
R
–
100k*
15V
R
A4
R
33.44R
OUTPUT
10mV/°C
–
+
7.918R
A8
A3
R
LT1021
10V
–
20.45R
A1
+
A6
16.30R
100k*
A7
AN28 F21
OP AMPS = 2s LT1014 QUAD
* = 0.1% METAL FILM
“R” ANNOTATED VALUES ARE IDEAL TARGET VALUES
R = 10k
= 1N4148
Figure 21. Breakpoint-Based Linearization (See Reference 3)
an28f
AN28-14
Application Note 28
C1
0.1μF
THERMOCOUPLE
TYPE
SEEBECK
COEFFICIENT
(μV/°C)
IC1
PIN
E
J
K, T
R, S
60.9
51.7
40.6
5.95
1
8
7
6
R2
98.8k
R1
1k
VR
2
+
Tm
(0°C TO
650°C)
+
–
CHROMEL
3
COPPER
V+
–
7
6
IC2
LT1097
+
VT
100Vm
4
CONSTANTAN
V–
COPPER
V+
V+
2
VCC
E
IC1
LT1025C
GND R–
NOTES:
1. ALL FIXED RESISTORS
ARE METAL FILM
2. 150 < (R4 + R5) < 200
4
1μF
1
+
1μF
–
5
VC
7
8
V–
VC
R6
1k
2
VX
VZ
3
VB
10V 4
IC3
2V 5 AD538 14
6
13
VY
R7
1k
1N914
12
11
10
1V
R5
180Ω
AN28 F22
VY = 1.513V
R3
5k
R4
16.2Ω
10 V
VOUT = m Tm
°C
VOUT = 1.513 VT0.917
THERMOCOUPLE AMPLIFIER
Figure 22. Continuous Function Linearization (See Reference 8)
9V
2
2
LT1019A-5
8
J
4
LT1025A
J TYPE
20k
+
+
10μF
VIN
–
6
0.1μF
GND
1N4148
COMMON
4
5
C0
3
7
+
1μF
6
LTC1052*
2
8
–
1μF
4
1
0.1μF
47Ω
10k
MC68HC05
LT1091A
CS
VCC
CH0
CLK
CH1
DOUT
DIN
GND
0.1μF
5V
SCK
MIS0
MOSI
SS
AN28 F23
1k
0.1%
*LTC1050 CAN BE USED
0.33μF
3.4k
1%
178k
0.1%
Figure 23. Processor-Based Linearization
an28f
AN28-15
Application Note 28
MES92L
LINEAR
DOAGAIN
SEGMENT
CHECK
ORG
FDB
ORG
FDB
ORG
FDB
ORG
FCB
ORG
OPT
STA
LDA
STA
LDA
STA
LDA
STA
JSR
NPO
JSR
LDA
STA
JSR
LDX
LDA
STA
DECX
LDA
STA
JSR
BPL
JSR
DECX
JMP
LDA
STA
INCX
LDA
STA
JSR
LDA
STA
DECX
LDA
STA
JSR
LDA
STA
JSR
LDA
STA
LDA
STA
JSR
BPL
*
TYPE J THERMOCOUPLE LINEARIZATION PROGRAM
*
WRITTEN BY GUY HOOVER LINEAR TECHNOLOGY CORPORATION
*
REV 1 10/4/87
*
N IS NUMBER OF SEGMENTS THAT THERMOCOUPLE RESPONSE IS DIVIDED INTO
*
TEMPERATURE (°C)=M•X+B
*
M IS SLOPE OF THERMOCOUPLE RESPONSE FOR A GIVEN SEGMENT
*
X IS A/D OUTPUT—SEGMENT END POINT
*
B IS SEGMENT START POINT IN DEGREES (°C • 2)
$1000
$00,$39,$74,$B0,$EE,$12B,$193,$262,$330,$397
TABLE FOR X
$1020
$85DD,$823A,$7FB4,$7DD4,$7CAF,$7BC3,$7B8A,$7C24,$7C1F,$7B3A TABLE FOR M
$1040
$00,$3C,$78,$B4,$F0,$12C,$190,$258,$320,$384
TABLE FOR B
$10FF
$13
N•2
$0100
]
$0A
LOAD CONFIGURATION DATA INTO $0A
#$00
CONFIGURATION DATA FOR PORT A DDR
$04
LOAD CONFIGURATION DATA INTO PORT A
#$FF
CONFIGURATION DATA FOR PORT B DDR
$05
LOAD CONFIGURATION DATA INTO PORT B
#$F7
CONFIGURATION DATA FOR PORT C DDR
$06
LOAD CONFIGURATION DATA INTO PORT C
HOUSEKP
INITIALIZE ASSORTED REGISTERS
CHECK
#$6F
$50
READ91
$10FF
$1000,X
$55
$1000,X
$54
SUBTRCT
SEGMENT
ADDB
DIN WORD FOR LTC1091 CH0, W/RESPECT TO GND, MSB FIRST
STORE IN DIN BUFFER
READ LTC1091
LOAD SEGMENT COUNTER INTO X
LOAD LSBs OF SEGMENT N
STORE LSBs IN $55
DECREMENT X
LOAD MSBs OF SEGMENT N
STORE MSBs IN $54
DECREMENT X
DOAGAIN
$1020,X
$54
$1020,X
$55
TBMULT
$1040,X
$55
$1040,X
$54
ADDB
#S7F
$50
READ91
#$02
$54
#$CC
$55
SUBTRCT
NOPROB
LOAD MSBs OF SLOPE
STORE MSBs IN $54
INCREMENT X
LOAD LSBs OF SLOPE
STORE LSBs IN $55
RETURNS RESULT IN $61 AND $62
LOAD LSBs OF BASE TEMP
STORE LSBs IN $55
DECREMENT X
LOAD MSBs OF BASE TEMP
DIN WORD FOR CH1
LOAD DIN WORD INTO $50
READ BATTERY VOLTAGE
LOAD MSB OF MIN BATT VOLTAGE
PUT IN MSB OF SUBTRACT BUFFER
LOAD LSB OF MIN BATT VOLTAGE
PUT IN LSB OF SUBTRACT BUFFER
COMPARE BATT VOLTAGE WITH MINIMUM
IF BATT OK GOTO NOPROB
Figure 24. Code for Processor-Based Linearization
an28f
AN28-16
Application Note 28
NOPROB
READ91
BACK91
BACK92
SUBTRCT
ADDB
TBMULT
JSR
LDA
STA
RTS
JSR
CLR
RTS
LDA
STA
LDA
BCLR
STA
TST
BPL
LDA
STA
AND
STA
TST
BPL
BSET
LDA
STA
RTS
ADDB
#$01
$56
SET BATTERY LOW FLAG
ADDB
$56
CLEAR LOW BATTERY FLAG
#$50
$0A
$50
2,$02
$0C
$0B
BACK91
$0C
$0C
#$03
$61
$0B
BACK92
2,$02
$0C
$62
BIT 0 PORT C GOES LOW (CS GOES LOW)
LOAD DIN INTO SP1 DATA REG. START TRANSFER
TEST STATUS OF SPIF
LOOP TO PREVIOUS INSTRUCTION IF NOT DONE
LOAD CONTENTS OF SPI DATA REG. INTO ACC
START NEXT CYCLE
CLEAR 6 MSBs OF FIRST DOUT
STORE MSBs IN $61
TEST STATUS OF SPIF
LOOP TO PREVIOUS INSTRUCTION IF NOT DONE
SET BIT 0 PORT C (CS GOES HIGH)
LOAD CONTENTS OF SPI DATA INTO ACC
STORE LSBs IN $62
LDA
SUB
STA
LDA
SBC
STA
RTS
LDA
ADD
STA
LDA
ADC
STA
RTS
CLR
CLR
CLR
CLR
STX
LSL
ROL
LDA
LDX
MUL
STA
STX
LDA
LDX
MUL
ADD
STA
TXA
ADC
STA
LDA
LDX
$62
$55
$62
$61
$54
$61
LOAD LSBs
SUBTRACT LSBs
STORE REMAINDER
LOAD MSBs
SUBTRACT W/CARRY MSBs
STORE REMAINDER
$62
$55
$62
$61
$54
$61
LOAD LSBs
ADD LSBs
STORE SUM
LOAD MSBs
ADD W/CARRY MSBs
STORE SUM
$68
$69
$6A
$6B
$58
$62
$61
$62
$55
$6B
$6A
$62
$54
$6A
$6A
$69
$69
$61
$55
CONFIGURATION DATA FOR SPCR
LOAD CONFIGURATION DATA
STORE CONTENTS OF X IN $58
MULTIPLY LSBs BY 2
MULTIPLY MSBs BY 2
LOAD LSBs OF LTC1091 INTO ACC
LOAD LSBs OF M INTO X
MULTIPLY LSBs
STORE LSBs IN $6B
STORE IN $6A
LOAD LSBs OF LTC1091 INTO ACC
LOAD MSBs OF M INTO X
ADD NEXT BYTE
STORE BYTE
TRANSFER X TO ACC
ADD NEXT BYTE
STORE BYTE
LOAD MSBs OF LTC1091 INTO ACC
LOAD LSBs OF M INTO X
Figure 24. Code for Processor-Based Linearization (Continued)
an28f
AN28-17
Application Note 28
NNN
HOUSEKP
MUL
ADD
STA
TXA
ADC
STA
LDA
LDX
MUL
ADD
STA
TXA
ADC
STA
LDA
BPL
LDA
ADD
STA
LDA
ADC
STA
LDA
STA
LDA
STA
LDX
RTS
BSET
BSET
RTS
$6A
$6A
$69
$69
$61
$54
$69
$69
$68
$68
$6A
NNN
$69
#$01
$69
$68
#$00
$68
$68
$61
$69
$62
$58
0,$02
2,$02
ADD NEXT BYTE
STORE BYTE
TRANSFER X TO ACC
ADD NEXT BYTE
STORE BYTE
LOAD MSBs OF LTC1091 INTO ACC
LOAD MSBs OF M INTO X
ADD NEXT BYTE
STORE BYTE
TRANSFER X TO ACC
ADD NEXT BYTE
STORE BYTE
LOAD CONTENTS OF $6A INTO ACC
LOAD CONTENTS OF $69 INTO ACC
ADD 1 TO ACC
STORE IN $69
LOAD CONTENTS OF $68 INTO ACC
FLOW THROUGH CARRY
STORE IN $68
LOAD CONTENTS OF $68 INTO ACC
STORE MSBs IN $61
LOAD CONTENTS OF $69 INTO ACC
STORE IN $62
RESTORE X REGISTER
RETURN
SET B0 PORT C
SET B2 PORT C
Figure 24. Code for Processor-Based Linearization (Continued)
APPENDIX A
Error Sources in Thermocouple Systems
Obtain good accuracy in thermocouple systems mandates
care. The small thermocouple signal voltages require
careful consideration to avoid error terms when signal
processing. In general, thermocouple system accuracy
better than 0.5°C is difficult to achieve. Major error sources
include connection wires, cold junction uncertainties,
amplifier error and sensor placement.
Connecting wires between the thermocouple and conditioning circuitry introduce undesired junctions. These
junctions form unintended thermocouples. The number of
junctions and their effects should be minimized, and kept
isothermal. A variety of connecting wires and accessories
are available from manufacturers and their literature should
be consulted (reference 4).
Thermocouple voltages are generated whenever dissimilar materials are joined. This includes the leads of
IC packages, which may be kovar in TO-5 cans, alloy 42
or copper in dual-in-line packages, and a variety of other
materials in plating finishes and solders. The net effect
of these thermocouples is “zero” if all are at exactly the
same temperature, but temperature gradients exist within
IC packages and across PC boards whenever power is
dissipated. For this reason, extreme care must be used to
ensure that no temperature gradients exist in the vicinity
of the thermocouple terminations, the cold junction compensator (e.g., LT1025) or the thermocouple amplifier. If a
gradient cannot be eliminated, leads should be positioned
isothermally, especially the LT1025 R– and appropriate
output pins, the amplifier input pins, and the gain setting
resistor leads. An effect to watch for is amplifier offset
voltage warm-up drift caused by mismatched thermocouple materials in the wire-bond/lead system of the IC
an28f
AN28-18
Application Note 28
package. This effect can be as high as tens of microvolts
in TO-5 cans with kovar leads. It has nothing to do with
the actual offset drift specification of the amplifier and can
occur in amplifiers with measured “zero” drift. Warm-up
drift is directly proportional to amplifier power dissipation.
It can be minimized by avoiding TO-5 cans, using low
supply current amplifiers, and by using the lowest possible supply voltages. Finally, it can be accommodated by
calibrating and specifying the system after a five minute
warm-up period.
A significant error source is the cold junction. The error
takes two forms. The subtractive voltage produced by the
cold junction must be correct. In a true cold junction (e.g.,
ice point reference) this voltage will vary with inability to
maintain the desired temperature, introducing error. In a
cold junction compensator like the LT1025, error occurs
with inability to sense and track ambient temperature. Minimizing sensing error is the manufacturer’s responsibility
(we do our best!), but tracking requires user care. Every
effort should be made to keep the LT1025 isothermal with
the cold junction. Thermal shrouds, high thermal capacity
blocks and other methods are commonly employed to
ensure that the cold junction and the compensation are
at the same temperature.
A final source of error is thermocouple placement. Remember that the thermocouple measures its own temperature.
In flowing or fluid systems, remarkably large errors can
be generated due to effects of laminar flow or eddy currents around the thermocouple. Even a “simple” surface
measurement can be wildly inaccurate due to thermal
conductivity problems. Silicone thermal grease can reduce
this, but attention to sensor mounting is usually required.
As much of the sensor surface as possible should be
mated to the measured surface. Ideally, the sensor should
be tightly mounted in a drilled recess in the surface. Keep
in mind that the thermocouple leads act as heat pipes,
providing a direct thermal path to the sensor. With high
thermal capacity surfaces this may not be a problem,
but other situations may require some thought. Often,
thermally mating the lead wire to the surface or coiling
the wire in the environment of interest will minimize heat
piping effects.
As a general rule, skepticism is warranted, even in the
most “obviously simple” situations. Experiment with sereral sensor positions and mounting options. If measured
results agree, you’re probably on the right track. If not,
rethink and try again.
Amplifier offset uncertainties and, to a lesser degree,
bias currents and open-loop gain should be considered.
Amplifier selection criteria is discussed in the text under
“Amplifier Selection.”
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN28-19
Application Note 28
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AN28-20
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