Bias Voltage and Current Sense Circuits for Avalanche Photodiodes

Application Note 92
November 2002
Bias Voltage and Current Sense Circuits for
Avalanche Photodiodes
Feeding and Reading the APD
Jim Williams, Linear Technology Corporation
INTRODUCTION
Avalanche photodiodes (APDs) are widely utilized in laser
based fiberoptic systems to convert optical data into
electrical form. The APD is usually packaged with a signal
conditioning amplifier in a small module. An APD receiver
module and attendant circuitry appears in Figure 1. The
APD module (figure right) contains the APD and a transimpedance (e.g., current-to-voltage) amplifier. An optical
port permits interfacing fiberoptic cable to the APD’s
photosensitive portion. The module’s compact construction facilitates a direct, low loss connection between the
APD and the amplifier, necessary because of the extremely
high speed data rates involved.
The receiver module needs support circuitry. The APD
requires a relatively high voltage bias (figure left) to
operate, typically 20V to 90V. This voltage is set by the bias
supply’s programming port. This programming voltage
may also include corrections for the APD’s temperature
dependent response. Additionally, it is desirable to monitor the APD’s average current (figure center), which indi-
cates optical signal strength. This information can be
combined with feedback techniques to maintain optical
signal strength at an optimal level. The feedback loop’s
operating characteristics can also determine if deleterious
degradation of optical components has occurred, permitting corrective measures to be taken. APD current is
typically between 100nA and 1mA, a dynamic range of
10,000:1. This measurement, which should be taken with
an accuracy inside 1%, normally must occur in the APD’s
“high side,” complicating circuit design. This restriction
applies because the APD’s anode is committed to the
receiver amplifier’s summing point.
The APD module, an expensive and electrically delicate
device, must be protected from damage under all conditions. The support circuitry must never produce spurious
outputs which could destroy the APD module. Particular
attention must be devoted to the bias supply’s dynamic
response under programming and power-up/down conditions. Finally, it is desirable to power the support circuitry
from a single 5V rail.
APD RECEIVER MODULE
TYPICALY
20V TO 90V
5VIN
APD
BIAS POWER
SUPPLY
APD
CURRENT
MONITOR
APD
BIAS
INPUT
APD
AMPLIFIER
OUTPUT
AMPLIFIER
PROGRAMMING
INPUT
OUTPUT IS GROUND
REFERRED APD BIAS CURRENT
MONITOR. APD CURRENT
TYPICALLY 100nA TO 1mA
OPTICAL
INPUT
AN92 F01
Figure 1. Avalanche Photodiode (APD) Module (Figure Right) Contains APD, Amplifier and
Optical Port. Power Supply (Figure Left) Provides APD Bias Voltage. APD Current Monitor
(Figure Center) Operates at High Common Mode Voltage, Complicating Signal Conditioning
, LTC and LT are registered trademarks of Linear Technology Corporation.
AN92-1
Application Note 92
The bias voltage and current measurement requirements
described above constitute a significant design challenge
and are addressed in the following text.
Simple Current Monitor Circuits (with Problems)
Figure 2’s straightforward approaches attempt to address
the current monitor problem. Figure 2a uses an instrumentation amplifier powered by a separate 35V rail to
measure across the 1kΩ current shunt. Figure 2b is
similar but derives its power supply from the APD bias line.
Although both approaches function, they do not meet APD
current sensing requirements. APD bias voltages can
range to 90V, exceeding the amplifier’s supply and common mode voltage limits. Additionally, the measurement’s
wide dynamic range requires the single rail powered
amplifier to swing within 100µV of zero, which is impractical. Finally, it is desirable for the amplifiers to operate
from a single, low voltage rail.
Figure 3’s circuit divides down the high common mode
current shunt voltage, theoretically permitting the 5V
powered amplifier to extract the current measurement
over a 20V to 90V APD bias range. In practice, this
arrangement introduces prohibitive errors, primarily because the desired signal is also divided down. The current
measurement information is buried in the divider resistor’s
tolerance, even with 0.01% components. The desired 1%
accuracy over a 100mA to 1nA range cannot be achieved.
Finally, although the amplifier operates from a single 5V
supply, it cannot swing all the way to zero.
VIN
10V TO 33V
1k
1%
–
990k
0.01%
A=1
BIAS OUTPUT
TO APD
5V
990k
0.01%
2.23k
1%
+
CURRENT
MONITOR OUTPUT
0mA TO 1mA = 0V TO 1V
LT1789
–
10k
0.01%
A = 100
10k
0.01%
AN92 F03
Figure 3. Dividing Down High Common Mode Voltage
Introduces Huge Errors, Even with Precision Components.
Desired 1% Accuracy Over 100nA to 1mA Current Monitor
Range Is Buried by Resistor Mismatch, Even with 0.01%
Resistors. Single Rail Powered Amplfier Cannot Swing
Close Enough to Zero. Approach Is Impractical
Carrier Based Current Monitor
Figure 4 utilizes AC carrier modulation techniques to meet
APD current monitor requirements. It features 0.4% accuracy over the sensed current range, runs from a 5V supply
and has the high noise rejection characteristics of carrier
based “lock in” measurements.
The LTC1043 switch array is clocked by its internal oscillator. Oscillator frequency, set by the capacitor at Pin 16,
is about 150Hz. S1 clocking biases Q1 via level shifter Q2.
Q1 chops the DC voltage across the 1kΩ current shunt,
modulating it into a differential square wave signal which
1N4684
3.3V
CURRENT
MONITOR OUTPUT
0mA TO 1mA = 0V TO 1V
LT1789
1k
1%
VIN
20V TO 90V
BIAS OUTPUT
TO APD
35V
+
It is clear from the preceding circuits that common circuit
approaches will not meet APD signal conditioning requirements. More sophisticated techniques are necessary.
VIN
10V TO 35V
1k
1%
BIAS OUTPUT
TO APD
10M
–
CURRENT
MONITOR OUTPUT
0mA TO 1mA = 0V TO 1V
LT1789
AN92 F02a
+
A=1
AN92 F02b
(2a)
(2b)
Figure 2. Instrumentation Amplifiers Extract Current Measurement from Modest Common Mode Voltages.
Figure 2a Requires Separate Amplifier Power and Bias Supply Connections; Figure 2b Derives Both Connections
from Single Point. Zener Level Shift Accomodates Amplifier Input Common Mode Range. Circuits Cannot
Operate from Single, Low Voltage Rail, Swing Close to Zero or Accomodate High Bias Voltages
AN92-2
Application Note 92
nating the 1kΩ shunt resistor’s voltage drop.1 Verifying
accuracy involves loading the APD bias line with 100nA to
1mA and noting output agreement.2
feeds A1 through 0.2µF AC coupling capacitors. A1’s
single-ended output biases demodulator S2, which presents a DC output to buffer amplifier A2. A2’s output is the
circuit output.
DC Coupled Current Monitor
Switch S3 clocks a negative output charge pump which
supplies the amplifier’s V – pins, permitting output swing
to (and below) zero volts. The 100k resistors at Q1
minimize its on-resistance error contribution and prevent
destructive potentials from reaching A1 (and the 5V rail) if
either 0.2µF capacitor fails. A2’s gain of 1.1 corrects for the
slight attenuation introduced by A1’s input resistors. In
practice, it may be desirable to derive the APD bias voltage
regulator’s feedback signal from the indicated point, elimi-
APD
HIGH VOLTAGE
BIAS INPUT
FOR OPTIONAL “ZERO CURRENT” FEEDBACK TO
APD BIAS REGULATOR, SEE APPENDIX A
1k*
1%
1µF
100V
1µF
100V
100k*
100k*
Figure 5’s DC coupled current monitor eliminates the
previous circuit’s trim but pulls more current from the
APD bias supply. A1 floats, powered by the APD bias rail.
The 15V zener diode and current source Q2 ensure A1
never is exposed to destructive voltages. The 1kΩ current
shunt’s voltage drop sets A1’s positive input potential. A1
balances its inputs by feedback controlling its negative
input via Q1. As such, Q1’s source voltage equals A1’s
VOUT = 20V TO 90V
TO APD
Q1
1N4690
5.6V
1M*
0.2µF
5V
–
10k
A1
LT1789
30k
+
Q2
MPSA42
0.2µF
5V
1µF
6
20k
+
2
S2
5
–
20k*
1M* –3.5V
–3.5V
20k
200k*
12
13
OUTPUT
0V TO 1V =
0mA TO 1mA
A2
LT1006
1µF
14
S1
18
5V
5V
3
15
+
* = 0.1% METAL FILM RESISTOR
1µF 100V = TECATE CMC100105MX1825
# CIRCLED NUMBERS = LTC1043 PIN NUMBER
+
S3
–3.5V TO
AMPLIFIERS
22µF
22µF
= 1N4148
= TP0610L
16
17
4
0.056µF
5V
AN92 F04
Figure 4. Lock-In Amplifier Technique Permits 1% Accurate APD Current Measurement Over 100nA to
1mA Range. APD Current Is AC Modulated by Q1, Single-Ended at A1 and Demodulated to DC by S2-A2
Note 1. See Appendix A, “Low Error Feedback Signal Derivation
Techniques,” for details.
Note 2. Appropriate high value load resistors, perhaps augmented with
a monitoring current meter, are available from Victoreen and other
suppliers. Tight resistor tolerance, while convenient, is not strictly
required, as output target value is set by current meter indication.
AN92-3
Application Note 92
positive input voltage and its drain current sets the voltage
across its source resistor. Q1’s drain current produces a
voltage drop across the ground referred 1k resistor identical to the drop across the 1kΩ current shunt and, hence,
APD current. This relationship holds across the 20V to 90V
APD bias voltage range. The 5.6V zener assures A1’s
inputs are always within their common mode operating
range and the 10M resistor maintains adequate zener
current when APD current is at very low levels.
A second output option substitutes an A-to-D converter,
providing a serial format digital output. No V – supply is
required, as the LTC2400 A-to-D will convert inputs to
(and slightly below) zero volts.
Resistors at strategic locations prevent destructive failures. The 51kΩ unit protects A1 if the APD bias line shorts
to ground. The 10k resistor limits current to a safe value
if Q1 fails and the 100k resistor serves a similar purpose
if Q2 malfunctions. As in the previous figure, APD voltage
regulator feedback may be taken at the current shunt’s
output to maintain optimal regulation.4 As stated, this
circuit does not require trimming and maintains 0.5%
accuracy. It does, however, pull current approximately
equalling the current delivered to the APD, in addition to
Q2’s collector current. This can be an issue if the APD bias
supply has restricted current capability.
Two output options are shown. A2, a chopper stabilized
amplifier, provides an analog output. Its output is able to
swing to (and below) zero because its V – pin is supplied
with a negative voltage. This potential is generated by
using A2’s internal clock to activate a charge pump which,
in turn, biases A2’s V – pin.3
APD
HIGH VOLTAGE
BIAS INPUT
1N4690
5.6V
FOR OPTIONAL “ZERO CURRENT” FEEDBACK TO
APD BIAS REGULATOR, SEE APPENDIX A
1k*
CURRENT SHUNT
10M
1k*
VOUT = 20V TO 90V
TO APD
51K
+
+
51k
1N4702
15V
1µF
A1
LT1077
–
Q1
ZVP0545A
100k
Q2
MPSA42
5V
10k
5V
1k*
Hi-Z OUTPUT
0V TO 1V = 0mA TO 1mA
LT1460
2.5V
1k*
VIN
* = 0.1% METAL FILM RESISTOR
BUFFERED OUTPUT
0mA TO 1mA = 0V TO 1V
+
1k
–
39k
CLK OUT
10µF
+
A2
LTC1150
= BAT85
Q2
2N3904
V–
100k
≈ –3.5V HERE
FO
LTC2400 SCK
A-TO-D
SDO
CS
5V
+
5V
VREF
10µF
DIGITAL
INTERFACE
OPTIONAL
DIGITAL OUTPUT
OPTIONAL BUFFERED OUTPUT
AN92 F05
Figure 5. A1-Q1 Float at High Voltage Rail, Measuring APD Current Via 1kΩ Shunt. Q1’s Ground Referred Drain
Current Provides Hi-Z Output. Buffer Options Include Analog (Figure Bottom Left) and Digital (Figure Bottom Right)
Note 3. Circuit veterans will exercise extreme wariness when
confronted with a bootstrapped biasing scheme such as this. Appendix
D, “A Single Rail Amplifier with True Zero Volt Output Swing,” should
soothe anxieties.
AN92-4
Note 4: See Appendix A, “Low Error Feedback Signal Derivation
Techniques.”
Application Note 92
APD Bias Supply
resistors prevent excessive switch current. C8 and C9,
series connected for high voltage capability, minimize
output noise. A 0V to 4.5V programming voltage results in
a corresponding 90V to 30V output (3% accuracy) with
about 2mA of current capacity.
All previous examples have been current monitors. Figure␣ 6, developed by Michael Negrete, is a high voltage APD
bias supply.5 The LT1930A switching regulator and L1
form a flyback based boost stage. The flyback events
pump a diode-capacitor network tripler, producing a high
voltage DC output. Feedback from the output via the R1R2 combination stabilizes the regulator’s operating point.
D6 and D7 protect the switch and feedback pins, respectively, from parasitic negative excursions and the 10Ω
Circuit output noise is quite low. Figure 7, taken with
500µA loading at VOUT = 50V, shows about 200µV ripple
and harmonic residue in a 10MHz bandwidth. This is
adequate for most APD receivers.6
R5
100Ω
D5
C4
1µF
50V
10Ω D4
C6
0.15µF
50V
C2
1µF
50V
VIN
SW
D6
LT1930A
SHDN
GND
VPROGRAM
0V TO 4.5V =
90VOUT TO 30VOUT
R6
1k
R2*
35.7k
+
C1
4.7µF
6.3V
R4*
5.49k
C8
0.15µF
C9
0.15µF
R1*
35.7k
FB
D7
47k
C3
1µF
50V
10Ω D2
+
5V
VOUT
30V TO 90V
D3
C5
0.15µF
50V
D1
L1
2.2µH
C7
0.1µF
100V
*1% METAL FILM RESISTORS
C1: TAIYO YUDEN JMK212BJ475MG
C2 TO C4: MURATA GRM42-2X7R105K050
C7: TAIYO YUDEN HMK316BJ104ML
C8, C9: VISHAY 695D154X9050AZ
D1 TO D7: CENTRAL SEMI CMDSH2-3TR
L1: MURATA LQH32C2R2M24
R3*
1.24k
AN92 F06
Figure 6. Boost Regulator/Charge Pump Supplies 30V to 90V APD Bias with Only 200µVP-P Noise
100µV/DIV
AC COUPLED
200ns/DIV
AN92 F07.tif
Figure 7. Figure 6’s APD Bias Supply Shows 200µVP-P Ripple and Harmonic Residue in 10MHz Bandwidth
Note 5: See Reference 7.
Note 6: Faithful noise measurements at these low levels requires
considerable care. See Appendices B and C for practical details.
AN92-5
Application Note 92
APD Bias Supply and Current Monitor
Figure 8, the Martin Configuration, combines the previous
circuit with Figure 5’s current monitor, providing a complete APD signal conditioner.7 The programmable APD
bias supply is as before, except that feedback comes via
A2. A2, sensing after the 1kΩ current shunt, isolates the
R1-R2 path loading, preventing it from influencing the
shunt’s voltage drop. A2’s action also insures tight output
regulation, despite the current shunt’s presence.8
The current monitor, borrowing from Figure 5, measures
across the 1kΩ current shunt, presenting its output in
Q1’s drain line. As shown, the output has about 1kΩ
output impedance, although either of Figure 5’s output
options may be employed.
When considering circuit operation, note that both amplifiers are powered by the charge pump’s high voltage
output, with their V – pin returned to the “2/3 VOUT” point.
This biasing permits the amplifiers to process high voltage
signals, although the voltage across them never exceeds
30V.
*1% METAL FILM RESISTORS
** 0.1% METAL FILM RESISTORS
C1: TAIYO YUDEN JMK212BJ475MG
C2 TO C4: MURATA GRM42-2X7R105K050
C7: TAIYO YUDEN HMK316BJ104ML
C8, C9: VISHAY 695D154X9050AZ
D1 TO D7: CENTRAL SEMI CMDSH2-3TR
L1: MURATA LQH32C2R2M24
VOUT
C5
0.15µF
50V
D1
–
Q1
ZVP0545A
R2*
35.7k
R1*
32.3k
+
R4*
5.49k
A1
1/2 LT1078
1/3 VOUT
FB
D7
51k
1k**
51k
C2
1µF
50V
R6
1k
10M
BIAS OUT
TO APD
30V TO 85V
+
D6
LT1930A
VPROGRAM
0V TO 4.5V =
85VOUT TO 30VOUT
C7
0.1µF
100V
1K**
C3
1µF
50V
10Ω D2
SW
GND
2/3 VOUT
D3
1N4690
5.6V
R5
100Ω
1N4148
10Ω D4
C6
0.15µF
50V
VIN
SHDN
C4
1µF
50V
+
C1
4.7µF
6.3V
Figure 9’s circuit, another complete APD bias supply and
current monitor, uses different techniques than the previous example. Advantages include 0.25% bias voltage and
current monitoring accuracy, small size and fewer high
voltage components for greater reliability. The LT1946
switching regulator and T1 form a flyback type boost
configuration. T1’s turn ratio provides voltage gain and the
high voltage flyback events are rectified and smoothed to
DC by the diode and capacitor in T1’s secondary. This DC
potential is divided down and fed back to A1. A1 compares
this signal to the APD bias programming input and sets the
LT1946’s operating point, closing a control loop. Loop
compensation is furnished by local rolloff at A1 and a lead
network across the 10M feedback resistor. This loop
establishes and maintains the APD bias output in accordance with the programming input’s value. C1, active at
VSUPPLY = 1.2V, prevents output overshoot at power turn
on by grounding the programming input command while
D5
L1
2.2µH
5V
Transformer Based APD Bias Supply and Current
Monitor
C8
0.15µF
50V
C9
0.15µF
50v
–
A2
1/2 LT1078
+
10k
51k
1k**
0V TO 1V = 0mA TO 1mA
ZOUT ≈ 1kΩ
SEE FIGURE 5 FOR
OUTPUT OPTIONS
R3*
1.24k
AN92 F06
Figure 8. Figure 5’s Current Monitor Combines with Figure 6’s Bias Supply, Providing APD Bias and
Current Measurement. A2 Buffers LT1930A’s Feedback Path Loading from Bias Supply Output, Eliminating
Current Error. Amplifiers Process 85V Signals, Although Voltage Across Them Never Exceeds 30V
Note 7: This circuit is based on work by Alan Martin.
AN92-6
Note 8: See Appendix A, “Low Error Feedback Signal Derivation
Techniques,” for further discussion.
Application Note 92
simultaneously forcing A1’s output low. This shuts off the
switching regulator and no high voltage is produced.
When power at turn on reaches ≈4V, C1 changes state and
A1’s positive input ramps to the programming voltage.
The switching regulator’s output follows this turn-on
profile and no overshoot occurs. The LT1004 clamps
spurious programming inputs beyond 2.5V, preventing
excessive high voltage outputs.9
T1’s Pin 3 will undergo increasing negative excursion with
greater APD current. Inverter A2 converts the shunt’s
negative voltage to a buffered positive output. Its gain,
scaled 1% above unity, compensates its input resistor’s
shunt loading error. Swing to zero is facilitated by returning A2’s V – pin to a small negative potential derived from
the LT1946’s VSW pin switching. The 10M-287k divider’s
current loading error is prevented from appearing in A2’s
output by a compensatory current from the APD bias
programming input. This compensating current, arriving
at A2 via the 100k-3.65k-1M network, is scaled to precisely balance out the shunt’s output portion due to the
10M-287k path’s loading error. See Appendix A for detailed discussion of this technique.
The circuit’s current monitor portion takes full advantage
of T1’s floating secondary. Here, the 1kΩ current shunt
resides in T1’s secondary return path (Pin 3), eliminating
the high common mode voltages encountered in the
previous “high side” sensed examples. Circuit ground is
declared at the shunt’s uncommitted terminal, meaning
5V
–V TO LT1077
BAV21
T1
0.01µF
1N970
1, 2
100Ω
4
1µF
100V
100k
5, 6
3
MBR0540
+
1µF
10M*
100k
0.1µF
100V
APD BIAS OUTPUT
20V TO 90V
0.01µF
1k*
+
10µF
5V
39k
VSW
10k
VSW
287k*
FB
LT1946
SD
5V
101k*
VC
SS
0.1µF
GND
100k
0.1µF
100k*
APD BIAS
PROGRAMMING
INPUT
0.55V TO 2.5V =
20V TO 90V OUT
200Ω
LT1004
2.5V
200k
3.65k*
50k
+
A1
1/2 LT1635
0.1µF
–
A2
LT1077
+
APD CURRENT
MONITOR OUTPUT
0V TO 1V = 0mA TO 1mA
–V
TO –V
AT LT1946
0.1µF
5V
100k
* = 0.1% METAL FILM RESISTOR
** = 1% METAL FILM RESISTOR
T1 = TDK SRW5EE-V01H003
NC
–
39k**
2k**
C1
1/2 LT1635
= 1N4148
= 2N3904
1M*
5V
–
2k
+
0.2V
REFERENCE
AN92 F09
Figure 9. A1 Controls LT1946 Boost Regulator to Supply 20V to 90V Bias. C1 Prevents
Output Overshoot at Power Turn-On. A2 Senses APD Current Across 1kΩ Shunt in T1’s
Output Return. Programming Input Feedforward to A2 Cancels 10M-287k Feedback
Divider’s Loading Error, Preserving Current Monitor Accuracy
Note 9: Optional circuitry allows input clamping at any desired voltage.
See Appendix E, “APD Protection Circuits.”
AN92-7
Application Note 92
Output noise for this circuit, shown in Figure 10, is about
1mVP-P in a 10MHz bandwidth. This is characteristic of
flyback regulators and somewhat higher than Figure 8’s
charge pump based arrangement. It is still acceptable for
most APD receivers, although special switching regulator
techniques (read on!) can considerably reduce this figure.
stabilizes the output, which may be varied by appropriate
biasing at the VPROGRAM input. Components at the LT1946
VC pin compensate the loop. Over a 20V to 90V output
range, the circuit remains within 2% of the VPROGRAM
input dictated output voltage. Figure 12 shows switching
related output noise is about 1.3 millivolts peak-to-peak in
a 10MHz bandwidth.
5V
L1
6.8µH
Q1
5V
500µV/DIV
BAV21
100Ω
1µF
100V
APD BIAS
OUTPUT
20V TO 90V
1µF
100V
1µF
1N4148
MBR0540
5V
1M*
V+
SD
5V
200ns/DIV
LT1946
SS
AN92 F08.tif
0.1µF
Figure 10. Figure 9’s Output Noise
Measures 1mVP-P in 10MHz Bandwidth
VSW
VC
Figure 11 borrows from Figure 9’s flyback technique to
form a simple, small area APD bias supply. Figure 9’s
current monitor function has been deleted—this circuit
provides only the bias supply. Additionally, Figure 9’s
transformer has been replaced with a 2-terminal inductor.
The circuit is a basic inductor flyback boost regulator with
a single important deviation. Q1, a high voltage device, has
been interposed between the LT1946 switching regulator
and the inductor. This permits the regulator to control Q1’s
high voltage switching without undergoing high voltage
stress. Q1, operating as a “cascode” with the LT1946’s
internal switch, withstands L1’s high voltage flyback
events.10 Diodes associated with Q1’s source terminal
clamp L1 originated spikes arriving via Q1’s junction
capacitance. The high voltage is rectified and filtered to
DC, forming the circuit’s output. Feedback to the regulator
GND
43k
1500pF
Inductor Based APD Bias Supply
FB
14k*
VPROGRAM
0V TO 1V = 90VOUT TO 20VOUT
= 1% METAL FILM RESISTOR
= ZETEX ZTN4424
= SUMIDA CDRH4D28-6R8
= TECATE CMC100105MX1825
AN92 F09
Figure 11. Q1 Cascoded with LT1946 Switches L1,
Providing 20V to 90V APD Bias Output. Q1’s Source Diodes
Clamp Parasitic Conducted Spikes to Safe Levels
500µV/DIV
200ns/DIV
AN92 F12.tif
Figure 12. Cascode Based Bias Supply Noise
in 10MHz Bandwidth Is About 1.3mVP-P
Note 10: See Reference 11.
AN92-8
*
Q1
L1
1µF 100V
Application Note 92
controlled by resistors at the RCSL and RVSL pins, respectively. In all other respects, the circuit behaves as a
classical push-pull, transformer based, step-up converter.
The VPROGAM input biases a feedback loop, setting the
output anywhere between 20V and 90V.
200µV Output Noise APD Bias Supply
Some APD receiver applications require extremely low
noise in an extended bandwidth. Figure 13’s APD bias
supply uses special switching regulator techniques to
achieve 200µV noise in a 100MHz bandwidth. The LT1533
is a “push-pull” output switching regulator with controllable switch transition times. Output harmonic content
(“noise”) is notably reduced with slower switch transition
times.11 Switch current and voltage transition times are
The controlled transition times result in a dramatic decrease in output noise. Figure 14 shows ripple and switching related residue of 200µV in a 100MHz bandwidth. This
is far below conventional regulators, meeting the most
stringent noise requirement.
5V
16k
RT
VIN
T1
COL A
510pF
1
CT
L3
22nH
LT1533
0.02µF
PGND
5V
VC
5
L2
33µH
2
L1
33µH
+
22µF
4
24k
1µF
100V
8
20V TO 90V
BIAS OUT TO APD
1µF
100V
RVSL COL B
24k
133k*
RCSL
FB
GND
VPROGRAM INPUT
0VIN TO 4VIN =
90VOUT TO 20VOUT
*
1µF 100V
L1, L2
L3
T1
2.49k*
7.5k*
= 1% METAL FILM RESISTOR
= TECATE CMC100105MX1825
= COOPER SD12-330
= COILCRAFT B07T
= COOPER CTX-02-16004
= BAV21
AN92 F13
Figure 13. Transformer Coupled 20V to 90V APD Bias Supply Controls Switch Transition Time for Extremely Low Output Noise
200µV/DIV
2µs/DIV
AN92 F14.tif
Figure 14. LT1533’s Controlled Transition Times Achieve Spectacularly Low Output Harmonic Residue.
Switching Related Noise Is Below 100µV, Fundamental Ripple About 200µV. Measurement Bandwidth is 100MHz
Note 11: Noise contains no regularly occurring or coherent
components. As such, switching regulator output “noise” is a
misnomer. Unfortunately, undesired switching related components in
the regulated output are almost always referred to as “noise.”
Accordingly, although technically incorrect, this publication treats all
undesired output signals as “noise.” See Reference 2.
AN92-9
Application Note 92
Low Noise APD Bias Supply and Current Monitor
Figure 15 builds on the previous circuit’s performance,
forming a complete, high performance APD signal conditioner. The bias supply is identical to Figure 13’s low noise
example, with the addition of the A1 based feedback
buffer. This stage, similar to the one in Figure 8, isolates
the regulator’s feedback path current from the 1kΩ shunt,
preserving current monitor accuracy. A1’s zener-current
source power biasing scheme permits it to process high
voltage signals even though it is a low voltage device.12
The current monitor, shown in block form, may be selected from the choices indicated depending upon requirements.
0.02% Accuracy Current Monitor
Some APD current monitor applications call for high
accuracy and stability. Figure 16’s unusual optical switching based approach achieves 0.02% accuracy over a
sensed 100nA to 1µA range. This scheme measures shunt
current by switching (S1A, S1B) a capacitor across the
shunt (“ACQUIRE”). After a time the capacitor charges to
the voltage across the shunt. S1A and S1B open and S2A
and S2B close (“READ”). This grounds one capacitor plate
and the capacitor discharges into the grounded 1µF unit at
S2B. This switching cycle is continuously repeated, resulting in A1’s ground referred positive input assuming the
same voltage that is across the floating 1kΩ shunt. The
LED driven MOSFET switches specified do not have junction potentials and the optical drive contributes no charge
injection error. A nonoverlapping clock prevents simultaneous conduction in S1 and S2, which would result in
charge loss, causing errors and possible circuit damage.
The 5.1V zener prevents switched capacitor failure if the
bias output is shorted to ground.
A1, a chopper stabilized amplifier, has a clock output. This
clock, level shifted and buffered by Q3, drives a logic
divider chain. The first flip-flop activates a charge pump,
pulling A1’s V – pin negative, permitting amplifier swing to
(and below) zero volts.13 The divider chain terminates into
a logic network. This network provides phase opposed
charging of the 0.02µF capacitors (Traces A and B, Figure␣ 17). The gating associated with these capacitors is
arranged so the logic provides nonoverlapping, complementary biasing to Q1 and Q2. These transistors supply
this nonoverlapping drive to the S1 and S2 actuating LEDs
(Traces C and D).
5V
16k
RT
VIN
COL A
510pF
1
CT
L3
22nH
LT1533
0.02µF
APD CURRENT MONITOR
SEE FIGURES 4, 5, 16, 18
AND 19 FOR OPTIONS
T1
PGND
5V
VC
5
L2
33µH
2
4
1µF
100V
8
20V TO 90V
APD BIAS
OUT
1k
+
22µF
24k
1N4670
5.6V
L1
33µH
10M
1µF
100V
51k
RVSL COL B
24k
1N969
22V
133k*
RCSL
0.1µF
FB
GND
VPROGRAM INPUT
0VIN TO 4VIN =
90VOUT TO 20VOUT
–
2.49k*
7.5k*
+
A1
LT1077
*
1µF 100V
L1, L2
L3
T1
= 1% METAL FILM RESISTOR
= TECATE CMC100105MX1825
= COOPER SD12-330
= COILCRAFT B07T
= COOPER CTX-02-16004
= BAV21
100k
Q1
MPSA42
5V
16k
AN92 F15
Figure 15. Figure 13 Augmented with Feedback Divider Buffer and
Current Monitor Provides Complete 100µV Noise APD Signal Conditioner
Note 12: The feedback buffer is considered in detail in Appendix A,
“Low Error Feedback Signal Derivation Techniques.”
AN92-10
Note 13: This scheme, a variant of the one described back in Figure 5,
is detailed in Appendix D, “A Single Rail Amplifier with True Zero Volt
Output Swing.”
Application Note 92
The extremely small parasitic error terms in the LED driven
MOSFET switches results in nearly theoretical circuit
performance. However, residual error (≈0.1%) is caused
by S1A’s high voltage switching pumping S2B’s 3pF to
4pF junction capacitance. This results in a slight quantity
of unwanted charge being transferred to the 1µF capacitor
at S2B. The amount of charge transferred varies with the
APD bias voltage (20V to 90V) and, to a lesser extent, the
varactor-like response of S2B’s off-state capacitance.
These terms are partially cancelled by DC feedforward to
A1’s negative input and AC feedforward from Q1’s gate to
S2B. The corrections compensate error by a factor of five,
resulting in 0.02% accuracy.
Optical switch failure could expose A1 to high voltage,
destroying it and possibly presenting destructive voltages
to the 5V rail. This most unwelcome state of affairs is
prevented by the 47k resistors in A1’s positive input.
FOR OPTIONAL “ZERO CURRENT” CONNECTION TO
APD BIAS REGULATOR, SEE APPENDIX A
1k
0.01%
20VIN TO 90VIN
BIAS OUT TO APD
1N4689
5.1V
5.1k
5V
Q3
2N3906
1µF
“ACQUIRE” TO S1A AND S1B
S1A
S1B
100k
47k
“READ” TO S2A AND S2B
S2A
S2B
1µF
47k
1µF
+
10k
CLKOUT
IOUT
0V TO 1V =
0mA TO 1mA
A1
LTC1150
–
10Ω
750k
10µF
+
47k
10µF
5V
+
Q1
TP0610L
“ACQUIRE”
(S1)
≈ –3.5V
HERE
BAT85s
150pF
Q
130k
130k
0.02µF
Q
5V
1/2
74C74
÷2
≈140Hz
1/2
74C74
÷2
74C90
÷ 10
Q2
TP0610L
0.02µF
“READ”
(S2)
1µF = PANASONIC ECP-U1C105MA5
S1, S2 = OPTICALLY DRIVEN MOSFETS. AROMAT AQW227NA (DUAL)
FOR VOLTAGES <80V, USE AQS225SX (QUAD, SO-16 PACKAGE)
= 1/4 74C02
510Ω
AN92 F16
Figure 16. A 0.02% Accurate APD Current Monitor Utilizes Optically Driven FETs and Flying Capacitor. Logic
Driven Q1-Q2 Provides Nonoverlapping Clocking to S1-S2 LEDs. Clock Derives from A1’s Internal Oscillator
A = 5V/DIV
B = 5V/DIV
C = 5mA/DIV
D = 5mA/DIV
2ms/DIV
AN92 F17.tif
Figure 17. Clocked, Cross Coupled Capacitors (Traces A and B) in 74C02 Based
Network Result in Nonoverlapping Drive (Traces C and D) to S1-S2 Actuating LEDs
AN92-11
Application Note 92
Digital Output 0.09% Accuracy Current Monitor
This circuit’s 0.09% accuracy does not equal the previous
analog ouput’s version because of the LT1460 reference’s
0.075% tolerance, which is not trimmable. The circuit can
be adjusted to 0.02% accuracy by trimming the 1kΩ shunt
so measured output current directly corresponds to A-to-D
output.
Figure 18 modifies the optically based current monitor to
supply a digital output. The schematic is essentially identical to Figure 16’s, with two significant differences. Here,
a digital output is supplied via the LTC2431 A-to-D converter. The converter’s differential inputs allow the same
feedforward based error correction used in the previous
example. The divider chain countdown ratio has changed
to accomodate a higher speed clock, sourced by the
LTC1799 oscillator. This higher speed clock, which times
A-to-D operation, centers the A-to-D’s internal notch filter
at the optical switches commutation frequency, maximizing rejection.14
Digital Output Current Monitor
Previous current monitor examples furnish digital output
from ground referenced A-to-D converters fed from analog level shifting stages. Figure 19 directly digitizes shunt
current by floating the A-to-D converter in the APD bias
line. The A-to-D output is level shifted in the digital
FOR OPTIONAL “ZERO CURRENT” FEEDBACK CONNECTION TO
APD BIAS REGULATOR, SEE APPENDIX A
1k
0.01%
20VIN TO 90VIN
BIAS OUT TO APD
1N4689
5.1V
5.1k
1µF
“ACQUIRE” TO S1A AND S1B
S1A
S1B
5V
33k
“READ” TO S2A AND S12B
S2A
S2B
1µF
33k
750k
10Ω
47k
5V
V+
VIN +
LT1460
2.5V
VREF
LTC2431
A-TO-D
SDO
SCK
VIN –
CS
GND
150pF
DIGITAL
INTERFACE
FO
≈179kHz
5V
TP0610L
Q
“ACQUIRE”
(S1)
130k
130k
0.02µF
Q
5V
≈140Hz
LTC1799
74C90
÷ 10
CD4024
÷ 128
OUT
RSET VIN GND
56.2k*
TP0610L
“READ”
(S2)
510Ω
1/2
74C74
÷2
5V
0.02µF
* = 1% METAL FILM RESISTOR
1µF = PANASONIC ECP-U1C105MA5
S1, S2 = OPTICALLY DRIVEN MOSFETS. AROMAT AQW227NA (DUAL)
FOR VOLTAGES <80V, USE AQS225SX (QUAD, SO-16 PACKAGE)
= 1/4 74C02
AN92 F18
Figure 18. Figure 16’s Optically Driven FET Based Current Monitor Modified for Digital Output.
LTC1799 Clocks A-to-D and Optical Switch LEDs. 0.09% Accuracy, Trimmable to 0.02%
Note 14: The LTC2431’s internal digital filter’s first null occurs at
1/2560 of the frequency applied to its FO pin. For details, see the
LTC2431 data sheet.
AN92-12
Application Note 92
domain, presenting ground referred digital data. This
simple approach is attractive, although the available APD
bias supply must supply about 3mA to the A-to-D and its
attendant circuitry.
ground referred data. One of the identical stages is shown;
the other indicated in conceptual form. The stage is
designed for low quiescent and dynamic current consumption while maintaining data fidelity. This is necessary
to minimize current drain from the APD bias supply and to
avoid modulating it with transient loading artifacts. High
voltage common emitter Q1 sources current to Q2, which
provides a ground referred logic compatible output. Capacitive feedforwards maintain data edge speed while
minimizing standing current requirements.
The LTC2410 and its LT1029 reference are powered
directly from the high voltage APD bias supply input.
Current sink Q3 and the LT1029 bias the LTC2410 V – pin,
maintaining 5V across the A-to-D over the 20V to 90V bias
rail range. The A-to-D’s differential inputs measure across
the 1kΩ current shunt. Resistors and a zener clamp
protect the A-to-D from excessive voltages if the APD bias
line is shorted to ground. The A-to-D’s digital outputs,
floating at high voltage, drive level shifts which provide
This circuit’s 0.25% untrimmed accuracy is due to shunt
and LT1029 tolerances. Trimming the LT1029 (see schematic note) permits 0.05% accuracy.
FOR OPTIONAL “ZERO CURRENT” FEEDBACK CONNECTION TO
APD BIAS REGULATOR, SEE APPENDIX A
1k
0.05%
VIN
20V TO 90V
BIAS OUT TO APD
5.1k
5.1k
1N4689
5.1V
–
VREF– VREF
V+
LTC2410
CS
10µF
LT1029
5V
TRIM
(SEE
NOTES)
1M
+
SDO
SCK
OUT
V
–
47k
SCK
5V
LEVEL SHIFT
IDENTICAL
TO Q1-Q2
STAGE
Q3
MPSA42
2.2k
470pF
10k
5V
Q1
2N5400
1k
SDO
OUT
LT1029 TRIM PERMITS
HIGHER ACCURACY.
SEE DATA SHEET
33k
5pF
33k
Q2
2N2369
33k
AN92 F19
Figure 19. A-to-D Converter Floats at High Voltage, Forming Digital Output Current Monitor.
Q1-Q2 Level Shift Provides Ground Referenced Digital Output. 0.25% Accuracy Is Trimmable to 0.05%
AN92-13
Application Note 92
Digital Output Current Monitor and APD Bias Supply
operating power from the APD supply. Resistive current
limiting and the 5.1V zener protect the A-to-D from high
voltage if the APD bias output is shorted to ground. Low
power optoisolators provide ground referred digital output while eliminating floating supply “starve out” due to
cross regulation interaction with the APD regulation loop.
Specifically, very low power APD bias outputs could result
in insufficient transformer flux to furnish floating supply
loading requirements. Common optoisolators require significant current, mandating the low power types specified.
The previous circuit’s discrete level shift stage would draw
even less power but the optoisolators are simple and
adequate.
Figure 20 also floats an A-to-D converter across the shunt,
while including an APD bias supply. The bias supply is
derived from the LT1946 switching regulator and Q1,
operating in nearly identical fashion to Figure 11’s circuit.
The primary difference is that Figure 11’s inductor is
replaced here with a transformer. The transformer’s primary winding furnishes high voltage step-up, similar to
Figure 11. The floating secondary drives an isolated LT1120
based 3.75V regulator. This floating regulator’s output,
stacked on top of the APD bias line, powers the LTC2400
A-to-D converter. The isolated 3.75V supply permits the Ato-D to measure across the 1kΩ shunt without pulling
0.001µF
1M*
510k*
FB
VIN
5V
T1
2
1
OUT
LT1120A
1N4148
GND
+
4
20Ω
REF 2.5V
SEE NOTES
1µF
1µF
100V
1µF
100V
APD BIAS OUTPUT
15V TO 70V
5.1k
1N4148
INPUT
33k
10µF
1N4689
5.1V
1k
1%
100Ω
MBR0520
+
10µF
BAV21
5V
+
10µF
35V
3
Q1
ZVN4424
3.75V REFERRED TO
T1, PIN 3
1M*
V+
GND
LTC2400
A-TO-D
200pF
SDO
VSW
GND
FB
LT1946
5V
SHDN SS
0.1µF
CS
REF
FO
SCK
5V
18.3k*
SCK
VC
43k
VPROGRAM
0V TO 1V = 70V TO 15V
1500pF
4.7k
5V
4.7k
*
1µF 100V
OPTOISOLATORS
T1
=
=
=
=
1% METAL FILM RESISTOR
TECATE CMC100105MX1825
AGILENT HCPL-2300
COOPER CTX02-16003X1
FOR 0.1% ACCURACY, USE LT1460 2.5V REFERENCE FOR LTC2400 AND
CHANGE 1k SHUNT TOLERANCE TO 0.025%
FOR OPTIONAL “ZERO CURRENT” FEEDBACK CONNECTION, SEE APPENDIX A
SDO
AN92 F20
Figure 20. APD Bias Supply with Digital Output Current Monitor. T1’s Primary Supplies APD High Voltage Source, Similar to
Figure␣ 11; Secondary Furnishes Power to Floating Circuitry. 1kΩ Shunt Voltage Drop Is Compensatible Using Optional Feedback
Circuitry. Optoisolators Provide Ground Referred Digital Output. Current Monitor Accuracy is 2%, Trimmable to 0.1%
AN92-14
Application Note 92
The LT1120 2.5V reference and 1kΩ shunt tolerances
dictate 2% circuit accuracy. If the tighter tolerance components noted in the schematic are used, 0.1% accuracy
is practical.
BIAS
FIGURE
SUPPLY
NUMBER CAPABILITY
ANALOG
OUTPUT
CURRENT
MONITOR
(100nA to 1mA)
DIGITAL
OUTPUT
CURRENT
MONITOR
(100nA to 1mA)
Summary
Figure 21’s chart is an attempt to summarize the circuits
presented, although such brevity breeds oversimplification. As such, although the chart reviews salient features,
there is no substitute for a thorough investigation of any
particular application’s requirements.
COMMENTS
4
No
Yes
No
0.4% Accuracy. High Noise Rejection
5
No
Yes
Yes
0.5% Accuracy. Draws Current from APD Bias Supply Approximately Equalling Current
Delivered to the APD in Addition to Housekeeping Current
6
Yes
30V to 90V
No
No
200µV Noise in 10MHz Bandwidth. 3% Accuracy
8
Yes
30V to 85V
Yes
No
3% Bias Voltage Accuracy. 0.5% Current Monitor Accuracy.
Current Monitor Has 1kΩ Output Impedance
9
Yes
20V to 90V
Yes
No
0.25% Bias Voltage Accuracy. 1mV Output Noise in 10MHz Bandwidth.
0.25% Current Monitor Accuracy. Small Size. Few Large Value, High Voltage
Capacitors Improves Reliability. Low Current Drain from APD Rail Permits Smaller
High Voltage Capacitors for a Given Ripple Level
11
Yes
20V to 90V
No
No
2% Bias Voltage Accuracy. 1.5mV Output Noise in 10MHz Bandwidth.
Small Size, Simple
13
Yes
20V to 90V
No
No
2% Bias Voltage Accuracy. 200µV Ripple and Noise in 100MHz Bandwidth.
Relatively Large Solution Size Due to 250kHz Oscillator Frequency
15
Yes
20V to 90V
Yes
Yes
2% Bias Voltage Accuracy. 200µV Ripple and Noise in 100MHz Bandwidth.
Current Monitor Accuracy Depends on Option Selected. Relatively Large Solution Size
Due to 250kHz Oscillator Frequency
16
No
Yes
No
0.02% Accuracy. Low Current Drain from APD Rail Permits Smaller High Voltage
Capacitors for a Given Ripple Level
18
No
No
Yes
0.09% Accuracy. 0.02% Achievable with Shunt Trimming. Low Current Drain from
APD Rail Permits Smaller High Voltage Capacitors for a Given Ripple Level
19
No
No
Yes
0.25% Accuracy. Trimmable to 0.05% by Adjusting Reference
20
Yes
15V to 70V
No
Yes
2% Bias Voltage Accuracy. 2% Current Monitor Accuracy.
0.1% Accuracy Obtainable with Optional LT1460 Reference. Low Current Drain from
APD Rail Permits Smaller High Voltage Capacitors for a Given Ripple Level
Figure 21. Summarized Characteristics of Techniques Presented. Applicable Circuit Depends on Application Specifics
Note: This application note was derived from a manuscript originally
prepared for publication in EDN magazine.
AN92-15
Application Note 92
REFERENCES
1. Meade, M.L., “Lock-In Amplifiers and Applications,”
London, P. Peregrinus, Ltd.
2. Williams, J., “A Monolithic Switching Regulator with
100µV Output Noise,” Linear Technology Corporation, Application Note 70, October 1997.
3. Williams, J., “Measurement and Control Circuit Collection,” “∆VBE Based Thermometer,” Linear Technology Corporation, Application Note 45, June 1991,
pp.␣ 7–8
4. Williams, J., “Applications for a Switched Capacitor
Instrumentation Building Block,” Linear Technology
Corporation, Application Note 3, July 1985.
5. Williams, J., “Monolithic CMOS-Switch IC Suits Diverse Applications,” EDN Magazine, October 4, 1984.
6. Williams, J., “A Fourth Generation of LCD Backlight
Technology,” “Floating Lamp Circuits,” Linear Technology Corporation, Application Note 65, November
1995, pp. 40–43, Figure 48.
7. Negrete, M., “Fiberoptic Communication Systems
Benefit from Tiny, Low Noise Avalanche Photodiode
Bias Supply,” Linear Technology Corporation, Design
Note 273, December 2001.
8. Martin, A., “Charge Pump Based APD Circuits,” Private Communication, May 2002.
9. Williams, J., “Applications of New Precision Op Amps,”
“Instrumentation Amplifier with VCM = 300V and 160dB
CMRR,” Linear Technology Corporation, Application
Note 6, January 1985, pp. 1–2.
10. Williams, J., “Bridge Circuits,” “Optically Coupled
Switched Capacitor Instrumentation Amplifier,” Linear Technology Corporation, Application Note 43,
June 1990, pp. 9–10
11. Hickman, R. W. and Hunt, F. V., “On Electronic Voltage
Stabilizers,” “Cascode,” Review of Scientific Instruments, January 1939, pp. 6–21, p. 16.
APPENDIX A
LOW ERROR FEEDBACK SIGNAL DERIVATION
TECHNIQUES
Various text circuits either detail or make reference to
counteracting loading effects of the APD bias supply’s
output feedback divider. If the divider is located before the
1kΩ current shunt, its current drain is not included in the
current monitor’s output and no error is incurred. A
potential difficulty with this approach is that the 1kΩ shunt
appears in series with the bias supply output, degrading
load regulation. The maximum 1mA shunt current produces a 1V output regulation drop. In some cases this is
permissible and no further consideration is required.
Circumstances dictating tighter load regulation require
compensation techniques.
Divider Current Error Compensation—“Low Side”
Shunt Case
When the shunt is in a transformer’s return path (“low side
shunt”), divider error is cancelled by introducing a compensatory term into the APD current monitor circuitry.
AN92-16
Figure A1 shows details. The output voltage divider’s
current loading error is prevented from appearing in A1’s
output by feeding forward a compensatory current from
the APD bias programming input. This compensating
current, arriving at A1 via RLARGE, is scaled to precisely
balance out the portion of shunt output contributed by the
voltage divider’s loading error.
Divider Current Error Compensation—“High Side”
Shunt Case
Figure A2 addresses situations where the shunt resides in
the “high side.” Such arrangements involve high common
mode voltages, seemingly mandating a high voltage buffer
amplifier to isolate the divider’s current loading. Figure A2
shows a way around this, using standard low voltage
amplifiers to process high voltage signals. A1, sensing
after the 1kΩ shunt, isolates the feedback divider’s loading while permitting the APD bias regulator to include the
shunt within its feedback loop. A1 is powered directly from
the bias regulator’s high voltage output but its V – pin is
Application Note 92
zener clamped with respect to its V + pin. Current sink Q1
maintains this bias over the wide range of possible APD
regulator outputs. Although A1 processes high voltage
signals, the voltage across it is held to safe levels. The 5.6V
zener in the APD bias line ensures A1’s inputs are always
inside their common mode operating range. The 10M
resistor maintains adequate zener bias when APD currents
are extremely low. A 51k resistor protects A1 from destructive high voltage if the APD bias output is shorted to
ground. Similarly, the 100k resistor prevents high voltage
from appearing on the 5V supply if Q1 fails.
THIS POINT
GOES NEGATIVE WITH
INCREASING CURRENT
V+
OUTPUT
VOLTAGE
DIVIDER
IAPD
TO APD
10M
1k
CURRENT
SHUNT
DIVIDER ERROR CURRENT
EAPD
I=
RDIVIDER
(E.G., AT EAPD = 90V
DIVIDER ERROR CURRENT = 8.75µA)
287k
FB
I = APD +
DIVIDER ERROR
PWM
CONTROLLER
APD BIAS
PROGRAMMING
INPUT
100k
RLARGE
101k
–
FEEDFORWARD
COMPENSATION CURRENT
APD CURRENT
MONITOR OUTPUT
(IAPD – DIVIDER ERROR)
A1
+
RLARGE SCALED SO
VPROGRAM INDUCED CURRENT =
DIVIDER ERROR CURRENT
AN92 FA01
Figure A1. Output Voltage Divider Current Loading Error Is Compensated with Feedforward from Programming Input.
A1 Algebraically Sums Feedforward Term and Current Shunt Information, Presents Corrected Output
APD
CURRENT
MONITOR
1N4690
5.6V
FROM APD BIAS
REGULATOR
25VIN TO 95VIN
1k
VOUT
TO APD
10M
51k
V+
1N969
22V
TO APD BIAS
REGULATOR
FEEDBACK
INPUT
+
0.1µF
A1
LT1077
FEEDBACK
DIVIDER
–
V
–
100k
Q1
MPSA42
5V
16k
AN92 FA02
Figure A2. A1 Follower Floats from High Voltage Rail,
Eliminates Feedback Divider Current Loading Error. Q1
Current Source and 22V Zener Maintain Low Voltage Across
Amplifier; 5.6V Zener Accomodates A1’s Input Range
AN92-17
Application Note 92
APPENDIX B
PREAMPLIFIER AND OSCILLOSCOPE SELECTION
The low level measurements described require some form
of preamplification for the oscilloscope. Current generation oscilloscopes rarely have greater than 2mV/DIV sensitivity, although older instruments offer more capability.
Figure B1 lists representative preamplifiers and oscilloscope plug-ins suitable for noise measurement. These
units feature wideband, low noise performance. It is
particularly significant that many of these instruments are
no longer produced. This is in keeping with current instrumentation trends, which emphasize digital signal acquisition as opposed to analog measurement capability.
INSTRUMENT
TYPE
MANUFACTURER
Amplifier
Hewlett-Packard
The monitoring oscilloscope should have adequate bandwidth and exceptional trace clarity. In the latter regard high
quality analog oscilloscopes are unmatched. The exceptionally small spot size of these instruments is well-suited
to low level noise measurement.1 The digitizing uncertainties and raster scan limitations of DSOs impose display
resolution penalties. Many DSO displays will not even
register the small levels of switching-based noise.
MODEL
MAXIMUM
NUMBER BANDWIDTH SENSITIVITY/GAIN
461A
150MHz
Gain = 100
AVAILABILITY
Secondary Market
COMMENTS
50Ω Input, Standalone
Differential Amplifier
Tektronix
1A5
50MHz
1mV/DIV
Secondary Market
Requires 500 Series Mainframe
Differential Amplifier
Tektronix
7A13
100MHz
1mV/DIV
Secondary Market
Requires 7000 Series Mainframe
Differential Amplifier
Tektronix
11A33
150MHz
1mV/DIV
Secondary Market
Requires 11000 Series Mainframe
Differential Amplifier
Tektronix
P6046
100MHz
1mV/DIV
Secondary Market
Standalone
Differential Amplifier
Preamble
1855
100MHz
Gain = 10
Current Production
Standalone, Settable Bandstops
Differential Amplifier
Preamble
1822
10MHz
Gain = 1000
Current Production
Standalone, Settable Bandstops
Figure B1. Some Applicable High Sensitivity, Low Noise Amplifiers. Trade-Offs Include Bandwidth, Sensitivity and Availability
Note 1: In our work we have found Tektronix types 453, 453A, 454,
454A, 547 and 556 excellent choices. Their pristine trace presentation
is ideal for discerning small signals of interest against a noise floor
limited background.
AN92-18
Application Note 92
APPENDIX C
PROBING AND CONNECTION TECHNIQUES FOR
LOW LEVEL, WIDEBAND SIGNAL INTEGRITY1
The most carefully prepared breadboard cannot fulfill its
mission if signal connections introduce distortion. Connections to the circuit are crucial for accurate information
extraction. The low level, wideband measurements
demand care in routing signals to test instrumentation.
Ground Loops
Figure C1 shows the effects of a ground loop between
pieces of line-powered test equipment. Small current flow
between test equipment’s nominally grounded chassis
creates 60Hz modulation in the measured circuit output.
This problem can be avoided by grounding all line powered test equipment at the same outlet strip or otherwise
ensuring that all chassis are at the same ground potential.
Similarly, any test arrangement that permits circuit current flow in chassis interconnects must be avoided.
100µV/DIV
Pickup
Figure C2 also shows 60Hz modulation of the noise
measurement. In this case, a 4-inch voltmeter probe at the
feedback input is the culprit. Minimize the number of test
connections to the circuit and keep leads short.
Poor Probing Technique
Figure C3’s photograph shows a short ground strap affixed to a scope probe. The probe connects to a point
which provides a trigger signal for the oscilloscope. Circuit output noise is monitored on the oscilloscope via the
coaxial cable shown in the photo.
500µV/DIV
2ms/DIV
AN92 C01
Figure C1. Ground Loop Between Pieces of Test
Equipment Induces 60Hz Display Modulation
5ms/DIV
AN92 C02
Figure C2. 60Hz Pickup Due to Excessive
Probe Length at Feedback Node
Note 1: Veterans of LTC Application Notes, a hardened crew, will
recognize this Appendix from AN70 (see Reference 2). Although that
publication concerned considerably more wideband noise measurement, the material is directly applicable to this effort. As such, it is
reproduced here for reader convenience.
AN92-19
Figure C3. Poor Probing Technique. Trigger Probe Ground Lead Can Cause Ground Loop-Induced Artifacts to Appear in Display
Application Note 92
AN92-20
Application Note 92
Figure C4 shows results. A ground loop on the board
between the probe ground strap and the ground referred
cable shield causes apparent excessive ripple in the display. Minimize the number of test connections to the
circuit and avoid ground loops.
Figure C10’s trace shows this to be true. The former
example’s aberrations and excessive noise have disappeared. The switching residuals are now faintly outlined in
the amplifier noise floor. Maintain coaxial connections in
the noise signal monitoring path.
Violating Coaxial Signal Transmission—Felony Case
Direct Connection Path
In Figure C5, the coaxial cable used to transmit the circuit
output noise to the amplifier-oscilloscope has been
replaced with a probe. A short ground strap is employed
as the probe’s return. The error inducing trigger channel
probe in the previous case has been eliminated; the ’scope
is triggered by a noninvasive, isolated probe.2 Figure C6
shows excessive display noise due to breakup of the
coaxial signal environment. The probe’s ground strap
violates coaxial transmission and the signal is corrupted
by RF. Maintain coaxial connections in the noise signal
monitoring path.
A good way to verify there are no cable-based errors is to
eliminate the cable. Figure C11’s approach eliminates all
cable between breadboard, amplifier and oscilloscope.
Figure C12’s presentation is indistinguishable from Figure
C10, indicating no cable-introduced infidelity. When
results seem optimal, design an experiment to test them.
When results seem poor, design an experiment to test
them. When results are as expected, design an experiment
to test them. When results are unexpected, design an
experiment to test them.
Test Lead Connections
Violating Coaxial Signal Transmission—
Misdemeanor Case
Figure C7’s probe connection also violates coaxial signal
flow, but to a less offensive extent. The probe’s ground
strap is eliminated, replaced by a tip grounding attachment. Figure C8 shows better results over the preceding
case, although signal corruption is still evident. Maintain
coaxial connections in the noise signal monitoring path.
Proper Coaxial Connection Path
In Figure C9, a coaxial cable transmits the noise signal to
the amplifier-oscilloscope combination. In theory, this
affords the highest integrity cable signal transmission.
In theory, attaching a voltmeter lead to the regulator’s
output should not introduce noise. Figure C13’s increased
noise reading contradicts the theory. The regulator’s output impedance, albeit low, is not zero, especially as
frequency scales up. The RF noise injected by the test lead
works against the finite output impedance, producing the
200µV of noise indicated in the figure. If a voltmeter lead
must be connected to the output during testing, it should
be done through a 10kΩ-10µF filter. Such a network
eliminates Figure C13’s problem while introducing minimal error in the monitoring DVM. Minimize the number of
test lead connections to the circuit while checking noise.
Prevent test leads from injecting RF into the test circuit.
Note 2: To be discussed. Read on.
100µV/DIV
(INVERTED)
5µs/DIV
AN85 C04
Figure C4. Apparent Excessive Ripple Results from Figure C3’s Probe Misuse.
Ground Loop on Board Introduces Serious Measurement Error
AN92-21
Application Note 92
Figure C5. Floating Trigger Probe Eliminates Ground Loop, But Output Probe
Ground Lead (Photo Upper Right) Violates Coaxial Signal Transmission
500µV/DIV
5µs/DIV
AN92 C06
Figure C6. Signal Corruption Due to Figure C5’s
Noncoaxial Probe Connection
AN92-22
Application Note 92
Figure C7. Probe with Tip Grounding Attachment Approximates Coaxial Connection
100µV/DIV
5µs/DIV
AN92 C08
Figure C8. Probe with Tip Grounding Attachment
Improves Results. Some Corruption Is Still Evident
AN92-23
Application Note 92
Figure C9. Coaxial Connection Theoretically Affords Highest Fidelity Signal Transmission
100µV/DIV
5µs/DIV
AN92 C10
Figure C10. Life Agrees with Theory. Coaxial Signal
Transmission Maintains Signal Integrity. Switching
Residuals Are Faintly Outlined in Amplifier Noise
AN92-24
Application Note 92
Figure C11. Direct Connection to Equipment Eliminates Possible Cable-Termination
Parasitics, Providing Best Possible Signal Transmission
100µV/DIV
5µs/DIV
AN92 C12
Figure C12. Direct Connection to Equipment Provides
Identical Results to Cable-Termination Approach.
Cable and Termination Are Therefore Acceptable
AN92-25
Application Note 92
200µV/DIV
5µs/DIV
AN92 C13
Figure C13. Voltmeter Lead Attached to Regulator Output
Introduces RF Pickup, Multiplying Apparent Noise Floor
Isolated Trigger Probe
The text associated with Figure C5 somewhat cryptically
alluded to an “isolated trigger probe.” Figure C14 reveals
this to be simply an RF choke terminated against ringing.
The choke picks up residual radiated field, generating an
isolated trigger signal. This arrangement furnishes a ’scope
trigger signal with essentially no measurement corruption. The probe’s physical form appears in Figure C15. For
good results, the termination should be adjusted for
minimum ringing while preserving the highest possible
amplitude output. Light compensatory damping produces
Figure C16’s output, which will cause poor ’scope triggering. Proper adjustment results in a more favorable output
(Figure C17), characterized by minimal ringing and welldefined edges.
Trigger Probe Amplifier
The field around the switching magnetics is small and may
not be adequate to reliably trigger some oscilloscopes. In
such cases, Figure C18’s trigger probe amplifier is useful.
It uses an adaptive triggering scheme to compensate for
variations in probe output amplitude. A stable 5V trigger
output is maintained over a 50:1 probe output range. A1,
operating at a gain of 100, provides wideband AC gain. The
output of this stage biases a 2-way peak detector (Q1
through Q4). The maximum peak is stored in Q2’s emitter
capacitor, while the minimum excursion is retained in Q4’s
AN92-26
emitter capacitor. The DC value of the midpoint of A1’s
output signal appears at the junction of the 500pF capacitor and the 3MΩ units. This point always sits midway
between the signal’s excursions, regardless of absolute
amplitude. This signal-adaptive voltage is buffered by A2
to set the trigger voltage at the LT1394’s positive input.
The LT1394’s negative input is biased directly from A1’s
output. The LT1394’s output, the circuit’s trigger output,
is unaffected by >50:1 signal amplitude variations. An
X100 analog output is available at A1.
Figure C19 shows the circuit’s digital output (Trace B)
responding to the amplified probe signal at A1 (Trace A).
Figure C20 is a typical noise testing setup. It includes the
breadboard, trigger probe, amplifier, oscilloscope and
coaxial components.
L1
PROBE
SHIELDED
CABLE
BNC CONNECTION
TO TERMINATION BOX
L1: J.W. MILLER #100267
TERMINATION BOX
1k DAMPING
ADJUST
4700pF
BNC
OUTPUT
AN92 FC14
Figure C14. Simple Trigger Probe Eliminates Board
Level Ground Loops. Termination Box Components
Damp L1’s Ringing Response
Figure C15. The Trigger Probe and Termination Box. Clip Lead Facilitates Mounting Probe, Is Electrically Neutral
Application Note 92
AN92-27
Application Note 92
10mV/DIV
10mV/DIV
10µs/DIV
10µs/DIV
AN92 C16
Figure C16. Misadjusted Termination Causes Inadequate
Damping. Unstable Oscilloscope Triggering May Result
AN92 C17
Figure C17. Properly Adjusted Termination
Minimizes Ringing with Small Amplitude Penalty
50Ω
ANALOG BNC OUTPUT
TO ’SCOPE TRIGGER INPUT
5V
2k
3
Q1
6
4
+
3M
500pF
0.005µF
A1
LT1227
5V
+
0.005µF
–
750Ω
Q2
2
5V
2k
5
1
–
13
1k
15
10Ω
3M
10
14
Q3
5V
12
A2
LT1006
Q4
11
470Ω
+
+
10µF
0.1µF
100µF
0.1µF
+
2k
0.1µF
470Ω
–
LT1394
Q1, Q2, Q3, Q4 = CA3096 ARRAY: TIE SUBSTRATE (PIN 16) TO GROUND
= 1N4148
TRIGGER PROBE
AND TERMINATION BOX
(SEE FIGURE C14 FOR DETAILS)
Figure C18. Trigger Probe Amplifier Has Analog and Digital Outputs. Adaptive
Threshold Maintains Digital Output Over 50:1 Probe Signal Variations
A = 1V/DIV
AC COUPLED
B = 5V/DIV
10µs/DIV (UNCALIB)
AN92 C19
Figure C19. Trigger Probe Amplifier Analog (Trace A)
and Digital (Trace B) Outputs
AN92-28
DIGITAL
TRIGGER
OUT BNC
TO ’SCOPE
AN92 C18
Figure C20. Typical Noise Test Setup Includes Trigger Probe, Amplifier, Oscilloscope and Coaxial Components
Application Note 92
AN92-29
Application Note 92
APPENDIX D
A SINGLE RAIL AMPLIFIER WITH TRUE ZERO VOLT
OUTPUT SWING
Performance requirements necessitate analog output current monitors to swing within 100µV of zero. This is
difficult because the circuits run from a single, positive
rail. No single rail amplifier can swing this close to zero
while maintaining accurate outputs. Figure D1’s power
supply bootstrapping scheme achieves the desired characteristics with minimal component addition.
A1, a chopper stabilized amplifier, has a clock output. This
output switches Q1, providing drive to the diode-capacitor
charge pump. The charge pump output feeds A1’s V –
terminal, pulling it below zero, permitting output swing to
(and below) ground. If desired, negative output excursion
can be limited by either clamp option shown.
Reliable start-up of this bootstrapped power supply scheme
is a valid concern, warranting investigation. In Figure D2,
the amplifier’s V – pin (Trace C) initially rises at supply
turn-on (Trace A) but heads negative when amplifier
clocking (Trace B) commences at about midscreen.
The circuit provides a simple way to obtain output swing
to zero volts, permitting a true “live at zero” output.
100Ω
5V
V+
5V
+
A1
LTC1150
–
1k 10µF
≈ –3.5V
HERE
10µF
Q1
2N3904
+
39k
V–
+
CLKOUT
100k
1k
DASHED LINE CIRCUITRY = CLAMP OPTIONS.
SEE TEXT
= BAT85
AN92 FD01
Figure D1. Single Rail Powered Amplifier Has True Zero Volt
Output Swing. A1’s Clock Output Switches Q1, Driving DiodeCapacitor Charge Pump. A1’s V – Pin Assumes Negative Voltage,
Permitting Zero (and Below) Volt Output Swing
A = 5V/DIV
B = 5V/DIV
C = 0.2V/DIV
5ms/DIV
AN92 D02
Figure D2. Amplifier Bootstrapped Supply Start-Up.
Amplifier V – Pin (Trace C) Initially Rises Positive at
5V Supply (Trace A) Turn-On. When Amplifier Internal
Clock Starts (Trace B, 5th Vertical Division), Charge
Pump Activates, Pulling V – Pin Negative
AN92-30
Application Note 92
APPENDIX E
APD PROTECTION CIRCUITS
APD receiver modules are electrically delicate and expensive devices. Because of this, Figure E1’s protection circuits may be of interest. They are designed to protect the
APD module from bias programming overvoltage error
(Figure E1a), excessive current (E1b) or destructive voltage (Figure E1c). In Figure E1a, Q1 is normally off and
programming voltage passes to the bias regulator voltage
programming input. Abnormally high inputs, defined by
the potentiometer’s setting, cause A1 to swing low, biasing Q1 and closing A1’s feedback loop. This causes Q1’s
emitter to clamp at the potentiometer wiper’s voltage,
safely limiting the bias regulator’s programming input.
Figure E1b is an APD current limiter. This particular circuit
is designed for use with “low side” shunts in transformer
coupled APD supplies, such as text Figure 9, although the
technique is generally applicable. As long as the shunt
INPUT FROM
PROGRAMMING VOLTAGE
current’s absolute value is below the current limit point, A2
is saturated high and the associated APD bias regulator
functions normally. Shunt overcurrent forces A2’s output
lower, pulling the regulator’s control pin (VC) lower and
limiting current. The 100pF-1MΩ combination stabilizes
A2 and the bias regulator assumes the characteristics of a
current source.
Figure E1c is an overvoltage crowbar. It is intended as the
last line of defense against uncontrolled APD bias supply
high voltage outputs. Normally, the LTC1696 crowbar IC
is below its 0.88V trigger threshold and the SCR is off. If
the APD bias rises too high the LTC1696 triggers, firing the
SCR. SCR turn-on “crowbars” the APD bias line, arresting
the high voltage and maintaining a short across the line via
its latch characteristic. If the APD bias supply has significant output impedance, prolonged SCR loading is not
deleterious; if not, the bias supply should be fused.
1K
TO BIAS
REGULATOR
VOLTAGE
PROGRAMMING
INPUT
5V
–
4.7k
A1
LT1006
50k
VOLTAGE
CLAMP
ADJUST
100Ω
Q1
2N3906
+
AN92 FE01a
(E1a) Programming Voltage Clamp
TO APD BIAS
OUTPUT
TO UNGROUNDED
SIDE OF 1kΩ SHUNT
AT TRANSFORMER
1M*
TO APD BIAS
REGULATOR VC PIN
SHUNT DERIVED
CURRENT
0.001µF
5V
+
LIMIT
CURRENT
A2
LT1077
1M*
–
7.5M*
TRIG AT
0.88V
100pF
CURRENT LIMIT INPUT
0V TO 1V = 0mA TO 1mA
+V
FB1
88.7k*
R
LTC1696
2N5063
OUT
GND
VTRIG = 75.3V
1M
AN92 FE01c
AN92 FE01b
(E1b) Current Limiter
(E1c) Bias Voltage Crowbar
* = 1% METAL FILM RESISTOR
= 1N4148
Figure E1. Protection Circuits Prevent APD Destruction Due to Hardware or Software Failures. Options Include
Programming Voltage Clamp (Figure E1a), Current Limiter (Figure E1b) and Bias Voltage Crowbar (Figure E1c)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN92-31
Application Note 92
AN92-32
Linear Technology Corporation
an92f LT/TP 1102 2K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2002
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