Dec 2005 Fast CMOS Op Amp Challenges Bipolar Amps on All Key Specs

DESIGN FEATURES
Fast CMOS Op Amp Challenges
Bipolar Amps on All Key Specs
by John Wright and Glen Brisebois
Introduction
The LTC6241 dual and LTC6242
quad CMOS op amps compete headon with bipolar op amps in noise,
speed, offset voltage, and offset drift,
while maintaining superior low input
bias and noise current. Crucial advances in these amplifiers’ parameters
translate to tighter system specs,
lower complexity, and a wider supply
voltage operating range than previous CMOS op amps. These extremely
low input bias current op amps are
optimized for high impedance transducer applications such as photodiode
transimpedance amplifiers, TIAs,
though they are also well suited to a
variety of precision applications.
The LTC6241 and LTC6242 do
not employ complicated post-package schemes to reduce offset voltage,
yet their 125µV offset voltage and
2.5µV/°C offset drift are among the
best CMOS amplifiers available. The
18MHz gain bandwidth and very low
noise further distinguishes them from
a field of mediocre amplifiers. They are
fully specified on 3V, and 5V, with an
HV version that guarantees operation
to ±5.5V. Supply current consumption
is 2.2mA/amplifier maximum. Table
1 summarizes the conservative specs
for these op amps.
The LTC6241 is available in the
SO8, and for compact designs it is
packaged in the tiny dual fine pitch
leadless (DFN) package. The LTC6242
is available in a 16-Pin SSOP as well
as a 5mm × 3mm DFN package.
CMOS with Low 1/f Noise?
What about Noise Current?
CMOS op amps have traditionally had
much higher 1/f noise than bipolar
amplifiers. It is common to find CMOS
amplifiers with a 1/f corner above
several kilohertz, but the LTC6241
rivals the best bipolar op amps with
a 1/f noise corner of only 40Hz. This
exceptionally low noise translates
Linear Technology Magazine • December 2005
to just 550nVP–P in a 0.1Hz to 10Hz
bandwidth, and represents the lowest
1/f noise available in a non-autozero
CMOS op amp.
In I-to-V applications such as
photodiode amplifiers, where the
amplifier is operated inverting, noise
current dominates at high frequency.
CMOS op amp noise current has
two sources. The first is the input
device channel thermal noise coupling through the gate-to-source
and gate-to-drain capacitances. The
second noise current is derived from
the op amp’s input capacitance, and
capacitance associated with the input transducer. This input referred
noise current (CV noise) is due to the
amplifier’s noise voltage, VN, impressed
across the total input capacitance,
CT, causing a current of magnitude
2πfCTVN to flow through the feedback
resistor.
The way to make CMOS or bipolar
low noise amplifiers is with large input
transistors. The problem is that big input structures carry the burden of high
input capacitance. High input capaci-
Table 1. LTC6241/LTC6242 Performance: Ta = 25°C, VS = 5V/0V unless otherwise specified.
The ● denotes specifications that apply over –40°C to 85°C.
Parameter
Conditions
Offset Voltage
VCM = 0
S8, LTC6241
GN16, LTC6242
DD, DHC, LTC6241/42
Min
Typ
Max
Units
40
50
100
125
150
550
µV
µV
µV
TC VOS
●
0.6
2.5
µV/°C
Input Bias Current
●
1
10
75
pA
pA
Noise Voltage
f = 1kHz
f = 0.1Hz to 10Hz
7
550
Noise Current
f = 100kHz
110
fA/�
Input Capacitance
f = 100kHz
CDM
CCM
0.5
3
pF
pF
Large Signal Gain
RL = 1kΩ to VS/2
90
215
V/mV
80
105
dB
CMRR
VCM = –V to +V – 1.5V ●
2.8
2.8
10
Operating Supply
Range
LTC6241/42
LTV6241HV/42HV
●
●
VOUT Low
ISINK = 5mA
●
190
VOUT High
ISOURCE = 5mA
●
4.81 4.675
Supply Current
per amplifier
●
1.8
Slew Rate
AV = –2, RL = 1kΩ,
●
5
10
Gain Bandwidth
Product
RL = 1kΩ
●
13
18
nV/�
nVP–P
6
11
V
V
325
mV
2.2
V
mA
V/µs
5
DESIGN FEATURES
ITAIL
V–
VIN
VIN
–
I1
V+
I2
M3
CM
V+
V+
DESD2
DESD1
+
I1
DESD5
RT2
RT1
M1
DIFFERENTIAL
DRIVE
GENERATOR
M2
VO
DESD6
C1
V–
V
V–
DESD4
DESD3
V+
Q2
Q1
BIAS
R1
–
M4
R2
V–
Figure 1. Simplified schematic
tance increases high frequency noise
current, as well as reduces overall op
amp speed. An uncommon feature
of the LTC6241 is its low differential
input capacitance of just 0.5pF, which
is a major benefit in I-to-V amplifier
designs. This input capacitance is 8
to 10 times lower than than that of
other CMOS amps.
Simple Architecture Yields
Low Noise and DC Precision
Figure 1 is a simplified schematic of
one half of the LTC6241, which has
a pair of low noise input transistors
M1 and M2. A simple folded cascode
Q1, Q2, and R1, R2 allow the input
stage to swing to the negative rail,
while performing level shift to the differential drive generator. Transistors
M1 and M2 along with current sources
90
16
VS = ±2.5V
80 SO-8 PACKAGE
12
60
NUMBER OF UNITS
NUMBER OF UNITS
70
50
40
30
10
8
6
20
4
10
2
0
VS = ±2.5V
2 LOTS
–55°C TO 125°C
14
–70
–50 –30 –10 10
30
50
INPUT OFFSET VOLTAGE (µV)
0
70
–1.0 –0.6 –0.2 0.2 0.6 1.0
DISTRIBUTION (µV/°C)
1.4
1.8
I1 and I2 have been optimized for low
noise and consume over 30% of the
die area. Low offset is achieved by
laser trimming resistors R T1 and R T2.
Stresses that occur during package
assembly have minimal affect on this
simple, stable architecture, and consequently, complicated post-package
trim schemes that adjust offset voltage
and drift are unnecessary.
The LTC6241 and LTC6242 were
intentionally designed without a
rail-to-rail input stage as to not compromise their noise specs. Many CMOS
rail-to-rail input amplifiers show large
offset shift and higher noise when the
common mode voltage is operating in
this top side transition region, limiting
their usefulness.
The LTC6241 and LTC6242 have
reverse-biased ESD protection diodes
on all inputs and outputs as shown
in Figure 1. These diodes protect the
amplifiers from ESD strikes up to
1.7kV. No current flows into the gate
on a DC basis, but these ESD protection diodes are the source of input bias
current specified on the data sheet.
These diodes have leakage current that
doubles approximately every 7°C, but
input current typically remains below
10pA up to 85°C ambient.
Capacitor C1 reduces the unity
cross frequency and improves the
frequency stability without degrading
the gain bandwidth of the amplifier.
Capacitor CM sets the overall amplifier
gain bandwidth. The differential drive
generator supplies signal to transistors M3 and M4 that swing the output
from rail-to-rail.
Figure 2. VOS distribution and VOS temperature coefficient distribution
60
VS = 5V, 0V
100
TA = 85°C
10
TA = 25°C
1
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
COMMON MODE VOLTAGE (V)
1000
TA = 25°C
VS = ±2.5V
VCM = 0V
50
NOISE CURRENT (fA/√Hz)
TA = 125°C
NOISE VOLTAGE (nV/√Hz)
INPUT BIAS CURRENT (pA)
1000
40
30
20
TA = 25°C
VS = ±2.5V
VCM = 0V
100
10
1
10
0
1
10
100
1k
FREQUENCY (Hz)
10k
100k
0.1
100
1k
10k
FREQUENCY (Hz)
100k
Figure 3. Input bias current vs common mode voltage and voltage and current noise vs frequency
6
Linear Technology Magazine • December 2005
DESIGN FEATURES
VIN
C
0.1µF
R
1M
–
VS +
1/2 LTC6241
+
–
VOUT = –∫ VIN/2πRCdt
t=0
VIN
R1
1M
Figure 4. A textbook integrator is inverting
Figure 2 shows the distribution of
offset voltage and offset voltage drift.
Figure 3 shows the input bias current
vs common mode voltage as well as the
noise voltage and current spectrum.
Applications
Non-Inverting Integrator
Integrators are used widely in feedback
control systems and filters. CMOS
input amplifiers like the LTC6241 are
preferred for this function because
the low input bias current allows the
use of large value resistors and small
capacitors for a given integration time
constant. The most common form
of integrator is the inverting form,
shown in Figure 4. It has a transfer
function of:
⌠ VIN
–
dt

⌡ 2πRC
t =0
If inversion is not desired in the
feedback control loop using the circuit
in Figure 4, a designer must add another op amp to invert again. A simpler
overall solution produces a non-inverting integrator using just one op amp.
Figure 5 shows the circuit.
At low frequencies, R1 • C1 does
not attenuate, and the non-inverting integration function is provided
by the op amp gain and its feedback
components C2 and R2. At higher
frequencies, C2 becomes a short
circuit so the op amp goes to a gain
of one, and the integration function
is provided by R1 and C1. If the time
constants are matched, the integrator
conformance is excellent. Matching is
not easy. In most loops, to guarantee
that the phase of the integrator does
not exceed 90 degrees, the time constants can be intentionally skewed so
that R1 • C1 < R2 • C2. For an example
Linear Technology Magazine • December 2005
VS+
VOUT = ∫ VIN/2πRCdt
t=0
1/2 LTC6241
+
C1
0.1µF
VS –
VOUT =
C2
0.1µF
R2
1M
LET R1 • C1 = R2 • C2
VS–
Figure 5. A non-inverting integrator can be very simple. Ideally, R1 • C1 = R2 • C2, but mismatch
is inevitable. To avoid any phase buildup from a mismatch, the time constants may be skewed so
that R1 • C1 < R2 • C2.
of a specific closed loop utilization of
a non-inverting integrator, see LTC
Design Note DN254.
under physical acceleration. Figure 6
shows the classical “charge amplifier”
approach. The op amp is in the inverting configuration so the sensor looks
into a virtual ground. All of the charge
generated by the sensor is transferred
across the feedback capacitor by the
op amp action. Because the feedback
capacitor is 100 times smaller than
the sensor, the output is forced to a
voltage 100 times what would have
Piezoelectric Accelerometers:
Inverting vs Non-Inverting
Figures 6 and 7 show two different
approaches to amplifying signals
from a capacitive sensor using the
LTC6241. The sensor in both cases
is a 770pF piezoelectric shock sensor
accelerometer, which generates charge
SHOCK SENSOR
VS +
MURATA-ERIE
PKGS-00LD
770pF
+
1/2
LTC6241HV
–
VS
CABLE CAPACITANCE
VARIATIONS OKAY
VOUT
–
7.7pF
BIAS RESISTOR
1GΩ
VISHAY-TECHNO
CRHV2512AF1007G
(OR EQUIVALENT)
VOUT = 110mV/G
VS = ±1.4V TO ±5.5V
Figure 6. Classical inverting charge amplifier. Variations in cable capacitance (i.e. length) do
not affect the signal gain. Use this circuit when the accelerometer is remote from the amplifier
and the cable length is unspecified. Drawbacks are that gain is set by the low valued feedback
capacitor and low frequency performance is set by the bias resistor working into the same.
SHOCK SENSOR
+
MURATA-ERIE
PKGS-00LD
770pF
–
VS +
1/2
LTC6241HV
100
1k
10k
VOUT
1k
+
1/2
LTC6241HV
BIAS RESISTOR
1GΩ
VISHAY-TECHNO
CRHV2512AF1007G
(OR EQUIVALENT)
–
100
VS –
10k
VOUT = 110mV/G
VS = ±1.4V TO ±5.5V
BW = 0.2Hz TO 10kHz
Figure 7. Non-inverting charge amplifier offers several advantages.
Stages can be paralleled for lower voltage noise. Bias resistor works
into higher capacitance for better low frequency response.
7
DESIGN FEATURES
CF
8pF
drawback to this circuit is that the
parasitic capacitance at the input reduces the gain slightly. This circuit is
favored in cases where parasitic input
capacitances such as traces and cables
are relatively small and invariant.
Consider making the bias resistor
larger than bandwidth calculations
would suggest. This actually reduces
the noise floor at low frequency. For
example, to support frequencies down
to 10Hz at –3dB, the bias resistor
would calculate to:
RF
1M
IPD
–
HAMAMATSU
LARGE AREA
PHOTODIODE
S1227-1010BQ
CPD = 3000pF
+
5V
1/2
LTC6241HV
VOUT = 1M • IPD
–5V
Figure 8. Large area photodiode amplifier
provides about 25kHz bandwidth. DCs are
good but output is noisy.
1
= 20MΩ
2π • 10Hz • 770pF
been the sensor’s open circuit voltage.
Thus, the circuit gain is 100.
The benefit of this approach is
that the signal gain of the circuit is
independent of any cable capacitance
introduced between the sensor and the
amplifier, making this a good solution
for remote accelerometers where the
cable length may vary. Difficulties
with the circuit are inaccuracy of the
gain setting with the small capacitor,
and low frequency cutoff due to the
bias resistor working into the small
feedback capacitor.
Figure 7 shows a non-inverting
amplifier approach. This approach
has many advantages. First, the gain
is set accurately with resistors rather
than with a small capacitor. Second,
the low frequency cutoff is dictated
by the bias resistor working into the
large 770pF sensor, rather than into
a small feedback capacitor, for lower
frequency response. Third, the noninverting topology can be paralleled
and summed (as shown) for scalable
reductions in voltage noise. The only
5V
–5V
PHILIPS
BF862
JFET
D
4.99k
S
At 10Hz, the 20M resistor would
contribute 580nV/√Hz of noise, and
be 3dB down just like the signal.
Making the resistor 1GΩ as shown,
its 4000nV/√Hz voltage noise would
be attenuated down to effectively
80nV/√Hz by the accelerometer
capacitance, while the signal would
barely be attenuated at all. That’s an
easy seven-fold improvement in the
signal-to-noise ratio.
Large Area Photodiode Amplifiers
Figure 8 shows the LTC6241 used as
a transimpedance amplifier for a high
capacitance large area photodiode. The
circuit has unity noise gain at DC, so
resolution is entirely noise limited. The
bandwidth rolls due to the fact that
the photodiode impedance drops with
frequency raising the effective gain
(the noise gain), which the op amp
looks into. This severely limits the
bandwidth and increases the output
noise. The –3dB bandwidth for this
CF
0.5pF
IPD
HAMAMATSU
LARGE AREA
PHOTODIODE
S1227-1010BQ
CPD = 3000pF
–
The LTC6241 and LTC6242 combine
the low noise, offset, and drift of the
best bipolar op amps with low input
bias and noise current of CMOS op
amps. These amplifiers operate from
2.7V to ±5.5V and represent all-in-one
solutions for fast, low noise signal
processing.
5V
1/2 LTC6241
+
VOUT = 1M • IPD
300nV/�
per DIV
–5V
Figure 9. A simple bootstrap circuit drastically improves the ACs while leaving the DCs excellent.
Output noise is now 221nV/√Hz at 10kHz, and bandwidth is 220kHz. Rise time is 1.58µs from a
3000pF photodiode at 1MΩ of gain!
8
Conclusion
3µV/�
RF
1M
G
circuit was measured at 25kHz, and
the output noise density at 10kHz was
measured at 1.6µV/√Hz. That may be
good enough for many applications. If
it’s not good enough, keep reading.
The main problem with the previous
circuit is the large capacitance of the
photodiode. The perfect thing to do
is to bootstrap that capacitance with
a low noise JFET. Figure 9 shows the
circuit. The low noise JFET source follower runs about 1mA down through
the 4.99k resistor, with the source
sitting about 0.6V above ground. Now
the effective input voltage noise placed
across the photodiode capacitance is
the 1nV/√Hz of the JFET rather than
the 8nV/√Hz of the op amp. The op
amp is looking into its own 3pF of input
capacitance plus the 2pF of gate-drain
capacitance, plus parasitics. That’s a
much better situation than looking
into 3000pF!
The effects of this simple modification are drastic. The compensation
capacitor CF can be reduced, and bandwidth is improved to 220kHz (1.58µs
rise time). Output noise density at
10kHz is reduced to 221nV/√Hz, as
shown in Figure 10. DC performance
remains excellent because the JFET is
not involved; it simply provides a slight
reverse bias to the photodiode.
0nV/�
f = 1kHz to 100kHz, 10kHz/DIV
Figure 10. Output noise spectral density
of the bootstrap circuit of Figure 9
Linear Technology Magazine • December 2005
Similar pages