Dec 2005 Fully Differential Amplifier with Rail-to-Rail Outputs Offers 16-Bit Performance at 1MHz on a Single 2.5V Supply

DESIGN FEATURES
Fully Differential Amplifier with
Rail-to-Rail Outputs Offers 16-Bit
Performance at 1MHz on a
by Arnold Nordeng
Single 2.5V Supply
Introduction
With increasing levels of IC integration,
and shrinking transistor geometries,
A/D converter supply voltages have
decreased and their inputs have been
designed to process signals differentially to maintain good dynamic range.
These ADCs typically run from a single
low voltage supply with an optimal
common mode input somewhere near
mid-supply. The LT1994 facilitates
interfacing to these ADCs by providing
differential conversion and amplification, common mode translation of wide
band, ground referenced, single-ended
or differential input signals. It comes
in an 8-pin MSOP or DFN package,
which is pin-for-pin compatible with
other commercially available fully-differential amplifiers.
What sets the LT1994 apart from
other fully- differential amplifiers are
its low noise, low distortion, rail-to-rail
output, and an input common mode
range that extends to ground on power
supplies as low as 2.5V. This eliminates
the need for a negative power supply,
and makes the LT1994 uniquely able
to interface to differential input ADCs
while sharing the same power supply.
This saves the user system cost, and
power.
RI
499
3V
1.5V
0.1 F
– +
2
VOCM
8
2.5V
2.75V
1.5V
0.25V
4
LT1994
VOUT
5
+ –
7
RI
499
0.1 F
3
1
6
2.75V
1.5V
0.25V
RF
499
VIN
5VP-P
0V
–2.5V
Figure 1. Common mode translation of VIN using the LT1994
Performance of LT1994
The first advantage of the LT1994
is that it can convert and level-shift
ground referenced, single-ended or
differential signals to VOCM pin referenced, differential output signals.
Figure 1 shows how. A single-ended
5VP–P ground referenced signal (which
swings 2.5V below the supply of both
the ADC and the LT1994) is translated
by the LT1994 from being a ground
referenced signal to a differential
mid-supply referenced signal. This
is accomplished within the LT1994
by two feedback loops: a differential
feedback loop, and a common mode
Table 1. LT1994 key specifications
Parameter
Typical Specification
Supply Current at 3V
13.3mA
en – Input referred Voltage Noise
3nV/�
HD2 at VIN =2VP–P, 1MHz
–94dBc
HD3 at VIN =2VP–P, 1MHz
–108dBc
Gain-Bandwidth
70MHz
Slew Rate
65V/µs
0.01% Settling on a 2V step
120ns
Linear Technology Magazine • December 2005
RF
499
feedback loop. Both loops have high
open loop gain, around 100dB. The
common mode feedback loop forces
the instantaneous average of the two
outputs to be equal to the voltage on the
VOCM pin. Its feedback loop is internal
to the LT1994. The differential feedback loop works similarly to traditional
op amps forcing the difference in the
summing node voltages to zero. As a
result, the differential output is simply
governed by the equation:
VOUT = VOUT+ – VOUT – ≈
RF
• VIN
RI
By eliminating the need for a
negative supply, the LT1994 gives
the user maximum dynamic range
at minimal power. Since each output
of the LT1994 is capable of swinging
rail-to-rail, and with the LT1994’s
3nV/√Hz input referred voltage noise
(see Figure 2 for the LT1994’s noise
spectral density plot), applications
such as the one shown in Figure 1 have
a signal-to-noise ratio approaching
96dB in a 10MHz noise bandwidth.
This represents a 6dB increase in
dynamic range compared to single
ended output rail-to-rail amplifiers
13
100
100
VS = 3V
TA = 25°C
10
INPUT CURRENT NOISE DENSITY (pA/√Hz)
INPUT REFERRED VOLTAGE NOISE DENSITY (nV/√Hz)
DESIGN FEATURES
10
en
in
1
10
100
1k
10k
FREQUENCY (Hz)
1
1M
100k
Linearity is enhanced using a fully
differential architecture allowing the
cancellation of even order harmonics.
To see how this works, a pure single
tone sine wave input is applied to the
LT1994 as shown in Figure 1. The outputs of the LT1994 can be represented
by a Taylor series expansion:
across the power supplies with short
traces with the V– tied directly to a
low-impedance ground plane. On split
supplies, additional 0.1µF high quality, low ESR, surface-mount bypass
caps should be used to bypass each
supply separately to a low-impedance
ground plane.
V 
 V 2
VOUT+ = K 1  IN  + K 2  IN  +
 2 
 2 
Interfacing to ADCs
3
V 
V 
K 3  IN  + K 4  IN  +
 2 
 2 
Figure 2. LT1994 input referred
noise spectral density
 V 
 V 2
VOUT – = K 1  – IN  + K 2  – IN  +
 2 
 2 
–40
VS = 3V
VIN = 2VP-P (SINGLE ENDED)
–50 R = R = 499Ω
F
I
 V 3
 V 4
K 3  – IN  + K 4  – IN  +
 2 
 2 
DISTORTION (dB)
–60
–70
3RD
HARMONIC
–80
VOUT is the difference:
2ND
HARMONIC
–90
VOUT = VOUT+ – VOUT –
V 
 V 3
= 2K 1  IN  + 2K 3  IN  +
 2 
 2 
–100
–110
100k
4
1M
FREQUENCY (Hz)
10M
Figure 3. LT1994 disortion vs frequency
with the similar noise floors. Some of
the LT1994’s key specifications are
tabulated in Table 1.
Another benefit of fully-differential
signal processing is that interference
such as ground noise or power supply
noise appear as common mode signals
and are rejected by the internal matching and balance of the amplifier. Power
supply rejection and common-mode
rejection becomes limited primarily
by internal transistor matching and
are typically around 100dB.
,
leaving just the odd harmonic terms.
Figure 3 shows a plot of distortion vs
frequency with the LT1994 configured
in the closed-loop unity gain configuration shown in Figure 1. With a 2VP–P,
1MHz, single-ended input, the 2nd
harmonic measures –94dBc, and the
3rd harmonic measures –108dBc.
Getting the best distortion out of the
LT1994 requires careful layout, paying
close attention to symmetry and balance. In single supply applications, it
is recommended that high quality, low
ESR, surface mount 1µF and 0.1µF
caps be paralleled and tied directly
The sampling process of ADCs create
a sampling glitch caused by switching
in the sampling capacitor on the ADC
front end which momentarily “shorts”
the output of the amplifier as charge
is transferred between the amplifier
and the sampling cap. The amplifier
must recover and settle from this load
transient before this acquisition period
ends for a valid representation of the
input signal.
In general, the LT1994 settles faster
from these periodic load impulses than
from a 2V input step, but it is a good
idea to place a small RC filter network
between the output of the LT1994 and
the input of the ADC to help absorb
the charge injection that comes out of
the ADC from the sampling process
(Figure 4 shows an example of this).
The capacitance of this decoupling
network serves as a charge reservoir
to provide high frequency charging
during the sampling process, while the
two resistors of the decoupling network
are used to dampen and attenuate any
charge kickback from the ADC.
The selection of the RC time constant is trial and error for a given ADC,
but the following general guidelines
are recommended: Too large a resistor in the decoupling network leaving
50Ω
475Ω
499Ω
3V
50Ω
1
2
0.1µF
8
– +
VOCM
24.9Ω
4
LT1994
5
+ –
6
0.1µF
0.1µF
3
7
499Ω
3V
AIN+
47pF
24.9Ω
VDD
LT1403A-1
AIN–
GND
SDO
CONV
SCK
VREF
50.4MHz
10µF
499Ω
Figure 4. ADC buffering with common mode translation and differential conversion
14
DIFFERENTIAL OUTPUT MAGNITUDE (dB)
LOW DISTORTION
SIGNAL SOURCE
0
f
= 2.8Msps
–10 fSAMPLE
IN = 1.001MHz
–20 INPUT = 2VP-P SINGLE ENDED
–30 SFDR = 93dB
–40
–50
–60
–70
–80
–90
–100
–110
–120
0 0.175 0.35 0.525 0.7 0.875 1.05 1.225 1.4
FREQUENCY (MHz)
Figure 5. 4096 sample FFT of
the LT1994 driving a 14-bit ADC
Linear Technology Magazine • December 2005
DESIGN FEATURES
R3
464Ω
3V
1
C1
270pF
2
0.1µF
8
R3
464Ω
0.1µF
3
– +
VOCM
4
LT1994
5
+ –
7
R1
232Ω
point FFT. The spurious free dynamic
range is about 93dB and is limited
by the non-linearities of the ADC
rather than the LT1994 (The SFDR of
the LTC1403A-1 is specified around
86dB at 1.4MHz). This shows that the
LT1994 has no problem settling and
accommodating the LTC1403A’s 39ns
acquisition times.
R2
232Ω
C2
68pF
6
Single 3V Supply, 2.5MHz,
2nd Order Fully Differential
Butterworth Filter
C2
68pF
R2
232Ω
Figure 6 shows a low noise, single
supply, butterworth active filter with
a 2.5MHz bandwidth suitable for antialiasing applications. The differential
output spot noise at 50kHz is about
7nV/√Hz, and the amplifier provides
about 40dB of stopband rejection at
25MHz. The filter’s frequency response
is shown in Figure 7. The filter’s low
frequency gain is set by the ratio of R2
to R1. If a different cutoff frequency
is desired, the capacitors C1, and C2
can easily be scaled inversely with
cutoff frequency.
Figure 6. Low noise differential active RC filter
insufficient settling time creates a
voltage divider between the dynamic
input impedance of the ADC and the
decoupling resistors. Too small of a
resistor possibly prevents the resistor from properly dampening the load
transient caused by the sampling
process, prolonging the time required
for settling.
Start with 25Ω on each output to
decouple the ADC input capacitance.
Then choose a capacitance (taking
account of the sampling capacitance),
which gives the amplifier time to settle
to desired accuracy during the acquisition period. In 16-bit applications,
this typically requires a minimum of
11 RC time constants. The capacitor
chosen should have a high quality dielectric (for example, C0G multi-layer
ceramic). Figure 4 shows the LT1994
driving the LTC1403A-1, a 14-bit
ADC, sampling at 2.8MHz on a single
3V supply. Figure 5 shows its 4096-
Gain-of-2 Amplifier
(No resistors required)
Figure 8 shows the LT1994 configured
in circuit configuration in which the
output consists of an in-phase and
an out-of-phase representation of
the input signal. The circuit has the
benefit of having high input impedance. The input-to-output transfer
function is governed by the equation:
VOUT = 2 • VIN
0.1µF
1V
2
8
1V
0V
3
0V
– +
VOCM
4
LT1994
5
+ –
7
VIN
VS = 3V
–10
–20
–30
–40
–50
–60
0.001
0.01
0.1
1
FREQUENCY (MHz)
10
100
Figure 7. Differential filter response
The circuit works well enough, but
the consequence of such a configuration is that it reflects the performance
of the common mode path, rather than
the differential path. Because of this,
the output does not have the benefit
of the differential noise (3nV/√Hz), but
rather is swamped by the common
mode noise of 15nV/√Hz gained up by
a factor of two (30nV/√Hz). This is a
consequence of mismatch in feedback
factors from the LT1994 outputs to
their respective inputs. In fact, whenever the two feedback paths from the
output to the input mismatch, and to
the degree they mismatch, common
mode noise is converted to differential
noise at the output.
eNO(DIFF ) = 2eN( VOCM)
(βF1 – βF 2 )
(βF1 + βF 2 )
where βF1, and βF2 are the two feedback factors from each output to their
respective input.
Conclusion
5V
1
0
GAIN (dB)
R1
232Ω
6
VOUT
–1V
1V
0V
0.1µF
–1V
–1V
–5V
The LT1994’s low noise, low distortion,
and high performance make it an ideal
amplifier for interfacing with single
supply ADCs. Its rail-to-rail outputs,
low distortion, and 3nV/√Hz input
referred voltage noise maximize dynamic range, and its ability to common
mode to ground eliminates the need
for a negative supply in single supply
systems, saving cost and power.
Figure 8. Gain of two (no resistors required)
For more information on parts featured in this issue, see
http://www.linear.com/designtools
Linear Technology Magazine • December 2005
15
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