INTERSIL ISL6227CAZ

ISL6227
®
Data Sheet
December 21, 2006
FN9094.4
Dual Mobile-Friendly PWM Controller with
DDR Option
Features
The ISL6227 dual PWM controller delivers high efficiency
precision voltage regulation from two synchronous buck DC/DC
converters. It was designed especially to provide power
regulation for DDR memory, chipsets, graphics and other
system electronics in Notebook PCs. The ISL6227’s wide
input voltage range capability allows for voltage conversion
directly from AC/DC adaptor or Li-Ion battery pack.
• Operates from an input battery voltage range of 5V to 24V
or from 3.3V/5V system rail
Automatic mode transition of constant-frequency synchronous
rectification at heavy load, and hysteretic (HYS) diodeemulation at light load, assure high efficiency over a wide range
of conditions. The HYS mode of operation can be disabled
separately on each PWM converter if constant-frequency
continuous-conduction operation is desired for all load levels.
Efficiency is further enhanced by using the lower MOSFET
rDS(ON) as the current sense element.
Voltage-feed-forward ramp modulation, current mode control,
and internal feedback compensation provide fast response to
input voltage and load transients. Input current ripple is
minimized by channel-to-channel PWM phase shift of 0°, 90° or
180° (determined by input voltage and status of the DDR pin).
The ISL6227 can control two independent output voltages
adjustable from 0.9V to 5.5V, or by activating the DDR pin,
transform into a complete DDR memory power supply solution.
In DDR mode, CH2 output voltage VTT tracks CH1 output
voltage VDDQ. CH2 output can both source and sink current, an
essential power supply feature for DDR memory. The reference
voltage VREF required by DDR memory is generated as well.
In dual power supply applications the ISL6227 monitors the
output voltage of both CH1 and CH2. An independent PGOOD
(power good) signal is asserted for each channel after the
soft-start sequence has completed, and the output voltage is
within PGOOD window. In DDR mode CH1 generates the only
PGOOD signal.
Built-in overvoltage protection prevents the output from going
above 115% of the set point by holding the lower MOSFET on
and the upper MOSFET off. When the output voltage re-enters
regulation, PGOOD will go HIGH and normal operation
automatically resumes. Once the soft-start sequence has
completed, undervoltage protection latches the offending
channel off if the output drops below 75% of its set point value.
Adjustable overcurrent protection (OCP) monitors the voltage
drop across the rDS(ON) of the lower MOSFET. If more precise
current-sensing is required, an external current sense resistor
may be used.
1
• Provides regulated output voltage in the range 0.9V–5.5V
• Complete DDR1 and DDR2 memory power solution with
VTT tracking VDDQ/2 and a VDDQ/2 buffered reference
output
• Flexible PWM or HYS plus PWM mode selection with HYS
diode emulation at light loads for higher system efficiency
• rDS(ON) current sensing
• Excellent dynamic response with voltage feed-forward and
current mode control accommodating wide range LC filter
selections
• Undervoltage lock-out on VCC pin
• Power-good, overcurrent, overvoltage, undervoltage
protection for both channels
• Synchronized 300kHz PWM operation in PWM mode
• Pb-free plus anneal available (RoHS compliant)
Applications
• Notebook PCs and Desknotes
• Tablet PCs/Slates
• Hand-held portable instruments
Ordering Information
PART #
PART
MARKING
ISL6227CA
ISL6227CA
TEMP.
RANGE (°C)
PACKAGE
PKG.
DWG. #
-10 to +100 28 Ld QSOP M28.15
ISL6227CAZ ISL6227CAZ
(Note)
-10 to +100 28 Ld QSOP M28.15
(Pb-free)
ISL6227IA
ISL6227IA
-40 to +100 28 Ld QSOP M28.15
ISL6227IAZ
(Note)
ISL6227IAZ
-40 to +100 28 Ld QSOP M28.15
(Pb-free)
ISL6227HRZ ISL6227HRZ -10 to +100 28 Ld QFN
(Note)
(Pb-free)
L28.5x5
ISL6227IRZ
(Note)
L28.5x5
ISL6227IRZ
-40 to +100 28 Ld QFN
(Pb-free)
Add -T to part number for Tape and Reel
NOTE: Intersil Pb-free plus anneal products employ special Pb-free
material sets; molding compounds/die attach materials and 100% matte
tin plate termination finish, which are RoHS compliant and compatible
with both SnPb and Pb-free soldering operations. Intersil Pb-free
products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2004-2006. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6227
20 BOOT2
BOOT1 6
23 BOOT2
ISEN1 7
22 ISEN2
BOOT1 3
21 EN2
VSEN1 10
19 VSEN2
18 OCSET2
SOFT1 12
17 VOUT2
VOUT1 6
16 VSEN2
VSEN1 7
17 SOFT2
15 OCSET2
16 PG2/REF
DDR 13
VIN 14
18 EN2
EN1 5
15 PG1
8
9 10 11 12 13 14
DDR
20 VOUT2
SOFT1
VOUT1 9
OCSET1 11
19 ISEN2
GND
29
ISEN1 4
OCSET1
EN1 8
PGND2
21 UGATE2
UGATE1 2
SOFT2
24 UGATE2
LGATE2
UGATE1 5
28 27 26 25 24 23 22
PHASE1 1
PG2/REF
26 PGND2
25 PHASE2
VCC
PGND1 3
PHASE1 4
PG1
28 VCC
27 LGATE2
VIN
GND 1
LGATE1 2
GND
TOP VIEW
LGATE1
28 LD 5X5 QFN
PGN1
28LD SSOP
TOP VIEW
PHASE2
Pinouts
Generic Application Circuits
OCSET1
L1
Q1
VOUT1
PWM1
C1
Q2
VIN
+1.80V
+
EN1
+5V to +24V
EN2
Q3
VCC
DDR
+5V
L2
VOUT2
PWM2
OCSET2
C2
Q4
+
+1.20V
ISL6227 APPLICATION CIRCUIT FOR TWO CHANNEL POWER SUPPLY
OCSET1
Q1
L1
VDDQ
PWM1
Q2
VIN
C1
+2.50V
+
EN1
EN2
+5V to 24V
Q3
VCC
L2
DDR
VTT
PWM2
+5V
PG2/VREF
OCSET2
Q4
C2
+
+1.25V
VREF
+1.25V
ISL6227 APPLICATION CIRCUIT FOR COMPLETE DDR MEMORY POWER SUPPLY
2
FN9094.4
December 21, 2006
ISL6227
Absolute Maximum Ratings
Thermal Information
Bias Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +6.5V
Input Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . GND - 0.3V to +25.0V
PHASE, UGATE . . . . . . . . . . . . . . . . . . . .GND-5V (Note 1) to 33.0V
BOOT, ISEN . . . . . . . . . . . . . . . . . . . . . . . . . . . GND-0.3V to +33.0V
BOOT with Respect to PHASE . . . . . . . . . . . . . . . . . . . . . . . . + 6.5V
All Other Pins . . . . . . . . . . . . . . . . . . . . . . GND -0.3V to VCC + 0.3V
Thermal Resistance (Typical)
θJA (°C/W) θJC (°C/W)
80
N/A
SSOP Package (Note 2) . . . . . . . . . . . . . .
QFN Package(Notes 3, 4) . . . . . . . . . . . .
36
6
Maximum Junction Temperature (Plastic Package). . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . .-65°C to +150°C
Recommended Operating Conditions
Bias Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5.0V ± 5%
Input Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . .+5.0V to +24.0V
Ambient Temperature Range . . . . . . . . . . . . . . . . . -10°C to +100°C
Junction Temperature Range . . . . . . . . . . . . . . . . . -10°C to +125°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTES:
1. 250ns transient. See Confining The Negative Phase Node Voltage Swing in Application Information Section
2. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
3. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See Tech
Brief TB379.
4. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted.
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
-
1.8
3.0
mA
ICCSN
-
-
1
μA
Rising VCC Threshold
VCCU
4.3
4.45
4.5
V
Falling VCC Threshold
VCCD
4
4.14
4.34
V
IVIN
-
-
35
μA
IVINS
-
-
1
μA
Commercial, ISL6227C
255
300
345
kHz
Industrial, ISL6227I
240
300
345
kHz
VCC SUPPLY
Bias Current
ICC
Shut-down Current
LGATEx, UGATEx Open, VSENx forced above
regulation point, VIN > 5V
VCC UVLO
VIN
Input Voltage Pin Current (Sink)
Shut-Down Current
OSCILLATOR
PWM1 Oscillator Frequency
Fc
Ramp Amplitude, pk-pk
VR1
VIN = 16V, by design
-
2
-
V
Ramp Amplitude, pk-pk
VR2
VIN = 5V, by design
-
0.625
-
V
Ramp Offset
VROFF
By design
-
1
-
V
Ramp/VIN Gain
GRB1
VIN ≥ 4.2V, by design
-
125
-
mV/V
Ramp/VIN Gain
GRB2
VIN ≤ 4.1V by design
-
250
-
mV/V
-
0.9
-
V
-1.0
-
+1.0
%
-
-4.5
-
μA
-
1.5
-
V
REFERENCE AND SOFT-START
Internal Reference Voltage
VREF
Reference Voltage Accuracy
Soft-Start Current During Start-Up
Soft-Start Complete Threshold
ISOFT
VST
3
By design
FN9094.4
December 21, 2006
ISL6227
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Noted. (Continued)
PARAMETER
SYMBOL
TEST CONDITIONS
MIN
TYP
MAX
UNITS
-2.0
-
+2.0
%
-
80
-
nA
PWM CONVERTERS
0.0mA < IVOUT1 < 5.0A; 5.0V < VBATT < 24.0V
Load Regulation
By design
VSEN Pin Bias Current
IVSEN
Minimum Duty Cycle
Dmin
-
4
-
%
Maximum Duty Cycle
Dmax
-
87
-
%
VOUT Pin Input Impedance
IVOUT
VOUT = 5V
-
134
-
kΩ
Undervoltage Shut-Down Level
VUVL
Fraction of the set point; ~2μs noise filter
70
75
80
%
VOVP1
Fraction of the set point; ~2μs noise filter
110
115
-
%
Overvoltage Protection
GATE DRIVERS
Upper Drive Pull-Up Resistance
R2UGPUP
VCC = 5V
-
4
8
Ω
Upper Drive Pull-Down Resistance
R2UGPDN
VCC = 5V
-
2.3
4
Ω
Lower Drive Pull-Up Resistance
R2LGPUP
VCC = 5V
-
4
8
Ω
Lower Drive Pull-Down Resistance
R2LGPDN
VCC = 5V
-
1.1
3
Ω
POWER GOOD AND CONTROL FUNCTIONS
Power Good Lower Threshold
VPG-
Fraction of the set point; ~3μs noise filter
84
89
92
%
Power Good Higher Threshold
VPG+
Fraction of the set point; ~3μs noise filter.
110
115
120
%
IPGLKG
VPULLUP = 5.5V
-
-
1
μA
VPGOOD
IPGOOD = -4mA
-
0.5
1
V
By design
-
-
260
μA
OCSET Sourcing Current Range
2
-
20
μA
EN - Low (Off)
-
-
0.8
V
EN - High (On)
2.0
-
-
V
-
-
0.1
V
0.9
-
-
V
DDR - Low (Off)
-
-
0.8
V
DDR - High (On)
3
-
-
V
0.99*
VOC2
VOC2
1.01*
VOC2
V
-
10
12
mΑ
PGOODx Leakage Current
PGOODx Voltage Low
ISEN Sourcing Current
Continuous-Conduction-Mode(CCM)
Enforced (HYS Operation Inhibited)
VOUTX pulled low
Automatic CCM/HYS Operation
Enabled
VOUTX connected to the output
DDR REF Output Voltage
VDDREF
DDR = 1, IREF = 0...10mA
DDR REF Output Current
IDDREF
DDR = 1. Guaranteed by design.
4
FN9094.4
December 21, 2006
ISL6227
100
100
90
90
80
80
70
70
EFFICIENCY (%)
EFFICIENCY (%)
Typical Operation Performance
60
50
[email protected] 5V
[email protected] 12V
[email protected] 19.5V
[email protected] 5V, PWM
[email protected] 12V, PWM
[email protected] 19.5V, PWM
40
30
20
10
0
0.01
0.1
1
LOAD CURRENT (A)
40
30
0
0.01
10
0.1
1
LOAD CURRENT (A)
10
FIGURE 2. EFFICIENCY OF CHANNEL 2, 1.8V,
HYS/PWM MODE
1.83
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
2.52
[email protected] 5V
[email protected] 12V
[email protected] 19.5V
[email protected] 5V, PWM
[email protected] 12V, PWM
[email protected] 19.5V, PWM
10
Vout @ 5V
Vout @ 12V
Vout @ 19.5V
Vout @ 5V, PWM
Vout @ 12V, PWM
Vout @ 19.5V, PWM
2.53
50
20
FIGURE 1. EFFICIENCY OF CHANNEL 1, 2.5V,
HYS/PWM MODE
2.54
60
2.51
2.5
2.49
Vout @ 5V
Vout @ 12V
Vout @ 19.5V
Vout @ 5V, PWM
Vout @ 12V, PWM
Vout @ 19.5V, PWM
1.82
1.81
1.8
1.79
2.48
2.47
0
1
2
3
LOAD CURRENT (A)
4
1.78
5
0
2
3
LOAD CURRENT (A)
4
5
FIGURE 4. OUTPUT VOLTAGE OF CHANNEL 2 vs LOAD
308
0.9025
306
0.902
304
0.9015
302
0.901
300
VREF (V)
FREQUENCY (KHz)
FIGURE 3. OUTPUT VOLTAGE OF CHANNEL 1 vs LOAD
1
298
296
0.9005
0.9
0.8995
294
0.899
292
0.8985
75% Quantile
Frequency Mean
25% Quantile
290
288
75% Quantile
Vref Mean
25% Quantile
0.898
286
0.8975
-20
0
20
40
60
80
TEMPERATURE (°C)
100
120
FIGURE 5. SWITCHING FREQUENCY OVER TEMPERATURE
5
-20
0
20
40
60
80
TEMPERATURE (°C)
100
120
FIGURE 6. REFERENCE VOLTAGE ACCURACY OVER
TEMPERATURE
FN9094.4
December 21, 2006
ISL6227
Typical Operation Performance
(Continued)
Vo1
Vo1
Vphase1
Vphase1
Ilo1
Ilo1
Vo2
Vo2
FIGURE 7. LOAD TRANSIENT (0 - 3A AT CHANNEL 1)
(DIODE EMULATION MODE)
FIGURE 8. LOAD TRANSIENT (0 - 3A AT CHANNEL 1)
(FORCED PWM MODE)
Vo1
Vo1
Vphase2
Vphase2
Ilo2
Ilo2
Vo2
Vo2
FIGURE 9. LOAD TRANSIENT (0 - 2A AT CHANNEL 2)
(DIODE EMULATION MODE)
FIGURE 10. LOAD TRANSIENT (0 - 3A AT CHANNEL 2)
(FORCED PWM MODE)
Vin1
Vin1
Vo1
Vo1
Vo2
Vo2
FIGURE 11. INPUT STEP-UP TRANSIENT AT PWM MODE
6
FIGURE 12. INPUT STEP-UP TRANSIENT AT HYS MODE
FN9094.4
December 21, 2006
ISL6227
Typical Operation Performance
(Continued)
Vin1
Vin1
Vo1
Vo1
Vo2
Vo2
FIGURE 13. INPUT STEP-DOWN TRANSIENT AT PWM MODE
FIGURE 14. INPUT STEP-DOWN TRANSIENT AT HYS MODE
EN1
EN1
PG1
PG1
SOFT1
SOFT1
Vo1
Vo1
FIGURE 15. SOFT-START INTERVAL AT ZERO INITIAL
VOLTAGE OF VO
FIGURE 16. SOFT-START INTERVAL WITH NON-ZERO INITIAL
VOLTAGE OF VO
Vo1
Vo1
Vphase1
Vphase1
Ilo1
Ilo1
Vo2
Vo2
FIGURE 17. OPERATION AT LIGHT LOAD OF 100mA,
CHANNEL 1
7
FIGURE 18. OPERATION AT HEAVY LOAD OF 4A,
CHANNEL 1
FN9094.4
December 21, 2006
ISL6227
Typical Operation Performance
(Continued)
Vo1
Vo1
PG1
PG1
Ilo1
Ilo1
Vo2
Vo2
FIGURE 19. OVERCURRENT PROTECTION AT CHANNEL 1
Vo1
FIGURE 20. SHORT-CIRCUIT PROTECTION AT CHANNEL 1
Vo1
Vphase1
Vphase2
Ilo1
Ilo2
Vo2
Vo2
FIGURE 21. MODE TRANSITION OF HYS→ _PWM
FIGURE 22. MODE TRANSITION OF PWM→ HYS
EN1
VTT
PG1
OCSET
SOFT1
VDDQ
Vo1
VCC
FIGURE 23. SOFT SHUTDOWN INTERVAL
8
FIGURE 24. VCC POWER-UP IN DDR MODE
FN9094.4
December 21, 2006
ISL6227
Typical Operation Performance
(Continued)
VDDQ
VDDQ
PGOOD1
PGOOD1
VTT
VTT
IL1
IL1
FIGURE 25. VIN = 19V, VDDQ 3A STEP LOAD, VTT 0A LOAD
FIGURE 26. VIN = 19V, VDDQ 3A STEP LOAD, VTT 3A LOAD
VDDQ
VDDQ
VTT
VTT
OCSET2
OCSET2
IL2
IL2
FIGURE 27. VIN = 19V, LOAD STEP ON VTT,
VDDQ HYS MODE, 0.14A
FIGURE 28. VIN = 19V, LOAD STEP ON VTT,
VDDQ PWM MODE, 0.14A
VDDQ
VTT
Vin
EN1
VTT
VDDQ
IL1
OCSET2
FIGURE 29. INPUT LINE TRANSIENT IN DDR MODE
9
FIGURE 30. VTT FOLLOWS VDDQ, ENABLE 2 IS HIGH
FN9094.4
December 21, 2006
ISL6227
Functional Pin Description
OCSET1
Signal ground for the IC.
This pin is a buffered 0.9V internal reference voltage. A
resistor from this pin to ground sets the overcurrent
threshold for the first controller.
LGATE1, LGATE2
SOFT1, SOFT2
These are the outputs of the lower MOSFET drivers.
These pins provide soft-start function for their respective
controllers. When the chip is enabled, the regulated 4.5µA
pull-up current source charges the capacitor connected from
the pin to ground. The output voltage of the converter follows
the ramping voltage on the SOFT pin in the soft-start
process with the SOFT pin voltage as reference. When the
SOFT pin voltage is higher than 0.9V, the error amplifier will
use the internal 0.9V reference to regulate output voltage.
GND
PGND1, PGND2
These pins provide the return connection for lower gate
drivers, and are connected to sources of the lower
MOSFETs of their respective converters.
PHASE1, PHASE2
The PHASE1 and PHASE2 points are the junction points of
the upper MOSFET sources, output filter inductors, and
lower MOSFET drains. Connect these pins to the respective
converter’s upper MOSFET source.
In the event of undervoltage and overcurrent shutdown, the
soft-start pin is pulled down though a 2k resistor to ground to
discharge the soft-start capacitor.
UGATE1, UGATE2
DDR
These pins provide the gate drive for the upper MOSFETs.
These pins are used to monitor the voltage drop across the
lower MOSFET for current feedback and Overcurrent
protection. For precise current detection these inputs can be
connected to the optional current sense resistors placed in
series with the source of the lower MOSFETs.
When the DDR pin is low, the chip can be used as a dual
switcher controller. The output voltage of the two channels
can be programmed independently by VSENx pin resistor
dividers. The PWM signals of channel 1 and channel 2 will
be synchronized 180 degrees out-of-phase. When the DDR
pin is high, the chip transforms into a complete DDR memory
solution. The OCSET2 pin becomes an input through a
resistor divider tracking to VDDQ/2. The PG2/REF pin
becomes the output of the VDDQ/2 buffered voltage. The
VDDQ/2 voltage is also used as the reference to the error
amplifier by the second channel. The channel phase-shift
synchronization is determined by the VIN pin when DDR = 1
as described in VIN below.
EN1, EN2
VIN
These pins enable operation of the respective converter
when high. When both pins are low, the chip is disabled and
only low leakage current is taken from VCC and VIN. EN1
and EN2 can be used independently to enable either
channel 1 or channel 2.
This pin has multiple functions. When connected to battery
voltage, it provides battery voltage to the oscillator as a feedforward for the rejection of input voltage variation. The ramp
of the PWM comparator is proportional to the voltage on this
pin (see Table 1 and Table 2 for details). While the DDR pin
is high in the DDR application, and when the VIN pin voltage
is greater than 4.2 volt when connecting to a battery, it
commands 90° out-of-phase channel synchronization, with
the second channel lagging the first channel, to reduce
inter-channel interference. When the pin voltage is less than
4.2V, this pin commands in-phase channel synchronization.
BOOT1, BOOT2
These pins power the upper MOSFET drivers of the PWM
converter. Connect this pin to the junction of the bootstrap
capacitor with the cathode of the bootstrap diode. The anode
of the bootstrap diode is connected to the VCC voltage.
ISEN1, ISEN2
VOUT1, VOUT2
These pins, when connected to the converter’s respective
outputs, set the converter operating in a mixed hysteretic
mode or PWM mode, depending on the load conditions. It
also provides the voltage to the chip to clamp the PWM error
amplifier in hysteretic mode to achieve smooth HYS/PWM
transition. When connected to ground, these pins command
forced continuous conduction mode (PWM) at all load levels.
VSEN1, VSEN2
PG1
PGOOD1 is an open drain output used to indicate the status
of the output voltage. This pin is pulled low when the first
channel output is out of -11% to +15% of the set value.
These pins are connected to the resistive dividers that set
the desired output voltage. The PGOOD, UVP, and OVP
circuits use this signal to report output voltage status.
10
FN9094.4
December 21, 2006
ISL6227
PG2/REF
Typical Application
This pin has a double function, depending on the mode of
operation.
Figures 31 and 32 show the application circuits of a dual
channel DC/DC converter for a notebook PC.
When the chip is used as a dual channel PWM controller
(DDR = 0), the pin provides an open drain PGOOD2 function
for the second channel the same way as PG1. The pin is
pulled low when the second channel output is out of
-11% to +15% of the set value.
The power supply in Figure 31 provides +2.5V and +1.8V
voltages for memory and the graphics interface chipset from
a +5.0-24VDC battery voltage.
In DDR mode (DDR = 1), this pin is the output of the buffer
amplifier that takes VDDQ/2 voltage applied to OCSET2 pin
from the resister divider. It can source a typical 10mA
current.
OCSET2
Figure 32 illustrates the application circuit for a DDR memory
power solution. The power supply shown in Figure 32
generates +2.5V VDDQ voltage from a battery. The +1.25V
VTT termination voltage tracks VDDQ/2 and is derived from
+2.5V VDDQ. To complete the DDR memory power
requirements, the +1.25V reference voltage is provided
through the PG2 pin.
In a dual channel application with DDR = 0, a resistor from
this pin to ground sets the overcurrent threshold for the
second channel controller. Its voltage is the buffered internal
0.9V reference.
In the DDR application with DDR = 1, this pin connects to the
center point of a resistor divider tracking the VDDQ/2. This
voltage is then buffered by an amplifier voltage follower and
sent to the PG2/REF pin. It sets the reference voltage of
channel 2 for its regulation.
VCC
This pin powers the controller.
11
FN9094.4
December 21, 2006
ISL6227
VIN
VCC (5V)
D1
BAT54W
Cdc
4.7µF
GND
Cin1
10 µF
Cfb1
0.01µ F
Rbt1
0Ω
Lo1
V1 (2.5V)
Rfb11
17.8k
Cb
Cbt1
0.15µF
4.7µH
Co11
220 µF
Q1
Rs1
2.0k
Co12
4.7µ F
FDS6912A
D2
BAT54W
VIN VCC
BOOT1
DDR
BOOT2
UGATE1
UGATE2
PHASE1
PHASE2
ISEN1
ISEN2
LGATE1
LGATE2
PGND1
PGND2
VOUT1
VOUT2
VSEN1
VSEN2
PG1
Rfb12
10k
Cbt2
0.15µF
Cin2
10µF
0Ω
Lo2
Rs2
2.0k
Q2
V2
V (1.8V)
4.7µH
Co21 Co22
220 µF 4.7µF
U1
Rfb21
10k
Rfb22
10k
EN2
SOFT1
SOFT2
OCSET2
OCSET1
Rset1
100k
Cfb2
0.01µF
FDS6912A
PG2
EN1
Csoft1
0.01 µ F
Rbt2
Csoft2
Rset2
100k
ISL6227
0.01µF
FIGURE 31. TYPICAL APPLICATION CIRCUIT AS DUAL SWITCHER, VOUT1 = 2.5V, VOUT = 1.8V
Vin
VCC (5V)
D1
BAT54W
Cdc
4.7µ F
GND
Cin1
10 µ F
Cbt1
0.15µF
0Ω
Lo1
4.6
4.6µH
VDDQ (2.5V)
Q1
Rfb1
17.8k
Cfb1
0.01
0.01µ F
Rbt1
Co11
Co13
220 µF 4.7µ F
Rs1
2.0k
FDS6912A
VIN VCC
BOOT1
DDR
BOOT2
UGATE1
UGATE2
PHASE1
PHASE2
ISEN1
ISEN2
LGATE1
LGATE2
PGND1
PGND2
VOUT1
VSEN2
VSEN1
PG2_REF
Rbt2
Cbt2
0.15µF
U1
EN1
EN2
Cin2
4.7µF
0Ω
Lo2
Rs2
1.0k
Q2
Vref (VDDQ/2)
Cref
4.7µF
OCSET2_VDDQ/2
VDDQ
Rd1
10k
VDDQ/2
OCSET1
Rset1
100k
Co22
4.7µF
FDS6912A
SOFT1
Csoft1
0.01µF
VTT (1.25V)
1.5µH
1.5
Co21
220µF
VOUT2
PG1
Rfb12
10K
D2
BAT54W VDDQ
SOFT2
ISL6227
Cf
0.1
0.1µF
Csoft2 (N/U)
0.01µF
Rd2
10k
FIGURE 32. TYPICAL APPLICATION AS DDR MEMORY POWER SUPPLY, VDDQ = 2.5V, VTT = 1.25V
12
FN9094.4
December 21, 2006
Block Diagram
BOOT1
PG1
VCC GND
EN1 VOUT1
VOUT2
EN2
BOOT2
REF/PG2
UGATE2
UGATE1
PHASE1
DDR=0
ADAPTIVE DEAD-TIME
DIODE EMULATION
V/I SAMPLE TIMING
PWM/HYS TRANSITION
PGND1
DDR=1
PHASE2
ADAPTIVE DEAD-TIME
DIODE EMULATION
V/I SAMPLE TIMING
PWM/HYS TRANSITION
PGND2
LGATE1
LGATE2
13
VCC
VCC
POR
MODE CHANGE COMP 1
MODE CHANGE COMP 2
ENABLE
HYSTERETIC COMPARATOR 1
SAME STATE FOR
8 CLOCK CYCLES
REQUIRED TO CHANGE
PWM OR HYS MODE
SAME STATE FOR
8 CLOCK CYCLES
REQUIRED TO CHANGE
PWM OR HYS MODE
BIAS SUPPLIES
REFERENCE
HYSTERETIC COMPARATOR 2
FAULT LATCH
ΔVHYS=15mV
ΔVHYS=15mV
SOFT START
15pF
1MΩ
15pF
1MΩ
500kΩ
1.25pF
VSEN2
1.25pF
OC1 DDR OC2
PWM1
PWM2
(200kΩ, DDR=1)
SOFT2
ERROR AMP 2
ERROR AMP 1
+ 0.9V
REF
4.4kΩ
DDR=0
DUTY CYCLE RAMP GENERATOR
PWM CHANNEL PHASE CONTROL
SOFT1
DDR EN1 EN2
140Ω
CURRENT
SAMPLE
CURRENT
SAMPLE
0
1
1
1
1
1
DDR=1
+
VIN
CH1CH2 φ
0 ⇔ 24.0V
180º
4.2 < VIN < 24.0V
90º
VIN < 4.2
0º
0.9V
REF
140Ω
CURRENT
SAMPLE
CURRENT
SAMPLE
OCSET1
DDR=0
+
0.9V REFERENCE
ISEN2
0.9V REFERENCE
OCSET2
+
DDR=1
OC1
FN9094.4
December 21, 2006
1/2.9
OCSET1
1/33.1
ISEN1
SAME STATE FOR
8 CLOCK CYCLES
REQUIRED TO LATCH
OVERCURRENT FAULT
OC2
VIN
DDR
VCC
SAME STATE FOR
8 CLOCK CYCLES
REQUIRED TO LATCH
OVERCURRENT FAULT
1/2.9
OCSET2
1/33.1
ISEN2
DDR VREF
BUFFER AMP
DDR VTT
REFERENCE
ISL6227
VOLTS/SEC
CLAMP
300kΩ
300kΩ
4.4kΩ
ISEN1
OV UV
PGOOD
DDR MODE
CONTROL
VOLTS/SEC
CLAMP
500kΩ
VSEN1
OV UV
PGOOD
ISL6227
Theory of Operation
Operation
The ISL6227 is a dual channel PWM controller intended for
use in power supplies for graphic chipsets, SDRAM, DDR
DRAM, or other low voltage power applications in modern
notebook and sub-notebook PCs. The IC integrates two
control circuits for two synchronous buck converters. The
output voltage of each controller can be set in the range of
0.9V to 5.5V by an external resistive divider.
The synchronous buck converters can operate from either
an unregulated DC source, such as a notebook battery, with
a voltage ranging from 5.0V to 24V, or from a regulated
system rail of 3.3V or 5V. In either operational mode the
controller is biased from the +5V source.
The controllers operate in the current mode with input
voltage feed-forward which simplifies feedback loop
compensation and rejects input voltage variation. An
integrated feedback loop compensation dramatically
reduces the number of external components.
When the SOFT pin voltage reaches 0.9V, the output voltage
comes into regulation, (see block diagram). When the SOFT
voltage reaches 1.5V, the power good (PGOOD) and the
mode control is enabled. The soft-start process is depicted in
Figure 33.
EN
1
0.9V
1.5V
2
VOUT
3
4
PGOOD
Ch1 5.0V
Ch2 2.0V
Ch4 5.0V
Ch3 1.0V
Depending on the load level, converters can operate either
in a fixed 300kHz frequency mode or in a HYS mode.
Switch-over to the HYS mode of operation at light loads
improves converter efficiency and prolongs battery life. The
HYS mode of operation can be inhibited independently for
each channel if a variable frequency operation is not
desired.
The ISL6227 has a special means to rearrange its internal
architecture into a complete DDR solution. When the DDR
pin is set high, the second channel can provide the capability
to track the output voltage of the first channel. The buffered
reference voltage required by DDR memory chips is also
provided.
Initialization
The ISL6227 initializes if at least one of the enable pins is
set high. The Power-On Reset (POR) function continually
monitors the bias supply voltage on the VCC pin, and
initiates soft-start operation when EN1 or EN2 is high after
the input supply voltage exceeds 4.45V. Should this voltage
drop lower than 4.14V, the POR disables the chip.
Soft-Start
When soft-start is initiated, the voltage on the SOFT pin of
the enabled channel starts to ramp up gradually with the
internal 4.5μA current charging the soft-start capacitor. The
output voltage follows the soft-start voltage with the
converter operating at 300kHz PWM switching frequency.
14
SOFT
M1.00ms
FIGURE 33. START-UP
Even though the soft-start pin voltage continues to rise after
reaching 1.5V, this voltage does not affect the output
voltage. During the soft-start, the converter always operates
in continuous conduction mode independent of the load level
or VOUT pin connection.
The soft-start time (the time from the moment when EN
becomes high to the moment when PGOOD is reported) is
determined by the following equation:
1.5V × Csoft
T SOFT = ---------------------------------4.5μA
(EQ. 1)
The time it takes the output voltage to come into regulation
can be obtained from the following equation.
T RISE = 0.6 × T SOFT
(EQ. 2)
During soft-start stage before the PGOOD pin is ready, the
undervoltage protection is prohibited. The overvoltage and
overcurrent protection functions are enabled.
If the output capacitor has residue voltage before startup,
both lower and upper MOSFETs are in off-state until the softstart capacitor charges equal the VSEN pin voltage. This will
ensure the output voltage starts from its existing voltage
level.
FN9094.4
December 21, 2006
ISL6227
Output Voltage Program
The output voltage of either channel is set by a resistive divider
from the output to ground. The center point of the divider is
connected to the VSEN pin as shown in Figure 34. The
output voltage value is determined by the following equation.
0.9V • ( R1 + R2 )
V O = ---------------------------------------------R2
(EQ. 3)
where 0.9V is the value of the internal reference. The VSEN
pin voltage is also used by the controller for the power good
function and to detect undervoltage and overvoltage
conditions.
Q1
UGATE
L1
VO
ISEN
C1
Q2
CZ
R1
LGATE
VOUT
VSEN
ISL6227
OCSET
R2
ROC
FIGURE 34. OUTPUT VOLTAGE PROGRAM
Operation Mode Control
VOUTx pin programs the two channels of ISL6227 in two
different operational modes:
1. If VOUTx is connected to ground, the channel will be put
into a fixed switching frequency of 300kHz CCM, also
known as forced PWM mode regardless of load
conditions.
2. If the VOUTx is connected to the output voltage, the
channel will operate in either fixed 300kHz PWM mode or
HYS mode, depending on the load conditions. It operates
in the PWM mode when the load current exceeds the
critical discontinuous conduction value, otherwise it will
operate in a HYS mode, as shown in the following table.
VOUT PIN
INDUCTOR
CURRENT
OPERATION
MODE
GND
Any value
Forced PWM
Connects to output voltage
≤ IHYS
HYS
Connects to output voltage
>IHYS1
PWM
15
The critical discontinuous conduction current value for the
PWM to HYS mode switch-over can be calculated by the
following expression.
( V IN – V O ) • V O
I HYS = ---------------------------------------------------2 • F SW • L O • V IN
(EQ. 4)
The HYS mode to PWM switch-over current IHYS1 is
determined by the activation time of the HYS mode
controller. It is affected by the ESR, the inductor value, the
input and output voltage.
VIN
RCS
The two channels can be programmed to operate in different
modes depending on the VOUTx connection and the load
current. Once both channels operate in the PWM mode,
however, they will be synchronized to the 300kHz switching
clock. The 180° phase shift reduces the noise couplings
between the two channels and reduces the input current ripple.
The HYS mode control can improve converter efficiency with
reduced switching frequency. The efficiency is further
improved by the diode emulation scheme in discontinuous
conduction mode. The diode emulation scheme does not
allow the inductor sink current from the output capacitor,
thereby reducing the circulating energy. It is achieved by
sensing the free-wheeling current going through the
synchronous MOSFET through Phase node voltage polarity
change after the upper MOSFET is turned off. Before the
current reverses direction, the lower MOSFET gate pulses
are terminated.
The PWM-HYS and HYS-PWM switch-over is provided
automatically by the mode control circuit, which constantly
monitors the inductor current through phase voltage polarity,
and alters the way the gate driver pulse signal is generated.
Mode Transition
For a buck regulator, if the load current is higher than critical
value IHYS1, the voltage drop on the synchronous MOSFET
in the free-wheeling period is always negative, and vice
versa. The mode control circuit monitors the phase node
voltage in the off-period. The polarity of this voltage is used
as the criteria for whether the load current is greater than the
critical value, and thus determines whether the converter will
operate in PWM or HYS mode.
To prevent chatter between operating modes, the circuit
looks for eight sequentially matching polarity signals before it
decides to perform a mode change. The algorithm is true for
both CCM-HYS and HYS-CCM transitions.
In the HYS mode, the PWM comparator and the error
amplifier, that provided control in the CCM mode, are put in a
clamped stage and the hysteretic comparator is activated. A
change is also made to the gate logic. The synchronous
MOSFET is controlled in diode emulation fashion, hence the
current in the synchronous MOSFET will be kept in one
direction only. Figures 35 and 36 illustrate the mode change
by counting eight switching cycles.
FN9094.4
December 21, 2006
ISL6227
VOUT
waiting for the transition to occur. The transition decision
point is aligned with the PWM clock. When the need for
transition is detected, there is a 500ns delay between the
first/last pulse of the PWM controller from the last/first pulse
of the hysteretic mode controller.
t
IIND
t
PHASE
COMP
t
1 2 3 4 5 6 7 8
MODE
OF
OPERAT ION
Current Sensing
Hysteretic
PWM
t
FIGURE 35. CCM—HYSTERETIC TRANSITION
VOUT
t
IIND
1
2 3 4 5
6 7
t
8
PHASE
COMP
MODE
OF
OPERAT ION
t
Hysteretic
PWM
t
FIGURE 36. HYSTERETIC—CCM TRANSITION
If load current slowly increases or decreases, mode
transition will occur naturally, as described above; however,
if there is an instantaneous load current increase resulting in
a large output voltage drop before the hysteretic mode
controller responds, a comparator with threshold of 20mV
below the reference voltage will be tripped, and the chip will
jump into the forced PWM mode immediately. The PWM
controller will process the load transient smoothly.
Once the PWM controller is engaged, 8 consecutive
switching cycles of negative inductor current are required to
transition back to the hysteretic mode. In this way, chattering
between the two modes is prevented. Current sinking during
the 8 PWM switching cycle dumps energy to input,
smoothing output voltage load step-down.
As a side effect to this design, the comparator may be
triggered consistently if the ESR of the capacitor is so big
that the output ripple voltage exceeds the 20mV window,
resulting in a pure PWM pulse.
The PWM error amplifier is put in clamped voltage during the
hysteretic mode. The output voltage through the VOUT pin
and the input voltage through the VIN pin are used to
determine the error amplifier output voltage and the duty
cycle. The error amplifier stays in an armed state while
16
The current on the lower MOSFET is sensed by measuring
its voltage drop within its on-time. In order to activate the
current sampling circuitry, two conditions need to be met.
(1) the Lgate is high and (2) the phase pin sees a negative
voltage for regular buck operation, which means the current
is freewheeling through lower MOSFET. For the second
channel of the DDR application, the phase pin voltage needs
to be higher than 0.1V to activate the current sensing circuit
for bidirectional current sensing. The current sampling
finishes at about 400ns after the lower MOSFET has turned
on. This current information is held for current mode control
and overcurrent protection. The current sensing pin can
source up to 260µA. The current sense resistor and OCSET
resistor can be adjusted simultaneously for the same
overcurrent protection level, however, the current sensing
gain will be changed only according to the current sense
resistor value, which will affect the current feedback loop
gain. The middle point of the Isen current can be at 75µA,
but it can be tuned up and down to fit application needs.
If another channel is switching at the moment the current
sample is finishing, it could cause current sensing error and
phase voltage jitter. In the design stage, the duty cycles and
synchronization have to be analyzed for all the input voltage
and load conditions to reduce the chance of current sensing
error. The relationship between the sampled current and
MOSFET current is given by:
I SEN ( R CS + 140 ) = r DS ( ON ) I D
(EQ. 5)
Which means the current sensing pin will source current to
make the voltage drop on the MOSFET equal to the voltage
generated on the sensing resistor, plus the internal resistor,
along the ISEN pin current flowing path.
Feedback Loop Compensation
Both channel PWM controllers have internally compensated
error amplifiers. To make internal compensation possible
several design measures were taken.
• The ramp signal applied to the PWM comparator has been
made proportional to the input voltage by the VIN pin. This
keeps the product of the modulator gain and the input
voltage constant even when the input voltage varies.
• The load current proportional signal is derived from the
voltage drop across the lower MOSFET during the PWM
off time interval, and is subtracted from the error amplifier
output signal before the PWM comparator input. This
effectively creates an internal current control loop.
FN9094.4
December 21, 2006
ISL6227
The resistor connected to the ISEN pin sets the gain in the
current sensing. The following expression estimates the
required value of the current sense resistor, depending on
the maximum continuous load current, and the value of the
MOSFETs rDS(ON), assuming the ISEN pin sources 75µA
current.
I MAX • r DS ( ON )
R CS = ------------------------------------------ – 140Ω
75μA
Lo
DCR
+
Co
Gm*Vc
+
-
Ro
Vo
ESR
FIGURE 37. THE EQUIVALENT CIRCUIT OF THE POWER
STAGE WITH CURRENT LOOP INCLUDED
The value of the injected resistor can be estimated by:
V IN r DS ( ON )
R i = ----------------- ---------------------------- • 4.4kΩ
V ramp R CS + 140
(EQ. 7)
Ri is in kΩ, and RDS and RCS are in Ω . VIN divided by
Vramp, is defined as Gm, which is a constant 8 or 18 dB for
both channels in dual switcher applications, when VIN is
above 3V. Refer to Table 1 for the ramp amplitude in different
VIN pin connections. The feed-forward effect of the VIN is
reflected in Gm. Vc is defined as the error amplifier output
voltage.
TABLE 1. PWM COMPARATOR RAMP AMPLITUDE FOR
DUAL SWITCHER APPLICATION
VIN PIN CONNECTIONS
Ch1 and Ch2 Input Voltage
VIN PIN CONNECTION
Ch1
(EQ. 6)
Because the current sensing circuit is a sample-and-hold
type, the information obtained at the last moment of the
sampling is used. This current sensing circuit samples the
inductor current very close to its peak value. The current
feedback essentially injects a resistor Ri in series with the
original LC filter as shown in Figure 37, where the sampleand-hold effect of the current loop has been ignored. Vc and
Vo are small signal components extracted from its DC
operation points.
Ri
TABLE 2. PWM COMPARATOR RAMP VOLTAGE AMPLITUDE
FOR DDR APPLICATION
VRAMP
AMPLITUDE
Input voltage >4.2V
Vin/8
Input voltage <4.2V
1.25V
GND
1.25V
17
Ch2
Input Voltage
VRAMP
AMPLITUDE
Input voltage >4.2V
Vin/8
Input voltage <4.2V
1.25V
GND
1.25V
Input voltage >4.2V
0.625V
GND
1.25V
The small signal transfer function from the error amplifier
output voltage Vc to the output voltage Vo can be written in
the following expression:
s
⎛ -------- + 1⎞
Ro
⎝ Wz
⎠
G ( s ) = G m --------------------------------------- --------------------------------------------------------R i + DCR + R o ⎛ s
s
⎞
⎛
------------- + 1 ------------- + 1⎞
⎝ Wp1
⎠ ⎝ Wp2
⎠
(EQ. 8)
The dc gain is derived by shorting the inductor and opening
the capacitor. There is one zero and two poles in this transfer
function. The zero is related to ESR and the output
capacitor.
The first pole is a low frequency pole associated with the
output capacitor and its charging resistors. The inductor can
be regarded as short. The second pole is the high frequency
pole related to the inductor. At high frequency the output
capacitor can be regarded as a short circuit. By
approximation, the poles and zero are inversely proportional
to the time constants, associated with inductor and capacitor,
by the following expressions:
1
Wz = -----------------------ESR*C o
(EQ. 9)
1
Wp1 = ------------------------------------------------------------------------------( ESR + ( Ri + DCR ) || R o )*C o
(EQ.10)
R i + DCR + ESR || R o
Wp2 = ---------------------------------------------------------Lo
(EQ.11)
Since the current loop separates the LC resonant poles into
two distant poles, and ESR zero tends to cancel the high
frequency pole, the second order system behaves like a first
order system. This control method simplifies the design of
the internal compensator and makes it possible to
accommodate many applications having a wide range of
parameters.
FN9094.4
December 21, 2006
ISL6227
The schematics for the internal compensator is shown in
Figure 38.
1.25pF
500K
TO PWM
COMPARATOR
4.4K
resistor RCS, output LC filter, and voltage feedback network,
the system loop gain can be accurately analyzed and
modified by the system designers based on the application
requirements.
60
50
1M 15pF
300K
Vc
+
40
VSEN
LC FILTER
30
0.9V
20
GAIN (dB)
ISEN
FIGURE 38. THE INTERNAL COMPENSATOR
10
COMPENSATOR
VO/VC
0
-10
LOOP GAIN
-20
-30
Its transfer function can be written as the following:
5
s
s
1.857 • 10 ⎛ --------------- + 1⎞ ⎛ --------------- + 1⎞
⎝ 2πf
⎠ ⎝ 2πf
⎠
z1
z2
Gcomp ( s ) = --------------------------------------------------------------------------------------------s
s ⎛ --------------- + 1⎞
⎝ 2πf
⎠
p1
-40
-50
-60
100
fz1 = 6.98kHz, fz2 = 380kHz, and f p1 = 137kHz
Outside the ISL6227 chip, a capacitor Cz can be placed in
parallel with the top resistor in the feedback resistor divider,
as shown in Figure 34. In this case the transfer function from
the output voltage to the middle point of the divider can be
written as:
(EQ.13)
The ratio of R1 and R2 is determined by the output voltage
set point; therefore, the position of the pole and zero
frequency in the above equation may not be far apart;
however, they can improve the loop gain and phase margin
with the proper design.
The Cz can bring the high frequency transient output voltage
variation directly to the VSEN pin to cause the PGOOD drop.
Such an effect should be considered in the selection of Cz.
From the analysis above, the system loop gain can be
written as:
Gloop ( s ) = G ( s ) • Gcomp ( s ) • Gfd ( s )
(EQ.14)
Figure 39 shows the composition of the system loop gain. As
shown in the graph, the power stage becomes a well
damped second order system as compared to the LC filter
characteristics. The ESR zero is so close to the high
frequency pole that they cancel each other out. The power
stage behaves like a first order system. With an internal
compensator, the loop gain transfer function has a cross
over frequency at about 30kHz. With a given set of
parameters, including the MOSFET rDS(ON), current sense
18
1•104
1•105
1•106
FREQUENCY (Hz)
FIGURE 39. THE BODE PLOT OF THE LC FILTER,
COMPENSATOR, CONTROL TO OUTPUT
VOLTAGE TRANSFER FUNCTION, AND SYSTEM
LOOP GAIN
where
sR 1 C z + 1
R2
Gfd ( s ) = --------------------- ---------------------------------------------R 1 + R 2 s ( R 1 || R 2 )C z + 1
1•103
(EQ.12)
Gate Control Logic
The gate control logic translates generated PWM signals
into gate drive signals providing necessary amplification,
level shift, and shoot-through protection. It bears some
functions that help to optimize the IC performance over a
wide range of the operational conditions. As MOSFET
switching time can vary dramatically from type to type, and
with the input voltage, the gate control logic provides
adaptive dead time by monitoring real gate waveforms of
both the upper and the lower MOSFETs.
Dual-Step Conversion
The ISL6227 dual channel controller can be used either in
power systems with a single-stage power conversion, when
the battery power is converted into the desired output
voltage in one step, or in the systems where some
intermediate voltages are initially established. The choice of
the approach may be dictated by the overall system design
criteria, or the approach may be a matter of voltages
available to the system designer, as in the case of PCI card
applications.
When the output voltage is regulated from low voltage such
as 5V, the feed-forward ramp may become too shallow,
creating the possibility of duty-factor jitter; this is particularly
relevant in a noisy environment. Noise susceptibility, when
operating from low level regulated power sources, can be
improved by connecting the VIN pin to ground, by which the
feed-forward ramp generator will be internally reconnected
from the VIN pin to the VCC pin, and the ramp slew rate will
be doubled.
FN9094.4
December 21, 2006
ISL6227
Voltage Monitor and Protections
The converter output is monitored and protected against
extreme overload, short circuit, overvoltage, and
undervoltage conditions. A sustained overload on the output
sets the PGOOD low and latches off the offending channel of
the chip. The controller operation can be restored by cycling
the VCC voltage or toggling both enable (EN) pins to low to
clear the latch.
first eight clock cycles, normal operation is restored and the
overcurrent circuit resets itself at the end of sixteenth clock
cycles; see Figure 40.
PGOOD
1
8 CLK
IL
SHUTDOWN
2
Power Good
VOUT
In the soft-start process, the PGOOD is established after the
soft pin voltage is at 1.5V. In normal operation, the PGOOD
window is 100mV below the 0.9V and 135mV higher than
0.9V. The VSEN pin has to stay within this window for
PGOOD to be high. Since the VSEN pin is used for both
feedback and monitoring purposes, the output voltage
deviation can be coupled directly to the VSEN pin by the
capacitor in parallel with the voltage divider as shown in
Figure 4. In order to prevent false PGOOD drop, capacitors
need to parallel at the output to confine the voltage deviation
with severe load step transient. The PGOOD comparator
has a built-in 3µs filter. PGOOD is an open drain output.
Overcurrent Protection
In dual switcher application, both PWM controllers use the
lower MOSFETs on-resistance rDS(ON), to monitor the
current for protection against shorted outputs. The sensed
current from the ISEN pin is compared with a current set by
a resistor connected from the OCSET pin to ground:
10.3V
R SET = --------------------------------------------------------I OC • r DS ( ON )
-------------------------------------- + 8μA
R CS + 140Ω
(EQ.15)
where, IOC is a desired overcurrent protection threshold and
RCS is the value of the current sense resistor connected to
the ISEN pin. The 8µA is the offset current added on top of
the sensed current from the ISEN pin for internal circuit
biasing.
If the lower MOSFET current exceeds the overcurrent
threshold, a pulse skipping circuit is activated. The upper
MOSFET will not be turned on and the lower MOSFET
keeps conducting as long as the sampled current is higher
than the threshold value, limiting the current supplied by the
DC voltage source. The current in the lower MOSFET will be
sampled at the internal 300kHz oscillator frequency and
monitored. When the sampled current is lower than the OC
threshold value, the following UGATE pulse will be released
and it allows turning on the upper MOSFET based on the
voltage regulation loop. This kind of operation remains for
eight clock cycles after the overcurrent comparator was
tripped for the first time. If after the first eight clock cycles the
sampled current exceeds the overcurrent threshold again,
within a time interval of another eight clock cycles, the
overcurrent protection latches and disables the offending
channel. If the overcurrent condition goes away during the
19
3
Ch1 5.0V
Ch3 1.0AΩ
Ch2 100mV
M 10.0μs
FIGURE 40. OVERCURRENT PROTECTION
Due to the nature of the used current sensing technique,
and to accommodate a wide range of the rDS(ON) variation,
the value of the overcurrent threshold should set at about
180% of the nominal load value. If more accurate current
protection is desired, a current sense resistor placed in
series with the lower MOSFET source may be used. The
inductor current going through the lower MOSFET is sensed
and held at 400ns after the upper MOSFET is turned off;
therefore, the sensed current is very close to its peak value.
The inductor peak current can be written as:
( V IN – V o ) • V o
I peak = -------------------------------------------- + I load
2L o • F SW • V IN
(EQ.16)
As seen from the equation above, the inductor peak current
changes with the input voltage and the inductor value once
an output voltage is selected.
After overcurrent protection is activated, there are two ways
to bring the offending channel back: (1) Both EN1 and EN2
have to be held low to clear the latch, (2) To recycle the Vcc
of the chip, the POR will clear the latch.
Undervoltage Protection
In the process of operation, if a short circuit occurs, the output
voltage will drop quickly. Before the overcurrent protection
circuit responds, the output voltage will fall out of the required
regulation range. The chip comes with undervoltage protection.
If a load step is strong enough to pull the output voltage lower
than the undervoltage threshold, the offending channel latches
off immediately. The undervoltage threshold is 75% of the
nominal output voltage. Toggling both pins to low, or recycling
Vcc, will clear the latch and bring the chip back to operation.
Overvoltage Protection
Should the output voltage increase over 115% of the normal
value due to the upper MOSFET failure, or for other reasons,
the overvoltage protection comparator will force the
synchronous rectifier gate driver high. This action actively
pulls down the output voltage and eventually attempts to
blow the battery fuse. As soon as the output voltage is within
FN9094.4
December 21, 2006
ISL6227
regulation, the OVP comparator is disengaged. The
MOSFET driver will restore its normal operation. When the
OVP occurs, the PGOOD will drop to low as well.
This OVP scheme provides a ‘soft’ crowbar function, which
helps clamp the voltage overshoot, and does not invert the
output voltage when otherwise activated with a continuously
high output from lower MOSFET driver - a common problem
for OVP schemes with a latch.
DDR Application
High throughput Double Data Rate (DDR) memory ICs are
replacing traditional memory ICs in the latest generation of
Notebook PCs and in other computing devices. A novel
feature associated with this type of memory are the
referencing and data bus termination techniques. These
techniques employ a reference voltage, VREF, that tracks
the center point of VDDQ and VSS voltages, and an
additional VTT power source where all terminating resistors
are connected. Despite the additional power source, the
overall memory power consumption is reduced compared to
traditional termination.
The added power source has a cluster of requirements that
should be observed and considered. Due to the reduced
differential thresholds of DDR memory, the termination
power supply voltage, VTT, closely tracks VDDQ/2 voltage.
Another very important feature of the termination power
supply is the capability to operate at equal efficiency in
sourcing and sinking modes. The VTT supply regulates the
output voltage with the same degree of precision when
current is flowing from the supply to the load, and when the
current is diverted back from the load into the power supply.
The ISL6227 dual channel PWM controller possesses
several important enhancements that allow re-configuration
for DDR memory applications, and provides all three
voltages required in a DDR memory compliant computer.
To reconfigure the ISL6227 for a complete DDR solution, the
DDR pin should be set high permanently to the VCC rail.
This activates some functions inside the chip that are
specific to DDR memory power needs.
In the DDR application presented in Figure 32, the first
controller regulates the VDDQ rail to 2.5V. The output
voltage is set by external dividers Rfb1 and Rfb12. The
second controller regulates the VTT rail to VDDQ/2. The
OCSET2 pin function is now different, and serves as an
input that brings VDDQ/2 voltage, created by the Rd1 and
Rd2 divider, inside the chip, effectively providing a tracking
function for the VTT voltage.
The PG2 pin function is also different in DDR mode. This pin
becomes the output of the buffer, whose input is connected
to the center point of the R/R divider from the VDDQ output
by the OCSET2 pin. The buffer output voltage serves as a
1.25V reference for the DDR memory chips. Current
capability of this pin is 10mA (12mA max).
20
For the VTT channel where output is derived from the VDDQ
output, some control and protective functions have been
significantly simplified. For example, the overcurrent, and
overvoltage, and undervoltage protections for the second
channel controller are disabled when the DDR pin is set
high. The hysteretic mode of operation is also disabled on
the VTT channel to allow sinking capability to be
independent from the load level. As the VTT channel tracks
the VDDQ/2 voltage, the soft-start function is not required,
and the SOFT2 pin may be left open, in the event both
channels are enabled simultaneously. However, if the VTT
channel is enabled later than the VDDQ, the SOFT2 pin
must have a capacitor in place to ensure soft-start. In case of
overcurrent or undervoltage caused by short circuit on VTT,
the fault current will propagate to the first channel and shut
down the converter.
The VREF voltage will be present even if the VTT is
disabled.
Channel Synchronization in DDR Applications
The presence of two PWM controllers on the same die
requires channel synchronization, to reduce inter-channel
interference that may cause the duty factor jitter and
increased output ripple.
The PWM controller is at greatest noise susceptibility when
an error signal on the input of the PWM comparator
approaches the decision making point. False triggering may
occur, causing jitter and affecting the output regulation.
A common approach used to synchronize dual channel
converters is out-of-phase operation. Out-of-phase
operation reduces input current ripple and provides a
minimum interference for channels that control different
voltage levels.
When the DDR pin is connected to GND for dual switcher
applications, the channels operate 180° out-of-phase. When
used in a DDR application with cascaded converters (VTT
generated from VDDQ), several methods of synchronization
are implemented in the ISL6227. In the DDR mode, when
the DDR pin is connected to VCC, the channels operate
either with 0° phase shift, when the VIN pin is connected to
the GND, or with 90° phase shift if the VIN pin is connected
to a voltage higher than 4.2V.
The following table lists the different synchronization
schemes and their usage:
DDR PIN
VIN PIN
SYNCHRONIZATION
0
Vin pin >4.2V
180° out of phase
1
Vin pin voltage <4.2V
0° phase
1
Vin pin voltage >4.2V
90° phase shift
FN9094.4
December 21, 2006
ISL6227
Application Information
current of the ISEN pin to meet the overcurrent protection
and the change the current loop gain. The lower the current
sensing resistor, the higher gain of the current loop, which
can damp the output LC filter more.
Design Procedures
GENERAL
A ceramic decoupling capacitor should be used between the
VCC and GND pin of the chip. There are three major
currents drawn from the decoupling capacitor:
1. the quiescent current, supporting the internal logic and
normal operation of the IC
2. the gate driver current for the lower MOSFETs
3. and the current going through the external diodes to the
bootstrap capacitor for upper MOSFET.
In order to reduce the noisy effect of the bootstrap capacitor
current to the IC, a small resistor, such as 10Ω, can be used
with the decoupling cap to construct a low pass filter for the
IC, as shown in Figure 41. The soft-start capacitor and the
resistor divider setting the output voltage is easy to select as
discussed in the “Block Diagram” on page 13.
TO BOOT
VCC
5V
10Ω
FIGURE 41. INPUT FILTERING FOR THE CHIP
Selection of the Current Sense Resistor
The value of the current sense resistor determines the gain
of the current sensing circuit. It affects the current loop gain
and the overcurrent protection setpoint. The voltage drop on
the lower MOSFET is sensed within 400ns after the upper
MOSFET is turned off. The current sense pin has a 140Ω
resistor in series with the external current sensing resistor.
The current sense pin can source up to a 260µA current
while sensing current on the lower MOSFET, in such a way
that the voltage drop on the current sensing path would be
equal to the voltage on the MOSFET.
A higher value current-sensing resistor will decrease the
current sense gain. If the phase node of the converter is very
noisy due to poor layout, the sensed current will be
contaminated, resulting in duty cycle jittering by the current
loop. In such a case, a bigger current sense resistor can be
used to reduce both real and noise current levels. This can
help damp the phase node wave form jittering.
Sometimes, if the phase node is very noisy, a resistor can be
put on the ISEN pin to ground. This resistor together with the
RCS can divide the phase node voltage down, seen by the
internal current sense amplifier, and reduce noise coupling.
Sizing the Overcurrent Setpoint Resistor
The internal 0.9V reference is buffered to the OCSET pin
with a voltage follower (refer to the equivalent circuit in
Figure 42). The current going through the external
overcurrent set resistor is sensed from the OCSET pin. This
current, divided by 2.9, sets up the overcurrent threshold and
compares with the scaled ISEN pin current going through
RCS with an 8µA offset. Once the sensed current is higher
than the threshold value, an OC signal is generated. The first
OC signal starts a counter and activates a pulse skipping
function. The inductor current will be continuously monitored
through the phase node voltage after the first OC trip. As
long as the sensed current exceeds the OC threshold value,
the following PWM pulse will be skipped. This operation will
be the same for 8 switching cycles. Another OC occurring
between 8 to 16 switching cycles would result in a latch off
with both upper and lower drives low. If there is no OC within
8 to 16 switching cycles, normal operation resumes.
ISEN
PHASE RCS
140 Ω
_
_
+ Σ + 8uA
+
Rdson
I SOURCING ( R CS + 140Ω ) = I D r DS ( ON )
+
÷ 33.1
(EQ.17)
OCSET
ID can be assumed to be the inductor peak current. In a
worst case scenario, the high temperature rDS(ON) could
increase to 150% of the room temperature level. During
overload condition, the MOSFET drain current ID could be
130% higher than the normal inductor peak. If the inductor
has 30% peak-to-peak ripple, ID would equal to 115% of the
load current. The design should consider the above factors
so that the maximum ISOURCING will not saturate to 260µA
under worst case conditions. To be safe, ISOURCING should
be less than 100µA in normal operation at room
temperature. The formula in the earlier discussion assumes
a 75µA sourcing current. Users can tune the sourcing
21
+
Rset
Isense
_
Amplifier
+
_
0.9 V
Reference
OC
Comparator
÷ 2.9
FIGURE 42. EQUIVALENT CIRCUIT FOR OC SIGNAL
GENERATOR
FN9094.4
December 21, 2006
ISL6227
Based on the above description and functional block
diagram, the OC set resistor can be calculated as:
10.3V
R set = --------------------------------------------------I OC r DS ( ON )
--------------------------------- + 8μA
R CS + 140
(EQ.18)
IOC is the inductor peak current and not the load current.
Since inductor peak current changes with input voltage, it is
better to use an oscilloscope when testing the overcurrent
setting point to monitor the inductor current, and to
determine when the OC occurs. To get consistent test results
on different boards, it is best to keep the MOSFET at a fixed
temperature.
The MOSFET will not heat-up when applying a very low
frequency and short load pulses with an electronic load to
the output.
As an example, assume the following:
• The maximum normal operation load current is 1
• The inductor peak current is 1.15-1.3 times higher than
the load current, depending on the inductor value and the
input voltage
• The rDS(ON) has a 45% increase at higher temperature
IOC should set at least 1.8 to 2 times higher than the
maximum load current to avoid nuisance overcurrent trip.
Selection of the LC Filter
The duty cycle of a buck converter is a function of the input
voltage and output voltage. Once an output voltage is fixed,
it can be written as:
Vo
D ( V IN ) = --------V IN
selection. Another factor to consider when choosing the
inductor is its saturation characteristics at elevated
temperature. Saturated inductors could result in nuisance
OC, or OV trip.
Output voltage ripple and the transient voltage deviation are
factors that have to be taken into consideration when
selecting an output capacitor. In addition to high frequency
noise related MOSFET turn-on and turn-off, the output
voltage ripple includes the capacitance voltage drop and
ESR voltage drop caused by the AC peak-to-peak current.
These two voltages can be represented by:
I pp
ΔV c = ---------------------8C o F
(EQ.22)
ΔV esr = I pp ESR
(EQ.23)
sw
These two components constitute a large portion of the total
output voltage ripple. Several capacitors have to be
paralleled in order to reduce the ESR and the voltage ripple.
If the output of the converter has to support another load
with high pulsating current, more capacitors are needed in
order to reduce the equivalent ESR and suppress the
voltage ripple to a tolerable level.
To support a load transient that is faster than the switching
frequency, more capacitors have to be used to reduce the
voltage excursion during load step change. Another aspect
of the capacitor selection is that the total AC current going
through the capacitors has to be less than the rated RMS
current specified on the capacitors, to prevent the capacitor
from over-heating.
(EQ.19)
The switching frequency, Fsw, of ISL6227 is 300kHz. The
peak-to-peak ripple current going through the inductor can
be written as:
V o ( 1 – D ( V IN ) )
I pp = ----------------------------------------F sw L o
(EQ.20)
As higher ripple current will result in higher switching loss
and higher output voltage ripple, the peak-to-peak current of
the inductor is generally designed with a 20%-40% peak-topeak ripple of the nominal operation current. Based on this
assumption, the inductor value can be selected with the
above equation. In addition to the mechanical dimension, a
shielded ferrite core inductor with a very low DC resistance,
DCR, is preferred for less core loss and copper loss. The DC
copper loss of the inductor can be estimated by:
2
(EQ.21)
P copper = I load DCR
The inductor copper loss can be significant in the total
system power loss. Attention has to be given to the DCR
22
FN9094.4
December 21, 2006
ISL6227
Selection of the Input Capacitor
When the upper MOSFET is on, the current in the output
inductor will be seen by the input capacitor. Even though this
current has a triangular shape top, its RMS value can be
fairly approximated by:
lin rms ( V IN ) =
(EQ.24)
D ( V IN )*I load
2
P lower ( V IN ) ≈ ( 1 – D ( V IN ) )I load rDS ( ON )Lower
(EQ.26)
For the upper MOSFET, its conduction loss can be written
as:
2
This RMS current includes both DC and AC components.
Since the DC component is the product of duty cycle and
load current, the AC component can be approximated by:
li nac ( V IN ) =
For the lower MOSFET, its power loss can be assumed to be
the conduction loss only.
2
( D ( V IN ) – D ( V IN ) )I load
(EQ.25)
AC components will be provided from the input capacitor.
The input capacitor has to be able to handle this ripple
current without overheating and with tolerable voltage ripple.
In addition to the capacitance, a ceramic capacitor is
generally used between the drain terminal of the upper
MOSFET and the source terminal of the lower MOSFET, in
order to clamp the parasitic voltage ringing at the phase
node in switching.
Choosing MOSFETs
For a notebook battery with a maximum voltage of 24V, at
least a minimum 30V MOSFETs should be used. The design
has to trade off the gate charge with the rDS(ON) of the
MOSFET:
• For the lower MOSFET, before it is turned on, the body
diode has been conducting. The lower MOSFET driver will
not charge the miller capacitor of this MOSFET.
• In the turning off process of the lower MOSFET, the load
current will shift to the body diode first. The high dv/dt of
the phase node voltage will charge the miller capacitor
through the lower MOSFET driver sinking current path.
This results in much less switching loss of the lower
MOSFETs.
The duty cycle is often very small in high battery voltage
applications, and the lower MOSFET will conduct most of
the switching cycle; therefore, the lower the rDS(ON) of the
lower MOSFET, the less the power loss. The gate charge for
this MOSFET is usually of secondary consideration.
The upper MOSFET does not have this zero voltage
switching condition, and because it conducts for less time
compared to the lower MOSFET, the switching loss tends to
be dominant. Priority should be given to the MOSFETs with
less gate charge, so that both the gate driver loss, and
switching loss, will be minimized.
23
P uppercond ( V IN ) = D ( V IN )I load RDS ( ON )upper
(EQ.27)
and its switching loss can be written as:
V IN I vally T on F sw V IN I peak T off F sw
P uppersw ( V IN ) = ---------------------------------------------- + ----------------------------------------------2
2
(EQ.28)
The peak and valley current of the inductor can be obtained
based on the inductor peak-to-peak current and the load
current. The turn-on and turn-off time can be estimated with
the given gate driver parameters in the Electrical
Specification Table on page 3. For example, if the gate driver
turn-on path of MOSFET has a typical on-resistance of 4Ω,
its maximum turn-on current is 1.2A with 5V Vcc. This
current would decay as the gate voltage increased. With the
assumption of linear current decay, the turn-on time of the
MOSFETs can be written with:
2Q gd
T on = ----------------I driver
(EQ.29)
Qgd is used because when the MOSFET drain-to-source
voltage has fallen to zero, it gets charged. Similarly, the turnoff time can be estimated based on the gate charge and the
gate drivers sinking current capability.
The total power loss of the upper MOSFET is the sum of the
switching loss and the conduction loss. The temperature rise
on the MOSFET can be calculated based on the thermal
impedance given on the datasheet of the MOSFET. If the
temperature rise is too much, a different MOSFET package
size, layout copper size, and other options have to be
considered to keep the MOSFET cool. The temperature rise
can be calculated by:
T rise = θ jaPtotalpower loss
(EQ.30)
The MOSFET gate driver loss can be calculated with the
total gate charge and the driver voltage Vcc. The lower
MOSFET only charges the miller capacitor at turn-off.
P driver = V cc Q gs F sw
(EQ.31)
Based on the above calculation, the system efficiency can
be estimated by the designer.
FN9094.4
December 21, 2006
ISL6227
Confining the Negative Phase Node Voltage Swing
with Schottky Diode
At each switching cycle, the body diode of the lower MOSFET
will conduct before the MOSFET is turned on, as the inductor
current is flowing to the output capacitor. This will result in a
negative voltage on the phase node. The higher the load
current, the lower this negative voltage. This voltage will ring
back less negative when the lower MOSFET is turned on.
A total 400ns period is given to the current sample-and-hold
circuit on the ISEN pin to sense the current going through
the lower MOSFET after the upper MOSFET turns off. An
excessive negative voltage on the lower MOSFET will be
treated as overcurrent. In order to confine this voltage, a
schottky diode can be used in parallel with the lower
MOSFET for high load current applications. PCB layout
parasitics should be minimized in order to reduce the
negative ringing of phase voltage.
The second concern for the phase node voltage going into
negative is that the boot strap capacitor between the BOOT
and PHASE pin could get be charged higher than VCC
voltage, exceeding the 6.5V absolute maximum voltage
between BOOT and PHASE when the phase node voltage
became negative. A resistor can be placed between the
cathode of the boot strap diode and BOOT pin to increase
the charging time constant of the boot cap. This resistor will
not affect the turn-on and off of the upper MOSFET.
Schottky diode can reduce the reverse recovery of the lower
MOSFET when transition from freewheeling to blocking,
therefore, it is generally good practice to have a schottky
diode closely parallel with the lower MOSFET. B340LA, from
Diodes, Inc.®, can be used as the external schottky diode.
Tuning the Turn-on of Upper MOSFET
The turn-on speed of the upper MOSFET can be adjusted by
the resistor connecting the boot cap to the BOOT pin of the
chip. This resistor can confine the voltage ringing on the boot
capacitor from coupling to the boot pin. This resistor slows
down only the turn-on of the upper MOSFET.
If the upper MOSFET is turned on very fast, it could result in
a very high dv/dt on the phase node, which could couple into
the lower MOSFET gate through the miller capacitor,
causing momentous shoot-through. This phenomenon,
together with the reverse recovery of the body diode of the
lower MOSFET, can over-shoot the phase node voltage to
beyond the voltage rating of the MOSFET. However, a bigger
resistor will slow the turn-on of the MOSFET too much and
lower the efficiency. Trade-offs need to be made in choosing
a suitable resistor value.
24
System Loop Gain and Stability
The system loop gain is a product of three transfer functions:
1. the transfer function from the output voltage to the
feedback point,
2. the transfer function of the internal compensation circuit
from the feedback point to the error amplifier output voltage,
3. and the transfer function from the error amplifier output to
the converter output voltage.
These transfer functions are written in a closed form in the
Theory of Operation section on page 18. The external
capacitor, in parallel with the upper resistor of the resistor
divider, Cz, can be used to tune the loop gain and phase
margin. Other component parameters, such as the inductor
value, can be changed for a wider cross-over frequency of the
system loop gain. A body plot of the loop gain transfer function
with a 45 degree phase margin (a 60 degree phase margin is
better) is desirable to cover component parameter variations.
Testing the Overvoltage on Buck Converters
For synchronous buck converters, if an active source is used
to raise the output voltage for the overvoltage protection test,
the buck converter will behave like a boost converter and
dump energy from the external source to the input. The
overvoltage test can be done on ISL6227 by connecting the
VSEN pin to an external voltage source or signal generator
through a diode. When the external voltage, or signal
generator voltage, is tuned to a higher level than the
overvoltage threshold (the lower MOSFET will be on), it
indicates the overvoltage protection works. This kind of
overvoltage protection does not require an external schottky
in parallel with the output capacitor.
Layout Considerations
Power and Signal Layer Placement on the PCB
As a general rule, power layers should be close together,
either on the top or bottom of the board, with signal layers on
the opposite side of the board. For example, prospective
layer arrangement on a 4 layer board is shown below:
1. Top Layer: ISL6227 signal lines
2. Signal Ground
3. Power Layers: Power Ground
4. Bottom Layer: Power MOSFET, Inductors and other
Power traces
It is a good engineering practice to separate the power
voltage and current flowing path from the control and logic
level signal path. The controller IC will stay on the signal
layer, which is isolated by the signal ground to the power
signal traces.
FN9094.4
December 21, 2006
ISL6227
Component Placement
BOOT1 and BOOT2
The control pins of the two-channel ISL6227 are located
symmetrically on two sides of the IC; it is desirable to
arrange the two channels symmetrically around the IC.
These pins di/dt are as high as that of the UGATEx;
therefore, the traces should be as short as possible.
The power MOSFET should be close to the IC so that the
gate drive signal, the LGATEx, UGATEx, PHASEx, BOOTx,
and ISENx traces can be short.
The ISEN trace should be a separate trace, and
independently go to the drain terminal of the lower MOSFET.
The current sense resistor should be close to ISEN pin.
Place the components in such a way that the area under the
ISL6227 has fewer noise traces with high dv/dt and di/dt,
such as gate signals and phase node signals.
The loop formed by the bottom MOSFET, output inductor,
and output capacitor, should be very small. The source of
the bottom MOSFET should tie to the negative side of the
output capacitor in order for the current sense pin to get the
voltage drop on the RDSON.
Signal Ground and Power Ground Connection
At minimum, a reasonably large area of copper, which will
shield other noise couplings through the IC, could be used
as signal ground beneath the ISL6227. The best tie-point
between the signal ground and the power ground is at the
negative side of the output capacitor on each channel, where
there is less noise. Noisy traces beneath the ISL6227 are
not recommended.
GND and VCC
At least one high quality ceramic decoupling cap should be
used across these two pins. A via can tie Pin 1 to signal
ground. Since Pin1 and Pin 28 are close together, the
decoupling cap can be put close to the IC.
LGATE1 and LGATE2
These are the gate drive signals for the bottom MOSFETs of
the buck converter. The signal going through these traces
have both high dv/dt and high di/dt, with high peak charging
and discharging current. These two traces should be short,
wide, and away from other traces. There should be no other
weak signal traces in parallel with these traces on any layer.
PGND1 and PGND2
Each pin should be laid out to the negative side of the
relevant output cap with separate traces.The negative side
of the output capacitor must be close to the source node of
the bottom MOSFET. These traces are the return path of
LGATE1 and LGATE2.
ISEN1 and ISEN2
EN1 and EN2
These pins stay high in enable mode and low in idle mode
and are relatively robust. Enable signals should refer to the
signal ground.
VOUT1 and VOUT2
These pins connect either to the output voltage or to the
signal ground. They are signal lines and should be kept
away from noisy lines.
VSEN1 and VSEN2
There is usually a resistor divider connecting the output
voltage to this pin. The input impedance of these two pins is
high because they are the input to the amplifiers. The correct
layout should bring the output voltage from the regulation
point to the SEN pin with kelvin traces. Build the resistor
divider close to the pin so that the high impedance trace is
shorter.
OCSET1 and OCSET2
In dual switcher mode operation, the overcurrent set resistor
should be put close to this pin. In DDR mode operation, the
voltage divider, which divides the VDQQ voltage in half,
should be put very close to this pin. The other side of the OC
set resistor should connect to signal ground.
SOFT1 and SOFT2
The soft-start capacitors should be laid out close to this pin.
The other side of the soft-start cap should tie to signal ground.
PHASE1 and PHASE2
These traces should be short, and positioned away from other
weak signal traces. The phase node has a very high dv/dt with
a voltage swing from the input voltage to ground. No trace
should be in parallel with these traces. These traces are also
the return path for UGATE1 and UGATE2. Connect these pins
to the respective converter’s upper MOSFET source.
Pin 5 and Pin 24, the UGATE1 and UGATE2
These pins have a square shape waveform with high dv/dt. It
provides the gate drive current to charge and discharge the
top MOSFET with high di/dt. This trace should wide, short,
and away from other traces similar to the LGATEx.
25
PG1 and PG2/REF
For dual switcher operations, these two lines are less noise
sensitive. For DDR applications, a capacitor should be
placed to the PG2/REF pin.
DDR
This pin should connect to VCC in DDR applications, and to
signal ground in dual switcher applications.
VIN
This pin connects to battery voltage, and is less noise sensitive.
FN9094.4
December 21, 2006
ISL6227
Copper Size for the Phase Node
Decoupling Capacitor for Switching MOSFET
Big coppers on both sides of the Phase node introduce
parasitic capacitance. The capacitance of PHASE should be
kept very low to minimize ringing. If ringing is excessive, it
could easily affect current sample information. It would be
best to limit the size of the PHASE node copper in strict
accordance with the current and thermal management of the
application.
It is recommended that ceramic caps be used closely
connected to the drain side of the upper MOSFET, and the
source of the lower MOSFET. This capacitor reduces the
noise and the power loss of the MOSFET. Refer to Figure 43
for the power component placement.
-
VIN
+
Identify the Power and Signal Ground
The input and output capacitors of the converters, the source
terminals of the bottom switching MOSFET PGND1, and
PGND2, should be closely connected to the power ground.
The other components should connect to signal ground.
Signal and power ground are tied together at the negative
terminal of the output capacitors.
Vo
+
Lo
Lo
FIGURE 43. A GOOD EXAMPLE POWER COMPONENT
REPLACEMENT. IT SHOWS THE NEGATIVE OF
INPUT AND OUTPUT CAP AND THE SOURCE OF
THE MOSFET ARE TIED AT ONE POINT.
26
FN9094.4
December 21, 2006
ISL6227
Quad Flat No-Lead Plastic Package (QFN)
Micro Lead Frame Plastic Package (MLFP)
2X
9
MILLIMETERS
D/2
D1
D1/2
2X
N
6
INDEX
AREA
0.15 C B
1
2
3
E1/2
E1
E
9
0.15 C A
4X
A1
-
0.02
0.05
-
A2
-
0.65
1.00
9
0.30
5,8
A3
0.20 REF
0.18
9
0.25
D
5.00 BSC
-
D1
4.75 BSC
9
2.95
3.10
-
4.75 BSC
2.95
3.10
9
3.25
7,8
0.50 BSC
-
k
0.20
-
-
-
L
0.50
0.60
0.75
8
N
28
2
0
7
3
Ne
7
3
8
P
-
-
0.60
9
NX k
θ
-
-
12
9
7
Rev. 1 11/04
1
(DATUM A)
NOTES:
2
3
6
INDEX
AREA
1. Dimensioning and tolerancing conform to ASME Y14.5-1994.
(Ne-1)Xe
REF.
E2
2. N is the number of terminals.
7
E2/2
NX L
N e
3. Nd and Ne refer to the number of terminals on each D and E.
8
4. All dimensions are in millimeters. Angles are in degrees.
5. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
9
CORNER
OPTION 4X
(Nd-1)Xe
REF.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
BOTTOM VIEW
A1
NX b
7. Dimensions D2 and E2 are for the exposed pads which provide
improved electrical and thermal performance.
5
8. Nominal dimensions are provided to assist with PCB Land Pattern
Design efforts, see Intersil Technical Brief TB389.
SECTION "C-C"
9. Features and dimensions A2, A3, D1, E1, P & θ are present when
Anvil singulation method is used and not present for saw
singulation.
C
L
L1
7,8
Nd
D2
2 N
C
L
3.25
5.00 BSC
4X P
8
9
0.10 M C A B
D2
(DATUM B)
A1
5
4X P
-
e
/ / 0.10 C
0.08 C
NX b
NOTES
1.00
E2
A
A3
MAX
0.90
E1
A2
SIDE VIEW
NOMINAL
0.80
E
B
C
SEATING PLANE
MIN
A
D2
0.15 C B
TOP VIEW
SYMBOL
b
E/2
2X
2X
28 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
(COMPLIANT TO JEDEC MO-220VHHD-1 ISSUE I)
0.15 C A
D
A
L28.5x5
10
L
L1
e
10
L
e
C C
TERMINAL TIP
FOR ODD TERMINAL/SIDE
FOR EVEN TERMINAL/SIDE
27
FN9094.4
December 21, 2006
ISL6227
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
M28.15
N
INDEX
AREA
H
0.25(0.010) M
28 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
B M
E
1
2
INCHES
GAUGE
PLANE
-B-
SYMBOL
3
L
0.25
0.010
SEATING PLANE
-A-
A
D
h x 45°
-C-
α
e
A2
A1
B
C
0.10(0.004)
0.17(0.007) M
C A M
B S
NOTES:
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2
of Publication Number 95.
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
MIN
MAX
MILLIMETERS
MIN
MAX
NOTES
A
0.053
0.069
1.35
1.75
-
A1
0.004
0.010
0.10
0.25
-
A2
-
0.061
-
1.54
-
B
0.008
0.012
0.20
0.30
9
C
0.007
0.010
0.18
0.25
-
D
0.386
0.394
9.81
10.00
3
E
0.150
0.157
3.81
3.98
4
e
0.025 BSC
0.635 BSC
-
H
0.228
0.244
5.80
6.19
-
h
0.0099
0.0196
0.26
0.49
5
L
0.016
0.050
0.41
1.27
6
N
α
28
0°
28
8°
0°
7
8°
3. Dimension “D” does not include mold flash, protrusions or gate
burrs. Mold flash, protrusion and gate burrs shall not exceed
0.15mm (0.006 inch) per side.
Rev. 1 6/04
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch)
per side.
5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allowable dambar protrusion shall be 0.10mm (0.004 inch) total in excess of “B”
dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
28
FN9094.4
December 21, 2006