April 2010 - 140W Monolithic Switching Regulator Simplifies Constant-Current/Constant-Voltage Regulation

140W Monolithic Switching Regulator Simplifies
Constant-Current/Constant-Voltage Regulation
Eric Young
The LT3956 is a monolithic switching regulator that can
generate constant-current/constant-voltage outputs in buck,
boost or SEPIC topologies over a wide range of input and
output voltages. With input and output voltages of up to 80V,
a rugged internal 84V switch and high efficiency operation, the
LT3956 can easily produce high power in a small footprint.
The LT®3956 combines key amplifier and
comparator blocks with a high current/
high voltage switching regulator in a
tiny 5mm × 6mm package. See Figure 1
for an example of how little board space
is needed to produce a complete constant-current, constant-voltage boost
circuit ideal for LED driving, supercap
charging or other high power applications that require the added protection
of input or output current limiting.
WHAT MAKES THE LT3956 TICK?
The big mover in the LT3956 is an
84V-rated, 90mΩ low side N-MOSFET switch
with an internally programmed current
limit of 3.9A (typ). The switching regulator can be powered from a supply as
high as 80V because the N-MOSFET switch
driver, the PWMOUT pin driver, and most
internal loads are powered by an internal LDO linear regulator that converts
VIN to 7.15V, provided the VIN supply is
high enough. The switch duty cycle and
current is controlled by a current-mode
pulse-width modulator—an architecture
that provides fast transient response, fixed
switching frequency operation and an
easily stabilized feedback loop at variable
inputs and outputs. The switching frequency can be programmed from 100kHz
to 1MHz with an external resistor, which
allows designers to optimize component
16 | April 2010 : LT Journal of Analog Innovation
size and performance parameters, such
as min/max duty cycle and efficiency.
And at the heart of the LT3956 is a dual
input feedback transconductance (gm)
amplifier that combines a differential
constant current sense with a standard
low side voltage feedback. The handoff
between these two loops is seamless and
predictable. The feedback loop operating closest to its set point is auto-selected
to be the loop controlling the flow of
charge onto the compensation R-C network attached to the VC pin. The voltage
Figure 1. Complete high power, constant current,
constant voltage boost circuit
level at the VC pin in turn controls the
current and duty ratio of the switch. A
more thorough description of operation
can be found in the LT3956 data sheet.
C1
2.2µF
×2
R1
332k
R2
100k
INTVCC
VIN
0.33Ω
INTVCC
ISP
R3
1M
LT3956
CTRL
1k
Q2
Q3
20k
2k
D2
ISN
FB
100k
VMODE
PWM
SS
RT
VC
0.01µF
PGND
VREF
R6
57.6k
28.7k
375kHz
C2
2.2µF
×10
SW
EN/UVLO
R5
1M
750mA
D1
22µH
VIN
6V TO 60V
(80V TRANSIENT)
200Ω
6.8nF
Figure 2. This 50W boost LED driver provides wide
input range, PWM dimming and LED fault protection
and reporting.
M1
R4
16.2k
LED+
Q1
PWMOUT
GND INTVCC
INTVCC
4.7µF
1k
18 WHITE
LEDs
50W
(DERATED IF
VIN < 22V)
M1: VISHAY SILICONIX Si7113DN
D1: DIODES INC PDS5100
L1: COILTRONICS DR125-220
C1,C2: MURATA GRM42-2X7R225
Q1: ZETEX FMMT497
Q2,Q3: ZETEX FMMT589
D2: BAV116W
D3: DIODES INC B1100/B
D3
design features
The big mover in the LT3956 is an 84V-rated, 90mΩ low side N-MOSFET
switch with an internally programmed current limit of 3.9A (typ). The switching
regulator can be powered from a supply as high as 80V because the
N-MOSFET switch driver, the PWMOUT pin driver, and most internal loads are
powered by an internal LDO linear regulator that converts VIN to 7.15V.
100
0.8
VLED+
50V/DIV
CURRENT
EFFICIENCY (%)
0.6
EFFICIENCY
90
0.4
85
0.2
80
0
10
20
30
VIN (V)
40
50
OUTPUT CURRENT (A)
95
PWM
5V/DIV
60
SW
50V/DIV
ILED
2A/DIV
ILED
500mA/DIV
FB
5V/DIV
VIN = 24V
10µs/DIV
VIN = 24V
200ns/DIV
Figure 3. High 94% efficiency means less than 3W
dissipation in the converter shown in Figure 2.
Figure 4. Boost PWM dimming waveforms for 60V
of LEDs shows microsecond rise and fall times and
excellent constant current regulation even over
short intervals.
Figure 5. LED+ terminal of boost shorted to GND is
prevented from damaging switching components by
a novel circuit.
A RUGGED HIGH POWER BOOST LED
DRIVER
Analog Dimming
solution for the luminary—the LED cathode current can return on a common
GND. A scope photo of PWM dimming
waveform (Figure 4) shows sharp rise
and fall times, less than 200ns, and quick
stabilization of the current. Although
a low side N-MOSFET disconnect at the
cathode is the simpler and more obvious (and a bit faster) implementation
for this particular boost circuit using the
LT3956, the use of high side PWM disconnect is important to a boost protection strategy to be discussed below.
Figure 2 shows a 50W boost LED driver
that operates from a 24V input, showing off some of the unique capabilities of this product when applied as an
LED driver. This boost circuit tolerates
a wide input range—from 6V to 60V. At
the low end of this VIN range, the circuit
is prevented from operating too close to
switch current limit by scaling back the
programmed LED current as VIN declines—
set by the resistor divider (R5 and R6) on
the CTRL pin. Figure 3 shows efficiency
and LED current versus VIN. The high
efficiency (94%) means passive cooling of
the regulator is adequate for all but the
most extreme environmental conditions.
ANALOG AND PWM LED DIMMING
The LT3956 offers two high performance
dimming methods: analog dimming
via the CTRL pin and the ISP/ISN current
sense inputs, and PWM dimming through
the PWM input and PWMOUT output.
Analog dimming is achieved via the voltage at the CTRL pin. When the CTRL pin
is below 1.2V, it programs the current
sense threshold from zero to 250mV (typ)
with guaranteed accuracy of ±3.5% at
100mV. When CTRL is above 1.2V, the
current sense threshold is fixed at 250mV.
At CTRL = 100mV (typ), the current sense
threshold is set to zero. This built-in offset
is important to the feature if the CTRL pin
is driven by a resistor divider—a zero
programmed current can be reached with
a non-zero CTRL voltage. The CTRL pin
is high impedance so it can be driven
in a wide variety of configurations.
PWM Dimming
Pulse width modulation (PWM) of
LED current is the preferred technique
to achieve wide range dimming of the
light output. Figure 2 shows a level shift
transistor Q1 driving a high side disconnect P-MOSFET M1. This configuration
allows PWM dimming with a single wire
CONSIDERATIONS FOR PROTECTING
THE LED, THE DRIVER, AND THE
INPUT POWER SUPPLY
LED systems often require load fault
detection. Limiting the output voltage in
the case of an open LED string has always
been a basic requirement and is achieved
through a resistor divider (R3 and R4)
at the FB input. If the string opens, the
switching regulator regulates VFB to a
constant 1.25V (typ). In addition to the
gm amplifier that provides this constant
April 2010 : LT Journal of Analog Innovation | 17
The boost circuit in Figure 2 uses the
voltage feedback (FB) input in a unique
fashion—protecting the LED+ node from
a fault to GND while preserving all the
other desirable attributes of the LED driver.
A standard boost circuit has a direct
path from the supply to the output, and
therefore cannot survive a GND fault on
its output when the supply current is not
limited. There are a number of situations
where one might desire to protect the
switching regulator from a short to GND of
the LED anode—perhaps the luminary
is separated from the driver circuit by
a connector or by a long wire, and the
input supply is a high capacity battery.
The LT3956 has a feature to provide this
protection. The overvoltage FB (OVFB)
comparator is a second comparator on
the FB input with a setpoint higher than
the VFB regulation voltage. It causes
the PWMOUT pin to transition low and
switching to stop immediately when
the FB input exceeds 1.31V(typ).
18 | April 2010 : LT Journal of Analog Innovation
VIN
OVLO
53V RISING
51V FALLING
VIN
EN/UVLO
10k
R1
1M
INTVCC
4.7µF
LT3956
Q1
R2
143k
PWM
PWMOUT
GND
BAV116
PWM
332k
by the resistor divider from VIN) exceeds
6.5V (INTVCC minus a VBE). When PWM falls
below its threshold, PWMOUT goes low
as well. Hysteresis of ~2V is provided by
PWMOUT. Because of the high PWM threshold (0.85V minimum over temperature),
the blocking diode D1 can be added to
preserve the PWM dimming capability.
Q1: ZETEX FMMT593
Figure 6. VIN overvoltage circuit halts switching
and disconnects the load during high input voltage
transients.
The OVFB comparator can be used in
an output GND fault protection scheme
(patent pending) for the boost. The key
elements are the high side LED disconnect
P-MOSFET (M1) and its supporting driving
circuit responsive to the PWMOUT signal,
and the output GND fault sensing circuit
consisting of D2, Q2 and two resistors that
provide signal to the FB node. The circuit
works by sensing the current flowing
in D2 when the output is shorted, and
thereby triggering the OVFB comparator. In
response to the OVFB comparator, the high
side switch M1 is maintained in an offstate and the switching is stopped until the
fault condition is removed. Figure 5 shows
the current waveform in the M1 switch
during and output short circuit event.
Additional Considerations for
Protecting the LED
Some harsh operating environments produce transients on the input power supply
that can overdrive a boosted output, if
only for a short while, and potentially
damage the LEDs with excessive current.
To discontinue switching and disconnect the LEDs during such a transient, a
simple add-on circuit to the PWM input,
shown as a breakout in Figure 6, disconnects the LED string and idles the switcher
when VIN exceeds 50V. The circuit works
by sourcing current to the PWM input
of the LT3956 from the collector of Q1
when VIN is low enough, but cutting off
that current when the base of Q1 (set
The LT3956 provides solutions to thermal dissipation problems encountered
driving LEDs. With high power comes
the concern about reduced lifetime of
the LED due to continuous operation at
high temperatures. Increasing numbers
of LED module applications implement
thermal sensing for the LED, usually
employing an NTC resistor coupled to
the LED heat sink with thermal grease.
A simple circuit employing the CTRL and
VREF pins of the LT3956 and an NTC resistor sensing the LED temperature produces a thermal derating curve for the
LED current as shown in Figure 7.
A CONSTANT-CURRENT/VOLTAGE
REGULATOR SERVES A WIDE RANGE
OF APPLICATIONS
Driving LEDs makes excellent use of
the LT3956 features, but it isn’t the
only application that requires constant
Figure 7. CTRL and VREF pins provide thermal derating to enhance LED reliability.
ISP
+
CTRL
ISN
100k
NTC
RT1
MURATA NCP18WM104
–
VREF
16.9k
LT3956
V(ISP-ISN)
300
V(ISP-ISN) THRESHOLD (mV)
voltage regulation, the FB input also has
two fixed setpoint comparators associated
with it. The lower setpoint comparator
activates the VMODE open collector pulldown when FB exceeds 1.20V (typ). After
the disconnection of the LED and loss of
the current regulation signal, the output
rises until it reaches the constant voltage
regulation setpoint. During this voltage
ramp the VMODE pin asserts and holds,
indicating that the LED load is open. This
signal maintains its state when PWM goes
low and the regulator stops switching,
allowing for the likelihood that output
voltage may fall below the threshold
without an occasional refresh provided by
switching. The VMODE pin quickly updates
when PWM goes high. The VMODE signal
can also indicate that the regulation mode
is transitioning from constant current to
constant voltage, which is the appropriate
function for current limited constant voltage applications, such as battery chargers.
250
200
150
100
50
0
25
45
65
85
TEMPERATURE (°C)
105
125
design features
voltage at constant current. It can be
used for charging batteries and supercapacitors, or driving a current source
load such as a thermoelectric cooler,
just to name a few examples. It can be
used as a voltage regulator with current limited input or output, or a current regulator with a voltage clamp.
intermittently, but the available power
might be limited based on an overall
system budget. The output charging rate
of the circuit of Figure 8 is not based on
any timer, but rather on the output voltage
level as sensed by the CTRL pin. Below a
certain output voltage, 22V in this case, the
input current is limited so that the switching regulator is maintained within its
own current limit. At higher output voltages, the default internal current sensing
threshold of 250mV (typ) establishes that
the input current cannot exceed 1.2A, and
so the output current drops. At very low
output voltages less than 1.5V, the network
driving the SS pin of LT3956 reduces the
switching frequency and the current limit
to maintain good control of the charging
current. When the load is within 5% of its
target voltage, the VMODE pin toggles to
indicate the end of constant current mode
and entry into constant voltage regulation.
Pursuing this line of thinking, Figure 8
shows a SEPIC supercap charger that draws
power from a fixed 24V input, and has
an input current limit of 1.2A. The
SEPIC architecture is chosen for several
reasons: it can do both step-up and
step-down, and it has inherent isolation
of the input from the output. A coupled
inductor is chosen over a 2-inductor
approach because of the smaller, lowercost circuit. The magnetic coupling
effect allows the use of a single coupling
capacitor and the switch current levels of the LT3956 make strategic use of
the readily available coupled inductor
offerings of major magnetics vendors.
This circuit is intended for a situation
where VIN does not experience much
variation during normal operation. The
design procedure for this type circuit
begins with setting the maximum input
current limit with the RSENSE value and the
250mV default threshold. The next design
step is to determine the VOUT level below
A charging circuit for a large value capacitor (1F or more) might be found in a nonbattery based backup power system. These
chargers would draw power from some
inductive based DC supply that operates
which VIN current is to be reduced through
CTRL to maintain less than 2.5A average
switching current. Assuming slightly less
than 90% efficiency, set the resistor divider
R5 and R6 to give CTRL = 1.1V when
VOUT =
VIN • 0.9 • IIN(MAX )
2.5A − IIN(MAX )
, at CTRL = 1.1V
The values of R5 and R6 should be an
order of magnitude higher value than
resistor R7. The resistor divider R7 and R8
is set to provide a minimum voltage at
CTRL, greater than 125mV, which is needed
to set non-zero value for input current.
CONCLUSION
The LT3956 simplifies power conversion
applications needing both constant-current
and constant-voltage regulation, especially
if they are constrained by board area and/
or bill-of-materials length. Its features
are selected to minimize the number of
external analog blocks for these types of
applications while maintaining flexibility.
Careful integration of these components
into the switching regulator makes it possible to easily produce applications that
would otherwise require a cumbersome
combination of numerous externals. n
Figure 8. Supercapacitor charger with current limited input provides controlled charging current over a wide output range.
SW
VIN
PGND
EN/UVLO
FB
LT3956
1:1
C1
10µF
VMODE
CTRL
INTVCC
VREF
SS
VC
C2
4.7µF
RT
GND
28.7k
370kHz
10nF
L1B
C3
10µF
R5
1M
R6
40.2k INTVCC
10k
PWM
VOUT
0V TO 28V
R3
536k
R4
24.9k
PWMOUT
D1
R8
14k
1M
R1
59k
R7
2k
Q1
R2
30.1k
3
INPUT AND OUTPUT CURRENT (A)
ISN
ISP
1µF
C4
10µF
L1A
33µH
RSENSE
200mΩ
VIN
28V
≤ 1.2A
2.5
2
OUTPUT
1.5
1
INPUT
0.5
0
0
5
10
15
20
VOUT (V)
25
30
L1: WURTH ELEKTRONIK 744871330
D1: ON SEMICONDUCTOR MBRS360T
Q1: MMBTA42
C1,C3,C4: TAIYO YUDEN GMK316BJ106
April 2010 : LT Journal of Analog Innovation | 19
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