Two High Power Monolithic Switching Regulators Include Integrated 6A, 42V or 3.3A, 42V Power Switches, Built-in Fault Protection and Operation up to 2.5MHz Matthew Topp and Joshua Moore Power supply designers looking to shrink applications and simplify layout often turn to monolithic switching regulators. Monolithics simplify power supply layout by including the power switch on the die—no external FETs or precision sense resistors are needed. Monolithics can also operate at substantially higher switching frequencies than their controller-only counterparts, thus reducing the size and number of external passive components. The benefits of monolithic regulators are clear, but they traditionally have one major limitation: as the required power level goes up, the likelihood of finding a suitable monolithic regulator diminishes. Two new high power monolithics, the LT3579 and LT3581, solve this problem by integrating 6A (42V) and 3.3A (42V) power switches, respectively. The LT3579 and LT3581 are highly flexible parts and can be configured in boost, SEPIC, inverting, or flyback configurations. They also offer many unique performance and fault protection features. When configured as high power boost converters, these parts can survive output overloads with only a few additional external components. They can also be configured to provide hot-plug and reverse input voltage protection. In addition, a novel master and slave power switch design allows high voltage charge pump circuits to be made with low power dissipation and few components. Both parts can be programmed to free-run from 200kHz to 2.5Mhz or can be synchronized with an outside clock source. The parts also provide a clock output pin, enabling the ICs to synchronize other switching regulators. The LT3579 comes in a 4mm × 5mm QFN and 20-lead TSSOP package, and the LT3581 comes in a 4mm × 3mm DFN and 16-lead MSE package. L1 2.2µH VIN 5V D1 VOUT 12V 1.7A M1 COUT1 10µF 100k 200k CIN 22µF VIN SW1 SW2 FAULT LT3579 130k FB CLKOUT RT SYNC VC 86.6k GND D2 VIN GATE SHDN COUT 10µF 6.34k SS 47pF CIN: 22µF, 16V, X7R, 1210 COUT1, COUT: 10µF, 25V, X7R, 1210 D1: VISHAY SSB43L D2: CENTRAL SEMI CMDSH-3TR L1: WÜRTH WE-PD 744771002 M1: SILICONIX SI7123DN 8.06k 0.1µF 2.2nF Figure 1. This 5V to 12V boost converter can survive the infamous metal file test where a wire attached to the output is dragged across the jagged surface of a grounded metal file VOUT 10V/DIV TO ONE OR MORE LT3579/LT3581 TO FAULT PIN OF ONE OR MORE LT3579/LT3581 TO ONE OR MORE LT3579/LT3581 VIN CLKOUT 2V/DIV RGATE IL 5A/DIV GATE FAULT LT3579/LT3581 FAULT 5V/DIV VIN 10µs/DIV Figure 2. Operating waveforms for Figure 1 circuit during brutal metal file test 16 | January 2011 : LT Journal of Analog Innovation Figure 3. Input disconnect schematic design features L1 4.7µH VIN 5V D1 M1 VOUT 12V 0.5A (VIN = 3V) 0.9A (VIN = 5V) C1 10µF 100k 51.1k VIN SW1 SW2 FAULT CIN2 4.7µF CIN1 10µF 130k 6.34k FB D3 VIN GATE LT3581 SHDN CLKOUT RT SYNC VC GND COUT1 10µF SS 86.6k 16.9k 0.22µF 2.2nF 47pF Figure 4. A 3V–5V input to ±12V output converter C2 4.7µF L2 3.3µH L3 3.3µH VOUT –12V 0.5A (VIN = 3V) 0.9A (VIN = 5V) D2 VIN CIN1, CIN3, CIN4: 10µF, 10V, X5R, 1206 CIN2 : 4.7µF, 10V, X5R, 0805 C1, COUT1, COUT2: 10µF, 25V, X7R, 1210 C2: 4.7µF, 25V, X7R, 1206 D1, D2: DIODES INC SBR2A40P1 D3: CENTRAL SEMI CTLSH1-40M563 CIN3 L1: COILCRAFT XPL7030-472ML 10µF L2, L3: COOPER DRQ125-3R3 M1: SILICONIX SI7123DN FAULT PROTECTION FEATURE: CURRENT OVERLOADS Most high power boost converters cannot survive an output overload condition because of the inherent DC pathway that exists from the input to output through the inductor and rectifying diode. An output overload or short causes the current in this pathway to increase and run away, thus damaging anything in this pathway or connected to it. The 12VOUT 500mV/DIV AC COUPLED CIN4 10µF 86.6k LT3579 and LT3581 include features that protect against such fault events. Figure 1 shows an LT3579 configured as a 5V input to 12V output boost converter with output short protection. An external PFET, diode, and resistor are all it takes to implement robust output short protection. In fact, this circuit can survive the infamous “file test,” where a wire tied to the output is swiped across the surface of a metal woodworking file tied to ground. Figure 2 shows the IL1 2A/DIV –12VOUT 500mV/DIV AC COUPLED IL2 + IL3 2A/DIV IL2 + IL3 2A/DIV 100µs/DIV Figure 5. Load step from 0.25A to 0.75A between +12V and −12V rail, with 5V input 143k FB FAULT GATE LT3579-1 FAULT IL1 2A/DIV SW1 SW2 10µs/DIV Figure 6. Transient short between rails with 5V input, 0.9A load before short SHDN CLKOUT RT SYNC VC GND SS COUT2 10µF ×2 14.3k 0.22µF 2.2nF 47pF operating waveforms during this normally destructive test—the LT3579 survives this brutal test without any problems. These parts also protect against several other types of fault conditions, including overcurrent conditions, overvoltage on VIN, and over-temperature inside the part. In systems where multiple LT3579s/LT3581s are incorporated to produce multiple rails, a single PFET and resistor can be used on the input side to protect all the rails from a current overload. Figure 3 shows how to set this up. Simply tie the FAULT pins of all ICs together and connect to a single pull-up resistor. The fault control scheme is designed so that if one part goes into fault, it pulls its FAULT pin low, causing the other parts to go into fault as well. Switching activity in all parts stops and all enter into a time-out period. This time-out period allows the components in the system to cool down. Only after the last part exits the time-out period do all parts attempt to restart. To January 2011 : LT Journal of Analog Innovation | 17 The LT3579 and LT3581 include features that protect against a number of fault events including output overloads or shorts, overcurrent conditions, overvoltage on VIN, and over-temperature inside the part. C2 2.2µF L1 3.3µH M2 6.34k • SW1 GATE VIN C1 3.3µF 100k GATE LT3579/LT3581 CIN: 0.1µF, 25V, X7R, 0805 C1: 3.3µF, 25V, X7R, 1206 C2: 2.2µF, 50V, X7R, 1206 C3: 10µF, 25V, X7R, 1210 D1: CENTRAL SEMI CTLSH2-40M832 L1, L2: COILCRAFT MSD7342-332MLB M1, M2: SILICONIX SI7123DN VIN Figure 7. Recommended connections for hot plug, reverse input voltage, and input overvoltage events isolate a fault to only one part, simply do not connect the FAULT pins together. The LT3579 and LT3581 can be easily mixed within a system while maintaining all overload and protection features. Figure 4 shows the LT3579-1 configured as an inverting converter working together with an LT3581 configured as a boost converter. Together, these converters generate a VIN 10V/DIV 43.2k L2 3.3µH FB 130k SHDN FAULT C3 10µF CLKOUT RT VC SYNC SS GND 100pF 0.1µF 10k 2.2nF Figure 8. A 9V–16V input to 12V output SEPIC with hot plug, reverse input voltage, and input overvoltage protection regulated ±12V output at up to 0.9A running off a 3V–5V input, with overload and over-temperature protection. The LT3579-1 is used because it features low input ripple (see page 19 for more about this feature of the LT3579-1). Figure 5 shows the load step response. This system not only accommodates output shorts and overloads between each rail to ground, but it can also tolerate these conditions between the rails as shown in Figure 6. FAULT PROTECTION FEATURES: HOT PLUG, REVERSE INPUT VOLTAGE, AND INPUT OVERVOLTAGE The GATE pin, SS pin and related circuitry can also be used to protect against hot plug, reverse input voltage, and input overvoltage events. Figure 7 shows one way to set this up. Hot plug protection is VIN 10V/DIV GND VIN 10V/DIV IL1 + IL2 2A/DIV VOUT 12V 1A (VIN = 9V) 1.1A (VIN = 12V) 1.3A (VIN = 16V) SW2 LT3581 VIN CIN 0.1µF D1 • VIN 9V TO 16V M1 IL1 + IL2 2A/DIV VGATE 20V/DIV SS 1V/DIV VOUT 20V/DIV SS 1V/DIV VOUT 10V/DIV IL1 + IL2 2A/DIV VOUT 10V/DIV 200ms/DIV Figure 9. Operating waveforms for a hot plug event 18 | January 2011 : LT Journal of Analog Innovation 100ms/DIV Figure 10. Operating waveforms for a negative VIN transient 500ms/DIV Figure 11. Operating waveforms for a VIN overvoltage transient design features Packed with the latest features and some of the highest power levels of any monolithic converters in the industry, the LT3579 and LT3581 venture into applications once reserved for controllers with external FETs. C2 1µF L1 3.3µH L2 3.3µH VOUT –12V 625mA • • VIN 5V D1 CLKOUT 2V/DIV SW1 SW2 VIN C1 3.3µF 100k 43.2k LT3581 FB SHDN GATE FAULT CLKOUT RT VC SYNC SS 143k SW 10V/DIV C3 4.7µF 47pF GND 0.1µF 11k 1nF VOUT 50mV/DIV AC COUPLED C1: 3.3µF, 16V, X7R, 1206 C2: 1µF, 25V, X7R, 1206 C3: 4.7µF, 25V, X7R, 1206 D1: DIODES INC. PD3S230H-7 L1, L2: COILCRAFT MSD7342-332MLB IL1 + IL2 2A/DIV 200ns/DIV Figure 13. High operating frequency results in low output ripple, even at maximum load Figure 12. A 5V input to −12V output inverting DC/DC converter useful for limiting the surge current when the input to the power supply is suddenly stepped from low voltage to normal. In a boost converter, there is a DC path from the input to the output capacitors of the circuit. Since these capacitors are initially discharged, large surge currents are possible if this feature is not used. Figure 8 shows a circuit designed to handle all these potentially dangerous conditions. Figure 9 shows the operating waveforms during a hot plug event, Figure 10 shows the waveforms during a negative VIN transient, and Figure 11 shows the result of a VIN overvoltage transient. The LT3579/LT3581 survives all these fault conditions and when the fault is removed, resumes a normal start-up cycle. parts can synchronize to an external clock. The CLKOUT pin on the parts is designed to drive the SYNC pins of other switching regulators. The LT3579 and LT3581 also encode die temperature information into the duty cycle of the CLKOUT signal, making thermal measurements simple. small. The amount of output ripple is also very low, as shown in Figure 13. Figure 14 shows a 2.8V to 4.2V input to 5V output boost running at 2MHz using the LT3579. This circuit is configured to survive output overloads and can deliver up to 2A of output current. Figure 12 shows a 2MHz, 5V input to −12V output inverting converter with 625m A of output current capability using the LT3581. Due to the high switching frequency, external components are USE THE LT3579-1 FOR EVEN MORE POWER AND SPEED The LT3579-1 is nearly identical to the LT3579 with one exception: the CLKOUT pin has a 50% duty cycle that does not vary Figure 14. Li-ion battery to 5V output boost running at 2MHz can deliver 2A of output current. L1 0.47µH VIN 2.8V TO 4.2V VIN SW1 SW2 SHDN 100k CIN: 10µF, 16V, X7R, 1206 COUT1, COUT: 22µF, 16V, X7R, 1210 D1: CENTRAL SEMI CTLSH3-30M833 L1: VISHAY IHLP-2020BZ-01-R47 M1: SILICONIX SI7123DN VOUT 5V 2A M1 COUT1 22µF HIGH POWER AND HIGH SPEED The combination of high current capability and high switching frequency make the LT3579/LT3581 useful in a wide range of applications. Not only can the parts be set for an internal oscillator frequency between 200kHz and 2.5MHz, but the D1 45.3k GATE LT3579 COUT 22µF CLKOUT FAULT VC RT SYNC GND SS CIN 10µF 43.2k 10k FB 47pF 22nF 6.34k 2.2nF January 2011 : LT Journal of Analog Innovation | 19 L2 4.7µH Figure 15. Dual phase 8V–16V input to 24V boost converter uses two LT3579-1s and can deliver up to 5.1A of output current D2 CPWR1, CPWR2: 10µF, 25V, X7R, 1210 CVIN1, CVIN2: 4.7µF, 25V, X7R, 1206 COUT1M, COUT1S, COUT: 4.7µF, 50V, X5R, 1210 D1, D2: CENTRAL SEMI CTLSH5-40M833 D3: CENTRAL SEMI CTLSH1-40M563 L1, L2: VISHAY IHLP-2525CZ-01-4R7 M1: SILICONIX SI7461DP COUT1S 4.7µF ×2 CPWR2 10µF SW1 SW2 CLKOUT LT3579-1 FB SLAVE FAULT VIN CVIN2 4.7µF SHDN RT SYNC GND GATE VC SS 0.22µF 86.6k with die temperature and is 180° out of phase with its own internal clock whether the part free runs or is synchronized. This difference allows for the construction of a dual phase converter in the boost, SEPIC, or inverting configurations. VPWR 8V TO 16V CPWR1 10µF VIN 3.3V TO VPWR MASTER AND SLAVE SWITCHES Both the LT3579 and LT3581 have a novel master/slave switch configuration. To implement current mode control, the MASTER CLKOUT 2V/DIV D1 VIN VOUT1 86.6k SW1 SW2 CLKOUT LT3579-1 MASTER FB CVIN1 4.7µF SYNC GND 4.99k 86.6k D3** VPWR GATE VC 47pF SS 0.22µF VPWR = 8V VPWR = 12V VPWR = 16V 2.4A 3.7A 5.1A VIN = VPWR 2.2A 3.1A 3.9A **OPTIONAL FOR OUTPUT SHORT CIRCUIT PROTECTION master switch (SW1 pin) has a current comparator to monitor the current. The slave switch (SW2 pin) has no current comparator and simply operates in phase with the master. For most applications, simply tie SW1 and SW2 pins together to get a 6A or 3.3A total current limit for the LT3579 and LT3581, respectively. Since it may be desirable in some situations to have a lower current limit with an easy way to upgrade to a higher current in the future, these parts can operate using only the master switch. To do this, simply float the slave switch pins. As a result, the LT3579 becomes a 3.4A part and the LT3581 becomes a 1.9A part. Figure 16. Output ripple at maximum load for the dual phase circuit shown in Figure 15 20 | January 2011 : LT Journal of Analog Innovation EFFICIENCY (%) 100µs/DIV Figure 17. Transient load response for the dual phase circuit shown in Figure 15 100 8 90 7 80 6 VIN = 12V 70 60 5 4 VIN = 3.3V 50 3 40 2 30 1 20 0 0.5 1 1.5 2 2.5 3 3.5 LOAD CURRENT (A) 4 POWER LOSS (W) ILOAD 1A/DIV 500ns/DIV 2.2nF VIN = 3.3V TO 5V IL1 + IL2 5A/DIV IL1 + IL2 5A/DIV 6.98k *MAX OUTPUT CURRENT VOUT 1V/DIV AC COUPLED VOUT AC COUPLED 100mV/DIV COUT 4.7µF ×2 137k FAULT SHDN RT VOUT 24V 5.1A* 6.34k** COUT1M 4.7µF ×2 21.5k 499k A major benefit of out-of-phase operation is an inherent reduction in input and output ripple. Figure 15 shows an 8V–16V input to 24V output dual-phase boost converter capable of delivering up to 5.1A of output current. Each part operates at 1MHz, but because the outputs operate out of phase the effective switching frequency of the converter is 2MHz. Figure 16 shows the output ripple at maximum load, Figure 17 shows the transient load response, and Figure 18 shows the efficiency. This circuit features output short circuit protection, which is easily removed if not needed. M1** VOUT1 L1 4.7µH 0 4.5 Figure 18. Converter efficiency reaches 93% for the dual phase circuit shown in Figure 15 design features The master/slave architecture provides a clear advantage when creating high voltage charge pump circuits. It is common practice to create high voltage rails by building a boost converter and adding charge pump stages to double or even triple the boost converter’s output voltage. At higher power levels, it becomes necessary to dampen the current spikes inherent in these charge pump circuits. Figure 19 shows a traditional approach, which uses high power resistors within the charge pumps. Without these resistors, the current spikes would cause the switching regulator to false trip, causing erratic and unstable operation. The problem is that these resistors add to the component count and generate additional heat. Figure 20 shows a better solution in which the master/slave switch configuration eliminates the need for the high power resistors. All current spikes caused by the charge pump stages are only seen by the slave switch, eliminating the possibility of false tripping. controllers, resulting in solution sizes unachievable by controller solutions. Advanced fault protection features make it possible to produce compact and rugged solutions without additional ICs. switching regulators. Both parts feature a wide input operating voltage range. The LT3579 can operate from 2.5V to 16V and survive transients to 40V. The LT3581 can operate from 2.5V to 22V with transients to 40V. Both parts have built-in programmable soft-start and automatic frequency foldback. Single pin feedback enables both positive and negative output voltages. Each part has an accurate comparator/reference for the SHDN pin, allowing the pin to be used as a programmable undervoltage lockout. A new master and slave switch architecture not only allows adjustment of the current limit but also significantly eases the design of high voltage boost plus charge pump circuits. These new features are simple to implement, yet stay out the way if not required. The LT3579, with a 6A, 42V switch comes in a 4mm × 5mm QFN or 20-lead TSSOP package. The LT3581, with a 3.3A, 42V switch comes in a 3mm × 4mm DFN or 16-lead MSE package. n CONCLUSION Packed with the latest features and some of the highest power levels of any monolithic converter in the industry, the LT3579 and LT3581 venture into applications once reserved for controllers. Monolithic converters can operate at clock speeds far beyond the ability of D6 C4 2.2µF ×2 BEST IN CLASS SPECIFICATIONS C3 2.2µF ×2 D2 L1 10µH VIN 9V TO 16V D1 M1** C1 2.2µF ×3 D9** VOUT2 100k VOUT1 536k CIN 10µF VIN SW1 SW2 FAULT SHDN RT SYNC LT3579 383k DC/DC GND Figure 19. Traditional method for building high power boost plus charge pump circuits D8** 8.2V 6.49k** FB D7** C2 2.2µF ×3 VIN GATE CLKOUT VC GND SS 86.6k SW VOUT1 67V C5 500mA* 2.2µF ×2 D4 D3 With so many new features, it is easy to overlook that the LT3579 and LT3581 include all the standard features available in many modern Linear Technology VIN C6 2.2µF ×2 D5 VOUT2 100V 330mA* CIN: 10µF, 25V, X7R, 1210 C1-C6: 2.2µF, 50V, X7R, 1210 D1-D6: DIODES INC SBR2A40P1 D7: CENTRAL SEMI CMDSH-3TR D8: CENTRAL SEMI CMDZ5237B-LTZ D9: DIODES INC MBRM360 L1: WÜRTH WE-PD 7447710 M1: SILICONIX SI7461DP 27pF 2.2µF 34k 470pF *MAX TOTAL OUTPUT POWER 22W (VIN = 9V) 27W (VIN = 12V) 33W (VIN = 16V) **OPTIONAL FOR OUTPUT SHORT CIRCUIT PROTECTION Figure 20. Master and slave switches of the LT3579/LT3581 allow a cooler running, simpler method for building boost plus charge pump circuits. January 2011 : LT Journal of Analog Innovation | 21