V07N4 - NOVEMBER

LINEAR TECHNOLOGY
NOVEMBER 1997
IN THIS ISSUE…
COVER ARTICLE
New 16-Bit SO-8 DAC
Has 1LSB Max INL and DNL
Over Industrial Temperature ......... 1
Jim Brubaker
and William C. Rempfer
Issue Highlight .............................. 2
LTC® in the News ........................... 2
DESIGN FEATURES
The LT®1374: New 500kHz, 4.5A
Monolithic Buck Converter ............ 5
Karl Edwards
The LT1581 Low Dropout Regulator
Provides 10 Amps of Output Current
..................................................... 8
Todd Owen
The LT1370: New 500kHz, 6A
Monolithic Boost Converter ......... 11
Karl Edwards
New LTC1504: Flexible, Efficient
Synchronous Switching Regulator
Can Source or Sink 500mA ......... 14
Dave Dwelley
Low Dropout Regulator Driver
Handles Fast Load Transients and
Operates on Single 3V–10V Input
................................................... 16
VOLUME VII NUMBER 4
New 16-Bit SO-8 DAC
Has 1LSB Max INL and DNL
Over Industrial Temperature
by Jim Brubaker
and William C. Rempfer
New generations of industrial
systems are moving to 16 bits and
hence require high performance 16-bit
data converters. The new LTC1595/
LTC1596 16-bit DACs from LTC
provide the easiest to use, most cost
effective, highest performance solution
for industrial and instrumentation
applications. The LTC1595/LTC1596
are serial input, 16-bit, multiplying
current output DACs. Features of the
new DACs include:
(complete list on page 25)
❏ ±1LSB maximum INL and DNL
over the industrial temperature
range
❏ Ultralow, 1nV-s glitch impulse
❏ ±10V output capability
❏ Small SO-8 package (LTC1595)
❏ Pin-compatible upgrade for
industry-standard 12-bit DACs
(DAC8043/8143 and AD7543)
DESIGN INFORMATION
Reference Squeezes More Performance
from Less Package ...................... 33
Nice Features
of the 16-Bit DACs
Lenny Hsiu
LT1579 Battery-Backup Regulator
Provides Uninterruptible Power ... 18
Todd Owen
High Efficiency Distributed
Power Converter Features
Synchronous Rectification .......... 21
Dale Eagar
DESIGN IDEAS
............................................. 25–32
John Wright
LTC1659, LTC1448: Smallest Rail-toRail 12-Bit DACs Have Lowest Power
................................................... 34
Hassan Malik
LTC1197/LTC1199: New Micropower
MSOP 10-Bit ADC Samples at 500ksps
................................................... 35
Guy Hoover and Marco Pan
New Device Cameos ..................... 37
Design Tools ................................ 39
Sales Offices ............................... 40
The LTC1595/LTC1596 use precision
thin-film resistors in a modified R/2R
architecture to provide a CMOS current output DAC, as shown in Figure
1. The two DACs have SPI/MICROWIRE™ compatible serial interfaces
and draw only 10µA from a single 5V
supply. They generate precision 0V–
10V or ±10V outputs using a single or
dual external op amp. The LTC1596
has an asynchronous clear input and
both devices have power-on reset.
16-Bit Accuracy Over
Temperature without
Autocalibration
In the past, only autocalibrated DACs
could achieve 16-bit (1LSB) accuracy
over temperature. Not only did this
require cumbersome calibration overhead for the user but the DACs also
had to be recalibrated every time the
temperature changed because of the
DAC’s poor linearity drift. In addition,
the highly complicated autocalibration circuitry made the DACs very
large and very expensive.
Now there is a better choice. The
LTC1595/LTC1596’s ultralow linearity drift (well below ±0.2LSB from
–40°C to 85°C) allows the DACs to be
factory trimmed and to hold their
16-bit accuracy over time and temperature. This provides a cleaner,
easier, more cost-effective solution
for precision systems.
Figure 2 shows the typical integral
nonlinearity (INL) and differential
nonlinearity (DNL) of the LTC1595.
The outstanding 0.25LSB typical values and the very low drift allow a
maximum 1LSB specification to be
guaranteed over the extended industrial temperature range.
continued on page 3
, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,
FilterCAD, LinearView, Micropower SwitcherCAD and SwitcherCAD are trademarks of Linear Technology Corporation.
MICROWIRE is a trademark of National Semiconductor Corp. Other product names may be trademarks of the companies
that manufacture the products.
EDITOR’S PAGE
Issue Highlights
The subject of our cover article this
issue is the LTC1595/LTC1596 serial
input, 16-bit, multiplying current
output DACs. These parts feature
±1LSB maximum INL and DNL over
the industrial temperature range,
ultralow, 1nV-s glitch impulse and
±10V output capability. They are pincompatible with industry-standard
12-bit DACs.
The remainder of this issue’s Design
Features section introduces a variety
of new power products:
LT1374 is a new 4.5A buck converter designed to meet the needs of
higher current and voltage applications. With its 500kHz operating
frequency and integral 80mΩ switch,
only a few surface mount components are required to produce a
complete switching regulator. The
LT1374 features current mode control, external synchronization and a
low current (20µA typical) shutdown
mode.
Another new 500kHz switcher
introduced in this issue is the LT1370,
a 6A boost converter. At 65mΩ
on-resistance and 42V maximum
switch voltage, the LT1370 can be
used in a wide range of output voltage
and current applications. LT1370 features include current mode operation,
external synchronization and a low
current shutdown mode (12µA typical). The LT1370 will operate in all
standard switching configurations,
including boost, buck, flyback, forward, inverting and SEPIC.
For even higher currents, we introduce the LT1339, the buck/boost
converter that “needs no steroids.”
The LT1339 is ideal for power levels
ranging from tens of watts to tens of
kilowatts. It has an innovative slopecompensation function that allows
the circuit designer freedom in controlling both the slope and offset of
the slope-compensation ramp. Additionally, the LT1339 has an average
current limit loop that yields a constant output current limit, regardless
of input and/or output voltage.
2
Yet another new switcher is the
LTC1504 8-pin step-down switching
regulator. It consists of a 200kHz
fixed frequency, voltage-feedback,
buck-mode switching regulator controller and a pair of 1.5Ω power
switches in an 8-pin SO package. The
LTC1504 improves functionality by
integrating a synchronous rectifier
on-chip, maximizing efficiency and
minimizing external parts count while
allowing the output to both sink and
source current: it can source or sink
up to 500mA with input voltages from
3.3V to 10V and output voltages as
low as 1.26V.
A new low dropout NPN regulator,
the LT1581, can provide up to 10
amps of output current. By using
separate supplies for the control circuitry and the drive to the NPN output
transistor, the dropout on the pass
transistor is reduced to the saturation voltage of the transistor. This
makes it is easy to obtain dropout
voltages as low as 430mV at 10A of
load current. The part is fully protected
from overcurrent and overtemperature conditions.
Another low dropout regulator
debuting this month is the LT1579
“smart regulator.” The LT1579 is a
dual input, single output, low dropout regulator that provides an
uninterruptible output voltage from
two independent input voltage sources
on a priority basis. It is capable of
providing 300mA from either input at
a dropout voltage of 0.4V.
The LT1573 is the latest low dropout voltage regulator driver from
Linear Technology. The LT1573 is
designed to provide a low cost solution to applications requiring high
current, low dropout and fast
transient response. The LT1573 is
available as an adjustable regulator
with an output range of 1.27V to
6.8V, and with fixed output voltages
of 2.5V, 2.8V and 3.3V. Output accuracy is better than 1% to meet the
critical regulation requirements of fast
microprocessors.
LTC in the News…
LTC Achieves Record
Quarterly Bookings,
Sales and Profits
“This was a strong summer quarter
for us, as we achieved record quarterly
bookings, sales and profits,” explains
Robert Swanson, president and CEO,
based on Linear Technology’s latest
quarterly sales and earnings report.
“We grew cash by roughly $63 million
and now have over $500 million in
cash and short-term investments.
Demand was strong for our products,
particularly in the communications
and computer end markets. This
demand should help us continue to
grow in the upcoming quarter.”
These comments are based on
results announced on October 14,
1997. LTC had net sales for its first
quarter ended September 28 of
$109,802,000—an increase of 22%
over net sales of the $90,063,000 for
the first quarter of the previous year.
The Company also reported record net
income for the quarter of $40,643,000
or $0.51 per share, an increase of 30%
over last year of $31,358,000 or $0.40
per share, reported for the first quarter of last year. A cash dividend of
$0.06 will be paid on November 12,
1997 to shareholders of record on
October 24, 1997.
The Design Ideas section of this
issue features two battery-related
applications: a single-cell Li-Ion
battery supervisor based on the
LT1496 op amp and the LT1634 precision reference, a low profile 800mA
Li-Ion charger using the LT1510-5
switch-mode battery charger. Also
included are a high voltage/high current bench supply, a switcher that
generates two bias voltages without a
transformer, a single-supply random
code generator and a low distortion,
low power HDSL driver.
In addition, we present design
information on the LT1460 voltage
reference in the SOT-23 package, the
LTC1448 and LTC1659 12-bit rail-torail DACs and the LTC1197/LTC1199
10-bit ADCs.
We conclude with a quintet of new
device cameos.
Linear Technology Magazine • November 1997
DESIGN FEATURES
56k
VREF
56k
2 RFB
1
56k
56k
56k
56k
56k
56k
56k
112k
112k
112k
112k
7k
3 OUT1
VDD 8
4 GND
DECODER
D15
(MSB)
LOAD
LD 5
CLK 7
CLK
D14
D13
D12
D11
•••
DAC REGISTER
D0
(LSB)
INPUT 16-BIT SHIFT REGISTER
IN
6 SRI
1595_01.EPS
Figure 1. The LTC1595/LTC1596 16-bit CMOS DACs use a precision thin-film modified R/2R architecture to provide unsurpassed accuracy and
stability. They provide a pin-compatible upgrade for industry-standard 12-bit CMOS DACs (DAC8043/8143 and AD7543).
LTC1595, continued from page 1
Ultralow 1nV-s Glitch
0.5
INTEGRAL NONLINEARITY (LSB)
0.4
0.3
0.2
0.1
LTC1595/LTC1596’s new proprietary
deglitcher reduces the output glitch
impulse to 1nV-s, which is ten times
lower than any other 16-bit industrial DAC. (1nV-s is equivalent to a
glitch of 7LSBs and 1µs duration.) In
addition, the deglitcher makes the
glitch impulse uniform for any code.
Figure 3 shows the output glitch for a
midscale transition with a 0V to 10V
output range.
0
Tiny Footprint: SO-8
–0.1
–0.2
–0.3
–0.4
– 0.5
0
49152
16384
32768
DIGITAL INPUT CODE
65535
1595_2a.eps
DIFFERENTIAL NONLINEARITY (LSB)
0.5
0.4
0.3
0.2
Previous 16-bit DACs have had large
packages that required large board
space. The new LTC1595 breaks this
trend with an SO-8 pinout. With the
addition of a single output op amp
(MSOP or SO-8) and a voltage reference, a complete 16-bit voltage output
DAC can be constructed in a fraction
of the space of previous products.
Pin-for-Pin Upgrade to
Industry Standard 12-Bit DACs
0.1
0
–0.1
–0.2
–0.3
–0.4
– 0.5
0
16384
32768
49152
DIGITAL INPUT CODE
65535
Figure 2. The outstanding INL and DNL
(typically less than 0.25LSB) and very low
linearity drift allow a maximum 1LSB spec to
be guaranteed over industrial temperature.
Linear Technology Magazine • November 1997
An especially helpful feature for some
users is the ease of upgrade to these
new 16-bit devices. They directly
replace the popular 12-bit DAC8043/
8143 and AD7543. Because sensitivity of the LTC1595/LTC1596 INL to
op amp offset voltage is reduced by
five times compared to the 12-bit
devices, most systems can be easily
upgraded to true 16-bit resolution
and linearity without requiring more
precise op amps.
0V–10V and ±10V
Output Capability
Precision 0V–10V Outputs
with One Op Amp
Figure 4 shows the circuit for a 0V–
10V output range. The DAC uses an
external reference and a single op
amp in this configuration. This circuit can also perform 2-quadrant
multiplication where the reference
input is driven by a ±10V input signal
+10
OUTPUT VOLTAGE (mV)
Glitches in a DAC’s output when it
updates can be a big problem in precision applications. Usually, the
worst-case glitch occurs when the
DAC output crosses midscale. The
COMPETITOR'S DAC
0
1nV-s TYP LTC1595/LTC1596
–10
0
1
2
TIME (µs)
3
4
1595 03 .eps
Figure 3. The proprietary deglitcher reduces
the output glitch to less than 1nV-s, which is
five times less than any other 16-bit
industrial DAC. Further, the deglitcher makes
the glitch impulse uniform, independent of
code.
3
DESIGN FEATURES
and VOUT swings from 0V to –VREF.
The full-scale accuracy of the circuit
is very precise because it is determined by precision-trimmed internal
resistors. The power dissipation of
the circuit is set by the op amp dissipation and the current drawn from
the DAC reference input (7k nominal). The supply current of the DAC
itself is less than 10µA.
An advantage of the LTC1595/
LTC1596 is the ability to choose the
output op amp to optimize the accuracy, speed, power and cost of the
application. Using an LT1001 provides excellent DC precision, low noise
and low power dissipation (90mW
total for Figure 4’s circuit). For higher
speed, an LT1007 or LT1122 can be
used. The LT1122 will provide settling to 1LSB in 3µs for a full-scale
transition. Figure 5 shows the 3µs
settling performance obtained with
the LT1122. The feedback capacitor
in Figure 4 ensures stability. In higher
speed applications, it can be used to
optimize transient response. In slower
applications, the capacitor can be
increased to reduce glitch energy and
provide filtering.
Precision ±10V Outputs
with a Dual Op Amp
Figure 6 shows a bipolar, 4-quadrant
multiplying application. The reference input can vary from –10V to 10V
and VOUT swings from –VREF to +VREF.
If a fixed 10V reference is used, a
precision ±10V bipolar output will
result.
Unlike the unipolar circuit of Figure 4, the bipolar gain and offset will
depend on the matching of the external resistors. A good way to provide
good matching and save board space
is to use a pack of matched 20k
resistors (the 10k unit is formed by
placing two 20k resistors in parallel).
The LT1112 dual op amp is an
excellent choice for high precision,
low power applications that do not
require high speed. The LT1124 will
provide faster settling. Again, with op
amp selection the user can optimize
the speed, power, accuracy and cost
of the application.
Conclusion: Time to Upgrade
The new LTC1595/LTC1596 provide
an excellent opportunity for users to
upgrade lower resolution systems to
16 bits, cleanly, easily and cost effectively. They give outstanding accuracy
over temperature, ultralow glitch impulse, small footprint and low cost.
For these reasons, they will improve
the performance of industrial and
instrumentation systems.
VIN
5V
CLOCK
DATA
LOAD
7
6
5
8
1
VDD VREF
2
RFB
33pF
CLK
SRI
OUT1
LTC1595
VOUT
5V/DIV
–
3
LD
+
GND
4
LT1001
VOUT
GATED VOUT
500µV/DIV
1595_04.EPS
1µs/DIV
Figure 4. With a single external op amp, the DAC performs
2-quadrant multiplication with ±10V input and 0V to –VREF
output. With a fixed –10V reference, it provides a precision 0V–
10V unipolar output.
7
µP
6
5
8
1
VDD VREF
2
RFB
33pF
CLK
SRI
R3
20k
R2
20k
VREF
–10V TO 10V
5V
0.1µF
Figure 5. When used with an LT1122 (in the circuit of
Figure 4), the LTC1595/LTC1596 can settle in 3µs to a
full-scale step. The top trace shows the output swinging
from 0V to 10V. The bottom trace shows the gated
settling waveform settling to 1LSB (1/3 of a division)
in 3µs.
LTC1595
LD
OUT1
3
–
1/2 LT1112
GND
4
+
R1
10k
–
1/2 LT1112
+
VOUT
(–VREF TO VREF)
1595_05.EPS
Figure 6. With a dual op amp, the DAC performs 4-quadrant multiplication.
With a fixed 10V reference, it provides a ±10V bipolar output.
4
Linear Technology Magazine • November 1997
DESIGN FEATURES
The LT1374: New 500kHz, 4.5A
Monolithic Buck Converter
by Karl Edwards
Circuit Description
Introduction
Expanding on the current range of
500kHz switchers, Linear Technology introduces a new 4.5A buck
converter, the LT1374. The LT1374 is
the big brother of the 1.5A LT1376.
Designed to meet the needs of higher
current and voltage applications, it
maintains high efficiency using an
on-chip 80mΩ switch. With its 500kHz
operating frequency and integral
switch, only a few external, surface
mount components are required to
produce a complete switching regulator. All the features found on the
LT1376 have been retained, including current mode control, external
synchronization and a low current
(typically 20µ A) shutdown mode.
Improvements have been made to
reduce start-up headroom and switching noise. A novel power device layout
makes it possible to fit a high speed,
bipolar, 80mΩ switch into a surface
mount SO-8 package. The LT1374 is
also available in DD and TO-220 packages for higher power applications.
LT1374 Features
❏ Constant 500kHz switching
frequency
❏ 80mΩ high speed switch
❏ 20µA shutdown current
❏ Uses all surface mount
components
❏ Cycle-by-cycle current limiting
❏ Available in SO-8 package
The LT1374 is a constant-frequency,
current mode buck converter. As
shown in Figure 1, an internal clock
and two feedback loops control the
power switch. In addition to the normal output feedback error amplifier,
a current sense amplifier monitors
switch current on a cycle-by-cycle
basis. The feedback error amplifier
output, VC, is compared to the switchcurrent sense output to control the
trip point of the power switch. This
removes the additional 90° phase shift
that occurs with voltage-controlled
systems, easing frequency compensation and giving a faster transient
response. Nonlinear slope compensation is added to the current sense
signal to prevent subharmonic oscil-
0.01Ω
INPUT
+
2.9V BIAS
REGULATOR
BIAS
–
CURRENT
SENSE
AMPLIFIER
VOLTAGE GAIN = 20
INTERNAL
VCC
SLOPE COMP
Σ
BOOST
0.9V
500kHz
OSCILLATOR
SYNC
S
CURRENT
COMPARATOR
+
SHUTDOWN
COMPARATOR
+
RS
FLIP-FLOP
DRIVER
CIRCUITRY
R
–
–
Q1
POWER
SWITCH
VSW
0.4V
FREQUENCY
SHIFT CIRCUIT
SHDN
3.5µA
+
Q2
–
FOLDBACK
CURRENT
LIMIT
CLAMP
FB
–
2.38V
VC
ERROR
AMPLIFIER
gm = 2000µMho
+
LOCKOUT
COMPARATOR
2.42V
GND
1374 BD
Figure 1. LT1374 block diagram
Linear Technology Magazine • November 1997
5
DESIGN FEATURES
100
D2
1N914
INPUT
6V TO 25V
C3*
10µF TO
50µF
BOOST
+
R1
56k
OUTPUT**
5V/4.25A
VSW
VIN
L1**
5µH
LT1374-5 BIAS
SHDN
GND
R2
33k
FB
VC
CC
3.3nF
D1
MBRS330T3
+
* RIPPLE CURRENT RATING > IOUT/2
** L1 = COILTRONICS UP2-4R7; (561) 241-7876
INCREASE L1 TO 10 µH FOR LOAD CURRENTS ABOVE 3.5A AND TO 20µH ABOVE 4A
C1
100µF, 10V
SOLID
TANTALUM
6
90
85
80
75
70
0
0.5
1.0 1.5 2.0 2.5
LOAD CURRENT (A)
1374_02.EPS
Figure 2. 5V buck converter
lation at higher duty cycles. The slope
amplitude has been set to stabilize
inductance values as low as 2µH. The
feedback pin, FB, provides several
overload protection functions. As feedback falls below 1.7V, a voltage clamp
is gradually applied to the VC pin,
reducing current limit. At power-up,
this performs a soft-start function.
When FB is below 1V, as in the case of
a shorted output, the oscillator frequency is reduced to 100kHz. Because
the minimum on-time for the switch
remains the same, minimum duty
cycle is effectively reduced by a factor
of five. This lowers power dissipation
in both the LT1374 and the catch
diode, D1, as shown in Figure 2.
High switch efficiency is attained
by driving the switch with a voltage
that is higher than the input voltage,
allowing the switch to saturate. This
boosted voltage is generated on the
BOOST pin with an external capacitor, C2, and diode, D2 as shown in
Figure 2.
The LT1374 has been improved by
reducing minimum start-up voltage
at low output currents, as shown in
Figure 4. At low output currents,
there is not enough energy in the
inductor at switch-off to drive the
switch-node capacitance down to
ground. In this condition, the minimum start-up voltage is the unboosted
voltage drop across the output switch.
Improvements made to the drive circuitry of the LT1374 have reduced
this drop from 2.5V on the LT1376 to
1.2V.
Additionally, the minimum boost
voltage required to fully saturate the
switch has been reduced to 2.3V.
EFFICIENCY (%)
95
C2
0.27µF
Current used by the boost circuit
appears as an efficiency loss; supplying this from a lower voltage, via the
boost diode, D2, improves efficiency.
For most applications with outputs
above 3V, the configuration shown in
Figure 2 will be optimal. Converters
with lower output voltages should
use the input or an alternative supply
for the boost diode.
Switch-transition time has important effects on the usability and overall
efficiency of the converter. A fast
switch-transition time gives high efficiency, but it also creates EMI and
RFI. Slowing the transition time of the
switch reduces noise but lowers efficiency. The LT1374 operates at higher
currents than the LT1376, so the
same transition time would result in
more noise. Because the low onresistance of the LT1374 reduces I2R
losses in the switch, a small increase
in switching transition losses is an
acceptable trade-off to reduce noise.
The switching time for the LT1374 is
24ns, approximately 50% slower than
that of the LT1376. Most of the
LT1374’s circuitry operates from an
internally regulated 2.9V bias line.
By default, the bias regulator draws
power from the input pin, but if the
BIAS pin is connected to an external
source higher than 3V (normally VOUT),
bias current will be drawn from this
source, improving efficiency.
An external clock signal (up to
900kHz) can be fed into the SYNC pin
to increase the internal oscillator frequency or synchronize it to a system
clock. The SYNC function is defeated
during start-up, when FB is low, to
allow correct operation of the reduced
3.0
3.5
1374_03
Figure 3. Efficiency: 10V in, 5V out
frequency, shorted-output protection
circuit. The shutdown pin has two
functions, undervoltage lockout and
shutdown. Below 2.38V, the switching function is disabled. With the
addition of an external divider to the
input (R1 and R2 in Figure 2), an
undervoltage lockout can be implemented. Below 0.4V, a complete
shutdown of the internal circuitry
occurs, reducing supply current to
20µA.
4.5A in an SO-8
The LT1374 is available in several
packages, two of which, the 7-lead
DD and the SO-8, are surface mount.
Thermal considerations are always
important when operating surface
mounted power devices without additional heat sinking. The output switch
of the LT1374 has been designed to
minimize power dissipation from both
on-resistance and base-drive current.
The SO-8 package has a fused ground
lead. This lead has a solid connection
to the tab to which the die is attached,
providing low thermal resistance out
of the package. Adding copper circuit
board area around this pin can reduce
junction-to-ambient thermal resistance to 80°C/W. This allows the use
of the SO-8 packaged LT1374 in
applications that would have previously required a TO-220. For example,
a motor drive may require 4A at startup, but only 2.5A when running. The
SO-8 packaged LT1374 can provide
the full 4.5A of switch current required
to supply the 4A peak load. It could
supply this current for several seconds due to the long thermal time
Linear Technology Magazine • November 1997
DESIGN FEATURES
8.0
LT1376
INPUT VOLTAGE (V)
7.5
SWITCH NODE
L1
7.0
5V
6.5
LT1374
VIN
6.0
HIGH
FREQUENCY
CIRCULATING
PATH
D1
LOAD
5.5
5.0
1374 F06
1
10
100
LOAD CURRENT (mA)
1000
1374_04
Figure 4. Minimum input voltage with
5V output
constant of the SO-8 package. Calculating the temperature rise for 2.5A of
continuous current using the circuit
in Figure 2, a 12V to 5V converter, we
see that power losses due to the switch
resistance, switch drive current and
quiescent current are 0.57W, 0.1W
and 0.04W, respectively, totaling
0.71W. On a well laid-out PCB using
the SO-8 package, this power dissipation would result in only a 57°C
rise of die temperature, allowing the
part to operate at an ambient temperature of 77°C. If 4A of continuous
current is required, use the surface
mounted DD package (θJA = 30°C/W).
For even higher power, the TO-220
package (θJC = 4°C/W) may be used.
Application:
5V/4.25A Buck Converter
With its 25V input and 4.5A minimum switch current, the LT1374 will
fit into a wide range of applications.
Figure 2 shows a typical buck converter with a 6V to 25V input range, a
5V output and 4.25A of output current capability. Due to the low
on-resistance of the switch, efficiency
remains high over a wide range of
currents, as shown in Figure 3. To
reduce power dissipation, both the
BIAS pin and boost circuit are supplied from the 5V output.
Several factors, including maximum current, core and copper losses,
size and cost, affect the choice of
inductor, L1. A high value, high current inductor gives the highest output
current with the lowest ripple, at the
expense of a large physical size and
cost. Lower inductance values tend to
Linear Technology Magazine • November 1997
Figure 5. High speed switch path
be physically smaller, have higher
current ratings and are cheaper, but
output ripple current, and hence
ripple voltage, increases.
The input capacitor, C3, experiences very high ripple currents, up to
IOUT/2, so low ESR tantalum capacitors are needed. At 4.25A output
current, two capacitors in parallel are
required to meet the ripple current
requirement. The ripple current in
the output capacitor, C1, is lower, but
its ESR still needs to be low to limit
output voltage ripple. The voltage drop
across the catch diode, D1, has a
significant effect on overall converter
efficiency, especially at higher input
voltages when the switch duty cycle is
low. Its ability to survive short-circuit
conditions may increase its power
rating. For good electrical performance, D1 must be placed close to
the LT1374. The power dissipated in
D1 will raise the PC board’s temperature around the LT1374. This must
be taken into account when modeling
or taking bench measurements of die
temperature.
The loop compensation capacitor,
CC, produces a pole in the frequency
response at 240Hz. Unity-gain phase
margin can be further improved with
the addition of a resistor, typically 2k,
in series with CC, adding a zero to the
frequency response. This, however,
can cause a large-signal subharmonic
problem in the loop. The output ripple
voltage feeds back through the error
amplifier to the VC pin, changing the
current trip point of the next cycle.
This changes the voltage ripple at the
output, and the loop is closed. Adding
a second capacitor directly from the
VC pin to ground to form a pole at onefifth the switching frequency solves
the problem.
PCB Layout
All high current, high speed circuits
require careful layout to obtain optimum performance. When laying out
the PCB, keep the trace length around
the high frequency switching components, shown in Figure 5, as short as
possible. This minimizes the EMI and
RFI radiation from the loop created by
this path. These traces have a parasitic inductance of approximately
20nH/inch, which can cause an
additional problem at higher operating voltages. At switch-off, the current
flowing in the trace inductance causes
a voltage spike. This is in addition to
the input voltage across the switch
transistor. At higher currents, the
additional voltage can potentially
cause the output switching transistor to exceed its absolute maximum
voltage rating.
Conclusion
With its 80mΩ on-resistance, the
LT1374 makes a very compact, low
parts count, high current DC/DC
converter without the need for separate control and power devices. The
wide selection of package outlines
provides a solution for most miniature, low profile applications. A 25V,
500kHz, 4.5A switch gives the flexibility for a wide range of applications.
The LT1374 complements and extends
the current range of Linear’s DC/DC
converters.
7
DESIGN FEATURES
The LT1581 Low Dropout Regulator
Provides 10 Amps of Output Current
by Todd Owen
Introduction
Improvements in microprocessors and
logic devices continue to increase their
speed and power. Gate dimensions
continue to decrease along with the
supply voltages required to power
them. Unfortunately, the supply current requirements for these new logic
devices are heading in the opposite
direction: a larger number of gates
driven at higher clock frequencies
requires higher supply current. In
addition, the load transients are larger
and faster. Current steps from several hundred milliamperes to several
amperes in tens of nanoseconds are
common.
Lower supply voltages, higher supply currents and faster load transients
require tighter specifications for regulators. Input/output differential
voltages tend to be under 1V, supply
currents can be up to 10A and transitions from no load to full load can
occur in tens of nanoseconds. Low
dropout PNP-based regulators can
meet input/output differential specifications, but their transient load
performance is less than stellar. Conventional NPN-based regulators can
meet the higher load current specifications and transient needs, but
dropout voltage tends to be above 1V.
In addition, this higher dropout voltage specification requires the use of
higher input supply voltages, leading
to increased power dissipation in the
regulator.
The LT1581
Solves Many Problems
The new LT1581 NPN regulator can
meet the needs of many new designs.
By using separate supplies for the
control circuitry and the drive to the
NPN output transistor, the dropout
on the pass transistor is reduced to
the saturation voltage of the transistor. With this approach, it is easy to
obtain dropout voltages as low as
8
accuracy has been improved to a guaranteed 0.6% tolerance. The part can
easily hold 1% output accuracy over
temperature, guaranteed, while
operating with an input/output differential voltage of well under 1V.
Careful design has eliminated any
supply sequencing issues associated
with a dual-supply system. The output is inhibited until both input
supplies are operating. If the control
voltage reaches a quiescent state first,
Lower supply voltages,
the output current will be limited to a
higher supply currents and
few milliamperes until the power input
faster load transients
voltage stabilizes. If the power input
require tighter
comes up first, the output will not
specifications for
turn on at all until the control voltage
reaches a quiescent state. The output
regulators. Input/output
can never turn on unregulated. The
differential voltages tend to
LT1581 can also be operated as a
be under 1V, supply
single-supply device by tying the concurrents can be up to 10A
trol and power inputs together.
and transitions from no
Dropout in single supply operation is
load to full load can occur
essentially the same as would be
expected in a conventional NPN-based
in tens of nanoseconds.
low dropout regulator.
All of these features add together
control voltage is 13V. The part is to make a part with excellent static
fully protected from overcurrent and and dynamic specifications. Additionovertemperature conditions. Fixed ally, the low dropout voltage solves a
output voltage and adjustable ver- problem many designers have been
sions are available in a 7-pin TO-220 faced with: power dissipation. Lowerpackage.
ing the input/output differential
Both fixed and adjustable versions voltage translates directly to reduced
of the LT1581 have remote-sense pins, power dissipation in the regulator.
permitting very accurate regulation Care must be exercised regarding
of output voltage at the load, where it board layout and thermal managecounts, rather than at the regulator. ment if the full performance capability
Typical load regulation is only 1mV of the LT1581 is to be achieved.
for a 2.5V output over a 100mA to 10A
range. The adjust pin is brought out Layout Considerations
on the fixed voltage version of the Board layout becomes more critical
part. This allows for the addition of a at higher current levels. Running too
small bypass capacitor on the feed- much current through a narrow trace
back divider, greatly improving can turn the trace into a fuse. Even
transient response. A capacitor value with a wide trace, the resistance preof 0.1µF to approximately 1µF will sented can cause problems. For 1 oz.
generally provide optimum transient copper, a 0.10 inch wide trace that is
response. Additionally, reference 0.7 inches long can easily handle up
220mV at 4A, 310mV at 7A or 430mV
at 10A of load current. The current
requirement for the control circuitry
is approximately 1/150 of the load
current, or about 70mA for a 10A
load. This control voltage must be at
least 1.2V greater than the output to
achieve these low dropout specifications. Maximum supply voltage to the
power input is 7V and the maximum
Linear Technology Magazine • November 1997
DESIGN FEATURES
ESR
EFFECTS
ESL
EFFECTS
CAPACITANCE
EFFECTS
1581_01.eps
SLOPE,
V ∆I
=
t
C
POINT AT WHICH REGULATOR
TAKES CONTROL
Figure 1. Transient effects
to 10A of current. However, the same
trace has a resistance of 3mΩ. This
adds 30mV to the overall dropout
voltage of the regulator, and without
remote load sensing, adds a 30mV
load regulation error to the output.
Remote load sensing can virtually
eliminate the problems associated
with the parasitic trace resistance,
but cannot remove the added dropout
voltage. Adding 30mV to the dropout
voltage constitutes an increase of 7%.
This additional drop reduces the available headroom, and can reduce the
maximum available output current.
Additional consideration must be
given to power dissipation in the copper trace. With 10A of current and
30mV of drop across the trace, 0.3W
of power is being dissipated in the
trace alone. This can add to the overall problem of system-level thermal
management. It is recommended that
a good reference source be consulted
to determine minimum required trace
widths.1
High frequency current transients
also necessitate careful attention to
board layout. Load current steps
contain higher order frequency components that must be handled by the
output decoupling network until the
regulator throttles to the new load
current level. Capacitors are not ideal
elements and contain parasitic resistance and inductance that dominate
the change in output voltage at the
beginning of a load transient step.
Using capacitors with low ESR, low
ESL and good high frequency characteristics is critical in high speed
applications. The location of the
decoupling network is critical to transient performance.
For DC operation, the designer’s
main concern will be parasitic resistance. Consideration must be given
to all high current paths to minimize
the parasitic resistance and inductance present in the input, output
and ground paths for the regulator.
Maximize the trace widths for all high
VCONTROL = 5V
3
VPOWER = 3.3V
VCONT
VIN
LT1581
SENSE
ADJ
VOUT
2
5
VOUT = 2.5V
4
VCC
7
C3
22µF
25V
+
+
R1
110Ω
1%
C2
220µF
10V
C4
0.33µF
C1
R2
110Ω 100µF
10V
1%
+
100µF
10V
2X
+
Thermal Management
for the LT1581
1µF
25V
10X
VSS
RTN
1581_02.eps
Figure 2. The LT1581 delivers 2.5V from 3.3V at up to 9A.
Linear Technology Magazine • November 1997
currents paths in the layout. The best
solution requires a multilayer board.
With one layer devoted to power and
a separate layer for the return path,
narrow copper traces are replaced
with entire copper planes. With power
and ground planes, parasitic resistance is reduced to nearly zero. An
additional benefit is the elimination
of the power dissipation in copper
traces, easing system-level thermal
management.
When considering the transient
performance, higher order frequency
components must not be neglected.
Capacitors with low ESR and low ESL
are a necessity. Transient load currents through the ESR of a capacitor
shows up as an instantaneous droop
in output voltage (∆V = ∆I • ESR).
When load current steps occur, parasitic inductance tries to limit the rate
of change of the current. The result is
a voltage spike at the start of the load
transient. This spike is defined by V =
L • dI/dt, where L is the ESL of the
capacitors and their interconnecting
traces. These transient effects are
illustrated in Figure 1. Decoupling
capacitors should be placed as close
to the load as possible, with multiple
capacitors used to distribute and
reduce overall parasitics. Any narrow
traces will increase parasitic inductance. Using power planes can reduce
these effects, but will not eliminate
them entirely. Other possible problems can occur when using vias to
connect capacitors to inner plane layers, as they have parasitic inductance
and resistance as well. The use of
multiple vias is highly recommended
to reduce parasitics. Also, be sure to
minimize the distance between pads
and vias.
MICROPROCESSOR
SOCKET
Even at the minimum differential voltages, power dissipation in the LT1581
can reach several watts. Without
adequate heat sinking, maximum
rated junction temperature can easily be exceeded. Excessive junction
temperature will reduce the reliability of the part. By using an adequate
heat sink, junction temperature can
9
DESIGN FEATURES
be maintained well under maximum,
ensuring reliable operation.
Power dissipation is made up of
two main components: the power in
the output transistor and the power
in the drive circuit. The power in the
additional control circuitry is negligible at a maximum of 25mW.
To calculate the total power dissipation, use the following formula:
PTOTAL = POUTPUT + PDRIVE
where POUTPUT =
(VPOWER – VOUT)(IOUT)
and PDRIVE =
(VCONTROL – VOUT)(ICONTROL)
ICONTROL is a function of output
current and is equal to between
IOUT/150 (typ) and IOUT/58 (max).
The electrical characteristics section of the LT1581 data sheet provides
separate thermal resistances and
maximum junction temperatures for
both the control circuitry and the
power transistor. These specifications
have been developed to more accurately reflect device temperature and
ensure safe operating temperatures.
Maximum junction temperature must
be calculated for both sections to
ensure that both thermal limits are
met.
Junction-to-case thermal resistance is specified from the IC junction
to the bottom of the case directly
below the die. This is the lowest resistance path for heat flow. Proper
mounting is required to ensure the
best possible thermal flow from this
area of the package to the heat sink.
Please consult Linear Technology’s
“Mounting Considerations for Power
Semiconductors,” 1990 Linear Applications Handbook, Volume 1, pages
RR3–1 to RR3–20. Note that the case
of the LT1581 is electrically connected
to the output.
The following example illustrates
how to calculate the heat sink size
needed to ensure reliable operation of
the LT1581. The circuit in Figure 2 is
used with the following assumptions:
VCONTROL (max continuous) = 5.25V
(5V + 5%),
VPOWER (max continuous) = 3.465V
(3.3V + 5%),
VOUT = 2.5V, IOUT = 8A, TA = 40°C,
θCASE–HEAT SINK = 1°C/W
(with thermal compound).
The following specifications are
from the LT1581 data sheet:
RθJC(CONTROL) = 0.65°C/W,
RθJC(POWER) = 2.5°C/W,
TJ(MAX) = 125°C for the control
section, 150°C for the power
section.
Power dissipation under these conditions is equal to:
PDRIVE = (VCONTROL – VOUT)(ICONTROL)
ICONTROL = IOUT/58 = 8A/58 = 138mA
PDRIVE = (5.25V – 2.5V)(138mA) =
380mW
POUTPUT = (VPOWER – VOUT)(IOUT)
= (3.465V – 2.5V)(8A) = 7.72W
Total Power Dissipation = 8.1W
Junction temperature will be equal
to:
TJ = TA + PTOTAL (θHEAT SINK + θCASE –
HEAT SINK + θJC)
For the control section:
TJ(MAX) = 125°C = 40°C +
8.1W(θHEAT SINK + 1°C/W +
0.65°C/W)
θHEAT SINK + 1.65°C/W = (125°C –
40°C)/8.1W
θHEAT SINK(MAX) = 8.8°C/W
For the power section:
TJ(MAX) = 150°C = 40°C +
8.1W(θHEAT SINK + 1°C/W + 2.5°C/W)
θHEAT SINK + 3.5°C/W = (150°C –
40°C)/8.1W
θHEAT SINK(MAX) = 10°C/W
As shown, the maximum thermal
resistance for the heat sink must be
below 8.8°C/W to ensure reliable
operation from the LT1581. This
resistance value can be used to specify
the heat sink and airflow requirements needed to operate the LT1581
below specified thermal limits.
Conclusion
The LT1581 can be used for a variety
of applications where high current
and low dropout voltage are a must. It
can deliver up to 10A of output current at a typical dropout voltage of
430mV. A remote-sense pin virtually
eliminates load regulation problems.
Supply-sequencing issues have been
eliminated. The LT1581’s excellent
transient performance makes it suitable for high current applications
where load current transients occur
at high frequency. The low dropout
voltage reduces the amount of input/
output differential voltage required
and simultaneously reduces the power
dissipation in the regulator. When
consideration is given to board layout
and thermal management, performance can be maximized.
1
Fink, D.G. and D. Christiansen: Electronic
Engineers’ Handbook (Second Edition). McGraw
Hill, 1982. Figure 7–75.
for
the latest information
on LTC products,
visit
www.linear-tech.com
10
Linear Technology Magazine • November 1997
DESIGN FEATURES
The LT1370: New 500kHz, 6A
Monolithic Boost Converter
Introduction
Circuit Description
Complementing and expanding on the
current LT1371/LT1372 family of
500kHz switchers, Linear Technology introduces the LT1370, a 6A boost
converter. At 65mΩ on-resistance, 42V
maximum switch voltage and 500kHz
switching frequency, the LT1370 can
be used in a wide range of output
voltage and current applications.
The high efficiency switch is
included on the die, along with the
oscillator, control and protection circuitry necessary for a complete
switching regulator. This part combines the convenience and low parts
count of a monolithic solution with
the switching capabilities of a discrete power device and controller. All
the features found on the LT1371/
LT1372 have been retained on the
LT1370, including current mode
operation, external synchronization
and low current shutdown mode
(12µA typical). Only a few surface
mount components are needed to
complete a small, high efficiency DC/
DC converter. The LT1370 will operate in all the standard switching
configurations, including boost, buck,
flyback, forward, inverting and SEPIC.
The LT1370 is a current mode
switcher. This means that switch duty
cycle is directly controlled by the
switch current rather than by the
output voltage. Referring to Figure 1,
the switch is turned on at the start of
each oscillator cycle. It is turned off
when the switch current reaches a
predetermined level. The switch-off
current threshold is controlled by an
output voltage error amplifier. This
technique has several advantages.
First, it responds quickly to input
voltage variations, unlike voltage mode
switchers, which have notoriously
poor line-transient response. Second,
it reduces the 90° phase shift at midfrequencies in the energy storage
inductor. This greatly simplifies
closed-loop frequency compensation
under widely varying input voltage
and output load conditions. Finally,
it allows simple pulse-by-pulse current limiting to provide maximum
switch protection under output overload or short-circuit conditions. A low
dropout internal regulator provides a
2.3V supply to all internal circuitry.
This low dropout design allows the
input voltage to vary from 2.7V to 25V
with virtually no change in device
performance. An internal 500kHz
oscillator is the basic clock for all
timing. It turns the output on via logic
and driver circuitry. Special adaptive
antisaturation circuitry detects the
onset of saturation in the power switch
and adjusts driver current instantaneously to limit switch saturation.
This minimizes driver dissipation and
provides very rapid turn off of the
switch.
A 1.245V bandgap reference biases
the positive input of the error amplifier. The negative input of the amplifier
is brought out for positive output
voltage sensing. The error amplifier
has nonlinear transconductance to
reduce output overshoot on start-up
or overload recovery. When the feedback voltage exceeds the reference by
40mV, error amplifier transconductance increases by a factor of ten. This
allows the current-trip threshold, set
by VC, to be lowered faster, reducing
overshoot. The feedback input also
VIN
SHUTDOWN
DELAY AND RESET
LT1370 Features
❏ 6A Minimum switch-current limit
❏ Constant 500kHz switching
frequency
❏ 65mΩ high speed switch
❏ 12µA shutdown current
❏ 2.7V minimum input operating
voltage
❏ Cycle-by-cycle current limiting
❏ Available in DD and TO-220
packages
❏ 42V switch voltage
by Karl Edwards
SW
LOW DROPOUT
2.3V REG
ANTI-SAT
S/S
SYNC
LOGIC
OSC
DRIVER
SWITCH
5:1 FREQUENCY
SHIFT
+
100k
NFB
NFBA
–
COMP
50k
–
FB
+
1.245V
REF
+
EA
IA
VC
AV ≈ 20
0.005Ω
–
GND
GND SENSE
1370_01.EPS
Figure 1. LT1370 block diagram
Linear Technology Magazine • November 1997
11
DESIGN FEATURES
L1*
D1
MBRD835L
VIN
OFF
ON
S/S
100
VOUT†
12V
VSW
R1
53.6k
1%
FB
+
95
LT1370
+
C1**
22µF
25V
VC
GND
C2
0.047µF
R3
2k
R2
6.19k
1%
C4**
22µF
25V
×2
*COILTRONICS
(561) 241-7876
UP2-4R7 (4.7µH)
UP4-100 (10µH)
**AVX TPSD226M025R0200
†MAX I
OUT
C3
0.0047µF
1370_02.EPS
L1
IOUT
4.7µH 1.8A
10µH 2.0A
EFFICIENCY (%)
5V
90
85
80
75
70
0
0.5
1.0
1.5
OUTPUT CURRENT (A)
2.0
1370_03
Figure 2. 5V to 12V boost converter
invokes oscillator frequency shifting,
which helps protect components during overload conditions. When the
feedback voltage drops below 0.6V,
the oscillator frequency is reduced
5:1, to 100kHz. Lower switching frequency, with the same minimum on
time, reduces the minimum switch
duty cycle and short-circuit current
in the inductor.
Similar to the LT1371, error amplifier circuitry allows the LT1370 to
directly regulate negative output voltages. The NFB pin regulates at –2.49V
while the amplifier’s output internally drives the FB pin to 1.245V.
This architecture, which uses the
same main error amplifier, avoids
duplication of functions and maintains ease of use. The error signal
developed at the amplifier output is
brought out externally to the VC pin.
The VC pin has three different functions. It is used for frequency
compensation, current limit adjustment and soft starting. During normal
regulator operation, this pin is in the
voltage range of 1V to 1.9V. The error
amplifier is a current output (gm) type,
so this voltage can be externally
clamped to lower the current limit.
Likewise, a capacitor-coupled external clamp provides soft start. Switch
duty cycle goes to zero if the VC pin is
pulled below the control threshold
(typically 1V), placing the LT1370 in
an idle mode.
The S/S pin has two functions,
synchronization and shutdown. The
internal oscillator can be synchronized to a higher frequency by applying
a TTL square wave to this pin. This
allows the part to be synchronized to
12
a system clock. If the S/S pin is held
low, after a short delay the LT1370
will enter shutdown mode. In shutdown mode, all internal circuitry is
disabled, reducing supply current to
12µA. An internal pull-up ensures
start-up when the S/S pin is left
open.
5V to 12V Boost Converter
Figure 2 shows a typical 5V to 12V
boost application. The feedback
divider network has been selected to
give the desired output voltage. As
long as R2 is less than 7k, FB input
bias current can be ignored. The
inductor needs to be chosen carefully
to meet both peak and average current values. The output capacitor can
see high ripple currents—often, as in
this application, higher than the ripple
rating of a single capacitor. This
requires the use of two surface mount
tantalums in parallel; both capacitors should be of the same value and
manufacturer. The input capacitor
does not have to endure such high
ripple currents and a single capacitor
will normally suffice. The catch diode,
D1, must be rated for the output
voltage and average output current.
The compensation capacitor, C2, normally forms a pole in the 2Hz to 20Hz
range, with a series resistor, R3, to
add a zero at 1kHz to 5kHz.
A second capacitor, C3, is sometimes required to prevent erratic
switching. Ripple current in the output
capacitor’s ESR causes voltage ripple.
This feeds back through the error
amp to the VC pin, changing the current-trip threshold cycle-to-cycle. The
problem appears as subharmonic
Figure 3. 12V output efficiency
oscillation. Adding C3, typically onetenth the value of the main
compensation capacitor, reduces the
loop gain at the switching frequency,
preventing the oscillation.
The ground return from the compensation network must be separate
from the high current switch ground.
If drops in the ground trace due to
switch current cause the VC pin to
dip, premature switch-off will occur.
This effect appears as poor load regulation. A solution to this is to return
the compensation network to the FB
pin. The S/S pin in this example is
driven by a logical on/off signal, a low
input forcing the LT1370 into its 12µA
shutdown mode. Figure 3 shows the
overall converter efficiency. Note that
peak efficiency is over 90%; efficiency
stays above 86% at the device’s maximum operating current.
Positive-to-Negative
Converter
The NFB (negative feedback) pin allows
negative output regulators to be
designed with direct feedback. In the
circuit shown in Figure 4, a 2.7V to
13V input, –5V output converter, the
output is monitored by the NFB pin
and a simple divider network. No complex level shifting or unusual
grounding techniques are required.
The regular FB pin is left open circuit
and the divider network, R2, R3, is
calculated based on the –2.49V NFB
reference voltage and 30µA of input
current. The switch-clamp diodes, D2
and D3, prevent the leakage spike
from the transformer, T1, from
Linear Technology Magazine • November 1997
DESIGN FEATURES
VIN
2.7V TO 13V
+
ON
VSW
S/S
•4
2
D2
P6KE-15A
D3
1N4148 1 •
C1
100µF
VIN
OFF
+
–VOUT†
–5V
R2
2.49k
1%
NFB
C3
0.0047µF
C4
100µF
×2
3
D1
MBRD835L
LT1370
VC
rent range. With good layout, the DD
package can have a thermal resistance as low as 20°C/W to 30°C/W,
junction-to-ambient. In the boost converter shown in Figure 2, with an
input voltage of 5V and an output
voltage of 12V at 2A, power dissipation in the LT1370 is 2.2W. This causes
a 50°C to 60°C rise in die temperature
above ambient. Maximum junction
temperature is 125°C, so this application would be able to operate up to
a 70°C ambient.
T1*
R3
2.49k
1%
GND
*PULSE PE-53719
(619) 674-8100
†MAX I
OUT
IOUT VIN
1.75A 3V
2.25A 5V
3.0A 9V
C2
0.047µF
R1
2k
1370_04.EPS
Conclusion
Figure 4. Positive-to-negative converter with direct feedback
exceeding the switch’s absolute maximum voltage rating. The Zener voltage
of D2 must be higher than the output
voltage, but low enough that the sum
of input voltage and clamp voltage
does not exceed the switch voltage
rating.
5V SEPIC Converter
Figure 5 shows a SEPIC converter.
One of the advantages of the SEPIC
topology is that the input voltage can
range from below to above the output
voltage. In Figure 5, the input voltage
range is from 4V to 9V, with a 5V
output. The magnetic coupling of
inductors L1A and L1B is not critical
for operation, but generally they are
wound on the same core. C2 couples
With its low resistance switch, 6A
operating current and 500kHz
operation, the LT1370 is ideal for
small, low parts count, high current
applications. Its high switching frequency eliminates the need for bulky
magnetics and capacitors. Compared
to a separate control device and power
switch, the LT1370’s monolithic
approach simplifies the design effort
required to implement a complete
DC/DC converter. The choice of TO220 and surface mount DD package,
42V switch, and flexibility of switching topologies makes the LT1370 ideal
for a wide range of applications. The
LT1370 extends Linear Technology’s
LT1371 and LT1372 family of 500kHz,
monolithic switchers all the way up to
6A.
the inductors together and eliminates
the need for a switch snubber network. C2 must have a very low ESR,
because the ripple current is equal to
ISW/2. Its capacitance value is not
critical and has no significant effect
on loop stability. The voltage across
C2 is equal to the input. A 4.7µF, 50V
ceramic will work in most SEPIC
applications. The S/S pin is used as
a logical on/off signal. In the off state,
there is no leakage to the output, and
only 12µA leakage from the input.
Thermal Performance
Due to the very low on-resistance of
the LT1370 switch and its driver efficiency, the surface mount DD package
version can be used over its full curVIN
4V TO 9V
L1A*
6.8µH
VIN
OFF
ON
VSW
S/S
•
LT1370
+
C1
33µF
20V
FB
GND
D1
MBRD835L
R2
18.7k
1%
C2
4.7µF
•
VC
+
L1B*
6.8µH
R1
2k
C4
0.047µF
C5
0.0047µF
R3
6.19k
1%
VOUT†
5V
C3
100µF
10V
×2
1370_05.EPS
C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E475ZY5U-C304
C3 = AVX TPSD107M010R0100
* BH ELECTRONICS 501-0726
(612) 894-9590
** INPUT VOLTAGE MAY BE GREATER OR
LESS THAN OUTPUT VOLTAGE
†MAX I
OUT
IOUT VIN
1.8A 4V
2A 5V
2.6A 7V
2.9A 9V
Figure 5. Two Li-Ion cells to 5V SEPIC converter
Linear Technology Magazine • November 1997
13
DESIGN FEATURES
New LTC1504: Flexible, Efficient
Synchronous Switching Regulator Can
Source or Sink 500mA
by Dave Dwelley
Introduction
The LTC1504 is a new addition to
LTC’s family of 8-pin step-down
switching regulators. It consists of a
200kHz fixed frequency, voltage-feedback, buck-mode switching regulator
controller and a pair of 1.5Ω power
switches in an 8-pin SO package. The
LTC1504 also includes a synchronous rectifier on-chip, maximizing
efficiency and minimizing external
parts count while allowing the output
to both sink and source current: it
can source or sink up to 500mA with
input voltages from 3.3V to 10V and
output voltages as low as 1.26V. The
LT1504 can achieve 100% duty cycle
at the output switch, maximizing
dropout performance with low inputto-output voltage differentials. The
L TC1504 includes an onboard
precision reference and user programmable current-limit and
soft-start circuits, allowing implementation of full-featured power
conversion circuits with a minimum
of external components. The architecture is optimized for maximum
efficiency at loads above 50mA and
IMAX
+
VCC
CIN
22µF
Minimum
Component-Count Circuits
Figure 1 shows a fully functional
LTC1504 5V to 3.3V regulator,
including current limit and soft-start,
using the fixed-output LTC1504-3.3
and only six external components.
The on-chip synchronous rectifier
eliminates the need for an external
catch diode, and the internal current-limit circuit requires only a single
resistor to program the maximum
SHUTDOWN
RIMAX**
68k
VIN
5V
does not include a light-load Burst
Mode™ circuit. This penalizes efficiency at very light loads but allows
the device to seamlessly shift between
sourcing and sinking current, opening
up a whole new class of applications.
A micropower shutdown mode is
included. The diminutive SO-8 package minimizes the amount of space
the LTC1504 fills while allowing
adequate thermal dissipation for
500mA load current levels. The
LTC1504 allows previously impossible (or at least awkward) tasks to be
completed with ease.
IMAX
LEXT 47µH
SHDN
VOUT
3.3V/500mA
SW
SS
CSS*
O.1µF
SHUTDOWN
RIMAX
68k
LTC1504-3.3
GND
output current. Efficiency is above
90% with load currents between 50mA
and 200mA, peaking at 92% at 100mA
and remaining above 82% all the way
to the maximum 500mA load. Current limit is set at 500mA in this
example; it can be reduced by lowering the value of RIMAX. CSS sets the
start-up time at approximately 25ms.
Both the current limit resistor (RIMAX)
and soft-start capacitor (CSS) are
optional; either can be deleted and its
pin left floating if the functionality is
not needed. An LTC1504 circuit can
be constructed with just four components: input and output capacitors,
an external inductor and a compensation capacitor.
The circuit in Figure 1 relies on the
ESR of the output capacitor to maintain loop stability with just a single
capacitor at the COMP pin. Figure 1
uses a surface mount electrolytic
capacitor with about 400mΩ ESR. A
low ESR tantalum output capacitor
can improve the transient response at
the output but requires a more complex compensation network at the
SENSE
+
COMP
COUT
100µF
CC
1000pF
VIN
5V
+
VCC
CIN
22µF
LEXT 47µH
SHDN
VOUT
3.3V/500mA
SW
LTC1504-3.3
GND
SS
RC
7.5k
CSS
O.1µF
+
SENSE
COMP
COUT
100µF
CF
220pF
CC
0.01µF
C IN = AVX TPSC226M016R0375
C OUT = SANYO 16CV100GX
L EXT = SUMIDA CD54-470
*OPTIONAL: DELETE TO DISABLE SOFT START
**OPTIONAL: DELETE TO DISABLE CURRENT LIMIT
Figure 1. Minimum parts count 5V–3.3V converter
14
1504_02EPS
1504_01.EPS
C IN = AVX TPSC226M016R0375
C OUT = AVX TPSE107M016R0125
L EXT = SUMIDA CD54-470
Figure 2. Improved transient response
Linear Technology Magazine • November 1997
DESIGN FEATURES
110Ω
NC
NC
110Ω
IMAX
I
SHDN
MAX SHDN
VCC
VCC
TERMPWR
10µF
CERAMIC
LEXT 47µH
SW
SW
LTC1504
LTC1504
SENSE
GND
GND
FB
SS
COMP
SS COMP
NC
COUT = AVX TPSC107M006R0150
LEXT = SUMIDA CD54-470
110Ω
7.5k
15k
110Ω
+
18
TO
27
LINES
IMAX
I
SHDN
MAX SHDN
VCC
VCC
5V
10µF
CERAMIC
COUT
100µF
NC
SPLIT SUPPLY
2.5V ±500mA
SW
SW
7.5k
11.8k
+
COUT
47µF
12.1k
220pF
0.01µF
220pF
1540_04.EPS
0.01µF
1540_03.EPS
COUT = TAJC476M016R
LEXT = SUMIDA CDRH73-470 (LOWER RIPPLE/HIGHER EFFICIENCY)
*CDRH73-220 (FASTER TRANSIENT RESPONSE)
Figure 3. SCSI-2 active terminator
Linear Technology Magazine • November 1997
LEXT
47µH (22µH)*
LTC1504
LTC1504
SENSE
GND
GND
FB
SS
COMP
SS COMP
12k
COMP pin (Figure 2). There is a
tradeoff to be made here: the minimum component count solution is
the simplest and uses the least
expensive components but pays a penalty in transient response. The low
ESR circuit in Figure 2 has improved
transient response and actually uses
less board space: the tantalum output capacitor is smaller than the
electrolytic device used in Figure 1
and the additional compensation components are tiny 0603 surface mount
devices. Unfortunately, the tantalum
capacitor used in Figure 2 costs significantly more than the electrolytic
device in Figure 1. You don’t get something for nothing.
Note that the input bypass capacitor in both Figures 1 and 2 is an AVX
TPS type, a relatively costly surgetested tantalum capacitor. This is a
small, surface mount device that has
a surge current rating adequate to
support the 500mA maximum load
current of the LTC1504. Buck regulators (like the LTC1504) inherently
draw large RMS currents from the
input bypass capacitor, and the
capacitor type chosen must be capable
of withstanding this current without
overheating. Different input bypass
capacitors can be used if their ripple
current rating is sufficient to withstand the expected demand (as a rule
of thumb, one-half the maximum expected output current). Just as you
wouldn’t specify a 1/4W resistor to
dissipate 1W of power, neither should
you specify an inadequate input bypass capacitor for a buck regulator.
SD
Figure 4. 5V supply splitter
Tiny, expensive 10µF ceramic capacitors work well, as do larger, cheaper
electrolytic capacitors with specified
ripple-current ratings. Circuits using
electrolytic input bypass devices often require an additional 0.1µ F
ceramic capacitor right next to the
LTC1504 to keep the input capacitor
ESR under control. As with all switching regulator circuits, layout is critical
to obtaining maximum performance;
if in doubt, contact the LTC Applications Department for component
selection and layout advice.
Sink/Source Capability
Improves SCSI Terminators
and Supply Splitters
Figure 3 shows an adjustable-output
LTC1504 connected as a 2.85V regulator for use as a SCSI terminator.
The ability of the LTC1504 circuit to
sink current makes it ideal for use in
terminator applications, where the
load is just as likely to be putting
current into the regulator as taking it
out. The synchronous-buck architecture of the LTC1504 allows it to shift
cleanly between sourcing and sinking current, making it ideal for such
applications. The high efficiency minimizes the power drawn from the
TERMPWR supply and minimizes the
heat generated by the terminator
circuit. Finally, the small number of
tiny external components required
minimizes the space used by the terminator circuit. As in Figure 2, a low
ESR output capacitor is used along
with an optimized compensation network to improve output transient
response and maintain maximum
data fidelity.
Substituting a different set of feedback resistors (Figure 4) creates a 5V
supply splitter, which creates a 2.5V
“ground” to allow analog circuitry to
operate from split supplies. Op amp
circuits and data converters like to
operate from dual supplies, and the
sink/source capability of the LTC1504
allows load currents to be returned
directly to the 2.5V “ground” supply.
The fast transient response of the
L TC1504 keeps the generated
“ground” voltage clean even as the
load switches rapidly between sinking
and sourcing current. Additionally,
the circuit in Figure 4 is significantly
more efficient than the traditional
resistor-divider method of supply
splitting at higher power levels.
Conclusion
The LTC1504 brings the benefits of
synchronous rectification to LTC’s
family of 8-pin switching regulators.
It provides a full-featured, high
efficiency, 500mA supply in a minimum-size, minimum-parts-count
implementation. The LTC1504 is a
natural fit in terminator circuits, supply-splitters and other applications
that require the output to both source
and sink current. Other uses include
distributed power, microprocessor
secondary-voltage supplies, low voltage, battery powered devices and
anywhere else a high efficiency, low
parts count, 500mA step-down supply is required.
15
DESIGN FEATURES
Low Dropout Regulator Driver Handles
Fast Load Transients and Operates on
by Lenny Hsiu
Single 3V–10V Input
Introduction
nal resistor can be added to reduce
the available base-drive current and
thereby limit the regulator output
current. The LT1573 is equipped with
an active-high shutdown and a
thermal shutdown function. The shutdown function can be used to reset
the overcurrent latch. The thermal
shutdown function can be used to
protect the PNP power transistor if it
is thermally coupled to the LT1573.
The LT1573 is available as an
adjustable regulator with an output
range of 1.27V to 6.8V and with fixed
output voltages of 2.5V, 2.8V and
3.3V. Output accuracy is better than
1% to meet the critical regulation
requirement of fast microprocessors.
A special 8-pin fused-lead surface
mount package is used to minimize
regulator footprint and provide
adequate heat sinking.
The LT1573 is the latest low dropout
voltage regulator driver from Linear
Technology. The LT1573 is designed
to provide a low cost solution to applications requiring high current, low
dropout and fast transient response.
When combined with an external PNP
power transistor, this device provides
up to 5A of load current with dropout
voltages as low as 0.35V. The LT1573’s
circuitry is designed for extremely
fast transient response. This greatly
reduces the bulk storage capacitance
required when the regulator is used
in applications with fast, high current load transients.
To keep cost and complexity low,
the LT1573 uses a new, time-delayed
latching current protection technique
that requires no external current
sense resistor. Base-drive current to
the external PNP is limited for instantaneous protection and a time-delayed
latch protects the regulator from continuous short circuits. The latch
time-out period can be varied by an
external capacitor. Guaranteed minimum available base-drive current to
the external PNP is 250mA. An exterVIN
4.5V–5.5V
CIN
100µF
TANT
VIN
LT1573
CTIME***
GND
QOUT
MOTOROLA
D45H11
RC 1k
R1
1.6k
+
COUT1*
FB
LOAD
R2
1k
* FOR T <45˚C, COUT1 = 24 × 1µF Y5V CERAMIC SURFACE MOUNT CAPACITORS
FOR T >45˚C, COUT1 = 24 × 1µF X7R CERAMIC SURFACE MOUNT CAPACITORS
PLACE COUT1 IN THE MICROPROCESSOR SOCKET CAVITY
** COUT2 = 220µF CHIP TANTALUM
*** CTIME = 0.5µF FOR 100ms LATCH-OFF TIME AT ROOM TEMPERATURE
†
SHDN (ACTIVE HIGH) SHOULD BE TIED TO GROUND IF NOT USED
Figure 1. 3.3V, 5A microprocessor supply
16
VOUT = 1.265 • (1 + R1/R2)
VOUT
CC 100pF
LATCH
+
RD
24Ω
RB
50Ω
DRIVE
COMP
External PNP Transistor
Selection Criteria
The selection of an appropriate external PNP transistor depends on the
regulator application specifications.
The critical PNP transistor selection
criteria include:
The basic regulator circuit is shown
in Figure 1. The adjustable-output 1. The maximum output current of
the PNP transistor
LT1573 senses the regulator output
voltage from its feedback pin via the 2. The dropout voltage at the
output voltage divider and drives the
maximum output current
base of the external PNP transistor to 3. The gain-bandwidth product, f ,
T
of the transistor
+
SHDN†
Basic Regulator Circuit
maintain the regulator output at the
specified value. For fixed-output versions of the LT1573, the regulator
output voltage is sensed from the
feedback pin via an internal voltage
divider. The resistor RD is required for
the overcurrent latch-off function. RD
is also used to limit the drive current
available to the external PNP transistor and to limit the power dissipation
in the LT1573. Limiting the drive
current to the external PNP transistor
will limit the output current of the
regulator, thereby minimizing the
stress on the regulator circuit under
overload conditions.
+
COUT2**
The PNP transistor must be able to
supply the specified maximum regulator output current to be qualified
for the regulator application. The VCE
saturation voltage of the transistor at
the maximum output current determines the dropout voltage of the
circuit. The dropout voltage determines the minimum regulator input
voltage for a certain specified output
voltage. The gain-bandwidth product, fT, of the transistor determines
how fast the voltage regulator can
follow an output load change without
losing voltage regulation.
The D45H11 from Motorola and
the KSE45H11TU from Samsung can
Linear Technology Magazine • November 1997
DESIGN FEATURES
Table 1. Dropout voltage of the D45H11
Drive
Current
20mA
20mA
40mA
40mA
60mA
60mA
80mA
100mA
100mA
150mA
200mA
150mA
200mA
250mA
Output
Current
1A
2A
2A
3A
3A
4A
4A
4A
5A
5A
5A
6A
6A
7A
Typical
Droput
Voltage
0.20V
0.50V
0.25V
0.50V
0.25V
0.70V
0.45V
0.35V
0.70V
0.40V
0.35V
0.65V
0.45V
0.50V
be used in all LT1573 regulator circuits with current ratings up to 5A.
The D45H11 can supply 5A of output
current with dropout voltage as low
as 0.35V. The gain-bandwidth product, fT, of the D45H11 is typically
40MHz; this enables a regulator composed of this PNP transistor and an
LT1573 to handle load changes of
several amps in a few hundred nanoseconds with a minimum amount of
output capacitance.
Dropout Voltage
The dropout voltage of an LT1573based regulator circuit is determined
by the VCE saturation voltage of the
discrete external PNP transistor when
it is driven with a base current equal
to the available drive current of the
LT1573. The LT1573 is guaranteed to
sink 250mA of base current (440mA
typical). The available drive current of
the LT1573 can be reduced by adding
a resistor, RD in Figure 1, in series
with the drive pin. Table 1 lists some
useful operating points for the
D45H11 from Motorola. These points
were empirically determined using a
sample of devices.
Selecting RD
Resistor RD can be used to limit the
available drive current to the external
PNP transistor. In order to select RD,
Linear Technology Magazine • November 1997
the user should first choose the value
of drive current that will give the
required value of output current and
dropout voltage. For circuits using a
D45H11 as a pass transistor, this can
be done using Table 1. For circuits
using transistors other than the
D45H11, the user must characterize
the transistor to determine the drivecurrent requirements for the specified
output current and dropout voltage.
In general, it is recommended that
the user choose the lowest value of
drive current that will satisfy the output current requirements. This will
minimize the stress on circuit components during overload conditions.
The value of RD can be calculated
with the following formula:
RD = ( VIN – VBE – VDRIVE )/(IDRIVE + IRB)
where:
VIN = the minimum input voltage to
the circuit
VBE = the maximum emitter/base
voltage of the PNP pass transistor
IDRIVE = the minimum PNP base
current required
IRB = the current through RB
= VBE/RB
VDRIVE = the Drive pin saturation
voltage when the drive pin
current equals (IDRIVE + IRB)
Overcurrent Latch-Off
In addition to limiting the base-drive
current, RD is included in the circuit
for the overcurrent protection latchoff function. There is a minimum
value for this resistance. It is calculated by the formula above, with the
drive current IDRIVE set to the guaranteed minimum available drive current
(= 250mA) from LT1573. Under some
conditions, RD can be replaced with a
short. This is possible in circuits where
an overload is unlikely and the input
voltage and drive requirements are
low. If RD is not included in the circuit,
the regulator is protected against overcurrent conditions only by the
thermal-shutdown function. After the
value of RD is determined, a certain
amount of base-drive current is
available to the external PNP. An
overcurrent or output short-circuit
condition will demand a base-drive
current greater than the LT1573 can
supply. The internal drive transistor
will saturate. This overcurrent condition will trigger a time-out latch to
turn off the regulator system. The
time-out period is determined by an
external capacitor connected between
the LATCH and GND pins. The timeout period is equal to the time it takes
for the capacitor to charge from 0V to
the latch threshold. If the overcurrent
or output short condition persists
longer than the time-out period, the
regulator will be shut down. Otherwise, the regulator will function
normally. The latch can be reset by
recycling input power, by grounding
the latch pin or by putting the device
into shutdown.
Shutdown Function
The regulator can be shut down and
the overcurrent latch can be reset at
the same time by pulling the SHDN
pin voltage higher than the shutdown
threshold (about 1.3V). The regulator
will restart itself if the SHDN pin is
pulled below the shutdown threshold. The shutdown pin voltage can be
higher than the input voltage. When
the shutdown pin voltage is higher
than 2V, the shutdown pin current
increases and is limited by a 20k
resistor.
Compensation
Figure 1 shows a microprocessor
power supply based on the LT1573
and the D45H11 PNP transistor, using twenty-four 1µF surface mount
ceramic capacitors in parallel with
one 220µF surface mount tantalum
capacitor at the output. To improve
the transient response to regulator
output-load variation of this circuit, a
capacitor in series with a resistor can
be inserted between the OUTPUT and
COMP pins; a 100pF capacitor in
series with a 1k resistor is
recommended. In theory, the output
capacitor forms the a pole in the
regulator system. An internal compensation capacitor forms another
pole. The external compensation cacontinued on page 24
17
DESIGN FEATURES
LT1579 Battery-Backup Regulator
Provides Uninterruptible Power
by Todd Owen
Introduction
lation. The LT1579 is capable of providing 300mA from either input at a
dropout voltage of 0.4V. Total quiescent current is 50µA: 45µA from the
primary input source, 2µA from the
secondary input source, and an additional 3µA from the higher voltage of
the two.
A block diagram of the LT1579 is
shown in Figure 1. A single error
amplifier controls both output stages
so regulation remains tight regardless of which input is providing power.
Threshold levels for the error amplifier and low-battery detectors are set
by the internal 1.5V reference. Output voltage is set by an internal
resistor divider for fixed voltage parts
and by an external divider for adjustable parts. Internal bias circuitry
powers the reference, error amplifier,
VIN1
VIN2
VOUT
DROPOUT
DETECT
SHDN
INTERNAL
RESISTOR DIVIDER
FOR FIXED-VOLTAGE
DEVICES ONLY
BIASCOMP
OUTPUT
DRIVER
CONTROL
1.5V REFERENCE
LBI1
E/A
WARNING
FLAGS
+
ADJ
+
18
BIAS CURRENT
CONTROL
SS
Smart Regulator
Makes the Grade
Designed for a multitude of applications, the LT1579 is a dual input,
single output, low dropout regulator
that provides an uninterruptible output voltage from two independent
input voltage sources on a priority
basis. All power supplied to the load
is drawn from the primary input (VIN1)
until the device senses that the primary source is failing. At this point,
the LT1579 smoothly switches from
the primary input to the secondary
input (VIN2) to maintain output regu-
output driver controls, logic flags and
low-battery comparators.
The LT1579 is designed to maintain regulation even if one of the
inputs is instantaneously removed. If
the primary input is supplying load
current, removal and insertion of the
secondary input creates no noticeable transient at the output. In this
case, the LT1579 continues to supply
current from the primary; no switching is required. If the primary input
source is unplugged while it is supplying load current, the LT1579 must
quickly switch to the secondary
source. In this case, the LT1579 sees
the input capacitor as a rapidly discharging battery. If it discharges too
quickly, the LT1579 does not have
sufficient time to switch over without
a large transient occurring at the
–
Many products, such as critical data
acquisition systems and process controllers, need to operate continuously
even when primary power is interrupted. Primary power for these
systems may come from a wall adapter,
a battery-charging unit powered from
the line or a removable battery. For a
line-powered circuit, a backup battery may provide reserve power until
line power is restored. In the case of a
battery-charging unit, the battery
automatically provides power during
a line outage. With removable batteries, a secondary backup battery is
needed to maintain power when the
main battery is discharged or removed.
These circuits have several common requirements, regardless of the
power source. Most important is the
need for the regulated voltage to be
provided to critical circuitry at all
times. The switch from one input
source to another should be seamless, without significant change in
the output. The input source that
provides power to the circuit first can
also be important (for example, the
main battery should provide power
before the backup battery, regardless
of which has the higher terminal voltage). Additionally, some form of power
monitoring circuitry may be needed
to help in power management.
BACKUP
DROPOUT
LB01
LB1
–
LBI2
+
LB02
LB2
–
1579_01.eps
Figure 1. LT1579 block diagram
Linear Technology Magazine • November 1997
DESIGN FEATURES
INPUT #1: 6V
IN1
+
5V
1µF
6V
R1
2.7M
R2
1M
6V
1µF
REPLACE
INPUT
#1
R3
2.7M
R4
1M
100k
100k
+
5V
300mA
4.7µF
LBO1
LBO2
DROPOUT
SS
LBI2
REMOVE
INPUT
#1
100k
BACKUP
IN2
+
100k
LT1579-5
LBI1
OUTPUT
50mV/DIV
OUT
SHDN
BIASCOMP
GND
TO POWER MANAGEMENT
0.01µF
1579_03.eps
Figure 2. Output deviation during removal
and replacement of primary input source
output. The input capacitor must be
large enough to supply load current
during the transition from primary to
secondary input. Plugging the primary input back in creates a smaller
transient on the output because both
inputs are present to supply current
during the transition. Figure 2 shows
a typical output transient using 10µF
input and output capacitors and a
100mA load. Peak output deviation is
less than 50mV. Proportionally larger
values for input and output capacitors are needed to limit peak deviations
on the output when delivering larger
load currents.
The LT1579 incorporates two
independent low-battery comparators
and two status flags that provide
information to power management circuitry. The status flags show which
input is supplying power to the load
and provide an early warning against
the loss of output regulation. A Secondary Select pin can be used to force
the switchover from the primary input
to the secondary. This active low logic
pin reduces power draw from the primary input to only 2µA. This provides
excellent protection for batteries that
are sensitive to deep-discharge conditions, such as Li-Ion cells. The part
can be put into a low power shutdown
state where all bias currents and device
functions are turned off and all logic
flags are high impedance. In shutdown, the quiescent current drops to
a total of only 7µA: 2µ A from the
primary input, 2µA from the secondary input and 3µ A from the higher of
the two input voltages.
Internal protection circuitry in the
LT1579 guards against a number of
fault conditions. Protection circuitry
Linear Technology Magazine • November 1997
Figure 3. LT1579 basic application
includes thermal-overload protection,
reverse-battery protection, inputto-input current protection, reversecurrent protection and overcurrent
protection. If maximum junction temperature is exceeded, the LT1579 will
shut off to prevent internal damage. If
either input is reversed, reverse current flow will be limited to less than
1mA. Inputs are isolated from one
another. No current will flow between
inputs, regardless of their relative
voltages. If the output is held high
while the inputs are grounded, reverse
currents are limited to 7µA. Currentlimit protection is designed to protect
the device if the output is shorted to
ground. With the output shorted to
ground, current will be drawn from
the primary input until it is discharged. No current is drawn from
the secondary input until the primary
input is discharged.
all logic outputs can sink 5mA at a
maximum output voltage of 1.2V.
Figure 4 is the timing diagram for
the basic circuit. No time scale is
shown for the timing diagram because actual discharge rates are a
function of the load current and the
type of batteries used. The timing
diagram is meant as a tool to help in
understanding the LT1579’s basic
operation.
Five milestones are noted on the
timing diagram. Time A is where the
primary input voltage drops enough
to trip the low-battery detector, LB1.
A B
D
E
VIN1
5V
6V
VIN2
Circuit Examples
The basic application of the LT1579 is
shown in Figure 3. It uses two independent voltage sources for the inputs.
These voltage sources may be batteries, wall adapters or any other DC
source. The low-battery comparators
are configured to give a low output if
either input voltage drops below 5.5V.
The trip points can be adjusted by
changing the values of the divider
resistors (R1 and R2 for LB1, R3 and
R4 for LB2). All logic outputs (LBO1,
LBO2, BACKUP and DROPOUT) are
open-collector outputs that require
an external pull-up resistor. They are
capable of sinking 20µA at a maximum output voltage of 0.32V, which
is useful for driving both CMOS and
TTL logic families. For driving LED’s,
C
6V
5V
VOUT 5V
4.8V
100mA
IIN1
0
100mA
IIN2
0
LB01
1
0
1
BACKUP
LB02
DROPOUT
0
1
0
1
0
1579_04.eps
Figure 4. Basic application timing diagram
19
DESIGN FEATURES
OUT
IN1
C1
1µF
IN1
R2
2.7M
R1
1M
LBI1
D2
D1
R3
1M
BACKUP
LBO1
R4
10M
C3
4.7µF
R10
1M
DROPOUT
VOUT
5V/300mA
MAIN GOOD
NC
SS
IN2
D3
IN2
C2
1µF
R5
1M
R6
2.7M
LT1579-5
LBI2
R7
1M
R8
330k
BIASCOMP
C4
0.01µF
LBO2
C5
0.1µF
D4
5.1V
1N751A
VCC
1/4
74C02
1/4
74C02
SHDN
GND
GND
1/4
74C02
RESET
R9
1.5M
1579_05.eps
D1–D3 = 1N4148
Figure 5. Added SR latch shuts the LT1579 off when both low-battery detectors are tripped.
The trip threshold for LB1 is set at
5.5V, slightly above the dropout voltage of the primary input. At time B,
the BACKUP flag goes low, signaling
the beginning of the transition from
the primary source to the secondary
source. Between times B and C, the
input current makes a smooth transition from VIN1 to VIN2. By time C, the
primary battery has exhausted most
of its useful charge. The primary input
will still deliver a small amount of
current to the load, diminishing as
the primary input voltage drops. By
time D, the secondary battery has
dropped to a low enough voltage to
trip the second low-battery detector,
LB2. The trip threshold for LB2 is also
set at 5.5V, slightly above where the
secondary input reaches dropout. At
time E, both inputs are low enough to
cause the LT1579 to enter dropout,
with the DROPOUT flag signaling the
impending loss of output regulation.
Some interesting things can be
noted on the timing diagram. The
amount of current available from a
given input is determined by the
input/output voltage differential. As
the primary voltage drops, the amount
of current drawn from the input also
drops, slowing discharge of the bat20
tery. Dropout-detection circuitry will
maintain the maximum current draw
from the input for the given input/
output voltage differential, based on
the impedance of the pass transistor.
In the case shown, this causes the
current drawn from the primary to
approach zero, although it never
reaches that point. Note that the primary begins to supply significant
current again when the secondary
input drops low enough to cause a
loss in output regulation. This occurs
because the input/output voltage differential of the primary input increases
as the output voltage drops. The
LT1579 will automatically maximize
the power drawn from the inputs to
maintain the highest possible output
voltage.
A final circuit example is shown in
Figure 5. This circuit has a few notable
changes from the basic application.
First, the Secondary Select pin is connected directly to LBO1. When the
primary input voltage drops below
the threshold level for LB1, the comparator output will pull the Secondary
Select pin low. This forces the device
to switch completely over to the secondary input, limiting the discharge
voltage of the cells. Second, the logic
gates used form an SR latch. When
both batteries are below the threshold
level for their respective comparators, the latch will be set, forcing the
part into shutdown. The latch is reset
by pulling up on the RESET node,
allowing the part to come out of
shutdown.
The series resistance of a battery
can cause its terminal voltage to rise
as its current decreases. This effect
can reset the low-battery detector and
cause the LT1579 to oscillate between
the primary and secondary inputs. To
combat this, the low-battery comparators have up to 18mV of built-in
hysteresis at the input to the comparator (LBI1, LBI2). The hysteresis
is determined by the amount of load
current on the comparator output. At
no load, the comparator hysteresis is
zero, increasing to a maximum of
18mV for load currents above 20µA.
For the pull-up resistor shown, load
current on the output of the comparator is 5µA, so hysteresis will be
5mV. With the values shown for resistor divider R2/R3, this translates to
19mV of hysteresis at the primary
input of the LT1579. Additional hysteresis can be added by connecting
continued on page 24
Linear Technology Magazine • November 1997
DESIGN FEATURES
High Efficiency Distributed
Power Converter Features
Synchronous Rectification
Introduction
Introducing the LT1339
The advent of the switching regulator
has greatly reduced the size, weight
and volume of power conversion circuitry, while improving both the speed
of response and efficiency. With the
output voltage requirements going
ever lower and currents ever higher,
close scrutiny is applied to the loss
mechanisms of the power converter.
The loss mechanisms are caused by
the circuit resistances (input capacitor ESR, power switch on-resistance,
DC and AC resistance of the inductor,
resistance of any current-sense elements, resistance in the output diode
and ESR in the output capacitors)
multiplied by the squares of their
respective currents, and by fixed forward-voltage losses in the output
diode, multiplied by the diode current. As output current goes up, the
first thing to do is minimize all losses
caused by resistance (because the
power is proportional to I2R). It is easy
to minimize resistance because we
have available very low ESR capacitors, low on-resistance MOSFETs
and low series-resistance inductors.
We have controllers that place a very
small voltage across the current sense
resistors. Because we do such a good
job of dropping the resistive loss
mechanisms, the output diode forward-voltage drop becomes the
greatest loss mechanism. This is how
the mandate for synchronous rectification comes about. Synchronous
rectification is achieved by replacing
the output diode with a low on-resistance switch. With synchronous
rectification, efficiencies are higher,
and, more importantly, power dissipated in the switching power supply
is lower, often eliminating the need
for heat sinks and/or fans.
The LT1339 is the buck/boost converter that needs no steroids. As a
full-featured switching controller, the
LT1339 incorporates the features
needed for system-level solutions. The
unfortunate lack of such features in
most PWM converters forces designers to grope for handfuls of jellybean
components. The LT1339 has an
innovative slope-compensation function that allows the circuit designer
freedom in controlling both the slope
and offset of the slope-compensation
ramp. Additionally, the LT1339 has
an average current limit loop that
yields a constant output current limit,
regardless of input and/or output
voltage. The LT1339’s RUN pin is
actually the input to a precision comparator, giving the designer freedom
to select an undervoltage lockout point
and hysteresis appropriate for the
design. The SYNC and SS (soft-start)
pins allow simple solutions to system-level design considerations. Like
all Linear Technology controllers, the
LT1339 has anti-shoot-through circuitry that ensures the robustness
that is demanded in real-world applications for medium and high power
conversion.
10V TO
18V
+
CINPUT
1000µF
16V ×2
OS-CON
13
2
C1
1µF
RT
15k
CT 1500pF
CCOMP
2200pF
4
3
5
CAVG 2200pF
RCOMP
4.7k
7
10
C2
0.1µF
D1 1N914
*
R1
100k
17
TG
5VREF
TS
SLOPE
BG
The circuit shown in Figure 1 is limited to 20V because of the maximum
rating (Abs Max) of the LT1339 VIN
pin. The input voltage can be extended
above 20V by inserting a 10V Zener
diode where the asterisk (*) is shown
in Figure 1. This will extend the input
voltage of Figure 1’s circuit up to 30V
(the Abs Max rating of the MOSFETs).
11
VC
SENSE –
12
FB
SYNC SGND PGND
15
Q1
IRL3803
L1
10µH
50A
RSENSE
0.002Ω
RSENSE
ILIMIT
0.01Ω
0.005Ω
0.002Ω
10A
20A
50A
+
16
SENSE +
8
Higher Input Voltages
18
LT1339
1
Figure 1 details a typical low voltage
buck converter. This circuit has a VIN
range of 10V to 18V with configurable
output current and voltage. This
simple circuit delivers 250W of load
power into a 5V load while maintaining efficiencies in the mid-nineties.
19
IAVG
VREF
Distributed Power
D2
1N5817
BOOST
RUN
CT
For input voltages ranging from
12V to 48V and output voltages ranging from 1.3V to 36V, the LT1339 is a
simple, robust solution to your powerconversion problems. The LT1339 is
ideal for power levels ranging from
tens of watts to tens of kilowatts. The
LT1339 is straightforward and
remarkably easy to use. This is one
power converter that’s not afraid of
20A, 50A or even 150A of load current.
C3
1µF
20
12VIN
by Dale Eagar
Q2–Q5
IRL3103D2
×4
D3
1N5817
RFB
3K
1.66K
1.25K
450Ω
40Ω
9
RREF
1k
5V
COUTPUT 50A
2200µF
6.3V
OS-CON
RFB
3k
1339_01.EPS
VOUT
5V
3.3V
2.8V
1.8V
1.3V
*SEE TEXT
Figure 1. 10V–18V in, 5V/50A out buck converter
Linear Technology Magazine • November 1997
21
DESIGN FEATURES
12V
(SUPPLIED SEPARATELY)
D1
1N914
+
C5
47µF
13
2
C1
1µF
RT
15k
4
3
Q7
FMMT619
C3
1µF
RUN
TG
5VREF
TS
SLOPE
BG
CAVG 2200pF
7
12V
C4
1µF
Q10
FMMT720
IAVG
11
SENSE +
VC
SENSE –
C3 10
VREF
0.1µF
5V
50A
COUT
2200µF
6.3V
OS-CON
×4
Q3–Q6
IRFZ44
×4
D3 1N914
12
RFB
9
FB
SYNC SGND PGND
8
10A
20A
50A
+
R1
10k
As the input voltage approaches 30V,
the bottom MOSFETs will begin to
exhibit “phantom turn-on.” This
phenomenon is driven by the instantaneous voltage step on the drain, the
ratio of CMILLER to CINPUT, and yields
localized gate voltages above VT, the
threshold voltage of the bottom MOSFET. To defeat the physicists, we add
3.3V of negative offset to the bottom
gate drive, effectively making the
threshold of the bottom MOSFETs
3.3V harder to reach (see Figure 2).
This offset is provided by the 3.3V
Zener, 1µF capacitor, 10k resistor
and the 1N914 diode preceding the
gate of the bottom MOSFETs.
ILIMIT
D2 3.3V
16
CT
1
Q9
FMMT619
18
LT1339
5
Q1–Q2
IRFZ44
×2
D4–D14
3A ×10
BOOST
19
RSENSE
CINPUT
0.01Ω
1500µF
0.005Ω
63V
0.002Ω
×4
L1
10µH 50A RSENSE
0.002Ω
+
Q8
FMMT720
20
12VIN
CT
1500pF
RCOMP
4.7k
C2
2200pF
48V
R1
100k
17
Blame it
on the Physicists
3k
RREF
1k
15
RFB
18.2K
8.66K
3K
1.66k
1.25k
VOUT
24V
12V
5V
3.3V
2.8V
1339_02.EPS
Figure 2. 48V in, 5V/50A out, high power buck converter
VIN
15V–25V
CIN
1000µF 35V
×2
+
R13
2k
Q7 FZT849
C10
R1
100k 0.1µF
C1
0.1µF
2
R2
1k
RT
47k
C11 +
22µF
35V
11
SENSE+
5VREF
4
SLOPE
BOOST
R3
560Ω
6
CAVG
2.2nF
5
9
14
10
TG
TS
CT
C3
1µF
20
C3
3.3µF
18
16
FB
R9
10Ω
C6
1µF
Q1, Q2 = SUD50N03
L1 = 15 TURNS AWG20 77130-A7
* T1 = POWER TRANSFORMER
** T2 = GATE-DRIVE TRANSFORMER
(SEE FIGURE 4 FOR DETAILS)
SGND
8
COUT
220µF 10V
OS-CON
Q5
8
7
4
6
2
C5
1µF
9
R5
PGND
15
U3
CNY17-3
D5
1N914
R8
1k
D6 1N914
C7 1µF
C9 0.1µF
R7 1k
3
C4
220pF
RFB
2.49k
V+
R6 560Ω
R11
10Ω
R12 1k
10
Si4539DY
Q6
PHASE
SYNC
1
+
Q2
12V
T2**
VC
Q4 Si4450
19
R14 10Ω
BG
VOUT
5V/6A
R10
10Ω
Q3 Si4450
SS
VREF
L1 7µH
D4
MURS
120
3.3Ω
R4
33k
D3
MURS120
T1*
D2
IN4148
12
U1
LT1339
IAVG
SECONDARY
GROUND
12V
Q1
13
17
RUN 12VIN
SENSE–
3
CT
230pF
RSENSE
0.02Ω
1/2W
D1
12V
ISOLATION
BARRIER
PRIMARY
GROUND
1
COLL
8
U2
REF
LT1431
GND-S
5
GND-F
6
RREF
2.49k
Figure 3. Galvanically isolated synchronous forward converter (See Figure 4 for details of T1 and T2.)
22
Linear Technology Magazine • November 1997
DESIGN FEATURES
T2: COILTRONIX VP1-1400 (500V ISOLATION)
T1: PHILIPS EFD20-3F3 CORE
Lp = 93µH, Al = 1150nH/T2 (NO GAP)
10
4
2MIL
POLYESTER
FILM
SECONDARY, 9 TURNS TRIFILAR 26AWG
PRIMARY, 9 TURNS TRIFILAR 26AWG
1
7
5
11
2
8
1500VDC ISOLATION
TUCK TAPE ENDS
6
3
12
9
1339 04 .eps
Figure 4. Transformer details of Figure 3’s circuit
The Synchronous
Forward Converter
Figure 3 details a Galvanically isolated LT1339 synchronous forward
converter. Operating at its rated load
of 6V at 5A, this circuit achieves 87%
efficiency with a 15V input and 85%
efficiency with a 24V input. Figure 4
shows details of the transformers used
in Figure 3’s circuit.
The forward converter can be thought
of as a buck converter with a
transformer ratio. The transformer
performs one or more of the following
functions:
1. Increases the duty factor in
converters with high transformation ratios, improving efficiency
2. Provides Galvanic isolation from
input to output
3. Allows the converter to operate in
the buck-boost mode, where the
input voltage can be either
higher or lower than the output
voltage
The Synchronous
Boost Converter
VIN
5V/60A
The LT1339, like the many other
members of the Linear Technology
power conversion family, is full featured and well suited to the power
conversion needs found in the real
world. This robust workhorse is finding itself designed into countless
applications, including power distribution, telecom, automotive and even
military applications.
The LT1339 becomes a synchronous
boost controller when the PHASE pin
is grounded. Figure 5 details a 250W
boost converter that outputs 28V at
9A from a 5V supply.
Authors can be contacted
at (408) 432-1900
Q1–Q2
IRF3205 ×2
Q7 FMMT720
RSENSE
0.002Ω
Conclusion
Q8 FMMT619
L1
40µH
+
C3
1µF
19
CINPUT +
220µF 6.3V
×4
D1
1N914
18
+
+
20
17
BOOST
12VIN
TG
RUN
R1
100k
13
C7
47µF
16V
RFB
27k
C4
1µF
U1
LT1339
16
Q3–Q6
IRF3205 ×4
BG
FB
5VREF
SLOPE
9
C5
10pF
R3 100Ω
4
RT
10k
12
SENSE+
CT
11
SENSE–
VC
10
14
C6
0.1µF
VREF
PHASE
IAVG
SS
SYNC PGND SGND
15
1
8
RREF
1.2k
2
Q10
FMMT720
R2 100Ω
COUT
2200µF
35V ×6
TS
12V
Q9
FMMT619
VOUT
28V/8.5A
12V (SUPPLIED SEPARATELY)
CT
2200pF
C1
1µF
3
7
RCOMP
5
7.5k
6
+
CSS
10µF
CAVG
2.2nF
CCOMP
1.5nF
L1 = 12T 4× AWG12 ON 77437-A7
Figure 5. This 5V to 28V synchronous boost converter limits input current at 60A (DC).
Linear Technology Magazine • November 1997
23
DESIGN FEATURES
Typical Applications
and Experimental
Transient Response
LT1573, continued from page 17
pacitor and resistor form a zero that
adds phase margin to the regulator
system to prevent high frequency oscillation. The LT1573 has an internal
pole at approximately 5kHz. An external compensation zero between
10kHz and 100kHz is usually required
to stabilize the regulator. The zero
frequency is primarily determined by
the compensation capacitor and can
be roughly calculated by the following
equation:
fZERO = (40kHz)
30pF
10pF ≤ CCOMP ≤ 100pF
CCOMP (pF)
A compensation resistor between
1k and 10k is suggested. A compensation resistor of 5k works for most
cases. In some cases, a larger value
compensation resistor in the suggested range is needed to stop
oscillation above 1MHz. In some cases,
the output capacitor may have enough
equivalent series resistance (ESR) to
generate the required zero, and the
external compensation zero may not
be needed.
Output Capacitor
The LT1573 is designed to be used
with an external PNP transistor with
a high gain-bandwidth product, fT, to
make a regulator with very fast transient response; this can minimize the
size of the output capacitor. For a
regulator consisting of an LT1573
and a D45H11, a single 10µF surface
50mV/DIV
2.5A/DIV
10µs/DIV
Figure 2. Transient response for a 0.2A
to 5A step
mount ceramic capacitor at the output is enough to allow the regulator to
handle varying output loads of up to
5A in a few hundred nanoseconds
and to remain stable with a 30pF
external compensation capacitor in
series with a 7.5k resistor between
the OUTPUT and COMPENSATION
pins. If tighter voltage regulation is
needed, more capacitance can be
added to the regulator output. If more
capacitance is added to the output,
the bandwidth of the regulator will be
lowered. A larger compensation
capacitor may be needed to lower the
frequency of the compensation zero
to avoid high frequency oscillation.
Equal-value output capacitors with
different ESR can have different transient responses. High frequency
response will be strongly affected by
parasitics in the output capacitors
and board layout. Some experimentation with the external compensation
will be required for optimum results.
circuits from any of a number of different input sources. It will provide
up to 300mA of output current at a
dropout voltage of 0.4V. Should the
primary input fail, the device switches
seamlessly to the secondary input,
maintaining output regulation. A
single error amplifier controls both
output stages so regulation remains
tight regardless of which input is providing power. The LT1579 can handle
instantaneous removal of either one
of its inputs without losing regulation. System power management is
Conclusion
aided by two status flags, which proThe LT1579 can provide a continu- vide information about which input is
ous regulated output voltage to critical providing power and signal the loss of
LT1579, continued from page 20
D1 and R4. The values shown will give
an additional 200mV of hysteresis.
When LBO1 and LBO2 are high
impedance and either input is greater
than 6.5V, the logic-flag voltages can
be above the maximum voltage rating. Internal clamps on the logic flags
limit the output voltage to approximately 6.5V and the pull-up resistor
values shown will limit the current
into the logic flags to less than the
maximum current rating.
24
A voltage regulator made with an
LT1573 can be used as a microprocessor power source, post regulator
for switching power supplies, high
efficiency, high accuracy linear regulator or ultralow dropout regulator. A
typical application circuit for a microprocessor power supply is shown in
Figure 1. Figure 2 shows the output
voltage transient of the circuit when
the output load varies between 0.2A
and 5A. About 80mV of transient
deviation from 3.3V output (2.5%)
can be observed.
Conclusion
The LT1573 provides a low cost solution to high current, fast transient,
low dropout voltage regulator applications such as fast microprocessor
power supplies. As a low dropout
regulator, the LT1573 can provide
load currents of up to 5A with dropout voltages as low as 0.35V. Because
of its fast transient capability, the
LT1573 cuts down the regulator cost
by requiring much less bulk storage
capacitance for tight voltage
regulation.
Authors can be contacted
at (408) 432-1900
output regulation. Two independent
low-battery comparators can be used
to monitor input voltages. Also, an
external pin can be used to force the
switch to the secondary input. Total
quiescent current of the LT1579 is
50µA, dropping to a mere 7µA in its
low power shutdown state. Internal
circuitry guards against a number of
fault conditions, including current
limit, thermal limit and reverse voltages, protecting sensitive circuitry and
inputs. Whether the application is
simple or complex, the LT1579 is
truly a “smart” regulator.
Linear Technology Magazine • November 1997
DESIGN IDEAS
Single-Cell Li-Ion Battery Supervisor
by Albert Lee
Recently introduced precision
products from Linear Technology
allow designers to implement high
precision applications at supermicropower levels. Among these
devices are the LT1496 quad precision input/rail-to-rail output op amp
and the LT1634 precision shunt voltage reference, which operate at only
1.5µA and 10µA, respectively. Even
at such low power levels, precision
performance is not compromised. The
LT1496 features 475µV maximum
input offset voltage and 1nA maximum input bias current. The LT1634
achieves 0.05% initial accuracy and
25ppm/°C maximum temperature
drift.
Figure 1 shows a single-cell Li-Ion
battery supervisory circuit. The building blocks of this circuit are the
LT1496 precision op amp and LT1634
voltage reference. The useful region
of operation of a single-cell Li-Ion
battery is between 4.2V and 3V. The
cell voltage drops fairly quickly below
3V. System operation below this voltage can be erratic. Although Li-Ion
battery use is becoming widespread,
it is costly to damage the battery. The
supervisory circuit protects the battery from overcharging and/or
overdraining and prevents the battery voltage from falling out of its
operating region. For instance, the
LT1496 operates down to 2.2V,
ensuring that circuit operation is
maintained when the battery voltage
falls below 3V.
The Li-Ion battery is monitored via
a voltage divider off the battery voltage (node A). The divided voltage is fed
into the positive inputs of comparators A2 and A3 and compared to the
threshold voltages of 1.75V and 1.25V,
respectively. These voltages are
selected so that the minimum battery
charge voltage is 3V and the maximum is 4.2V. The LT1634 1.25V
reference is buffered by op amp A1.
The constant 1.25V across R2 creates
a 1µA constant current, so that the
output of A1 is amplified to 1.75V.
This output drives RS to provide constant bias current for the LT1634.
Depending on the battery voltage,
the circuit is in one of the three states,
as shown in Table 1.
The voltage at node A is compared
to the two threshold voltages to determine the state of the circuit. For
instance, when node A reaches or
exceeds 1.75V (battery voltage reaches
4.2V), the outputs of A2 and A3 will
swing to the positive rail, terminating
the charger and connecting the load
to the battery. When node A falls
between 1.25V and 1.75V (battery
voltage between 3V and 4.2V), the
output of A2 swings low, turning the
charger on, while the output of A3
stays high, leaving the load connected.
When node A falls below 1.25V (battery voltage less than 3.0V), the output
of A2 stays low, keeping the charger
OFF
CHARGER
SW*
VBAT
DESIGN IDEAS
Single-Cell Li-Ion Battery
Supervisor ............................... 25
Albert Lee
800mA Li-Ion Battery Charger
Occupies Less Volume than
Two Stacked Quarters ............. 27
1.75V
10µA
+
RSW
1M
5%
R3
1.75M
0.1%
BATTERY
+
A
A1
1/4 LT1496
Single-Supply
Random Code Generator .......... 29
R1
500k
0.1%
–
Richard Markell
LT1634
1.250V
1.25V
George Feliz and Adolfo Garcia
100V, 2A Constant-Voltage/
Constant-Current Bench Supply
................................................31
RH1
10M
5%
D1
Fran Hoffart
A Low Distortion, Low Power,
Single-Pair HDSL Driver Using
the LT1497 ............................. 30
–
A2
1/4 LT1496
RS
175k
5%
TO LOAD
1µA
R2
1.25M
0.1%
VBAT
+
RH2
10M
5%
D2
–
A3
1/4 LT1496
A4
1/4 LT1496
–
+
R4
1.25M
0.1%
Mitchell Lee
Switcher Generates Two Bias
Voltages without Transformer
................................................32
Jeff Witt
Linear Technology Magazine • November 1997
D1, D2 = 1N458
R1–R4 = CAR6 SERIES IRC (512) 992-7900
*TP0610L for 50mA LOAD
Figure 1. Single-cell Li-Ion battery supervisory circuit
25
DESIGN IDEAS
Node A
Output A2 Output A3 Output A4
1
< 3V
< 1.25V
Low
Low
High
2
3V< V
< 4.2V
1.25V< V
< 1.75V
Low
High
Low
3
> 4.2V
> 1.75V
High
High
Low
on. The output of A3 will also swing
low, which, in turn, will cause the
output of A4 to go high, turning off
FET SW that disconnects the load
from the battery.
If node A were to bounce around at
either threshold voltage, the circuit
would bounce between states. To
avoid this problem, hysteresis is
added via the resistor and diode networks connected between the outputs
of A2 and A3 and their positive inputs. Figure 2 shows the behavior of
VBAT vs node A entering the trip points
with hysteresis. When VBAT rises to
4.2V (node A increases to 1.75V), op
amp A2’s output will switch from low
to high, causing current to flow
through RH1. The additional current
will raise node A by an amount
∆VAHYS1, which will clearly put the
circuit in state 3. The circuit will not
exit state 3 until VBAT falls to ∆VHYS1
(310mV for the circuit shown) below
4.2V, which will cause node A to fall
back to the upper trip point of 1.75V
(point 1 of Figure 2). Similarly, when
VBAT drops below 3V (node A falls
below 1.25V), op amp A3’s output will
switch low, causing current to conduct through RH2. This will drag node
A an amount ∆VAHYS2 below 1.25V,
which will put the circuit in state 1.
The circuit will not exit state 1 until
the battery voltage is charged to an
amount ∆VHYS2 (149mV for circuit
shown) above 3V (point 2). This will
bring node A back up to the lower trip
Status
Load off,
charge
state
Load on,
charge
state
Load on,
charge
terminated
point, 1.25V, bringing the circuit out
of state 1. The amount of hysteresis
desired can be calculated using the
following formulas:
High Trip Point:
VBAT = (R3 • (VOHMIN + VBE +
1.75V)/RH1 + 1.75V • (R3/R4) +
1.75V)/(1 + R3/RH1)
∆VHYS1 = 4.2V – VBAT
Low Trip Point:
IRH2 = (1.25V – VOLMAX – VBE)/RH2
∆VHYS2 = IRH2 • R3
where:
VOHMIN = output voltage swing high
(LT1496)
VOLMAX = output voltage swing low
(LT1496)
VBE = diode voltage of 1N458
Using an automobile analogy, if
the LT1496 op amp is the transmission of the circuit (switching from one
state to the next), the LT1634 voltage
reference is the engine. It not only
generates the threshold voltages, but
also the amount of error that the
circuit will have. How much accuracy
and error you get depends on the car
you drive. Maximum input offset voltage and input bias current for the
LT1496 are 475µV and 1nA, respectively. The LT1634 is a 0.05% initial
accuracy, 25ppm/°C tempco, 10µA
precision shunt reference. Its 1.250V
output voltage will appear at the input
of A3 with an accuracy of 0.088%
∆VAHYS1
1
1.75
∆VHYS1
VBAT
1.25
∆VAHYS2
2
∆VHYS2
State
(initial accuracy + input offset voltage). R1 and R2 being 0.1% resistors,
the worst-case ratio error will be 0.2%.
The worst-case voltage error across
R1 will then be 0.2% or 1mV. This
error on the 1.75V threshold voltage
is 0.057%. Similarly, error at 1.75V
due to worst-case 2nA input bias
current is 0.057%. Total worst-case
error at 1.75V will be 0.202%.
VBAT error contributed by the voltage divider branch will consist of
three terms: resistor matching, op
amp input bias current and input
offset voltage. The amount of error is
different at the two trip points when
VBAT is 3V or 4.2V. Similar calculations as above result in 0.328% when
VBAT = 3V and 0.268% when VBAT =
4.2V. Therefore, total battery voltage
error at either trip points is better
than 0.47%. Since only the ratios of
R1 to R2 and R3 to R4 are critical,
precision matched resistors with ten
times better performance can be used
to reduce the overall error by 33%.
This supervisory circuit demonstrates unparalleled performance
achievable only with Linear
Technology’s supermicropower precision devices. The supervisory circuit
consumes only 20µA. Battery voltage
monitoring and control accuracy is
better than 0.5%.
VA (V)
Table 1. Circuit states
3.0
VBAT (V)
4.2
Figure 2. V BAT vs VA with hysteresis
Authors can be contacted
at (408) 432-1900
26
Linear Technology Magazine • November 1997
DESIGN IDEAS
800mA Li-Ion Battery Charger
Occupies Less Volume than
Two Stacked Quarters
Each new generation of cell phones,
PDAs, portable instruments and other
handheld devices is invariably more
powerful, smaller and, most likely,
thinner than the last. The circuit
shown in Figure 1 is designed to
charge one or two Lithium-Ion cells at
currents up to 800mA, with all components equal to or less than 2.2mm
(0.086 inches) tall. Using 0.031 inch
PC board material, the total circuit
thickness for this charger is 3.4mm
(0.136in) or the thickness of two
quarters. The complete 800mA
constant-current/constant-voltage
charger, including the PC board,
occupies less volume than two quarters. This compact, low profile
construction is ideal for cell phones
or other applications where circuit
height is restricted.
LT1510-5CGN High
Efficiency 500kHz Switch
Mode Battery Charger IC
The charger consists of an LT1510
constant-voltage/constant-current
PWM IC, which includes an onboard
1.5A switch. The LT1510 is available
in either 200kHz or 500kHz versions;
the higher frequency version allows
lower value, smaller-sized inductors
to be used. An internal 0.5% reference allows precision battery-voltage
programming and a current programming pin allows a single resistor, PWM
signal or a programming current from
a DAC to control the charging current. Also included are undervoltage
lockout and a low quiescent current
sleep mode that is activated when
input power is removed.
The internal NPN switch achieves
low saturation voltage by using a
bootstrapped base-drive technique.
A low boost voltage is bootstrapped to
the switch pin to generate a base-
drive voltage that is greater than the
input voltage. The ideal boost voltage
is from 3V to 5V. If a single lithium cell
is being charged, the battery voltage
can be used for the boost voltage, but
for two or more cells, a separate 3V
source is recommended for maximum
efficiency.
Fused-Lead Package Offers
Lower Thermal Resistance
The LT1510-5 is available in a specially constructed 16-lead plastic
SSOP package that has the die-attach
paddle connected (fused) directly to
the four corner leads and fits in the
same area as an SO-8 package. This
low profile fused-lead package provides a lower thermal resistance by
conducting much of the heat generated by the die through the copper
leads to the PC board copper. To take
advantage of the improved thermal
properties of this fused-lead package,
it is important to provide as much PC
board copper around the package
leads as is practical. Back-side copper and internal copper layers
by Fran Hoffart
interconnected by feed-through vias
all contribute to the overall effectiveness of the PC board used as a heat
sink. Other heat-producing surface
mount components, such as Schottky
diodes and the inductor, also rely on
the PC board copper to conduct heat
away from the components.
Although the actual component
area required for the charger can be
as low as 0.72in2, it is very likely that
additional copper area may be needed
to maintain component temperatures
at safe levels. The amount of copper
area required depends on factors such
as maximum ambient temperature,
airflow, input voltage and charging
current. The LT1510-5 can be
programmed for higher charging currents, but package and PC board
thermal constraints will determine
the maximum output current.
Selecting Thin Components
Some components are inherently thin
(resistors, ceramic capacitors, diodes
and transistors) but others are more
difficult to make thin (inductors;
VIN = 12V–20V
DI
MBRM
140T3
D2
MBRM140T3
1
2
C2
0.22µF
L1
TP3-100
10µH
3
4
5
6
D3
MMBD914LT1
7
8
(
VBAT = 2.465V 1 +
R5 + R6
R4
GND
GND
SW
VCC
BOOST
VCC
GND
OVP
PROG
LT1510-5
NC
VC
NC
SENSE
BAT
GND
GND
16
C1
15
14
10µF
13
12
11
10
9
IBAT = 2000
R2
300
R3
1k
2.465V
R1
)
R1
6.19k
1%
C3
1µF
C4
0.1µF
IBAT
)
+
TO VIN
R4
4.99k
0.5%
(
Q1
2N7002
R5
R6
11.0k
0.5%
1.02k
0.5%
+ C5
VBAT = 8.4V
Li-ION
BATTERY
(2 CELLS)
22µF
IBAT = 800mA
DI 1510 01.eps
Figure 1. Compact, low profile, constant-current/constant-voltage charger for Li-Ion batteries
Linear Technology Magazine • November 1997
27
DESIGN IDEAS
electrolytic and tantalum capacitors).
The inductor used in this design is a
very thin Coiltronics 10µH gapped
toroid made of ferrite material, measuring 2.2mm (0.086in) in thickness.
Other thicknesses, from 1.8mm to
3mm, are also available in the Coiltronics Thin-Pac™ line of inductors.
High dielectric-constant ceramic
chip capacitors are used for C1
through C4. C1 is a Y5U type, selected
because of its small size and high
RMS current capability; X7R type
material is selected for C2 through C4
because of its better temperature and
voltage coefficients. The output
capacitor, C5, is a special polymer
aluminum electrolytic surface mount
capacitor with a package height of
1.8mm. Available from Panasonic, this
solid electrolyte capacitor features
small size, low ESR, high RMS current rating and long life.
Even standard SO packages for
ICs are thicker than the 2.2mm limit
required for this design; fortunately,
the LT1510-5 is available in an SSOP
package (GN package), which has a
maximum height of only 1.75mm
(0.069in.).
D1 and D2 are Motorola PowerMite®
1A, 40V Schottky diodes with a maximum package height of 1.15mm.
Although these diodes are very small,
they are rated for 1A, provided there
is sufficient PC board copper surrounding the leads to provide a heat
path for the power dissipated in the
diodes. D3 is a small-signal silicon
diode in a SOT-23 package with a
maximum package height of 1.1mm.
Also available in a SOT-23 package is
Table 1. Low-profile components used in Figure 1’s circuit
Reference
Designator Quantity
28
C1
1
C2
1
C3
1
C4
1
C5
1
D1, D2
D3
L1
2
1
1
Q1
1
R1
1
R2
1
R3
1
R4
1
R5
1
R6
1
U1
1
Part Number
Description
10µF, 25V, 20%
THCR50E1E106ZT
Y5U Ceramic
0.22µF, 25V, 20%
12063C224MAT1A
X7R Ceramic
1µF, 10V, 20%
0805ZC105MAT
X7R Ceramic
0.1µF, 50V, 20%
08055G104MAT1A
X7R Ceramic
22µF, 12.5V, 20%
Polymer
EEFCD1B220R
Aluminum
Electrolytic
MBRM140T3
1A, 40V Schottky
MMBD914LT1
0.2A 100V Silicon
TP3-100
10µH Thin-Pac
SOT-23
2N7002
N-Channel
MOSFET
6.19k. 1%
Chip Resistor
300Ω, 5%
Chip Resistor
1k, 5%
Chip Resistor
4.99k, 0.5%
Chip Resistor
11.0k, 0.5%
Chip Resistor
1.02k, 0.5%
Chip Resistor
LT1510-5CGN
Battery Charger IC
Vendor
Phone
Marcon
(847) 696-2000
AVX
(207) 282-5111
AVX
(207) 282-5111
AVX
(207) 282-5111
Panasonic (408) 945-5660
Motorola (800) 441-2447
Motorola (800) 441-2447
Coiltronics (561) 241-7876
Zetex
(516) 543-7100
MOSFET Q1, which is used as an
output-divider disconnect switch to
prevent the divider current from discharging the battery when input power
is removed.
Charger Operation
A typical charge profile for a discharged Li-Ion battery is an initial
constant-current charge at 800mA
until the battery voltage rises to the
programmed voltage. It then changes
to a constant-voltage charge, with the
charging current gradually decreasing to near 0mA as the battery
approaches full charge. If complete
charge termination is required, pulling the VC pin low or sinking zero
current from the program pin stops
the charge current. These signals
could be supplied by an external timer
or microprocessor.
When the input power is removed,
the LT1510-5 goes into a low quiescent current (3µA) sleep mode, with
this current coming from the battery.
This low battery drain current allows
the battery to remain connected to
the charger for an extended period of
time without appreciably discharging
the battery. Additional battery-drain
current can result from reverse leakage current in the Schottky catch
diode D1. Many Schottky diodes have
relatively high leakage currents, so
care must be exercised in their
selection.
Refer to the LT1510 data sheet for
complete product specifications and
to design notes DN111 and DN124
and application note AN68 for additional application information.
Conclusion
IRC
(512) 992-7900
IRC
(512) 992-7900
IRC
(512) 992-7900
IRC
(512) 992-7900
IRC
(512) 992-7900
IRC
(512) 992-7900
LTC
(408) 432-1900
A compact, low profile Lithium-Ion
battery charger using all surface
mount components has been
described. With a total circuit height
of only 3.2mm (0.136in.) including
PC board, this circuit can charge one
or two Lithium-Ion cells at currents
up to 800mA. Other battery chemistries can also be charged, although
charge termination circuitry will be
required if fast charging is used.
Thin-Pac is a trademark of Coiltronics, Inc. PowerMite is
a registered trademark of Motorola Corp.
Linear Technology Magazine • November 1997
DESIGN IDEAS
Single-Supply
Random Code Generator
Introduction
That Fuzz is Noise
With the proliferation of satellite
receivers, cable systems and Internet
commerce, there is a need for secure
encryption. A pseudorandom code
generator that filled the bill several
years ago is now considered
“hackable” by many knowledgeable
people.
Presented here is a truly random
code generator that operates from a
single supply. The circuit allows
operation from a single 5V supply
with a minimum of adjustments.
The circuit produces random ones
and zeroes by comparing a stream of
random noise generated in a Zener
diode to a reference voltage level. If
the threshold is correctly set and the
time period is long enough, the noise
will consist of a random but equal
number of samples above and below
threshold.
The circuit shown in Figure 1 is the
random noise generator. Optimum
noise performance is obtained from a
1N753A Zener diode, which has a 6.2
volt Zener “knee.” The diode is used to
generate random noise. We have found
that optimum noise output for this
diode occurs at the “knee” of the I-V
curve, where the Zener just starts to
limit voltage to 6.2 volts.
Operating a 6.2V Zener from a 5V
supply required some thought. Obviously, some type of voltage boosting
scheme was needed to provide the
diode with the 8V or more that it
requires in this circuit. U1, an
LTC1340 low noise, voltage-boosted
varactor driver, provides 9.2V at 20µA
from an input of 5V. This Zener current is the optimal for noise output
from the diode (at 20µA the output is
about 20mVP-P).
by Richard Markell
The 1M and 249k resistors bias the
input to operational amplifier U2 to
1.25V to match the input common
mode range of comparator U3. The
1µF capacitor provides an AC path
for the noise. Note: be careful where
you place any additional capacitors
in this part of the circuit or the noise
may be unintentionally rolled off. This
is one circuit where noise is desirable.
U2 is an LT1215 23MHz, 50V/µs,
dual operational amplifier that can
operate from a single supply. It is
used as a wideband, gain-of-eleven
amplifier to amplify the noise from
the Zener diode; the second op amp in
U2 is unused. U3, an LT1116 high
speed, ground-sensing comparator,
receives the noise at its positive input.
A threshold is set at the negative
comparator input and the output is
adjusted via the 2k potentiometer for
an equal number of ones and zeroes.
continued on page 36
5V
U1
LTC1340
1
2
3
0.1µF
4
CP
AVCC
VCC
OUT
SHDN
AGND
PGND
IN
8
7
0.1µF
9.2V
6
10pF
5
5V
5V
5V
1000pF
~20mVP-P
NOISE
1N753A
6.2V
3
+
2
8
–
4
1k
1
U2
LT1215
1M
1.25V
1µF
1µF
1µF
3
1µF
TANT
470k
249k
2
+
20
VCC
1
+
8
U3
LT1116
5 LE
–
6
4
+
3
7
47k
10µF
0.1µF
1µF
0.1µF
CLOCK IN
DO
QO
2
2-LEVEL
OUTPUT
U4
74HC373
11 LE
GND
10
OE
1
5k
+
10µF
5V
2k
10 TURN
Figure 1. Single-supply random code generator
Linear Technology Magazine • November 1997
29
DESIGN IDEAS
A Low Distortion, Low Power,
Single-Pair HDSL Driver Using the LT1497
by George Feliz and Adolfo Garcia
Low Distortion Line Driver
High speed digital subscriber line
(HDSL) interfaces support full-duplex
data rates up to 1.544Mbps over
12,000 feet using two standard 135Ω
twisted-pair telephone wires. The high
data rate is achieved with a combination of encoding 2 bits per symbol
using two-binary, one-quaternary
(2B1Q) modulation, and sophisticated
digital signal processing to extract
the received signal. This performance
is possible only with low distortion
line drivers and receivers. In addition, the power dissipation of the
transceiver circuitry is critical because
it may be loop-powered from the central office over the twisted pair. Lower
power dissipation also increases the
number of transceivers that can
placed in a single, non-forced–air
enclosure. Single-pair HDSL requires
the same performance as two-pair
HDSL over a single twisted pair and
operates at twice the fundamental
2B1Q symbol rate. In HDSL systems
that use 2B1Q line coding, the signal
passband necessary to carry a data
rate of 1.544Mbps is 392kHz. This
signal rate will be used to quantify the
performance of the LT1497 in this
article.
The circuit of Figure 1 transmits signals over a 135Ω twisted pair through
a 1:1 transformer. The LT1497 dual
125mA, 50MHz current feedback
amplifier was chosen for its ability to
cleanly drive heavy loads, while consuming a modest 7mA maximum
supply current per amplifier in a thermally enhanced SO-8 package. The
driver amplifiers are configured in
gains of two (A1) and minus one (A2)
to compensate for the attenuation
inherent in the back-termination of
the line and to provide differential
drive to the transformer. The transmit power requirement for HDSL is
13.5dBm (22.4mW) into 135Ω, corresponding to a 1.74VRMS signal. Since
2B1Q modulation is a 4-level pulse
amplitude modulated signal, the crest
factor (peak to RMS) of this signal is
1.61. Thus, a 13.5dBm, 2B1Q modulated signal yields 5.6VP-P across the
135Ω load. The corresponding output
signal current is ±20.7mA peak. This
modest drive level increases for varying line conditions and is tested with
a standardized collection of test loops
that can have line impedances as low
as 25Ω. The LT1497’s high output
current and voltage swing drive the
135Ω line at the required distortion
560Ω
560Ω
5V
–
A1
1/2 LT1497
VIN
68.1Ω
+
1.1*
+
560Ω
135Ω
560Ω
–
A2
1/2 LT1497
–
68.1Ω
*MIDCOM 671-7807
(800) 643-2661
+
DI 1497 01.eps
–5V
Figure 1. LT1497 HDSL driver
30
VOUT
5.6VP-P
0
AMPLITUDE (dBm)
Introduction
–50
2HD
–100
100 200 300 400 500 600 700 800 900 1000 1100
DI 1497 02
FREQUENCY (kHz)
Figure 2. Harmonic distortion of Figure 1’s
circuit with a 400kHz sine wave and an
output level of 5.6VP-P into 135Ω
level of –72dBc. For a data rate of
1.544Mbps and 2-bit-per-symbol
encoding, the fundamental frequency
of operation is 392kHz.
The LT1497 provides such low distortion because it operates at only a
fraction of its output current capability and is well within its voltage swing
limitations. There are other LTC
amplifiers that can achieve this performance, but at the expense of higher
power dissipation or a larger package.
Performance
The circuit of Figure 1 was evaluated
for harmonic distortion with a 400kHz
sine wave and an output level of
5.6VP-P into 135Ω. Figure 2 shows
that the second harmonic is –72.3dB
relative to the fundamental for the
135Ω load. Third harmonic distortion
is not critical, because received signals are heavily filtered before being
digitized by an A/D converter. Performance with a 50Ω load (to simulate
more challenging test loops) is slightly
better at –75dB. The output signal
was attenuated to obtain maximum
sensitivity of the HP4195A network
analyzer used for the measurements.
With multicarrier applications such
as discrete multitone modulation
continued on page 32
Linear Technology Magazine • November 1997
DESIGN IDEAS
100V, 2A, Constant-Voltage/ConstantCurrent Bench Supply
by Mitchell Lee
Most engineering labs are well
stocked with low voltage, moderate
current power supplies, but higher
voltage supplies capable of several
amperes of output current are hard
to find. I solved this problem in my lab
by building the supply shown in
Figure 1.
The circuit is based on U1, an
LT1270 high efficiency switching regulator configured in a SEPIC topology,
which allows the output to be adjusted higher or lower than the input
voltage. Operation is similar to that of
a flyback converter, but the primary
and secondary windings are coupled
together by capacitor C1. This allows
the primary and secondary windings
to share current, reducing copper
loss; it also eliminates the snubbing
circuitry and losses found in flyback
converters.
The converter is designed to operate from an input of 40V to 60V,
supplied by a line transformer, diode
bridge and filter capacitor (not shown).
Output voltage is linearly adjustable
from zero to 100V via potentiometer
R20.
The current is limited by two independent loops. The first current limit
loop is user controlled over a range of
2.2k
L1
20µH
MUR1560
T1
VIN
40V–60V
+
3.3k
2W
10µF
200V
FILM CAP
MBR745
2N6387
10Ω
IRF450
+
VOUT
0V–100V
10µF
200V
FILM CAP
0.03Ω, 2W
0.1µF
1k
MUR120
100k
Jesus Rosales played a significant role in the
design, building and testing of this circuit.
T1 = PRIMARY: 57 TURNS 20AWG
SECONDARY: 57 TURNS 20AWG
MPP 55076 MAG INC CORE
L1 = 18 TURNS 18AWG
55380-A2 MAG INC CORE
1000pF
+
C1*
10µF, 100V
FILM CAP
zero to 8A by setting potentiometer
R21. This setting does not interact
with changes in output voltage. A
second current limit loop limits the
maximum available current as a function of voltage (components R1–R5
and U2), minimizing component
stress. Under any given operating
condition, the lower of the two loops
takes control. Maximum available
output current is highest at low output voltage settings (about 8A), and
decreases to 2A at 100V output.
4.7k
1N5817
100Ω
1N5817
8
–
4
U1
LT1270
+
5
U2A
1/2 LT1413
7
2
10Ω
1N4148
1k
3
R21, 1k
3.9k
6
5
0.1µF
4
R3
4.5k
1k
R4
3.9k
R5
2.7M
3.9k
3.9k
2.2k
1
1N4148
+
1
0.1µF
15k
U2B
1/2 LT1413
–
3
2
10k
2N2907
–
7
680µF
100V
×2
+
+
+
56µF
35V
U3B
1/2 LT1215
+
15V
0.33µF
3.9V
20k
2
R1
3.9k
3
R2
3.9k
R20
10k
15k
100Ω
0.01µF
U3A
1/2 LT1215
4
5
1µF
2.2k
1
+
100Ω
6
–
8
22V
1µF
10k
LT1034CZ-2.5
Figure 1. 100V/2A Constant Voltage/Constant Current Bench Supply
Linear Technology Magazine • November 1997
31
DESIGN IDEAS
Switcher Generates Two Bias Voltages
without Transformer
by Jeff Witt
LCD displays and CCD imaging
circuits in today’s portable products
require several bias voltages from a
10V to 20V input at a few mA. When
symmetric bipolar bias supplies are
needed, the negative supply can be
generated with a discrete charge pump
operating from the power switch of
the boost regulator that generates
the positive supply. However, an asymmetric bipolar supply is typically
required: for example 20V and –10V
for LCD displays or 15V and –7.5V for
CCDs. One possible solution is to add
a linear regulator to the negative output; this adds cost and greatly reduces
the efficiency of the switcher. Another
possibility is a 2-output flyback circuit, but the added cost and bulk of a
L1
47µF
VIN
3.3V
transformer make this solution unappealing. The circuit in Figure 1
avoids these penalties, producing 20V
at 5mA and –10V at 5mA from 3.3V
with 73% efficiency. The circuit uses
standard surface mount parts.
The LT1316, a micropower Burst
Mode switching regulator with an integrated 0.6A power switch, operates
in an ordinary boost circuit to generate the 20V (VOUT1) set by resistor
divider R1 and R2. An internal comparator at the FB pin regulates the
output by gating the LT1316’s oscillator. A charge pump (C2 and
associated diodes) coupled to the
LT1316’s switch pin generates the
negative output voltage. This negative output (VOUT2) is monitored by
BAT54
VOUT1
20V/5mA
150pF
7
5
SW
6
VIN
FB
SHDN
LT1316
C1
33µF
10V
LBO
LBI
RSET
3
R5
10k
8
C3
3.3µF
35V
R1
1M
150k
R3
1M
1
2
VOUT1 200mV/DIV
(AC COUPLED)
R2
64.9k
GND
4
82k
BAT54
C2
1µF
35V
L1 = COILCRAFT DO1608C-473
C1 = AVX TAJB336M035R
C2 = AVX TAJA105M035R
C3, C4 = AVX TAJB335M035R
BAT54
BAT54
SW PIN 20V/DIV
V OUT2 1V/DIV
(AC COUPLED)
R4
590k
Q1
2N7002
C4
3.3µF 35V
LBO PIN 5V/DIV
VOUT2
–10V/5mA
0.1ms/DIV
Figure 1. By gating the charge pump, this circuit generates a regulated
negative output with a magnitude different from that of the positive output.
HDSL Driver, continued from page 30
Figure 2. Voltage waveforms of Figure 1’s circuit
third-order intermodulation products
are well below –72dB. With a 50Ω
load, performance is within 1dB–2dB
of that with the 135Ω load.
0
AMPLITUDE (dBm)
(DMT) becoming as prevalent as
single-carrier applications, another
important measure of amplifier
dynamic performance is 2-tone intermodulation. This evaluation is a
valuable tool to gain insight to amplifier linearity when processing more
than one tone at a time.
For this test, two sine waves at
300kHz and 400kHz were used with
levels set to obtain 5.6VP-P across the
135Ω load. Figure 3 shows that the
32
the LT1316’s low-battery detector
through the resistor divider R3 and
R4, using the positive 20V output as
a reference. When the negative output falls below 10V, the low-battery
detector output (LBO pin and lowest
trace in Figure 2) turns Q1 on, enabling the charge pump and charging
output capacitor C4. Note that the
switch pin jumps between ground
and ~10V during this period. Once
the negative output has been charged
enough to overcome the low-battery
detector’s hysteresis, Q1 turns off
and the switch pin is free to fly to 20V,
charging the positive output.
This circuit can also operate directly from two alkaline or NiCd cells.
Slightly higher peak currents are necessary; change R5, which determines
the peak switch current of the LT1316,
to 6.8kΩ and change L1 to 15µH.
Conclusion
–50
3IMD
–100
100
200
3IMD
300
400
500
FREQUENCY (Hz)
600
DI 1497 03
Figure 3. 2-tone intermodulation for Figure
1’s circuit
The circuit presented provides outstanding distortion performance in
an SO-8 package with remarkably
low power dissipation. It is ideally
suited for single pair digital subscriber
line applications, especially for remote
terminals.
Linear Technology Magazine • November 1997
DESIGN INFORMATION
Reference Squeezes More Performance
by John Wright
from Less Package
Introduction
The small outline transistor (SOT)
package is used extensively for space
sensitive, surface mount, discretetransistor applications. Recently,
several manufacturers have put voltage references into SOT packages,
but the current crop sacrifices performance to fit into this stingy
package. Simply put, the package
and die area are so small that techniques used to design good references
need modification. Today’s applications demand more performance in
smaller PC board space and the size
of the SOT is ideal—the SOT footprint
is 4.7 times smaller than the 8-pin SO
package footprint. The challenge is to
make a quality reference small enough
to fit into the SOT package.
Voltage Reference or
Precision Regulator
The LT1460S3 is a series reference
that provides supply current and
power dissipation advantages over
shunt references, which must idle
their entire load current to operate.
This new reference uses curvature
compensation to obtain low temperature coefficient, and laser-trimmed
precision thin-film resistors to achieve
high output accuracy. The manufacturing process used is a
stepper-based, high speed bipolar process that was selected for its density.
The LT1460S3 uses a proprietary trim
algorithm at wafer sort that guarantees less than 20ppm/°C temperature
coefficient (typically 10ppm/°C). The
temperature coefficient (TC) is affected
by package stress during assembly,
widening the distribution slightly. The
result is three grades, two with
guaranteed TC ≤ 20ppm/°C. This performance is something of a milestone,
because the highest grade TC available in SOT heretofore was 50ppm/°C,
and that is the lowest grade available
on the LT1460S3. Low stress thinfilm resistors and special layout
techniques were used to achieve 0.2%
maximum tolerance without the use
of gel coat to protect the surface from
stress. This combination of low TC
and high accuracy makes the
LT1460S3 capable of 8 bits of absolute accuracy over temperature
without a system calibration.
The LT1460S3 supplies up to 20mA
of output drive, making it ideal for
precision regulator applications. It is
stable with any value output capacitor, including zero, and this can be
helpful in applications where PC board
space is at a premium, or when fast
settling is demanded. An output
capacitor used for stability slows down
the reference transient response. The
LT1460S3 settles in 1µs to 0.2% for a
100µA pulse with no output capacitor and in just 2µs for a 1mA pulse.
solder-related shift to typically 0.02%.
A graph of the output voltage after IR
reflow solder is shown in Figure 1.
After the part is soldered onto a PC
board, the output voltage should not
drift with time.
When an instrument manufacturer
sends out a PC board for duty in the
field, it has had extensive testing,
perhaps a calibration, and its voltage
reference is stable. Over time, however, the reference output voltage may
change. This long-term drift is caused
mainly by differential stress between
the IC package and the PC board.
This output shift is a nonlinear function of time, and is normally expressed
in ppm/√kh. What this means is that
the shift in the first one thousand
hours is more than that in the subsequent one thousand hours. Figure 2
shows a plot of long-term drift of the
LT1460S3 soldered to a PC board.
This part was measured in a 30°C
oven that was controlled to ±1°C. The
long-term drift is the trend line created by this plot; over one thousand
hours, this typical part moved 50ppm,
or about 125µV for the LT1460S3-2.5.
Hysteresis in the output voltage is
another package-stress phenomenon.
A reference output voltage does not
return to exactly the same 25°C value
after it has been at different temperatures. This is because the stress on
48
40
DISTRIBUTION (%)
36
LT1460HC LIMITS
32
28
24
20
16
12
8
4
0
2.490
2.500
OUTPUT VOLTAGE (V)
2.510
1460_01.EPS
Figure 1. Typical distribution of SOT-23
LT1460HC VOUT after IR reflow solder
Linear Technology Magazine • November 1997
Soldering a SOT reference onto a PC
board is a harsh thing to do. The
reference that leaves Linear Technology is within the data sheet
specification, but the thermal shock
of being soldered onto a PC board
causes the reference to shift. The
problem is that the short, stiff leads
do not absorb stress and it is transferred to the die. Low stress thin-film
resistors and careful layout techniques in the LT1460S3 reduce
DRIFT (PPM)
Measure It on a PC Board
44
80
70
60
50
40
30
20
10
0
–10
–20
–30
–40
–50
–60
–70
–80
TREND LINE
0 100 200 300 400 500 600 700 800 900 1000
TIME (HOURS)
1460_02
Figure 2. LT1460S3 long-term drift
33
DESIGN INFORMATION
the die changes with temperature and
never returns to exactly the same
state in which it started. Hysteresis is
roughly an exponential function of
the temperature excursion away from
25°C and depends on whether the IC
was previously at a higher or lower
temperature. The output voltage is
always measured at 25°C, but the IC
is cycled over its specified temperature
range before successive measurements. The LT1460S3 has a typical
hysteresis of 100ppm for a 0°C to
70°C excursion. Hysteresis increases
Summary
to 250ppm with a –40°C to 85°C temperature change because larger stress
is associated with a larger temperature range. The initial accuracy
specification of 0.2% includes a margin for hysteresis. This is because it is
not possible to know what temperature changes the part may be
subjected to on its way to the customer. Because hysteresis is increased
by wider temperature excursions, the
part may shift out of tolerance if is
stored outside of the specified temperature range.
The LT1460S3’s high output current,
low temperature coefficient and
excellent, yet conservative accuracy
specification set a new standard for
SOT voltage references. This new reference is available in 2.5V and 5V
output voltages. For even tighter tolerance and lower temperature
coefficient, the LT1460 is also available in MSOP, SO, PDIP and 3-lead
TO-92 packages.
Authors can be contacted
at (408) 432-1900
LTC1659, LTC1448: Smallest Rail-to-Rail
12-Bit DACs Have Lowest Power
In this age of portable electronics,
power and size are the primary concerns of most designers. The LTC1659
and the LTC1448 are rail-to-rail, 12bit, voltage output DACs that address
both of these concerns. The LTC1659
is a single DAC in an MSOP-8 package that draws only 250µA from a 3V
or 5V supply, whereas the LTC1448
is a dual DAC in an SO-8 package
that draws 450µA from a 3V or 5V
supply.
Figure 1 shows a convenient way
to use the LTC1659 in a digital control loop where 12-bit resolution is
required. The output of the LTC1659
will swing from 0V to VREF, because
there is a gain of one from the REF pin
to VOUT at full-scale. Because the output can only swing up to VCC, VREF
should be less than or equal to VCC to
prevent the loss of codes and degradation of PSRR near full-scale.
To obtain full dynamic range, the
REF pin can be connected to the
supply pin, which can be driven from
a reference to guarantee absolute
accuracy (see Figure 2). The LT1236
is a precision 5V reference with an
input range of 7.2V to 40V. In this
configuration, the LTC1659 has a wide
output swing of 0V to 5V. The LTC1448
can be used in a similar configuration
where dual DACs are needed.
34
by Hassan Malik
VREF ≤ VCC
2.7V to 5.5V
DIN
µP
VCC
REF
CLK
LTC1659
CS/LD
GND
VOUT
CONTROL
VOLTAGE
(0V TO VREF)
1659_01
Figure 1. 12-bit DAC for digital control loop
LT1236
VIN
(7.2V to 40V)
IN
OUT
GND
0.1µF
DIN
µP
CLK
CS/LD
VCC
LTC1659
REF
VOUT
CONTROL
VOLTAGE
(OV TO 5V)
GND
1659_02
Figure 2. 12-bit DAC with wide output swing
Linear Technology Magazine • November 1997
DESIGN INFORMATION
LTC1197/LTC1199: New Micropower
MSOP 10-Bit ADC Samples at 500ksps
by Guy Hoover and Marco Pan
Introduction
The LTC1197 and LTC1199 10-bit
serial ADCs, recently released by Linear Technology, offer small size, low
power operation and fast sample rates
with good AC and DC performance.
These parts are ideal for low power,
high speed and/or compact designs.
In this article, we will examine the
features and performance of the
LTC1197/LTC1199 that make these
parts excellent choices for such new
designs.
as 200mV. This translates to an LSB
size of only 200µV. Combined with
the high impedance of the analog
input, this allows direct digitization
of low level transducer outputs, which
can save board space and the cost of
a gain stage.
With its software-selectable
2-channel MUX, the LTC1199 is capable of measuring either one
differential or two single-ended inputs. Both parts have a built-in
sample-and-hold.
single-ended or differential inputs,
a d j u s t a b l e re f e r e n c e v o l t a g e ,
SPI-/MICROWIRE-compatible serial
I/O and 3V or 5V operation.
Performance
Micropower Performance with
Auto Shutdown at Full Speed
Figure 1. Supply current (I CC) vs sampling
frequency fS
Now let’s take a look at the performance of the LTC1197/LTC1199.
Running continuously, the LTC1197L
consumes only 2.2mW at the maximum sampling rate (25mW for the
Serial I/O
LTC1197). You can reduce the power
Using either 3- or 4-wire serial inter- consumption dramatically, as shown
faces, the LTC1197/LTC1199 are in Figure 1, simply by lowering the
hardware and software compatible sampling rate. The formula for calcuwith both SPI and MICROWIRE pro- lating power consumption is:
tocols. This compatibility is achieved
PD = VCC • ICC • tCONV • fS
with no additional circuitry, allowing
easy interface to many popular where PD is the power consumption,
VCC is the supply voltage, ICC is the
processors.
supply current while the conversion
3V or 5V Supplies
is occurring, tCONV is the conversion
The LTC1197/LTC1199 are 5V parts time and fS is the sample rate. As you
(VCC = 4V–9V for the LTC1197 and can see from the formula, lowering fS
4V–6V for the LTC1199). Also avail- reduces the power consumption linable for use in 3V systems are the early. It is also important to minimize
LTC1197L and the LTC1199L (VCC = tCONV by clocking the ADC at its maxi2.7V–4V). Designed for use in mixed- mum rate during the conversion.
supply systems, the digital inputs of Although the ADC draws slightly more
these devices can be taken above the power at high clock rates, the total
VCC voltage without damaging the power is less because the device is on
ADC. This is useful in systems where for a shorter period of time. For
the ADC is running at a lower supply example, the 2.2mW is cut to 10.4µW
voltage than the processor. If the ADC by reducing the sampling rate from
is running at a higher supply voltage 210kHz to 1kHz. The calculation for
than the processor, the ADC serial power dissipation is as follows:
data output voltage can easily be dePD = 2.7V • 0.8mA • 4.8µs •
creased to a level appropriate for the
1kHz
= 10.4µW
processor.
To summarize, the LTC1197/
LTC1199/LTC1197L/LTC1199L are High Speed Capability
a very flexible group of parts. They are Even though the LTC1197/LTC1199
capable of providing a designer with a are capable of micropower operation,
small footprint, one or two channels, they are able to convert at rates of up
Linear Technology Magazine • November 1997
35
Features
Smallest Size (MSOP)
The LTC1197/LTC1199 are among
the smallest ADCs available. The serial
inter face allows the LTC1197/
LTC1199 to be offered in 8-pin packages. The MSOP package (an SO is
also available) reduces the small footprint even further. These are some of
the first ADCs available in the MSOP
package, which is about half the size
of the SO-8.
Flexible Inputs
The LTC1197 has a single differential
input with an adjustable reference.
The adjustable reference input allows
the full scale to be reduced to as low
10000
SUPPLY CURRENT (µA)
1000
100
VCC = 5V,
fCLK = 7.2MHz
10
1
0.1
0.01
VCC = 2.7V,
fCLK = 3.5MHz
0.1
10
100
1
SAMPLING FREQUENCY (kHz)
1000
1197_01
CONCLUSIONS
to 500kHz (210kHz for LTC1197L).
These parts can also digitize fast input signals up to the Nyquist
frequency (250kHz for the LTC1197)
with over nine effective number of
bits (ENOBs).
Good DC and AC Specs
The DC specifications of these parts
are very good. Linearity (both INL and
DNL) is typically 0.3LSB with a maximum spec of 1LSB. Offset is specified
at 2LSBs (Max) and gain error is
specified at 4LSBs (Max). These speci-
fications are guaranteed over the full
temperature range of the part. Both
commercial and industrial temperature range versions are available.
AC performance is equally impressive. S/(N + D) is typically 60dB (58dB
for the L version). THD is typically
–64dB (–60dB for the L version) and
the peak harmonic or spurious noise
is typically –68dB (–63dB for the L
version). The L version has slightly
degraded specifications due to the
smaller dynamic range and the higher
internal noise of the ADC at its lower
operating current.
Conclusion
operation, an LT1004-2.5 is used as a
reference at the front end of a precision voltage divider string. A series of
voltages is generated along the divider
string and a jumper is used to connect this voltage to a buffer and then
to the negative input of the LT1116
comparator. As was the case with the
2k pot, the voltage at pin 2 (the negative input of the comparator) sets the
threshold for the comparator. The
selection of voltage taps on the resistor
string is arbitrary; they were selected
to allow a good adjustment range
(defined as allowing jumper adjustment to 50% ones and 50% zeros) for
a sample of ten 1N753A Zener diodes
used to produce noise. The jumper
could (and probably should) be replaced with analog switches controlled
by a microprocessor in medium- to
high-volume applications.
We have seen that the LTC1197 and
LTC1199 have a small footprint and
are capable of micropower operation.
These parts have a versatile serial
interface that is SPI/MICROWIRE
compatible. The adjustable reference
input, 2-channel, software-selectable
MUX and 5V or 3V operation add to
the versatility of this ADC family.
When this versatility is combined with
the high conversion rate and good DC
and AC performance, you can see
why these ADCs are good choices for
low power, high speed and/or compact designs.
Random Code, continued from page 29
The 5k resistor and the 10µF capacitor provide limited hysteresis so that
the adjustment of the potentiometer
is not as critical. Latch U4, a 74HC373,
ensures that the output remains
latched throughout one clock period.
The circuit’s output is taken from
U4’s Q0 output.
Some Thoughts on
Automatic Threshold
Adjustment
Several circuit designers have asked
about threshold adjustment without
manual knobs or potentiometers. One
way to implement this would be to
have the microprocessor count the
number of ones and zeroes over a
given time period and adjust the
threshold (perhaps via a digital pot) to
produce the required density of ones.
A more “analog” method of adjusting threshold might be to implement
an integrator with reset. This circuit
integrates the number of ones and
zeroes over time to produce a zero
result for an adjustment that produces equal numbers of ones and
zeroes. Again, a digital pot could be
used to adjust threshold, with the
threshold being decreased for the case
of “not enough ones” and increased
for the case of “too many ones.”
After many more conversations
with the “cyber illuminati,” the circuit
in Figure 2 was devised. This circuit
can be used to replace the pot shown
in the dashed box in Figure 1. In
36
5V
15k
1µF
0.01µF
LT1004
-2.5
12k 1%
(1.30V)
499 1%
(1.25V)
JUMPER SELECTS
THRESHOLD VOLTAGE
499 1%
(1.20V)
499 1%
(1.15V)
499 1%
(1.10V)
11k 1%
5V
1µF
3
+
8
LT1490
2
–
1
TO LT1116
PIN 2 (FIGURE 1)
4
GROUND PINS 5 AND 6
Figure 2. Jumper selects threshold for Figure 1’s circuit
Linear Technology Magazine • November 1997
NEW DEVICE CAMEOS
New Device Cameos
LTC1234: Low Dropout
Regulator with Comparator
The LTC1660 sets a new standard in and Shutdown
LTC1660 Micropower, Octal,
10-Bit DAC with Sleep Mode
DAC density by integrating eight high
quality addressable 10-bit DACs in a
single, tiny 16-pin package, no larger
than an SO-8: required board area is
only 0.006in2 per DAC.
Operating on a single 2.7V–5.5V
supply rail, the LTC1660 draws just
60µA per DAC (480µA for all eight),
for true micropower performance.
Sleep mode further reduces total supply-plus-reference current to just 1µA.
The LTC1660 is guaranteed monotonic over temperature—DNL error is
typically ±0.2LSB (±0.5LSB Max).
Each of the rail-to-rail output amplifiers can source or sink up to 5mA.
The outputs swing to within a few
millivolts of either supply rail when
unloaded and have an equivalent output resistance of 80Ω when driving a
load to the rails. The output amplifiers are stable driving capacitive loads
up to 1000pF.
The 3-wire serial interface uses a
16-bit input word comprising four
address/control bits, ten input-code
bits, and two don’t-care bits. Asynchronous CLR, power-on reset and
daisy-chain capability are also provided. It is possible to keep one or
more chips in a daisy chain in continuous Sleep mode by giving the
Sleep instruction to these chips each
time the active chips in the chain are
updated.
Ultralow supply current, powersaving Sleep mode and extremely
compact size make the LTC1660 ideal
for battery-powered applications; its
straightforward usability, high performance and low cost make it an
excellent choice as a general-purpose
converter.
Linear Technology Magazine • November 1997
The LTC1234 combines a micropower,
positive, low dropout linear regulator
and a low-battery comparator. With
only 20µA quiescent current, typical
dropout voltages are 250mV for 5V
output with 500mV of load, or 150mV
for 3.3V with 250mA of load. An internal 750mA P-channel pass transistor
draws no base current, allowing the
device to draw less than 35µA of
quiescent current over the extended
temperature range, independent of
load current.
Protection circuitry protects
against output short circuits, thermal overloads and reverse current
when the input potential falls below
the output.
A micropower comparator with an
open drain output can be used for
system or battery monitoring. The SD
pin allows the regulator to be shut
down while the comparator and reference remain alive. The quiescent
current drops down to 5µA in shutdown mode.
The LTC1234 is available in the
8-pin SO wide package.
LTC1068-XX:
New Family of Quad
Universal, Switched Capacitor
Filter Building Blocks
The LTC1068-XX consist of four identical, low noise, switched capacitor
filter building blocks. Each building
block, together with three to five
resistors and an external clock, provides 2nd order filter functions such
as lowpass, bandpass, highpass and
notch.
High precision quad 2nd order,
dual 4th order or single 8th order
filters can be designed with an
LTC1068-XX. All these devices are
fully supported by the new FilterCAD™ for Windows® filter design
software.
An external clock tunes the center
frequency of each filter building block.
The clock-to-center frequency ratio is
internally set to 25:1 (LTC1068-25),
50:1 (LTC1068-50) or 200:1
(LTC1068-200). The clock-to-center
frequency ratio can also be modified
with external resistors. The internal
sampling rate of all the LTC1068-XX
devices is twice the clock frequency.
This allows the frequency of input
signals to approach twice the clock
frequency before aliasing occurs. The
LTC1068-XX is designed to complement the LTC1068, which features a
100:1 clock-to-center frequency ratio.
The LTC1068-200, with its internal
400:1 sampling-rate-to-center-frequency ratio behaves almost like a
continuous time filter; it is recommended for non-band-limited filter
circuits such as highpass and notch.
Notch filters with 80dB notch depth
can be easily realized.
The LTC1068-50 is aimed at filter
applications, especially lowpass and
bandpass, requiring single 3V to 5V
supply and low power consumption.
The LTC1068-50 consumes 3mA
typical.
The LTC1068-25 lowers its internal clock-to-center frequency ratio so
it can achieve 200kHz cutoff frequencies with a 5MHz clock.
The LTC1068-XX family is available in 28-pin SSOP. Demo boards
are also available to qualified
customers.
A customized version of the
LTC1068-XX in a 16-lead SO with
internal thin-film resistors can be
ordered. Please contact LTC marketing for details.
37
NEW DEVICE CAMEOS
LTC1541/LTC1542:
Micropower Op Amp,
Comparator and Reference
typical gain bandwidth of 10kHz and
a typical slew rate of 8V/ms. The
comparator has ±3mV of internal hysteresis
to ensure clean output
The LTC1541/LTC1542 combine a
switching
even with slow moving input
micropower amplifier, comparator and
signals.
bandgap reference (LTC1541 only) in
The LTC1541/LTC1542 are availan 8-pin package. These parts operate
able
in 8-pin SO and MSOP packages.
from single supplies of 2.5V to 12.6V
or dual supplies of ±1.25V to ±6.3V,
with a typical supply current of 6µA.
Both the op amp and comparator LTC1562 Quad
feature a common mode input voltage Universal Active Filter
range that extends from the negative with SNR >100dB
supply to within 1.3V of the positive The LTC1562 is a new low noise, low
supply. The output stages swing from distortion, DC-accurate, continuousrail-to-rail. The comparator’s invert- time filter block with rail-to-rail inputs
ing input is internally connected to and outputs, optimized for 50kHz–
the reference output (LTC1541) or 200kHz applications (other cutoff
bonded out without the reference frequency ranges are available by custom order). It contains four matched,
(LTC1542).
The reference output voltage is 1.2V independent 2nd order filter sections,
±1% over the extended temperature which can be cascaded in any combirange (–40°C to 85°C). The output can nation, such as one 8th order or two
drive a bypass capacitor of up to 4th order filters. Each section’s re0.01µ F without oscillation. It can also sponse is programmed with three
source up to 2mA and sink up to 20µA. external resistors, for center freThe op amp is internally compen- quency, Q and gain, and each section
sated to be unity-gain stable, with a provides lowpass and bandpass out-
puts. A highpass response is available if one of the resistors is replaced
with an external capacitor. Center
frequency is internally trimmed to
±0.5% and is very stable with temperature. A shutdown pin puts the
chip into “zero-power” shutdown
state.
Operating from single or dual power
supplies of 4.75V to 10.5V total, the
LTC1562 is designed for applications
where dynamic range is critical, such
as receivers, data modems and DSP
antialiasing or reconstruction. With
±5V supplies and Q = 1, a 2nd order
section delivers a typical signal-tonoise ratio (SNR) of 103dB, total
harmonic distortion of –96dB and DC
offset of 2mV. In applications that
filter signals of wide amplitude range,
it is possible to use the LTC1562’s
gain-setting resistor for gain control,
which reduces the input noise at
higher gains and extends the dynamic
range (maximum signal to minimum
noise) to 118dB.
The LTC1562 is offered in 16-pin
DIP and 20-pin SSOP packages.
for
the latest information
on LTC products,
visit
www.linear-tech.com
For further information on any
of the devices mentioned in this
issue of Linear Technology, use
the reader service card or call
the LTC literature service
number:
1-800-4-LINEAR
Ask for the pertinent data sheets
and Application Notes.
38
Linear Technology Magazine • November 1997
DESIGN TOOLS
DESIGN TOOLS
Applications on Disk
Noise Disk — This IBM-PC (or compatible) program
allows the user to calculate circuit noise using LTC op
amps, determine the best LTC op amp for a low noise
application, display the noise data for LTC op amps,
calculate resistor noise and calculate noise using specs
for any op amp.
Available at no charge
SPICE Macromodel Disk — This IBM-PC (or compatible) high density diskette contains the library of LTC
op amp SPICE macromodels. The models can be used
with any version of SPICE for general analog circuit
simulations. The diskette also contains working circuit
examples using the models and a demonstration copy
of PSPICE™ by MicroSim.
Available at no charge
SwitcherCAD™ — The SwitcherCAD program is a powerful PC software tool that aids in the design and
optimization of switching regulators. The program can
cut days off the design cycle by selecting topologies,
calculating operating points and specifying component values and manufacturer’s part numbers. 144
page manual included.
$20.00
SwitcherCAD supports the following parts: LT1070
series: LT1070, LT1071, LT1072, LT1074 and LT1076.
LT1082. LT1170 series: LT1170, LT1171, LT1172 and
LT1176. It also supports: LT1268, LT1269 and LT1507.
LT1270 series: LT1270 and LT1271. LT1371 series:
LT1371, LT1372, LT1373, LT1375, LT1376 and
LT1377.
Micropower SwitcherCAD™ — The MicropowerSCAD
program is a powerful tool for designing DC/DC converters based on Linear Technology’s micropower
switching regulator ICs. Given basic design parameters, MicropowerSCAD selects a circuit topology and
offers you a selection of appropriate Linear Technology
switching regulator ICs. MicropowerSCAD also performs circuit simulations to select the other components
which surround the DC/DC converter. In the case of a
battery supply, MicropowerSCAD can perform a battery life simulation. 44 page manual included.
$20.00
MicropowerSCAD supports the following LTC micropower DC/DC converters: LT1073, LT1107, LT1108,
LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,
LT1300, LT1301 and LT1303.
Technical Books
1990 Linear Databook, Vol I —This 1440 page collection of data sheets covers op amps, voltage regulators,
references, comparators, filters, PWMs, data conversion and interface products (bipolar and CMOS), in
both commercial and military grades. The catalog
features well over 300 devices.
$10.00
1992 Linear Databook, Vol II — This 1248 page
supplement to the 1990 Linear Databook is a collection
of all products introduced in 1991 and 1992. The
catalog contains full data sheets for over 140 devices.
The 1992 Linear Databook, Vol II is a companion to the
1990 Linear Databook, which should not be discarded.
$10.00
Linear Technology Magazine • November 1997
1994 Linear Databook, Vol III —This 1826 page
supplement to the 1990 and 1992 Linear Databooks is
a collection of all products introduced since 1992. A
total of 152 product data sheets are included with
updated selection guides. The 1994 Linear Databook
Vol III is a companion to the 1990 and 1992 Linear
Databooks, which should not be discarded. $10.00
1995 Linear Databook, Vol IV —This 1152 page
supplement to the 1990, 1992 and 1994 Linear Databooks is a collection of all products introduced since
1994. A total of 80 product data sheets are included
with updated selection guides. The 1995 Linear Databook Vol IV is a companion to the 1990, 1992 and 1994
Linear Databooks, which should not be discarded.
$10.00
1996 Linear Databook, Vol V —This 1152 page supplement to the 1990, 1992, 1994 and 1995 Linear
Databooks is a collection of all products introduced
since 1995. A total of 65 product data sheets are
included with updated selection guides. The 1996
Linear Databook Vol V is a companion to the 1990,
1992, 1994 and 1995 Linear Databooks, which should
not be discarded.
$10.00
1990 Linear Applications Handbook, Volume I —
928 pages full of application ideas covered in depth by
40 Application Notes and 33 Design Notes. This catalog covers a broad range of “real world” linear circuitry.
In addition to detailed, systems-oriented circuits, this
handbook contains broad tutorial content together
with liberal use of schematics and scope photography.
A special feature in this edition includes a 22-page
section on SPICE macromodels.
$20.00
1993 Linear Applications Handbook, Volume II —
Continues the stream of “real world” linear circuitry
initiated by the 1990 Handbook. Similar in scope to the
1990 edition, the new book covers Application Notes
40 through 54 and Design Notes 33 through 69.
References and articles from non-LTC publications
that we have found useful are also included. $20.00
1997 Linear Applications Handbook, Volume III —
This 976 page handbook maintains the practical outlook
and tutorial nature of previous efforts, while broadening topic selection. This new book includes Application
Notes 55 through 69 and Design Notes 70 through
144. Subjects include switching regulators, measurement and control circuits, filters, video designs,
interface, data converters, power products, battery
chargers and CCFL inverters. An extensive subject
index references circuits in LTC data sheets, design
notes, application notes and Linear Technology magazines.
$20.00
Interface Product Handbook — This 424 page handbook features LTC’s complete line of line driver and
receiver products for RS232, RS485, RS423, RS422,
V.35 and AppleTalk ® applications. Linear’s particular
expertise in this area involves low power consumption,
high numbers of drivers and receivers in one package,
mixed RS232 and RS485 devices, 10kV ESD protection of RS232 devices and surface mount packages.
Available at no charge
Power Solutions Brochure — This 84 page collection
of circuits contains real-life solutions for common
power supply design problems. There are over 88
circuits, including descriptions, graphs and performance specifications. Topics covered include battery
chargers, PCMCIA power management, microprocessor power supplies, portable equipment power supplies,
micropower DC/DC, step-up and step-down switching
regulators, off-line switching regulators, linear regulators and switched capacitor conversion.
Available at no charge
High Speed Amplifier Solutions Brochure —
This 72 page collection of circuits contains real-life
solutions for problems that require high speed
amplifiers. There are 82 circuits including descriptions, graphs and performance specifications. Topics
covered include basic amplifiers, video-related applications circuits, instrumentation, DAC and photodiode
amplifiers, filters, variable gain, oscillators and current
sources and other unusual application circuits.
Available at no charge
Data Conversion Solutions Brochure — This 52 page
collection of data conversion circuits, products and
selection guides serves as excellent reference for the
data acquisition system designer. Over 60 products
are showcased, solving problems in low power, small
size and high performance data conversion applications—with performance graphs and specifications.
Topics covered include ADCs, DACs, voltage references and analog multiplexers. A complete glossary
defines data conversion specifications; a list of selected application and design notes is also included.
Available at no charge
Telecommunications Solutions Brochure — This 72
page collection of circuits, new products and selection
guides covers a wide variety of products targeted for
the telecommunications industry. Circuits solving real
life problems are shown for central office switching,
cellular phone, base station and other telecom applications. New products introduced include high speed
amplifiers, A/D converters, power products, interface
transceivers and filters. Reference material includes a
telecommunications glossary, serial interface standards, protocol information and a complete list of key
application notes and design notes.
Available at no charge
continued on page 40
Acrobat is a trademark of Adobe Systems, Inc. AppleTalk
is a registered trademark of Apple Computer, Inc. PSPICE
is a trademark of MicroSim Corp.
Information furnished by Linear Technology Corporation
is believed to be accurate and reliable. However, Linear
Technology makes no representation that the circuits
described herein will not infringe on existing patent rights.
39
DESIGN TOOLS, continued from page 39
CD-ROM
LinearView — LinearView™ CD-ROM version 2.0 is
Linear Technology’s latest interactive CD-ROM. It allows you to instantly access thousands of pages of
product and applications information, covering Linear
Technology’s complete line of high performance analog products, with easy-to-use search tools.
The LinearView CD-ROM includes the complete product specifications from Linear Technology’s Databook
library (Volumes I–V) and the complete Applications
Handbook collection (Volumes I–III). Our extensive
collection of Design Notes and the complete collection
of Linear Technology magazine are also included.
A powerful search engine built into the LinearView CDROM enables you to select parts by various criteria,
such as device parameters, keywords or part numbers.
All product categories are represented: data conversion, references, amplifiers, power products, filters
and interface circuits. Up-to-date versions of Linear
Technology’s software design tools, SwitcherCAD,
Micropower SwitcherCAD, FilterCAD, Noise Disk and
Spice Macromodel library, are also included. Everything you need to know about Linear Technology’s
products and applications is readily accessible via
LinearView. LinearView 2.0 runs under Windows ® 3.1,
Windows 95 and Macintosh ® System 7.0 or later.
Available at no charge.
World Wide Web Site
Linear Technology Corporation’s customers can now
quickly and conveniently find and retrieve the latest
technical information covering the Company’s products on LTC’s new internet web site. Located at
www.linear-tech.com, this site allows anyone with
internet access and a web browser to search through
all of LTC’s technical publications, including data sheets,
application notes, design notes, Linear Technology
magazine issues and other LTC publications, to find
information on LTC parts and applications circuits.
Other areas within the site include help, news and
information about Linear Technology and its sales
offices.
Linear Technology Corporation
1630 McCarthy Boulevard
Milpitas, CA 95035-7417
Phone: (408) 432-1900
FAX: (408) 434-0507
Linear Technology Corporation
Houston, TX 77478
Phone: (972) 733-3071
FAX: (972) 380-5138
U.S. Area
Sales Offices
Linear Technology Corporation
5510 Six Forks Road, Suite 102
Raleigh, NC 27609
Phone: (919) 870-5106
FAX: (919) 870-8831
Linear Technology Corporation
266 Lowell St., Suite B-8
Wilmington, MA 01887
Phone: (508) 658-3881
FAX: (508) 658-2701
The site is searchable by criteria such as part numbers,
functions, topics and applications. The search is performed on a user-defined combination of data sheets,
application notes, design notes and Linear Technology
magazine articles. Any data sheet, application note,
design note or magazine article can be downloaded or
faxed back. (Files are downloaded in Adobe Acrobat™
PDF format; you will need a copy of Acrobat Reader to
view or print them. The site includes a link from which
you can download this program.)
Acrobat is a trademark of Adobe Systems, Inc.; Windows
is a registered trademark of Microsoft Corp.; Macintosh is
a registered trademark of Apple Computer, Inc.
International
Sales Offices
World Headquarters
NORTHEAST REGION
Linear Technology Corporation
3220 Tillman Drive, Suite 120
Bensalem, PA 19020
Phone: (215) 638-9667
FAX: (215) 638-9764
Other web sites usually require the visitor to download
large document files to see if they contain the desired
information. This is cumbersome and inconvenient. To
save you time and ensure that you receive the correct
information the first time, the first page of each data
sheet, application note and Linear Technology magazine is recreated in a fast, download-friendly format.
This allows you to determine whether the document is
what you need, before downloading the entire file.
CENTRAL REGION
Linear Technology Corporation
2010 E. Algonquin Road, Suite 209
Schaumburg, IL 60173
Phone: (847) 925-0860
FAX: (847) 925-0878
Linear Technology Corporation
Kenosha, WI 53144
Phone: (414) 859-1900
FAX: (414) 859-1974
NORTHWEST REGION
Linear Technology Corporation
1900 McCarthy Blvd., Suite 205
Milpitas, CA 95035
Phone: (408) 428-2050
FAX: (408) 432-6331
SOUTHWEST REGION
Linear Technology Corporation
21243 Ventura Blvd., Suite 227
Woodland Hills, CA 91364
Phone: (818) 703-0835
FAX: (818) 703-0517
SOUTHEAST REGION
Linear Technology Corporation
17000 Dallas Parkway, Suite 219
Dallas, TX 75248
Phone: (972) 733-3071
FAX: (972) 380-5138
Linear Technology Corporation
15375 Barranca Parkway, Suite A-211
Irvine, CA 92718
Phone: (714) 453-4650
FAX: (714) 453-4765
Linear Technology Corporation
9430 Research Blvd.
Echelon IV Suite 400
Austin, TX 78759
Phone: (512) 343-3679
FAX: (512) 343-3680
© 1997 Linear Technology Corporation/Printed in U.S.A./41K
SINGAPORE
Linear Technology Pte. Ltd.
507 Yishun Industrial Park A
Singapore 2776
Phone: 65-753-2692
FAX: 65-754-4113
FRANCE
Linear Technology S.A.R.L.
Immeuble “Le Quartz”
58 Chemin de la Justice
92290 Chatenay Malabry
France
Phone: 33-1-41079555
FAX: 33-1-46314613
SWEDEN
Linear Technology AB
Sollentunavägen 63
S-191 40 Sollentuna
Sweden
Phone: (08)-623-1600
FAX: (08)-623-1650
GERMANY
Linear Technology GmbH
Oskar-Messter-Str. 24
D-85737 Ismaning
Germany
Phone: 49-89-962455-0
FAX: 49-89-963147
JAPAN
Linear Technology KK
5F NAO Bldg.
1-14 Shin-Ogawa-cho Shinjuku-ku
Tokyo, 162
Japan
Phone: 81-3-3267-7891
FAX: 81-3-3267-8510
KOREA
Linear Technology Korea Co., Ltd
Namsong Building, #403
Itaewon-Dong 260-199
Yongsan-Ku, Seoul 140-200
Korea
Phone: 82-2-792-1617
FAX: 82-2-792-1619
TAIWAN
Linear Technology Corporation
Rm. 602, No. 46, Sec. 2
Chung Shan N. Rd.
Taipei, Taiwan, R.O.C.
Phone: 886-2-521-7575
FAX: 886-2-562-2285
UNITED KINGDOM
Linear Technology (UK) Ltd.
The Coliseum, Riverside Way
Camberley, Surrey GU15 3YL
United Kingdom
Phone: 44-1276-677676
FAX: 44-1276-64851
LINEAR TECHNOLOGY CORPORATION
1630 McCarthy Boulevard
Milpitas, CA 95035-7417
(408) 432-1900 FAX (408) 434-0507
www.linear-tech.com
For Literature Only: 1-800-4-LINEAR
Linear Technology Magazine • November 1997