V18N1 - MARCH

LINEAR TECHNOLOGY
MARCH 2008
VOLUME XVIII NUMBER 1
Zhizhong Hou
Complete IF Receiver
Has 16-Bit, 130Msps ADC,
Fixed-Gain Amplifier
and Antialias Filter in
11.25mm × 11.25mm
µModule Package
Increase I2C or SMBus Data Rate and
Reduce Power Consumption with
Low Power Bus Accelerator .................8
Introduction
IN THIS ISSUE…
COVER ARTICLE
Complete IF Receiver Has
16-Bit, 130Msps ADC, Fixed-Gain
Amplifier and Antialias Filter in
11.25mm × 11.25mm
µModule™ Package .............................1
Todd Nelson
Linear in the News… ...........................2
DESIGN FEATURES
Voltage and Current Monitoring from
7V to 80V in 3mm × 3mm DFN-10 ........5
Sam Tran
6-Input Supervisors Offer Accurate
Monitoring and 125°C Operation ......10
Shuley Nakamura and Al Hinckley
High Power, Single Inductor,
Surface Mount Buck-Boost µModule
Regulators Handle 36VIN, 10A Loads
.........................................................16
Manjing Xie
1.5% Accurate Single-Supply
Supervisors Simplify Part Selection
and Operate to 125°C ........................20
Bob Jurgilewicz and Roger Zemke
Versatile Current Sense Amplifiers
Offer Rail-to-Rail Input, 150°C
Operating Temperature ....................24
William Jett and Glen Brisebois
Compact Hot Swap™ Solution
Simplifies Advanced Mezzanine
Card Design ......................................27
Chew Lye Huat
DESIGN IDEAS
....................................................31–36
(complete list on page 31)
New Device Cameos ...........................37
Design Tools ......................................39
Sales Offices .....................................40
In the design of high speed receivers for communications, test or
instrumentation equipment, several
specialized disciplines converge in
one place—the analog-to-digital converter (ADC). Unfortunately, the ADC
is not a simple black box where an
RF designer applies the signal and a
digital designer retrieves the accurate
output. Careful design of the signal
conditioning circuitry to drive the ADC
is critical. Something as seemingly
straightforward as board layout can
degrade the downstream signal by a
few precious decibels. The problem
is that the disciplines required for
the engineering on either side of the
ADC, namely RF/IF design and digital
design, do not include mastery of the
art of ADC interface design. Someone
has to put in the effort to properly drive
the ADC. But who? Instead of adding
more work to either designer’s plate,
what if the ADC were really a black
box, already loaded with integrated
signal conditioning components in an
optimized layout? Now, that would be
a better solution.
by Todd Nelson
The LTM9001 is built using
Linear Technology’s µModule
technology to create an IC
form factor System-in-aPackage (SiP) that includes
a high speed 16-bit ADC,
antialiasing filter and a low
noise, differential amplifier
with fixed gain. It can
digitize wide dynamic range
signals with an intermediate
frequency (IF) range
up to 300MHz.
The LTM9001 is exactly that black
box. It is built using Linear Technology’s µModule™ technology to create
an IC form factor System-in-a-Package
(SiP) that includes a high speed 16bit ADC, antialiasing ilter and a low
noise, differential ampliier with ixed
gain. It can digitize wide dynamic range
continued on page L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. Adaptive Power, Bat-Track, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView,
µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational
Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, True Color PWM,
UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the
companies that manufacture the products.
L LINEAR IN THE NEWS
Linear in the News…
New µModule Receiver Family Launched
EDN Innovation Award Finalists
Recently, Linear introduced a new family of signal chain
µModule receiver products. The LTM9001, the irst in a
series of System in Package (SiP) signal chain receiver
modules, uses Linear’s breakthrough µModule packaging
technology, now incorporated in a growing family of power
µModule DC/DC controllers. The LTM9001 is a semi-customizable IF/baseband receiver subsystem that includes
a high performance 16-bit Analog-to-Digital converter
(ADC) sampling
up to 160Msps,
an antialiasing
ilter, and ixed
gain differential ADC driver.
The LTM9001
µModule receiver is applicable
in high sensitivity wireless
basestations
and high resolution instrumentation. Systems designers beneit from
simpliied design and test, consistent high performance,
a compact footprint and the elimination of layout-related
performance problems.
Announced in early February, the LTM9001 µModule
receiver has already received coverage in several major
technical publications. The product will be featured in
upcoming cover articles in High Frequency Electronics in
the US, Electronic Product Design in the UK, Elektronik
Informationen in Germany, as well as in articles in key
publications throughout Asia.
The LTM9001 is gaining signiicant interest from receiver
manufacturers as a way reduce time-to-market while
improving functionality. It does so by delivering a high
level of integration without compromising performance.
The device combines RF, digital and mixed-signal technology in a tiny package, precluding the need to call on
applications specialists when a project is underway. All
internal components are optimized for the highest system
performance, with integration and layout issues resolved
in the package. The LTM9001 is a tested, individual package that can be picked and placed easily on the board,
thus reducing the required design time and complexity
normally associated with such functions.
In addition, the LTM9001 has the potential for customization. For orders meeting a minimum size, the LTM9001
can be conigured for various sampling rates and the
differential ADC driver can be substituted for ixed gain
versions ranging from 8dB up to 26dB. As a result, the
LTM9001 signiicantly eases the challenge in designing
high performance communications and instrumentation
systems.
EDN magazine in January announced inalists for the
annual EDN Innovation Awards, which includes several
Linear Technology nominees.
q For Innovator of the Year, EDN nominated Linear cofounder and Chief Technology Oficer Robert Dobkin.
q In the Power ICs category, EDN selected as a inalist
the LT3080 3-terminal adjustable LDO regulator,
which was designed by Bob Dobkin and his team at
Linear Technology.
q For the Analog IC category, the LTC6102 current
sense ampliier.
q For Best Contributed Article, Jim Williams’ article,
“Designing Instrumentation Circuitry with RMS/DC
Converters.”
Visit www.linear.com for complete descriptions and
data sheets for these products. See www.edn.com for Jim
Williams’ article.
2
On the Road in China
Linear Technology is on the road in China, exhibiting in the
4-city IIC Conference & Exhibition. Linear is participating
with a booth at all four conference locations:
q Chengdu—February 28–29, Booth 5D32
q Shenzhen—March 3–4, Booth 2H06
q Beijing—March 6–7, Booth B17
q Shanghai—March 10–11, Booth 4Q09
At the IIC Conference, where overall attendance is expected to exceed 30,000, Linear will highlight a broad range of
products and solutions. These include Power Management
ICs (PMICs), power µModule controllers, LT3080 3-terminal linear regulator, high speed ADCs, the new LTM9001
µModule receiver, high frequency RF products including the
LT5570 RMS
power detector,
the LTC6102
current sense
ampliier, ADC
drivers, DACs,
and LED drivers. L
Linear Technology
will highlight a
broad range of
products at the
IIC Conference
with a booth at all
four conference
locations in China.
Linear Technology Magazine • March 2008
DESIGN FEATURES L
VCC = 3.3V
LTM9001, continued from page 1
VDD = 3.3V
SENSE
LTM9001
0VDD = 0.5V TO 3.6V
D15
IN–
RF
ANTI-ALIAS
FILTER
SAW
•
•
•
16-BIT
130Msps ADC
D0
CLKOUT
IN+
LO
CMOS
OR
LVDS
OF
DIFFERENTIAL
FIXED GAIN
AMPLIFIER
OGND
GND
ENC+ ENC–
ADC CONTROL PINS
Figure 1. A typical application and simplified block diagram of the LTM9001
signals with an intermediate frequency
(IF) range up to 300MHz. Figure 1
shows a typical application.
How is a µModule component
different than a traditional IC? The
µModule construction allows the
LTM9001 to mix standard ADC and
ampliier components regardless of
their process technology and match
them with passive components for
a particular application. The result
is a high performance product with
no process technology compromises
and the potential for semi-custom
adaptations.
What’s Inside?
The µModule receiver consists of wirebonded die, packaged components and
passives mounted on a high performance, 4-layer, Bismaleimide-Triazine
(BT) substrate. BT is similar to other
laminate substrates such as FR4 but
has superior stiffness and a lower
coeficient of thermal expansion.
In time, several different versions
of the LTM9001 will be available. The
LTM9001-AA, as the irst release, is
conigured with a 16-bit, 130Msps
ADC. The ampliier gain is 20dB with
an input impedance of 200Ω and an
input range of ±250mV. The matching network is designed to optimize
the interface between the ampliier
outputs and the ADC inputs under
these conditions. Additionally, there
is a second order bandpass ilter
designed for 162.5MHz, ±25MHz to
prevent aliasing and to limit the noise
from the ampliier.
Linear Technology Magazine • March 2008
Extracting the full performance
from 16-bit, high speed ADCs requires
careful layout as well as good circuit
design. The substrate design carefully
shields sensitive analog traces, maximizes thermal conduction through
multiple ground pads and minimizes
coupled noise by including bypass
capacitors inside the module and close
to the ADC. A common problem with
traditional ADC board layouts is long
traces from the bypass capacitors to
the ADC. The bare die construction
with internal bypass capacitors provides the closest possible decoupling
and eliminates the need for external
bypass capacitors.
The passive ilter network implements an antialias ilter and matches
the ampliier outputs to the ADC inputs. Most communications receiver
applications utilize a highly selective
ilter between the mixer and the ADC
driver. The antialias ilter between
the ADC driver and the ADC inputs
limits the wideband ampliier noise
and helps preserve the high SNR of
the ADC. Printed circuit board (PCB)
layout has a signiicant impact on the
performance even if the circuit topology and component values are correct.
The signal paths must be symmetric
and isolated from the clock inputs and
digital outputs.
The low noise, low distortion ampliier stage provides gain without adding
signiicant noise or distortion to the
signal. Despite the low noise of the
ampliier, the noise is multiplied by the
same gain as the ampliier, so higher
gain unavoidably adds noise to the
system. However, the input range of
the ampliier is proportionately smaller
thanks to the gain and this smaller
input range allows for lower distortion
from the preceding components. The
ampliier inputs present a resistive
200Ω differential input impedance
which is simple to match to most
common, high speed, single-ended or
differential signal paths. This presents
a more straightforward interface than
a switched-capacitor ADC and simpliies the connection to the inal stage
of the RF signal chain.
Why 162.5MHz?
The ADC inside the LTM9001 has
a full power bandwidth of 700MHz
and the ampliier is suitable for input frequencies up to 300MHz, so
why was 162.5MHz chosen for this
irst version? Nyquist theory tells us
that the minimum sample rate for a
given input frequency is twice that
frequency. Working backwards, an
ADC sampling at 130Msps can capture a frequency range up to 65MHz
wide. Undersampling allows us to
move that frequency range. Hence
the irst Nyquist zone is DC – 65MHz,
the second is 65MHz to 130MHz, the
third is 130MHz to 195MHz, and so
on, see Figure 2.
The LTM9001-AA is intended for
instrumentation applications. In such
applications, the linearity and dynamic
range requirements are extremely
high. Traditional instruments utilize
preselectors and multiple down-con3
L DESIGN FEATURES
NYQUIST ZONE 1
DC TO 65MHz
DC
CENTER = 32.5MHz
NYQUIST ZONE 2
65MHz TO 130MHz
65MHz
CENTER = 97.5MHz
NYQUIST ZONE 3
130MHz TO 195MHz
fSAMPLE
CENTER = 162.5MHz
NYQUIST ZONE 4
195MHz TO 260MHz
195MHz
CENTER = 227.5MHz
260MHz
Figure 2. Nyquist zones for 130MHz sample rate
version stages to place the band of
interest at DC. With the advent of high
performance ADCs capable of undersampling, modern instruments are able
to eliminate the inal down-conversion
stage without sacriicing performance.
The LTM9001-AA coniguration selects the third Nyquist zone with the
bandpass ilter set squarely in the
middle of the zone.
More than Just
a Buffered ADC
The sample-and-hold front end of
discrete ADCs presents a complex
charge/discharge proile to the drive
circuitry. Ideally, the input circuitry
should be fast enough to fully charge
the sampling capacitor during the
sampling period (half of the clock
period), but this is not always possible and the incomplete settling may
degrade the SNR and SFDR. Some
manufacturers promote a “buffered”
ADC as a solution but this falls short
of addressing the system-level solution
since a low distortion ampliier is still
required to provide the full-scale input
to the ADC.
From the system view, the ADC
follows the RF and IF portions of the
receiver chain and converts the signal
to a digital format. The signal comes
from the antenna with very little power.
0
AMPLITUDE (dBFS)
–20
–40
–60
HD3
–80
HD2
–100
–120
0
12288
20480
28672
4096
FFT BIN NUMBER (32k TOTAL)
Figure 3. An FFT of the LTM9001 at 160MHz
input frequency with the randomizer on
4
The signal must be iltered and ampliied through each stage. Ampliication
(gain) increases the total noise and
reduces the headroom, which generally causes more distortion. The added
distortion may be addressed with a
higher supply voltage or a higher power
ampliier, neither of which is preferable. Therefore, from the system-level
point of view, an ADC with a small
input range is better.
The LTM9001 meets these systemlevel criteria. The resistive ampliier
inputs are easily matched and it has
an input range of ±250mV, enabling
the use of low OIP3 components or
higher loss SAW ilters. The noise of
the ampliier is low enough that the
SNR of the LTM9001 is good despite
the high gain (see Figure 3).
Working with a
µModule Receiver
The LTM9001 uses a land grid array
(LGA), which provides higher pin density than dual in-line or quad packages
and better thermal conduction than
BGA packages. The high integration
of the LTM9001 makes the PCB board
layout simple. The multilayer substrate
allows greater lexibility in pin placement on the package relative to pin
placement on the die. The LTM9001
has been optimized for a low-through
layout so that the interaction between
inputs, clock and digital outputs is
minimized. The analog and clock inputs are surrounded by ground pads
and a continuous row of ground pads
further separate the analog and digital
signal lines. However, to optimize its
electrical and thermal performance,
some layout considerations are still
necessary. See the actual evaluation
board in Figure 4.
Use large PCB copper areas for
ground. This helps to dissipate heat
through the board and also helps
to shield sensitive on-board analog
signals. Common ground (GND) and
output ground (OGND) are electrically
isolated on the LTM9001, but for most
digital output conigurations should
be connected on the PCB underneath
the part to provide a common return
path.
Use multiple ground vias. Using as
many vias as possible helps to improve
the thermal performance of the board
and creates necessary barriers separating analog and digital traces on the
board at high frequencies. Take care to
separate analog and digital traces as
much as possible, using vias to create
high frequency barriers. This reduces
digital feedback that can reduce the
signal-to-noise ratio (SNR) and dynamic range of the LTM9001.
Conclusion
µModule technology, introduced irst
by Linear Technology for DC/DC converters, now brings the advantages of
small size, higher integration and ease
of use for high speed ADC applications.
By integrating ine-line CMOS and
SiGe components with appropriate
passive networks, the challenging
task of matching a ixed gain ampliier to a high speed ADC is done. All is
reduced to an easy-to-use black box:
the LTM9001. L
Figure 4. An evaluation board shows the
small overall circuit. Note that no external
components are required.
Linear Technology Magazine • March 2008
DESIGN FEATURES L
Voltage and Current Monitoring from
7V to 80V in 3mm × 3mm DFN-10
by Zhizhong Hou
Introduction
Accurate power supply voltage and
current monitoring is increasingly
important in everything from industrial and telecom applications to
automotive and consumer electronics.
A complete power monitoring system
typically includes a sense resistor, a
precision ampliier, an analog to digital
converter (ADC) and a proper interface
to report data to a host controller. The
LTC4151 and LTC4151-1 combine all
of these components (except the sense
resistor) into one IC, resulting in a full
featured, rugged and simple-to-use
solution for accurate high side current sensing and voltage monitoring
(see Figure 1).
The LTC4151 and
LTC4151-1 offer the benefits
of high side current sensing
without any of the usual
complexity. Each integrates
a precision high voltage
amplifier and associated
level shift circuit for high
side current sensing, a
precision voltage divider for
supply voltage monitoring,
a 12-bit ADC and an I2C
interface—all in small MS10
or tiny 3mm × 3mm
DFN-10 packages.
High Side vs
Low Side Sensing
In a power monitoring system, the
sense resistor can be placed either
between the system ground and the
load (low side sensing) or between
the system supply and the load (high
side sensing). For many applications,
high side sensing is desirable, but
it is traditionally more dificult to
implement.
Low side sensing is relatively simple
in concept and design, but a low side
sense resistor loats the load above
system ground. Thus, the ground
potential seen by the load varies
with changing load current. This can
result in the load seeing signiicant
ground noise during transient spiking
load currents. Worse yet, a failed or
disconnected low side sense resistor
causes the load ground to be charged
to the full supply voltage, presenting
a potential safety hazard.
High side sensing avoids these problems, but requires a number of high
performance devices and interfaces.
For instance, a robust high side sense
ampliier is required to withstand
high supply voltage or high voltage
transients. Also, a precision level shift
circuit is needed to accurately translate the large supply-referred signals
to appropriate ground level signals
for the ADC.
Full Featured High Side,
High Voltage Digital Monitors
The LTC4151 and LTC4151-1 offer the
beneits of high side current sensing
without any of the usual complexity, plus they provide supply voltage
monitoring in the same package. Each
0.02Ω
VIN
7V TO 80V
2
1
VIN
10
SENSE+
SENSE–
LTC4151
3
+
ADR1
5V
8
–
SHDN
25X
20k
VDD
20k
VREF = 2.048V
6
1
50
MUX
12-BIT ADC
I2C
6V
µCONTROLLER
7
4
6V
SCL
SCL
SDA
SDA
ADR0
9
GND
5
ADIN
Figure 1. Full featured current and voltage monitor simplifies high voltage, high side sensing.
Linear Technology Magazine • March 2008
5
1.0
2
0.5
1
ADC INL (LSB)
ADC INL (LSB)
L DESIGN FEATURES
0
–0.5
–1.0
0
–1
0
1024
2048
3072
–2
4096
0
1024
CODE
2048
3072
4096
CODE
Figure 2. Typical INL error of ADIN
voltage is within ±0.5LSB.
Figure 3. Typical INL error of current
sense voltage is within ±1LSB.
integrates a precision high voltage
ampliier and associated level shift
circuit for high side current sensing,
a precision voltage divider for supply
voltage monitoring, a 12-bit ADC and
an I2C interface—all in small MS10 or
tiny 3mm × 3mm DFN-10 packages.
A dedicated ADIN pin is directly connected to the ADC input for monitoring
any external voltage. See Figure 1 for
a simpliied block diagram.
Using the I2C interface, the parts
can be conigured into either a continuous scan mode (default upon power
up) or a snapshot mode. In continuous scan mode, the parts repeatedly
measure three voltages in sequence:
the differential high side sense voltage
between the SENSE+ and SENSE– pins,
the supply voltage at the VIN pin and
an external voltage at the ADIN pin
at a refreshing frequency of 7.5Hz. In
snapshot mode, the host controller
can instruct the parts to perform a
one-time measurement of a speciic
signal. The conversion time of SENSE
voltage is 67ms and that of VIN and
ADIN voltages is 33ms. Thanks to the
oversampling Sigma-Delta ADC, any
ripples within each conversion cycle
are simply averaged out.
Easy to Use
Figure 1 shows just how easy it is to
put together a complete voltage and
high side current monitor. The only
required external components are a
sense resistor and two pull-up resistors (with the SHDN pin loat and ADIN
pin tied to GND).
The LTC4151 and the LTC4151-1
maintain high precision for supplies
from 7V to 80V, an ideal range for
applications with 12V, 24V or 48V
supply voltages. The absolute maximum voltages of the supply pin and
the two sense input pins are all rated
at 90V, which helps the part survive
high voltage transients. This wide
input voltage range allows the part to
be directly connected to high voltage
supplies without the need of a secondary supply, unlike many other supply
monitors.
The LTC4151 and the LTC4151-1
can be conigured with one of nine I2C
addresses via the ADR1 and ADR0
pins (high, low or open). These two
pins are also rated at an absolute
maximum voltage of 90V, again precluding the need for a separate low
voltage supply.
Wide Dynamic Range
and High Accuracy
LTC4151 and LTC4151-1 each
combine a precision high side sense
ampliier and a true 12-bit ADC. The
result is a current and voltage monitor that offers a unique combination
of high resolution and wide dynamic
range. The full scale of the current
sense voltage is 81.92mV with a resolution of 20µV/LSB. The full scale of
the supply voltage is 102.4V with a
resolution of 25mV/LSB. The full scale
at ADIN is 2.048V with a resolution of
500µV/LSB. As Figures 2 and 3 show,
the typical integral nonlinearity errors
RS
0.02Ω
VIN
48V
3.3V
1
R1
20k
10
SENSE+
SENSE–
SCL
2
4
R3
5.1k
R4
0.51k
MOCD207M
6
8
1
7
7
6
2
3
5
4
R5
0.51k
R6
10k
R7
10k
VIN
LTC4151-1
3
R2
20k
SDAI
ADR1
SDA0
ADR0
ADIN
SCL
8
5
VADIN
µCONTROLLER
MOCD207M
1
5
2
3
6
7
4
8
SDA
GND
9
VDD
Figure 4. The LTC4151-1 makes it easy to implement optoisolation.
6
Linear Technology Magazine • March 2008
DESIGN FEATURES L
(INLs) of the ADIN voltage and the
current sense voltage are both within
±1LSB. In addition, the current sense
voltage, the supply voltage and the
ADIN voltage are all measured with
high accuracy at the full scale (1.25%,
1% and 1%, respectively) over the full
industrial temperature range.
F1
VIN1
48V
RS
0.02Ω
F2
VIN2
48V
D2
D3
D4
R1
150k
R2
301k
SCL
LTC4151
R3
3.4k
I2C
SDA
ADR1
V+
LOAD
V–
ADR0
GND
GND
D1, D2, D3, D4: MBRM5100
CONDITION*
RESULT
NADIN ≥ 1.375 • NVIN
NORMAL OPERATION
0.835 • NVIN ≤ NADIN < 1.375 • NVIN
F2 IS OPEN
0.285 • NVIN ≤ NADIN < 0.835 • NVIN
F1 IS OPEN
(I2C NOT RESPONDING)
BOTH F1 AND F2 ARE OPEN
* VIN1 and VIN2 differ by less than 20%. NADIN and NVIN are digital
codes measured by the ADC at the ADIN and VIN pins, respectively.
Figure 5. A single LTC4151 monitors current, supply voltage and fuses.
SDAI and SCL, all pull-up resistors
on these three pins can be directly
connected to the high voltage supply,
eliminating the need for a separate low
voltage pull-up supply.
ADIN Pin is Useful for
Fuse Monitoring and
Temperature Sensing
The LTC4151 and the LTC4151-1 feature a dedicated ADIN pin that can be
used to monitor any external voltage.
Figure 5 shows a simple circuit that
not only measures current and supply voltage but also monitors a pair of
fuses on the high side.
The fuses are monitored by comparing the voltages at the VIN and ADIN
0.2Ω
VIN
48V
SENSE+ SENSE –
VIN
250mA
LOAD
I 2C
40.2k
1%
LTC4151
ADIN
SDA
ADR1
GND
pins. ADIN is connected to the two
inputs after the fuses through a Y
divider. Diodes D3 and D4 compensate
the diode-OR D1 and D2. The voltage
at ADIN varies as the status of the
fuses changes, as shown in the table
in Figure 5. Since the ADIN voltage is
approximately ratiometric to VIN, the
results are independent of the supply
seen at VIN. The limitation of this circuit
is that the two inputs must remain
within 20% of each other.
The ADIN pin can also be used to
monitor board temperature with an
NTC thermistor as shown in Figure 6.
In that circuit, VIN is connected on the
downstream side of the sense resistor
so that the quiescent current of the
LTC4151 is measured.
Conclusion
SCL
1.5k
1%
SENSE–
ADIN
The LTC4151 features a SHDN pin with
an internal 5µA pullup. When SHDN
is tied to GND, the part enters shut
down mode and the typical quiescent
current is reduced to 120µA at 12V,
about 10% of the normal operating
current (1.2mA). In applications with
battery supplies, one can use this pin
to save power consumption.
The LTC4151-1 trades in the SHDN
pin for an inverted SDAO pin to enable a simple optoisolation scheme.
Optoisolation is inevitably required in
applications where the host controller
sits at a different ground level from
the power monitor. The LTC4151-1
makes this job easy with split SDA
pins: the SDAI (data input) pin and a
unique SDAO (inverted data output)
pin. In addition, the SCL and the SDAI
pins each have an internal 6V clamp
(sinking up to 5mA current).
When using optoisolators with the
LTC4151-1, connect the SCL and SDAI
pins to the outputs of the incoming
optoisolators and connect the SDAO
pin to the anode of the outgoing optoisolator, as shown Figure 4. With
the outgoing optoisolator clamping the
SDAO and the internal 6V clamps on
100k AT 25°C
1%
SENSE+
VIN
Power-Saving Shutdown
or Easy Optoisolation?
You Choose.
VISHAY
2381 615 4.104
D1
ADR0
High side current sensing and voltage
monitoring could not be easier than
with the LTC4151 and the LTC4151-1
supply monitors. Their wide supply
range and high level of integration
simpliies design, while desirable features, such as 12-bit resolution, high
accuracy, I2C interface, optoisolation
support and small footprints make
them an easy it in a wide variety of
applications. L
Figure 6. Temperature monitoring is simple with LTC4151 and an NTC thermistor.
Linear Technology Magazine • March 2008
7
L DESIGN FEATURES
Increase I2C or SMBus Data Rate and
Reduce Power Consumption with
by Sam Tran
Low Power Bus Accelerator
Introduction
I2C and SMBus 2-wire buses use
simple open-drain pull-down drivers
with resistive or current source pullups. Communications protocols in
these systems allow multiple devices
to drive and monitor the bus without
bus contention, creating a robust
communications link. Unfortunately,
as systems trend towards higher complexity and lower supply voltages, the
advantages gained by the simplicity
of the open-drain pull-down protocol
are offset by the disadvantages of
increased rise times and greater DC
bus power consumption.
As designs require higher reliability
and a greater number of features, the
number of peripherals attached to the
I2C or SMBus system increases. Some
systems extend the bus to edge connectors where I/O cards with additional
peripherals are removed and inserted
onto the bus. The added peripherals
directly increase the equivalent capacitance on the bus, slowing rise times.
Slow rise times can seriously impact
data reliability and limit the maximum
practical bus speed to well below the
established I2C or SMBus maximum
transmission rate. Rise times can be
improved by using lower bus pull-up
resistor values or higher ixed current
source values, but the additional bus
pull-up current raises the low state
bus voltage, VOL, as well as the DC bus
power consumption. Another issue in
systems with swappable I/O cards is
ESD susceptibility.
The LTC4311 bus accelerator addresses all of these issues. It comes
in a tiny 2mm × 2mm DFN or SC70
package and operates over a wide
power supply range of 1.6V to 5.5V,
making it easy to it in any number
of applications.
Figure 1 shows a typical low voltage application circuit. The LTC4311
8
VCC
1.6V
VCC
C1
0.01µF
VCC
1.6V
LTC4311
BUS1
ENABLE
GND
10k
10k
BUS2
I2C
SCL
SDA
CLK
IN
DATA
IN
CLK
IN
DATA
IN
CLK
OUT
DATA
OUT
CLK
OUT
DATA
OUT
DEVICE 1
DEVICE N
Figure 1. Typical LTC4311 low voltage application circuit
provides strong slew rate controlled
pull-up currents on the bus for
smooth, controlled transitions during
rising edges to decrease rise times in
highly capacitive systems, as shown
in Figure 2. The LTC4311’s slew rate
controlled pull-up currents are strong
enough to allow I2C or SMBus systems
to achieve switching frequencies up
to 400kHz for bus capacitances in
excess of 1nF. In addition, because
the accelerator pull-up impedance
is signiicantly lower than the bus
pull-up resistance, the system has
greater immunity to noise on rising
edges.
1V/DIV
LTC4311
RPULL-UP = 15.8k
VCC = 5V
CLD = 200pF
fI2C = 100kHz
1µs/DIV
Figure 2. Comparison of I2C waveform
for the LTC4311 vs resistive pull-up
The LTC4311’s strong pull-up currents allow users to choose larger bus
pull-up resistor values to reduce VOL,
DC bus power consumption and fall
times, while still meeting rise time and
switching frequency requirements.
This is especially helpful for 2-wire
systems where devices require resistances in series with their pull-down
devices for ESD protection, since
VOL on these devices is reduced with
larger bus pull-up resistor values. The
larger bus pull-up resistor values are
also beneicial in systems operating
at bus supplies below 2.7V, where
VOL can be reduced well below the
I2C speciication, thereby increasing
noise margins.
For I2C or SMBus systems where
large numbers of I/O cards can be
inserted and removed, the LTC4311’s
slew rate controlled pull-up currents
properly address rise time issues
despite large variations in bus capacitance. The controlled slew rate
regulates the rise rate of the bus to
50V/µs–100V/µs, independent of bus
capacitance.
With very light loads, as occurs
when some or all cards are removed,
no relections occur on the bus due
Linear Technology Magazine • March 2008
DESIGN FEATURES L
to the slew rate controlled nature of
the pull-up currents. When the bus is
heavily loaded, the LTC4311 provides
strong, controlled pull-up currents
to signiicantly decrease rise times
on the bus for capacitive loads well
beyond 1nF.
All of these features, coupled with
high ±8kV HBM ESD ruggedness,
make the LTC4311 ideally suited, and
in many cases necessary, for I2C or
SMBus systems having large numbers
of removable I/O cards.
5mA
BUS1
VCC
SLEW RATE
DETECTOR
5mA
BUS2
Circuit Operation
Figure 3 shows a functional block
diagram of the LTC4311. The LTC4311
consists of two independent but identical circuits for each bus, consisting of
a slew rate detector, two voltage comparators, and a slew rate controlled
bus pull-up current.
The slew-rate detector monitors
the bus and activates the accelerators
only when the bus rise rate is greater
than 0.2V/µs. This ensures that the
accelerators never turn on when the
bus voltage is in a DC state or falling.
The irst voltage comparator is used to
hold off the accelerator until the bus
voltage exceeds a threshold voltage,
VTHR. For supply voltages below 2.7V,
VTHR is supply dependent, deined as
0.3 • VCC. At higher supply voltages,
VTHR is a constant 0.8V. This optimizes
the LTC4311 for use in low voltage
systems, while offering rise time acceleration over a larger voltage range
for I2C and SMBus systems operating
at bus voltages above 2.7V.
Once both conditions are met, the
slew limited bus accelerator is enabled
to quickly slew the bus. An internal
slew rate comparator monitors the bus
rise rate and controls the accelerator
pull-up current to limit the bus rise
rate to 50V/µs–100V/µs, independent
of the bus capacitance. A second voltage comparator disables the pull-up
current when the bus is within 400mV
of the bus pull-up supply.
For systems where a single bus accelerator is not suficient to meet the
rise time requirement, additional bus
accelerators can be added in parallel
to further decrease the rise time.
continued on page 2
Linear Technology Magazine • March 2008
SLEW RATE
DETECTOR
+
VTHR
–
CONTROL
LOGIC AND
INTERNAL SLEW
COMPARATOR
+
VTHR
–
+
VCC – 0.4
–
+
VCC – 0.4
–
+
ENABLE
GND
–
1V
Figure 3. LTC4311 functional block diagram
VCC
2.5V
VCC
C1
0.01µF
VCC
2.5V
LTC4311
OFF ON
BUS1
R1
10k
ENABLE
GND
R2
10k
BUS2
I2C
SCL
SDA
CLK
IN
DATA
IN
CLK
IN
DATA
IN
CLK
OUT
DATA
OUT
CLK
OUT
DATA
OUT
DEVICE 1
DEVICE N
Figure 4. Typical LTC4311 application with low current shutdown
9
L DESIGN FEATURES
6-Input Supervisors Offer Accurate
Monitoring and 125°C Operation
by Shuley Nakamura and Al Hinckley
Introduction
The latest trio of power supply supervisors from Linear Technology is ideal
for today’s multi-voltage systems that
require accurate supply monitoring.
The LTC2930, LTC2931, and LTC2932
are 6-input voltage monitors capable of
maintaining 1.5% threshold accuracy
from –40°C to 125°C. The combination of monitored supply voltages is
set by a single pin. Each part offers
16 threshold voltage combinations,
thus meeting the needs of almost
any multi-voltage system. This programmability eliminates the need to
qualify, source and stock unique part
numbers for different threshold voltage
combinations.
The overall architecture and operating speciications of these three
devices are similar, but each has
unique features (see Table 1). The
LTC2930 generates a reset after
any undervoltage event or when the
manual reset input (MR) pulls low. It
is ideal for space-constrained applications as it comes in a compact 3mm
× 3mm 12-lead DFN package. The
LTC2931 includes a watchdog input
(WDI), a watchdog output (WDO) and
user-adjustable watchdog periods to
enable microprocessor monitoring
and control. The LTC2932 can vary
its monitor thresholds from 5% to
12.5%, and a reset disable pin provides margining capability. Both the
LTC2931 and LTC2932 are packaged
in 20-pin TSSOP packages and have
separate comparator outputs, enabling individual supply monitoring
and/or sequencing.
LTC2931
VREF
VPG
GND
Table 1. LTC2930, LTC2931, LTC2932 feature summary
Feature
LTC2930
LTC2931
LTC2932
Conigurable Input
Threshold Combinations
16
16
16
Threshold Accuracy
1.5%
1.5%
1.5%
Adjustable Reset Time
L
L
L
Buffered Reference
L
L
L
L
L
Individual Comparator
Outputs
Manual Reset
Independent Watchdog
Circuitry
L
Supply Tolerance
Fixed, 5%
Fixed, 5%
User Selectable
5%, 7.5%, 10%, 12.5%
Package
12-lead
3mm × 3mm DFN
20-lead
F Package
20-lead
F Package
Single Pin Configuration
Makes Life Easy
These supervisors offer an elegant
method of coniguring the input voltage thresholds. Figure 1 shows how a
single resistive divider at the VPG pin
sets the supervisor into one of the 16
threshold options shown in Table 2.
See the data sheet for suggested modesetting resistor values.
The actual thresholds are set by
integrated precision dividers for 5V,
3.3V, 3V, 2.5V, 1.8V, and 1.5V supply
monitoring. For other supply values,
uncommitted comparators with 0.5V
thresholds allow virtually any positive
supply to be monitored using a resistive divider, as shown in Figure 2a.
The V4 input also monitors negative
voltages—with the same 1.5% accuracy—using the integrated buffered
reference for offset (see Figure 2b).
VTRIP
V3, V4,
V5 OR V6
LTC2931
R3
1%
VREF
R4
1%
R4
1%
LTC2931
V4
R3
1%
VTRIP
0.5V
R2
1%
2a.
Figure 1. Mode selection
10
L
Reset Disable
+
–
R1
1%
L
2b.
Figure 2. Using a resistive divider to set the voltage trip point
Linear Technology Magazine • March 2008
DESIGN FEATURES L
What Does Threshold
Accuracy Mean?
Table 2. Voltage threshold modes
Consider a 5V system with ±5% supply tolerance. The 5V supply may vary
between 4.75V to 5.25V. System ICs
powered by this supply must operate
reliably within this band (and a little
more, as explained below). A perfectly
accurate supervisor for this supply
generates a reset at exactly 4.75V.
However, no supervisor is this perfect.
The actual reset threshold of a supervisor luctuates over a speciied band;
the LTC2930, LTC2931 and LTC2932
vary ±1.5% around their nominal
threshold voltage over temperature
(Figure 3). The reset threshold band
and the power supply tolerance bands
should not overlap. This prevents false
or nuisance resets when the power
supply is actually within its speciied
tolerance band.
The LTC2930, LTC2931 and
LTC2932 boast a ±1.5% reset threshold accuracy, so a “5%” threshold is
usually set to 6.5% below the nominal
input voltage. Therefore, a typical
5V, “5%” threshold is 4.675V. The
threshold is guaranteed to lie in the
band between 4.750V and 4.600V
over temperature. The powered system must work reliably down to the
low end of the threshold band, or risk
malfunction before a reset signal is
properly issued.
A less accurate supervisor increases
the required system voltage margin
and increases the probability of system
malfunction. The tight ±1.5% accuracy
speciication of the LTC2930, LTC2931
5V
V1 (V)
V2 (V)
V3 (V)
V4 (V)
V5 (V)
V6 (V)
5.0
3.3
2.5
1.8
ADJ
ADJ
5.0
3.3
2.5
1.5
ADJ
ADJ
5.0
3.3
2.5
ADJ
ADJ
ADJ
5.0
3.3
1.8
ADJ
ADJ
ADJ
5.0
3.3
1.8
–ADJ
ADJ
ADJ
5.0
3.3
ADJ
ADJ
ADJ
ADJ
5.0
3.3
ADJ
–ADJ
ADJ
ADJ
5.0
3.0
2.5
ADJ
ADJ
ADJ
5.0
3.0
1.8
ADJ
ADJ
ADJ
5.0
3.0
ADJ
ADJ
ADJ
ADJ
3.3
2.5
1.8
1.5
ADJ
ADJ
3.3
2.5
1.8
ADJ
ADJ
ADJ
3.3
2.5
1.8
–ADJ
ADJ
ADJ
3.3
2.5
1.5
ADJ
ADJ
ADJ
3.3
2.5
ADJ
ADJ
ADJ
ADJ
3.3
2.5
ADJ
–ADJ
ADJ
ADJ
and LTC2932 improves the reliability
of the system over supervisors with
wider threshold speciications.
Glitch Immunity =
No Spurious Resets!
Monitored supply voltages are far from
being ideal, perfectly lat DC signals.
Riding on top of these supplies are
high frequency components caused
by a number of sources such as the
output ripple of the power supply or
SUPPLY TOLERANCE
MINIMUM
RELIABLE
SYSTEM
VOLTAGE
IDEAL
SUPERVISOR
THRESHOLD
4.75V
±1.5%
THRESHOLD
BAND
NOMINAL
SUPPLY
VOLTAGE
–5%
4.675V
–6.5%
4.6V
–8%
REGION OF POTENTIAL MALFUNCTION
Figure 3. Tight 1.5% threshold accuracy yields high system reliability
Linear Technology Magazine • March 2008
coupling from other signals. If the
monitored voltage is near or at the reset threshold voltage, this noise could
cause spurious resets. Fortunately,
the LTC2930, LTC2931 and LTC2932
have been designed with this potential
issue in mind, so spurious resets are
of little to no concern.
Some supply monitors overcome
spurious resets by adding hysteresis
to the input comparator. The amount
of applied hysteresis is stated as
a percentage of the trip threshold.
Unfortunately, this degrades monitor
accuracy because the true accuracy
of the trip threshold is now the percentage of added hysteresis plus the
advertised accuracy of the part. The
LTC2930, LTC2931 and LTC2932 do
not use hysteresis, but instead use
an integration scheme that requires
transients to possess enough magnitude and duration to switch the
comparators. This suppresses spurious resets without degrading the
monitor accuracy.
The COMP5 comparator output
response to a “noisy” input on the
LTC2931 is demonstrated in Figure 4.
11
L DESIGN FEATURES
–2mV DC STEP APPLIED HERE
V5
100mV/DIV
500mV
VRT
Vn
100mVP–P
tRST
tUV
300µs PROPAGATION DELAY
COMP5
RST
COMPn
Figure 4. Comparator output is
resistant to noisy input voltage
In the example shown, a 500kHz,
100mV P–P sine wave centered at
500mV is applied to the V5 input. The
threshold voltage of the adjustable
input, V5, is 500mV. Even though
the signal amplitude goes as low as
450mV, COMP5 remains high. Next,
the DC level of the input is dropped
2mV. In response, COMP5 pulls low
and remains low. As mentioned earlier,
only transients of long enough duration
and magnitude trigger the comparator
output to pull high or low.
Adjustable Reset Timeout Period
for Varied Application Needs
Each of the supervisors includes an
adjustable reset timeout period, tRST.
Once all the inputs are above their
threshold values, the reset timer is
started (Figure 5). RST stays low for
Figure 5. RST timing diagram
the duration of tRST and remains low
as long as the time between transients
is less than the reset timeout. In other
words, the reset timeout prevents supply transients with frequencies greater
than 1/tRST from causing undesired
toggling at the RST output. Keeping
RST low during these supply transients
suppresses spurious resets.
The reset timeout period is adjustable to accommodate a variety of
microprocessor applications. Conigure the reset timeout period, tRST, by
connecting a capacitor, CRT, between
the CRT pin and GND. The value of
this capacitor is determined by
CRT =
(
)
tRST
= 500 pF ms • tRST
2MΩ
Leaving the CRT pin unconnected
generates a minimum reset timeout of
approximately 25µs. Maximum reset
timeout is limited by the largest available low leakage capacitor.
Additional Glitch Filtering
Even though all six comparators
have built-in glitch iltering, adding
bypass capacitors on the V1 and V2
inputs is recommended, because of
these two, the input with the higher
voltage functions as VCC for the entire
chip. Additional ilter capacitors may
be added to the V3, V4, V5 and V6
inputs if needed to suppress troublesome noise.
Open-Drain Reset
The RST outputs on the LTC2930,
LTC2931 and LTC2932 are opendrain and contain weak pull-up
current sources to the V2 voltage.
5V
10k
5V
3V
12V
3.3V
1.8V
–5.2V
1V
0.9V
8V
6V
V1
68.1k
1%
86.6k
1%
487k
1%
0.1 F
1020k
1%
V2
V3
V4
RST
1400k
1%
V1
2150k
1%
V2
V3
0.1 F
V4
LTC2930
V5
121k
1%
100k
1%
100k
1%
R1A
16.2k
1%
R2A
86.6k
1%
RST
V5
V6
V6
MR
MR
VREF
VPG
RESET
LTC2930
10k**
GND
CRT
CRT
47nF
100k
1%
MANUAL RESET
PUSHBUTTON
100k
1%
R1B
100k40.2k
1% 1%
R2B
59k
1%
VREF
VPG
GND
CRT
CRT
47nF
tRST = 94ms
**OPTIONAL FOR EXTENDED ESD TOLERANCE
Figure 6. Wired-OR system reset
12
Linear Technology Magazine • March 2008
DESIGN FEATURES L
12V (9.6V THRESHOLD)
LTM4600
VIN
VOUT
5V
5V
RUN
VIN
VIN
LTC2950-1
VIN
INT
PB
LT3028
VOUT 1.8V
1.8V
SYSTEM
LOGIC
LTC3704
VIN
VOUT –5.2V
10k
3.3V
SHDN
10k
KILL
10k
SHDN
10k
EN
10k
LT3028
VOUT 3.3V
–5.2V
RUN/UVLO
VIN
10k
LT3028
VOUT 2.5V
2.5V
SHDN
DONE
12V
SUPPLY
STATUS
COMP1 COMP2 COMP3 COMP4 COMP5 COMP6
V1
10k
0.1 F
365k
1%
487k
1%
V2
T0
V3
T1
V4
1820k
1%
V5
LTC2932
RDIS
RST
V6
100k
1%
100k
1%
121k
1%
VREF
R1
16.2k
1%
VPG
CRT
GND
4.7nF
tRST = 9.4ms
R2
86.6k
1%
Figure 7. Five supply 12.5% tolerance power-up sequencer with pushbutton
The open-drain structure provides
many advantages. For instance, each
of these outputs can be externally
pulled-up to voltages higher than V2
using a pull-up resistor. This facilitates
the use of multiple devices operating
under different I/O voltages. In addition, multiple open-drain outputs can
be conigured in a “wired-OR” format
where the outputs are tied together.
Figure 6 showcases two LTC2930
supervisors, whose open-drain RST
outputs are tied together and pulledup to 5V via a 10k pull-up resistor.
If one RST output pulls low due to a
reset event, it sinks current and pulls
the other output low.
Linear Technology Magazine • March 2008
Comparator Outputs Enable
Individual Supply Monitoring
and Sequencing Support
Real-time comparator outputs on both
the LTC2931 and LTC2932 indicate
the status of the individual inputs.
Similar to the RST output, the comparator outputs are also open-drain
and have weak pull-up current sources
to the V2 voltage.
While RST pulls low when an
undervoltage event occurs on any of
the monitored supplies, a comparator
output pulls low only when its counterpart input is below its threshold
voltage. The ability to monitor the
status of each supply is useful in
multi-voltage systems where it is
important to know which particular
supply has failed.
The individual comparator outputs
also allow power supply sequencing.
Figure 7 shows the LTC2932 in a
5-supply power-up sequencer. As
an input reaches its threshold, the
respective comparator output pulls
high and enables the next DC/DC
converter.
The LTC2950-1 is used to provide
pushbutton control for the sequencer.
After the pushbutton is pressed, the
LTC2950-1 pulls the RUN pin of the
LTM4600 high. Subsequently, the
LTM4600 generates a 5V output which
13
L DESIGN FEATURES
supplies power to each of the four
DC/DC converters.
VRT
Vn
Three Supervisor Flavors
tMRD
tRST
tUV
tRST
RST
LTC2930: Manual Reset (MR)
Forces RST Low
Use the manual reset input (MR) on
the LTC2930 to issue a forced reset,
independent of input voltage levels. A
10µA (typical) internal current source
pulls the MR pin to VCC. A logic low on
this pin pulls RST low. When the MR
pin returns high, RST returns high
after the selected reset timeout period
has elapsed, assuming all six voltage
inputs are above their thresholds
(Figure 8). The input-high threshold
on the MR pin is 1.6V (max), allowing
the pin to be driven by low voltage
logic as well.
MR
tMRW
Figure 8. MR timing diagram
timeout period, tWD, is adjusted by
connecting a capacitor, CWT, between
the CWT pin and ground. The value of
this capacitor is determined by
C WT =
(
)
t WD
= 50 pF ms • t WD
20MΩ
Leaving the CWT pin unconnected
generates a minimum watchdog timeout of approximately 200µs. Maximum
watchdog timeout is limited by the largest available low leakage capacitor.
LTC2931: Monitor a Microprocessor
with the Watchdog Function
The LTC2931’s independent watchdog
circuitry monitors a microprocessor’s
activity. The microprocessor is required to change the logic state of the
WDI pin on a periodic basis in order to
clear the watchdog timer. The LTC2931
consists of a watchdog input (WDI), a
watchdog output (WDO) and a timing
pin (CWT), which allows for a user
adjustable watchdog timeout period.
Figure 9 illustrates the watchdog timer
and its relationship to the reset timer
and WDI.
The watchdog timeout period is
adjustable and can be optimized for
software execution. The watchdog
LTC2932: Margining Capabilities
and Wider Threshold Tolerances
In high reliability system manufacturing and testing, it is important to
verify that the components will conTable 3. LTC2932 Tolerance Selection
T0
T1
TOLERANCE
(%)
VREF
(V)
Low
Low
5
1.210
Low
High
7.5
1.175
High
Low
10
1.146
High
High
12.5
1.113
tinue to operate at or below the rated
power supply tolerance. Veriication
usually involves margining the power
supplies, running their outputs at or
beyond rated tolerances. The LTC2932
is designed to complement such testing
in two ways. First, the RST output can
be disabled by pulling RDIS low. In this
state, the RST output remains high
despite any undervoltage events which
may occur during margining tests. This
does not affect the individual supply
monitoring, which is independent of
the logic state of RDIS. Second, lowering the trip thresholds can increase
supply headroom to match the margining ranges. This is simply a matter of
changing the two tolerance selection
inputs, T0 and T1, to adjust the global
supply tolerance to 5%, 7.5%, 10%, or
12.5% (see Table 3).
Automotive Application
The ease of implementation, wide
operating temperature range, and
low supply current requirements for
the LTC2930, LTC2931 and LTC2932
supervisors make them ideal for
automotive applications. Figure 10
tRST
RST
WDI
tWP
tRST
t < tRST
tRST
WDO
tWD
tWD
A
B
C
D
E
tWD
F
G
A. UNDERVOLTAGE EVENT OCCURS, RST PULLS LOW, WDO PULLS HIGH, AND RST TIMER STARTS.
B. RST TIMES OUT (ALL INPUT VOLTAGES BECOME GOOD BEFORE RST TIMEOUT), tRST, THEN WATCHDOG TIMER STARTS.
C. WATCHDOG TIMES OUT, tWD, AND WDO PULLS LOW. RST TIMER STARTS.
D. WDI TRANSITION OCCURS BEFORE RST TIMEOUT. WDO PULLS HIGH AND WDO TIMER STARTS.
E. WDI TRANSITION OCCURS WHILE WDO IS HIGH. WATCHDOG TIMER CLEARS AND RESTARTS.
F. WATCHDOG TIMES OUT. WDO PULLS LOW AND RST TIMER STARTS.
G. RST TIMES OUT. WDO PULLS HIGH AND WATCHDOG TIMER STARTS.
Figure 9. Watchdog timing diagram
14
Linear Technology Magazine • March 2008
DESIGN FEATURES L
BATTERY
RSENSE
4mΩ
UNREGULATED12V
IRF3710 IRF3710
LTC3780
BUCK-BOOST CONVERTER
6V to 30V IN / 12V, 5A OUT
1N4148
5V
2150k
10Ω
2N3904
10Ω
5V
R5
10K
SNS
100k
V1
0.1µF
1M
GATE
OUT
100k
“ALWAYS ON”
ELECTRONICS
µP I/O
COMP5
COMP6
LT4356DE-1
FLT
V2
µP
V3
V4
5V
LTC2931
TMR
3.3V
2.5V
1.8V
POWER
SYSTEMS
5V
100k 845k
CWT
AOUT
V5
V6
CWT
100k
GND
VREF
LOWBAT
R1A
59k
MODE 6
VPG
CRT
5V
CRT
SHDN
1µF
100k
COMP1
IN+
IN
100k
COMP4
4.99k
VCC
100k
100k
COMP3
FB
383k
0.1µF
100k
COMP2
59k
SHDN
CTMR
0.1µF
REGULATED 12V
GND
5V
R2A
40.2k
OUT
LT3010-5
100k
100k
µP I/O
SENS
RST
1µF
GND
WDI
WDI
WDO
WDO
5V
5V
0.1µF
100k
V1
COMP1
V2
COMP3
100k
100k
100k
100k
“IN CABIN”
ELECTRONICS
µP I/O
COMP4
COMP5
COMP6
2150k
V4
V3
100k
V5
LTC2932
V6
2.5V
845k
5V
POWER
SYSTEMS
3.3V
VREF
VPG
100k
100k 511k
R1B
66.5k
MODE 5
R2B
34.8k
CRT
RST
CRT
RDIS
GND
µP I/O
T0
T1
Figure 10. The LTC2931 and LTC2932 in an automotive application
is a block diagram schematic of an
automotive application that uses
the LTC2931 and LTC2932. It was
designed to highlight and utilize the
features of these parts beyond simple
voltage monitoring. The voltage monitors are powered by the LT3010-5,
a ixed 5V micropower linear regulator. Voltage transient protection
is provided by the LT4356DE-1
Linear Technology Magazine • March 2008
overvoltage protection regulator and
inrush limiter.
In a typical automotive power system, a distinction is made between
“Always On” and “In Cabin” electronics. “Always On” systems include
critical electronics that deal with the
safety and security of an automobile
and, as the name implies, are always
on. “In Cabin” electronics pertain to
comfort and entertainment accessories used in automobiles. In the
event the battery is low, for instance,
the in cabin electronics are turned off
to preserve and siphon power to the
critical path.
In this automotive application, power for the always on critical electronics
is generated by the LTC3780 buck
continued on page 4
15
L DESIGN FEATURES
High Power, Single Inductor,
Surface Mount Buck-Boost µModule
Regulators Handle 36VIN, 10A Loads
by Manjing Xie
Introduction
VIN
5V TO 36V
One of the most daunting tasks for a
power supply designer is producing a
high power density supply where the
output voltage sits within the input
voltage range. High power density
buck-boost converters usually require
complex and bulky magnetics, run at
low eficiency, and place high electrical and thermal stresses on devices.
The LTM4605 and LTM4607 µModule
regulators can cure these buck-boost
headaches. The secret is in reducing
the component selection to an inductor, a sense resistor and bulk input
and output capacitors. The resulting
solution is as close to a plug-and-play
buck-boost converter you can get in
an IC form factor.
CLOCK SYNC
10µF
50V
VIN
PLLIN V
OUT
22µF
25V
×2
FCB
ON/OFF
RUN
LTM4607
4.7µH
330µF
25V
SW2
RSENSE
SENSE+
0.1µF
7.5mΩ
SENSE–
SS
SGND
PGND
VFB
7.15k
Figure 1. Remarkable simplicity in the face of an otherwise daunting task. Only a few
components are required to provide an efficient 12V, 5A output from a 5V-to-36V input.
VIN
C1
16
4.5V
TO
36V
CIN
M1
SW2
INTVCC
M2
PGOOD
L
SW1
RUN
ON/OFF
VOUT
100k
12V
5A
STBYMD
CO1
M3
COUT
0.1µF
100k
COMP
VFB
M4
The LTM4605 and LTM4607 incorporate most of the components needed
for a complete buck-boost solution in
their 15mm × 15mm × 2.8mm low proile packages, including the switching
controller, power FETs and support
components. Their synchronous 4switch architecture can achieve up
to 92% eficiency in boost mode and
97.7% eficiency in buck mode.
The LTM4605 has a 4.5V to 20V
input range and a 0.8V to 20V output
range, while the LTM4607 takes a 4.5V
to 36V input to a 0.8V to 24V output.
Such wide voltage ranges save signii-
VOUT
12V
5A
SW1
EXTVCC
The LTM4605 and LTM4607
incorporate most of the
components needed for a
complete buck-boost solution
in their 15mm × 15mm ×
2.8mm low profile packages,
including the switching
controller, power FETs and
support components.
+
INT
COMP
CONTROLLER
RFB
7.15k
RSENSE
SENSE+
SS
SS
0.1µF
PLLIN
INT
FILTER
RSENSE
SENSE–
PLLFLTR
PGND
INT
FILTER
FCB
SGND 1000pF
Figure 2. Simplified block diagram of the LTM4607
Linear Technology Magazine • March 2008
DESIGN FEATURES L
SW1
10V/DIV
SW1
/DIV
SW1
/DIV
SW2
10V/DIV
SW2
/DIV
SW2
/DIV
VOUT(AC)
200mV/DIV
OUT(AC)
OUT(AC)
/DIV
/DIV
VIN = 32V
VOUT = 12V
IOUT = 5A
2µs/DIV
VIN = 5V
VOUT = 12V
IOUT = 5A
Figure 3. Buck mode waveforms
cant design time as one part can serve a
wide variety of applications—reducing
the need to certify and stock different
parts for different applications. To
round out these devices as complete
power supplies, both include important safety features including true
soft-start, output overvoltage protection (OVP) and foldback protection in
buck, boost and buck-boost modes.
Inside the LTM4605
and LTM4607
Figure 2 is a simpliied internal diagram of the LTM4607, showing the two
switching legs: the boost leg connected
to the output, and the buck leg to the
100
VIN = 12V
VIN = 6V
VIN = 6V
VOUT = 12V
ILOAD = 5A
CURSOR 1 = INDUCTOR
CURSORS 2, 3, 4, 5 = LTM4607
EFFICIENCY (%)
VIN = 32V
8
7
6
5
4
80
3
70
VOUT = 12V
60
0
2
POWER LOSS
VIN = 6V
VIN = 12V
VIN = 32V
6
4
LOAD CURRENT (A)
8
POWER LOSS (W)
Figure 1 shows the LTM4607 providing
a 12V, 5A output from a 5V–36V input. The LTM4607 operates as a buck
converter at input voltages above the
output, as a boost converter at input
voltages well below the output, and a
combination of the two in the transition region where the input voltage is
close to the output.
VIN = 12V
VOUT = 12V
IOUT = 5A
2
1
0
10
Figure 6. Efficiency and power loss for the
LTM4607 at different input voltages
2µs/DIV
Figure 5. Transition mode waveforms
Figure 4. Boost mode waveforms
90
12V, 5A Supply
from a 5V–36V Input
2µs/DIV
input. Each synchronous switching
leg is formed by two MOSFETs. SW1
and SW2 tap the middle of the internal
boost leg and buck leg respectively.
Operation of the LTM4607 buck,
boost and transition regions can be
seen in Figures 3 to 5. In buck mode,
SW1 is connected to the output and the
internal buck leg switches to regulate
the output voltage. In boost mode, SW2
is connected to the input and the boost
leg is in action. During the transition
mode, when the input voltage is close
to the output, both buck leg and boost
legs are in action. To keep the internal
bootstrap circuits constantly alive for
both legs, the “inactive” leg refreshes
every tenth cycle of the “active” leg.
That is, the boost leg switches once
for every ten switching cycles in buck
mode while the buck leg switches
similarly in boost mode.
VIN = 32V
VOUT = 12V
ILOAD = 10A
CURSOR 1 = INDUCTOR
CURSORS 2, 3, 4, 5 = LTM4607
7a
7b
Figure 7. Thermal images with VOUT = 12V and 25°C ambient temperature.
Linear Technology Magazine • March 2008
17
L DESIGN FEATURES
VIN
5V TO 36V
CLOCK SYNC 0° PHASE
10µF
50V
R5
100k
PGOOD VIN
PLLIN V
OUT
RUN
FCB
LTM4607
COMP
SW1
INTVCC
LTC6908-1
C1
0.1µF
R4
324k
5.1V
RSENSE
SENSE+
OUT1
EXTVCC
GND
OUT2
SS
SET
R2
7mΩ
STBYMD
–
SENSE
C3
0.1µF
MOD
SGND
330µF
25V
VOUT
12V
10A
SW2
PLLFLTR
V+
C2
22µF
25V
×2
L1
3.3µH
+
VFB
PGND
RFB
3.57k
2-PHASE OSCILLATOR
CLOCK SYNC 180° PHASE
10µF
50V
PGOOD VIN
PLLIN V
OUT
FCB
RUN
LTM4607
COMP
SW1
INTVCC
+
330µF
25V
SW2
PLLFLTR
RSENSE
EXTVCC
SENSE+
STBYMD
SENSE–
SS
SGND
C4
22µF
25V
×2
L2
3.3µH
PGND
R3
7mΩ
VFB
Figure 8. Schematic of two LTM4607 in parallel to provide 12V at 10A output from 6V to 36V source
Efficiency Considerations
The internal MOSFETs are optimized
and able to deliver up to 12A output
current for the LTM4605 and 10A
for the LTM4607 in buck mode. The
LTM4607’s maximum output current
in boost mode is rated at 5A DC (typical). Either part can be easily paralleled
for higher current applications (see
next section). The current limitations
are imposed by power losses from the
internal MOSFETs.
Figure 6 shows the typical eficiency
of the LTM4607 at various input voltages. Worst-case eficiency usually
occurs at minimum and maximum VIN.
At minimum VIN, increased inductor
current and related conduction losses
take the biggest bite out of eficiency,
while at maximum VIN, switching
losses dominate. Derating is necessary under certain input, output and
thermal conditions. Thermal images
of the circuit shown in Figure 1 are
shown in Figure 7. Note that Figure
7b shows a single µModule regulator
supplying 10A in buck mode.
PolyPhase® Paralleling for
High Output Current
If an application requires higher
current than a single LTM4605 or
LTM4607 can supply, two or more
µModule regulators can be connected
in parallel (Figure 8) to increase the
available output current. The current
VOUT(AC)
200mV/DIV
IL1
5A/DIV
ILOAD
20A/DIV
IL2
5A/DIV
IL1
10A/DIV
IL2
10A/DIV
VOUT
5V/DIV
VIN = 5V
VOUT = 12V
IOUT = 10A
5ms/DIV
9a
VIN = 20V
VOUT = 12V
LOAD STEP = 15A
100µs/DIV
9b
Figure 9. Inductor current waveforms at start-up and load transient with two LTM4607s in parallel
18
Linear Technology Magazine • March 2008
DESIGN FEATURES L
VIN
COUT
COUT
SW1
L1
RS
SW2
CIN
GND
VOUT
CIN
CIN
10a
10b
Figure 10. Recommended parts placement and layout
mode architecture of the LTM4607 facilitates eficient PolyPhase operation
(interleaved switch operation) of the
parallel regulators. In a parallel setup,
a single VOUT set resistor is shared by
the regulators. The current control
signal and COMP pins are tied together,
thus resulting in balanced current
sharing, as shown in Figure 9
The 200kHz to 400kHz phase lock
loop of LTM4607 enables interleaved
switch operation, which reduces input
and output ripple current. Figure 8
shows two LTM4607s connected in
parallel to provide a 12V, 10A output.
Interleaved clock signals are generated
by the LTC6908-1.
PCB Layout Considerations
With over 12A of inductor current
and four switching MOSFETs, care
must be taken during the PCB layout
to minimize EMI and thermal stress.
Figure 10 shows the recommended
component placement of components
on both the top and bottom of the
board.
There are two critical loops. One is
formed by input capacitors, the SW2
pins, the sense resistor and GND;
the other is formed by output capacitor, the SW1 pins, the sense resistor
and GND. Because of the high dI/dt
pulsing current in both loops, their
area should be minimized. Thus, the
sense resistor should be put directly
beneath the module with as many
vias as possible on RSENSE and GND.
When components are restricted to one
layer, place the sense resistor as close
as possible to the module. Low ESR
Linear Technology Magazine • March 2008
and ESL ceramic capacitors should
be used at the input and output and
be placed as close to the module as
possible. The second layer should be
a solid ground plane.
Because both the LTM4605 and
LTM4607 use a versatile and responsive current mode control architecture,
an external sense resistor is required.
Accurate sensing of the inductor
current is required for both system
stability and an accurate current
limit. Because the sensed current is
pulsating, parasitic inductance along
the current path should be minimized.
A Kelvin connection is recommended
as shown in Figure 11, and the current sense traces should be close to
each other.1
Layout with the LTM4605 and
LTM4607 couldn’t be easier—there are
SW1
so few components—but it is important
to carefully consider thermal design.
SW1 and SW2 nodes should be
connected to a large sized copper conductor utilizing the inner and bottom
layers to help dissipate heat. Thermal
vias should be placed beneath the
module and on the copper planes as
shown in Figure 10.
Since both the SW1 and SW2 nodes
produce high dV/dt values, it is better to keep control signal traces away
from these nodes.
To improve reliability and thermal
performance, the thermal profile
should be tuned to minimize voids.
Also try to place control signals in inner layers to free the thicker top and
bottom layers for current conduction
and thermal dissipation.
continued on page 6
SW2
VIN
L1
RSENSE
VOUT
CIN
COUT
+ –
SGND
PGND
PGND
RSENSE
KELVIN CONNECTIONS TO RSENSE
Figure 11. Kelvin connection used to sense current.
19
L DESIGN FEATURES
1.5% Accurate Single-Supply
Supervisors Simplify Part Selection
and Operate to 125°C
by Bob Jurgilewicz and Roger Zemke
Introduction
A new line of full-featured single-input
supervisors are easy to place, easy to
bias and easy to conigure. They are
also highly accurate, an important
feature for keeping systems running
reliably.
The LTC2915, LTC2916, LTC2917
and LTC2918 provide as many as
twenty-seven integrated, user-selectable thresholds that are compatible
with many standard power supply
voltages. A user-adjustable input also
allows for customizable thresholds.
The reset timeout period is ixed at
200ms, or add a capacitor to generate
a custom timeout. The even-numbered
parts include an option to generate a
reset-on-demand using the manual
reset input, which is compatible with
mechanical or electrical switching. The
LTC2917 and LTC2918 have watchdog
circuits that monitor processor signal
activity within a user-adjustable window or non-windowed time period.
Electrical speciications are guaranteed to 125°C, so these supervisors
are perfect for high temperature
environments, such as automotive
applications. Operating voltage range
begins at a low 1.5V, and extends to any
12V
Table 1. LTC2915, LTC2916, LTC2917 and LTC2918 feature summary
Feature
LTC2915
LTC2916
LTC2917
LTC2918
9 Selectable Thresholds
L
L
L
L
Wide Temperature Range
–40°C to +125°C
L
L
L
L
Threshold Accuracy
±1.5%
±1.5%
±1.5%
±1.5%
Shunt Regulator for High
Voltage Operation
L
L
L
L
Quiescent Current
30µA
30µA
30µA
30µA
Low Voltage Reset
0.8V
0.8V
0.8V
0.8V
Reset Timeout: 200ms Fixed
or Externally Adjustable
L
L
L
L
L
L
L
L
Power Supply
Glitch Immunity
Selectable Supply Tolerance
–5%, –10%, –15%
L
L
L
Non–Windowed Watchdog
-A
-A
Windowed Watchdog
-B
-B
3.3V
CBYPASS
0.1µF
VCC
VCC
RPU
10k
VCC
VCC
LTC2915
LTC2915
µP
RST
SEL1
RST
5V
3.3V
GND
SEL2
TOL
GND
L
Watchdog Timeout:
1.6s Fixed or
Externally Adjustable
RPU
10k
CBYPASS
0.1µF
VM
L
Manual Reset
3.3V
RCC
11k
VM
µP
RST
SEL1
RST
GND
SEL2
TOL
GND
RT
RT
CRT
Figure 1. A 12V supply monitored from 12V, utilizing
internal shunt regulator with 3.3V logic out
20
L
3.3V
Figure 2. A 5V, –10% tolerance supply monitor
with 200ms internal reset timeout
Linear Technology Magazine • March 2008
DESIGN FEATURES L
3.3V
3.3V
9V
CBYPASS
0.1µF
R2
866k
CBYPASS
0.1µF
RPU
10k
VCC
VCC
µP
RST
LTC2915
RST
SEL1
R1
51.1k
VCC
VCC
LTC2915
VM
1V
GND
µP
RST
VM
GND
SEL2
TOL
GND
TOL
GND
RT
RT
CRT
0.01µF
Figure 3. A 9V, –15% tolerance supply monitor with 3.3V logic out
positive high voltage when biasing the
integrated 6.2V shunt regulator. With
all these features (and more discussed
below), users can qualify one product
to meet almost any supervisory need.
Table 1 summarizes the features for
all four products.
Easy Placement
Despite a general trend to integrate devices as much as possible, single-input
supervisors have certain advantages
over multi-voltage devices. The singleinput supervisor is not taxed with
the requirement to be engaged with
multiple supply voltages so that it is
much easier to place. Lead pitches on
modern device packages dictate that
multi-supply supervisors have their
monitor inputs physically close to
each other. Such covenants naturally
lead to signal routing and congestion
problems. Furthermore, due to the
close proximity of multiple supply
lines, undesirable noise coupling can
be a problem.
Specifying a physical system location for a multi-supply supervisor
involves tradeoffs since an optimal
distance between supplies, super-
visor and microprocessor may be
dificult to achieve. Systems using a
single supervisor do not suffer from
these problems; the supervisor may
be located as near to the monitored
supply or processor as desired. The
LTC2915 and LTC2916 are available
in low proile (1mm) TSOT-23 and
DFN (3mm × 2mm) packaging. The
LTC2917 and LTC2918 are available
in 10-lead MSOP and DFN (3mm ×
2mm) packaging.
In concept, the job of a good supply
monitoring supervisor is simple: when
a power supply voltage drops below a
speciied value, generate a reset. In
reality, the job of a supervisor is much
more complicated. Start-up and shutdown conditions, noise, and transients
all contribute to the complexity of a
real supervisor’s job. If the supervisor
generates a reset while the monitored
supply is actually within speciication,
the result is annoying and consumes
operating margin. Spurious resets
generated by typical supply noise are
equally vexing. Worse yet, not eliciting
a microprocessor reset at voltages too
RPU
10k
µP
RST
RPU
10k
VCC
VCC
RST
SEL1
µP
LTC2915
GND
SEL2
1.5V
VM
RST
RST
I/O GND
SEL1
MR
MANUAL RESET
PUSHBUTTON
CBYPASS
0.1µF
VCC
VCC
GND
low for proper system behavior can be
catastrophic.
Supervisor threshold accuracy is a
critical speciication and must be reckoned with during the system design
phase. Most power supplies are speciied to operate within a tolerance band.
Consider the example of monitoring
a 5V supply with a ±10% tolerance.
The lowest speciied output voltage is
therefore 4.5V. An ideal voltage monitor (perfect accuracy) would generate
a reset at precisely 4.5V and below,
regardless of operating conditions,
indicating an out-of-tolerance supply
voltage. The problem is that ideal,
perfectly accurate voltage monitors
do not exist. A randomly selected realworld voltage monitor has a threshold
that resides within a distributed band
of values. All 27 of the LTC2915,
LTC2916, LTC2917 and LTC2918
selectable thresholds have the same
relative threshold band of ±1.5% of
the selected nominal input voltage,
over the full temperature range (–40°C
to 125°C). The 5V monitor threshold
band is therefore 150mV wide.
The upper limit of the threshold
band should be coincident with the
Correct and Stable Operation
LTC2916
VM
Figure 4. A 1V, –15% tolerance supply monitor with 90ms timeout
3.3V
CBYPASS
0.1µF
10k*
RST
SEL1
SEL2
CRT
1.8V
RPU
10k
–15% –5%
SEL2
RT
GND
CRT
RT
CRT
*OPTIONAL RESISTOR RECOMMENDED TO EXTEND ESD TOLERANCE
Figure 5. 1.8V, –5% supply monitor with manual reset
Linear Technology Magazine • March 2008
Figure 6. 1.5V supply monitor with tolerance control
for margining, –5% operation with –15% margining
21
L DESIGN FEATURES
lowest speciied power supply output
voltage (4.5V in our example). Otherwise, operating range is potentially
consumed if the monitor threshold
reaches above 4.5V. Using the monitor
voltage select (SEL1, SEL2) and tolerance (TOL) inputs on the LTC2915 and
LTC2917 (for 5V supply, 10% reset
threshold), we can conigure the upper threshold limit to 4.5V. The lower
threshold limit is 150mV below, or
4.35V. Statistically, most devices will
have an actual threshold closer to
4.425V, which is the center of threshold band. Because of the threshold
spread, the powered system must work
reliably down to the lower threshold
limit, over temperature. It is easy to
see why less accurate monitors (larger
threshold spreads) can contribute to
system problems.
The monitor threshold discussion,
so far, deals only with the DC value of
1.8V
CBYPASS
0.1µF
RPU
10k
VCC
LTC2916
VM
RST
12V
SEL1
SEL2
SEL2
GND
1.8V
1.8V
RST
GND
TOL
GND
RT
µP
RST
VM
SEL1
MR
MANUAL RESET
PUSHBUTTON
VCC
VCC
LTC2915
RT
1.8V
Figure 7. Dual supply monitor (1.8V and 12V) with manual reset and 200ms Reset timeout
the monitored supply. Real supplies
also have AC components or noise
from sources such as load transients
and switching artifacts. These AC
components should be ignored by the
monitor, since they can cause undesirable spurious reset events. One way
to avoid noise-induced sporadic resets
is to add hysteresis to the monitor
(VTRIP = 10.64V)
comparators—many monitors on the
market use this method. There is a
problem with this approach, in that
the added hysteresis degrades the accuracy of the monitor and ushers in
the design problems discussed earlier.
The LTC2915, LTC2916, LTC2917
and LTC2918 single supervisors do
not apply hysteresis. Instead, the
3.3V
12V
3.3V
R2
1.15M
CBYPASS
0.1µF
RPU
10k
RCC
11k
CBYPASS
0.1µF
VCC
RPU
10k
VCC
VCC
VCC
LTC2917-B
LTC2917
R1
49.9k
µP
VM
RST
RST
SEL1
WDI
I/O
1V
GND
SEL1
WDI
I/O
GND
RT
RT
WT
3.3V
CRT
Figure 8. A 12V supply monitor powered from 12V,
utilizing the internal shunt regulator with 3.3V logic out
tWDU = 1.6s
tWDL = 50ms
tRST = 200ms
Figure 9. A 1V supply monitor with windowed
watchdog timeout and internal timers selected
3.3V
3.3V
CBYPASS
0.1µF
CBYPASS
0.1µF
RPU
10k
VCC
RST
RST
SEL1
WDI
I/O
GND
µP
VM
RST
SEL1
WDI
RST
GND
SEL2
SEL2
TOL
GND
VCC
LTC2917
µP
VM
RPU
10k
VCC
VCC
LTC2917-A
WT
TOL
GND
RT
CWT
0.047µF
CRT
0.0022µF
Figure 10. A 12V supply monitor with 20ms reset
timeout and 3.4s watchdog timeout, with 3.3V logic out
22
RST
TOL
GND WT
CWT
12V
RST
SEL2
SEL2
TOL
GND
µP
VM
WT
RT
CRT
Figure 11. A 3.3V, –10% tolerance supply
monitor with disabled watchdog
Linear Technology Magazine • March 2008
DESIGN FEATURES L
comparators incorporate anti-glitch
circuitry. Any transient at the input
of the monitor comparator must be
of suficient magnitude and duration
(energy) to switch the comparator. Designs utilizing these single supervisors
promote correct and glitch-free resets,
which leads to stable and ultimately
more reliable systems.
Processor Communication
Two of the monitors (LTC2917 and
LTC2918) communicate with host
processors through their watchdog
circuits. The basic requirement for
the processor is to “pet” the watchdog
periodically to avoid being “bitten” by
the dog. Processor resets are invoked
by the built-in watchdog hardware
when the watchdog petting frequency
has become too slow or too fast. Precise knowledge of the system’s timing
characteristics is required to set the
watchdog timeout period. Adjust the
watchdog timeout period by connecting a capacitor between the watchdog
timing input (WT) and ground. Connect
WT to VCC to achieve a default 1.6s
timeout, without the need for external
capacitance.
Simple and Compliant Bias
A unique feature common to all four
of these devices is the ability to provide operating bias from almost any
positive voltage. It does not matter
whether it is a 1.8V LDO, 5V switcher,
12V car battery, 24V wall-wart or 48V
telecom supply; the integrated 6.2V
shunt regulator can work with any
system. For input voltages above 5.7V
the only requirement is to size the
bias resistor (RCC) to the range of the
input voltage. Connect RCC between
the high voltage supply and the VCC
input. Below 5.7V, simply connect
the supply directly to the VCC input.
Deriving resistor sizing for worst-case
operation requires knowledge of the
minimum (VS(MIN)) and maximum
(VS(MAX)) input supply:
VS(MAX ) − 5.7 V
5mA
≤ RCC ≤
VS(MIN) − 7 V
250µA
Be sure to decouple the VCC input
using a 0.1µF (or greater) capacitor
to ground.
Qualify Once, Specify Forever
During product development cycles,
power supply requirements often
change. While supply requirements
are changing, your choice of supervisor doesn’t have to. The LTC2915,
LTC2916, LTC2917 and LTC2918
can relieve the burden of having to
ind the right supervisor for the job.
Qualify any one of these parts and you
can monitor any one of eight different
supply voltages, each with three different internally ixed thresholds. You
can also monitor any custom voltage
down to 0.5V using an external resistor divider. Multi-supply monitoring is
easily achieved by using two or more
devices and connecting their RST
outputs together.
Meet Your Match
The LTC2915, LTC2916, LTC2917 and
LTC2918 single supervisors are the
perfect match for a variety of applications. Browse the applications shown
in the igures and quickly ind the right
application for your system.
Conclusion
The LTC2915, LTC2916, LTC2917
and LTC2918 are feature-laden single
supervisors that can be comfortably
placed near your monitored supply
and/or microprocessor, leading to
easy printed circuit board layout and
reliable system operation.
Unprecedented configurability
makes it possible to qualify and stock
just one product that can meet all of
your supervisory needs. Integration
provides twenty-seven user-selectable monitor thresholds with ±1.5%
accuracy. Any non-standard threshold can be user-conigured with the
adjustable setting.
Other features include high voltage operation, conigurable reset and
watchdog timers, manual reset, and
low quiescent current. External components are seldom required to realize
fully functional designs. Electrical
speciications are guaranteed from
–40°C to 125°C. L
LTC411, continued from page 9
Auto Detect Standby Mode
and Disable Mode
To conserve power, when both bus
voltages are within 400mV of the bus
pull-up supply, the LTC4311 enters
standby mode, consuming only 26µA
of supply current. When ENABLE
is forced low, as shown in Figure 4,
the LTC4311 enters a disable mode
and consumes less than 5µA of supply current. Both bus pins are high
impedance when in disable mode or
when the LTC4311 is powered down,
regardless of the bus voltage.
Linear Technology Magazine • March 2008
Conclusion
The LTC4311 eficiently and effectively
addresses slow rise times, decreased
noise margins at low bus supplies, and
increased DC bus power consumption
found in 2-wire bus systems. Strong
slew rate controlled pull-up currents
quickly and smoothly slew the I2C
or SMBus bus lines, decreasing rise
times to allow up to 400kHz operation for bus capacitances in excess
of 1nF. The advantages of the strong
slew rate controlled currents extend
to reducing the low state bus voltage,
DC bus power consumption, and fall
times, since larger value bus pull-up
resistors can be used.
With a small 2mm × 2mm × 0.75mm
DFN or SC70 footprint, high ±8kV HBM
ESD performance and low power consumption in standby or disable mode,
the LTC4311 Low Voltage I2C or SMBus
accelerator is also ideally suited for
all I2C or SMBus systems. Examples
of such systems include notebooks,
palmtop computers, portable instruments, RAIDs, and servers where I/O
cards are hot-swapped. L
23
L DESIGN FEATURES
Versatile Current Sense Amplifiers
Offer Rail-to-Rail Input,
150°C Operating Temperature
by William Jett and Glen Brisebois
Introduction
24
3V TO 36V
RSENSE
0.02Ω
RIN
100Ω
+IN
–IN
+
–
LOAD
V–
V+
Q1
OUT
LT6106
ROUT
1k
VOUT
200mV/A
Figure 1. Typical current sense application using the LT6106. Sense resistor is 20mΩ and gain is
10, so transfer function is VOUT/ILOAD = 0.2V/A.
SOURCE
–0.3V TO 44V
RIN2
VS+
100Ω
0.02Ω
RIN1
100Ω
VS–
TO LOAD
LT6105
+IN
+
–IN
VOUT
VOUT = 1V/A
–
V+
ROUT
4.99k
V–
(
)
VOUT = VS + − VS − •
R OUT
RIN
RIN 1 = RIN 2 = RIN
R OUT
RIN
AV =
2.85V TO 36V
Figure 2. Typical LT6105 connection. Sense resistor is 20mΩ and gain is 50, so the transfer
function is VOUT/ILOAD = 1V/A. Note the wide input source.
250µV and the power supply rejection
of 106dB makes the accuracy almost
independent of supply.
The LT6107 is functionally identical
to the LT6106, but provides guaranteed performance and speciications
at junction temperatures up to 150°C.
This suits it to under hood automotive
and some industrial applications that
can exceed 125°C for relatively short
periods of time. Input offset voltage is
less than 400µV over the entire temperature range of –40°C to 150°C. As
with the LT6106, two external resistors
set the ampliier gain.
0.6
0.4
ACCURACY (% OF FULL SCALE)
Fast and accurate current measurement is required in an increasing
number of electronics subsystems.
The list of applications that call for
current sensing includes diagnostic
system assessment, fault detection,
load protection and scaling, battery
“gas gauge” monitoring, and impending component failure detection, to
name just a few. The challenge is that
there is no one-size-its-all solution for
current measurement. For example,
protection circuits often emphasize
measurement speed, while battery
applications usually emphasize accuracy and low power. Nevertheless,
design time can be reduced by using
an accurate current measurement ampliier with features suited to the task
at hand, such as the latest members
of Linear Technology’s current sense
ampliier family, the LT6105, LT6106
and LT6107.
The LT6105 is distinguished by its
rail-to-rail inputs. It is a perfect it
for automatic test equipment (ATE)
systems and other systems that use a
combination of ixed voltage supplies
and programmable or switched supplies. The LT6105 features an input
voltage range of –0.1V to 44V that
is independent of the power supply
voltage. The supply voltage for the
LT6105 can be obtained from any
convenient source within the supply
range of 2.85V to 36V. For instance, a
programmable supply with an output
of 0V to 24V can be monitored while
the LT6105 is powered by a ixed 5V
supply.
The LT6106 is distinguished by
ease-of-use and accuracy in single
supply environments. Just add two
resistors and tie the supply to the sense
resistor, and the device is conigured
to measure currents in supplies from
2.7V to 36V. Input offset voltage is a low
LIMIT OVER TEMPERATURE
0.2
0
TYPICAL PART AT TA = 25°C
–0.2
–0.4
–0.6
LIMIT OVER TEMPERATURE
–0.8
5A FULL SCALE RIN = 100Ω
–1.0 RSENSE = 0.02Ω ROUT = 1k
AV = 10
V+ = 3V
–1.2
0
1
3
2
LOAD CURRENT (A)
4
5
Figure 3. Measurement accuracy vs load
current for the LT6106 and LT1607
Linear Technology Magazine • March 2008
DESIGN FEATURES L
Table 1 summarizes the guaranteed
performance of the LT6105, LT6106
and the LT6107.
LT6106, LT6107
Theory of Operation
LT6105
25°C
LT6106
25°C
LT6107
–40°C to 150°C
Input Voltage Range
–0.3V to 44V
2.7V to 36V
2.7V to 36V
Supply Voltage Range
2.85V to 36V
2.7V to 36V
2.7V to 36V
VOS
300µV/1000µV*
250µV
400µV
Maximum Differential
Input Voltage
44V
0.5V
0.5V
CMRR
95dB
N/A
N/A
PSRR
100dB
106dB
106dB
Gain Error
±1%/±2.5%*
–0.65% to 0%
–0.65% to 0%
Output Current
1mA
1mA
1mA
Supply Current
340µA
95µA
125µA
Package
MS8, DFN
TSOT-5
TSOT-5
* Input common mode voltage = 12V, 0V
Referring to Figure 1, the current to
be measured, ILOAD, passes through
a sense resistor RSENSE, resulting in
a voltage drop of VSENSE. Feedback
from the ampliier causes a current
to low in RIN and Q1 such that the
ampliier inputs are equal, V–IN = V+IN.
The current in Q1 also lows through
ROUT. The output voltage is therefore
proportional to the load current and
is given by
VOUT = ILOAD • R SENSE
TO LOAD
RSENSE
VS–
LOAD
The LT6105 tolerates negative voltages on its inputs of up to –9.5V. In
addition, it can also be used to sense
across fuses or MOSFETs as shown
in Figure 4. The LT6105 has no problem when the fuse or MOSFET opens
because it has high voltage PNP’s and
a unique input topology that features
full high impedance differential input
swing capability to ±44V. This allows
Linear Technology Magazine • March 2008
C1
0.1 F
DC SOURCE
(≤44V)
+IN
–IN
V+
+
C2
0.1 F
5V
V–
OUT
OUTPUT
LT6105
ROUT
Figure 4. The LT6105 can monitor across a fuse or switch. It’s inputs are undamaged even when
split wide apart, and current is limited to about 3mA.
V+
Q2
Q1
The LT6105:
Robust and Easy to Use
VS+
RIN2
RIN1
R
• OUT
RIN
The overall accuracy graph shown
in Figure 3 combines the effects of
gain error and input offset voltage
to create a worst-case error band for
the application circuit in Figure 1. A
slight negative gain error, typically
–0.25%, is due to the inite current
gain of the PNP.
FUSE
–
The LT6105, LT6106 and LT6107
use traditional external gain-setting
resistors. This is actually an important
feature in a current sense ampliier.
Most current sense applications require a very small maximum sense
shunt voltage (to minimize power loss),
which must be ampliied to match
either a very speciic comparator
threshold or ADC input voltage span.
The ability to carefully control the gain
is paramount to optimize performance.
Figures 1 and 2 show typical applications of the LT6106 and LT6105.
Parameter
+
Flexible Gain Setting
Table 1. Guaranteed performance specifications
RIN2
VS+
VSENSE
Q3
Q6
Q7
+IN
VOUT
RSENSE
0.02Ω
+
RIN1
VS–
–IN
–
ROUT
Q4
Q5
ILOAD
Figure 5. Block diagram of the LT6105
25
L DESIGN FEATURES
24VDC
1N5818
VBAT = 3.6V
ICPO = 200µA
5V/DIV CCPO = 2.2ΩF
200Ω
1%
–
LT6105
–IN
0V/OFF
+
200Ω
1%
5V/ON
24V, 3W
SOLENOID
1Ω
1%
+IN
2N7000
10V/DIV
5VDC
V+
2V/DIV
V–
VOUT
VOUT = 25mV/mA
4.99k
1%
Figure 6. Simplest form of solenoid driver. The LT6105 monitors the current in both on and
freewheel states. Lowest common mode voltage is 0V, while the highest is 24V plus the forward
voltage of the Schottky diode. See waveforms, Figure 7.
direct sensing of fuse or MOSFET
voltage drops, without concern for
an open circuit condition in the fuse
or MOSFET.
Another beneit of the LT6105
is that you can leave it connected to
a battery even when it is unpowered.
When the LT6105 loses power, or is
intentionally powered down, both
sense inputs remain high impedance. In fact, when powered down,
the LT6105 inputs actually draw
less current than when powered up.
Powered up or down, it represents a
benign load.
The LT6105 extends the current
sense measurement concept used in
the LT6106 (and others) to accommodate an input voltage range that
includes ground. For both the LT6105
and LT6106, the voltage developed
across a sense resistor is translated
into a current that appears in the
output pin. The voltage gain is set
by the ratio of the input and output
resistors, (ROUT/RIN). The wide input
range in the LT6105 is obtained by
the use of two feedback paths to the
input pins.
24VDC
1Ω
1%
24V/OFF
200Ω
1%
–
1N914
24V, 3W
SOLENOID
2k
1%
–IN
5VDC
TP0610L
+
200Ω
1%
19V/ON
2k
1%
LT6105
1N5818
Figure 7. Waveforms for solenoid driver. Top
trace is the MOSFET gate, with high on.
Middle trace is the bottom of the solenoid/
inductor. Bottom trace is the LT6105 output,
representing solenoid current at 80mA/DIV.
Glitches are useful indicators of solenoid
plunger movement.
Referring to Figure 5, when the input voltage VS+ is between 0V and 1.6V,
devices Q1, Q2, and Q3 are active and
devices Q4–Q7 are off. Feedback from
the ampliier causes the current to low
in Q1, which equalizes the ampliier
input voltages. Devices Q1 and Q3 are
matched, so the collector current of
Q3 will equal the collector current of
Q1. The output voltage is then
VOUT = ILOAD • R SENSE •
V+
ROUT
RIN1
When the input voltage VS+ is greater
than 1.6V, devices Q4, Q5, Q6, and
Q7 are active and devices Q1–Q3 are
off. Again, feedback from the ampliier
causes the current to low in Q4 which
equalizes the ampliier input voltages.
The current in Q4 is mirrored to the
output through the matching of Q4 to
Q5 and Q6 to Q7. The output voltage
for this mode is given by
VOUT = ILOAD • R SENSE •
+IN
ROUT
RIN2
LT6105 Application:
Solenoid Current Monitor
V–
VOUT
VOUT = 25mV/mA
4.99k
1%
Figure 8. Similar circuit to Figure 6 but with solenoid grounded, so freewheeling forces inputs
negative. Providing resistive pullups keeps amplifier inputs from falling outside of their accurate
input range.
26
50ms/DIV
The large input common mode range
of the LT6105 makes it suitable for
monitoring currents in quarter, half,
and full bridge inductive load driving
applications. Figure 6 shows an example of a quarter bridge. The MOSFET
pulls down on the bottom of the solecontinued on page 8
Linear Technology Magazine • March 2008
DESIGN FEATURES L
Compact Hot Swap Solution Simplifies
Advanced Mezzanine Card Design
by Chew Lye Huat
Introduction
Advanced Mezzanine Cards (AdvancedMC) are 80W add-on modules
that are central to the extensibility
of MicroTCA and AdvancedTCA high
availability systems. AdvancedTCA targets high capacity, high performance
applications whereas MicroTCA serves
smaller, cost-sensitive applications
with less demanding requirements.
Unlike other mezzanine modules,
which are bolted on, AdvancedMC
modules can be inserted into a live
backplane improving serviceability
and lexibility.
The LTC4223 offers a compact and
comprehensive Hot Swap solution for
Advanced Mezzanine Cards, providing
inrush control, overcurrent protection
and supply monitoring. It detects
board insertion and extraction, allowing the 12V and 3.3V auxiliary power
supply to be delivered in a controlled
manner without damaging the connector. The LTC4223 provides a current
monitor output for the 12V supply,
allowing real-time current monitoring.
Short-circuit faults on the 12V and
3.3V output are rapidly isolated with
The LTC4223 offers a
compact and comprehensive
Hot Swap solution for
Advanced Mezzanine Cards,
providing inrush control,
overcurrent protection and
supply monitoring. It also
detects board insertion and
extraction, thus protecting
the connector.
RS
6mΩ
CARRIER AMC MODULE AMC
CONNECTOR CONNECTOR
Q1
Si7336ADP
PWR
12V
BULK SUPPLY
BYPASS CAPACITOR
2
5
3.3V
R2
51Ω
BULK SUPPLY
BYPASS CAPACITOR
C2
330nF
7
PWR ENABLE
4
3.3V
R6*
10k
MP GOOD
3.3V
R5*
10k
R4*
10k
PWR GOOD
FAULT
1
16
12VSENSE 12VGATE
AUXIN
12VOUT
AUXOUT
6
MP ENABLE
3.3V
CG
RG 15nF
47Ω
R3
10Ω
12VIN
12V
7.4A
15
12
MP
10
PS1
3.3V
150mA
VCC
AUXON
12ON
LTC4223-1
11
14
13
AUXPGOOD
12PGOOD
FAULT
3.3V
INTELLIGENT
PLATFORM
MANAGEMENT
CONTROLLER
1µF
3
7
6
1
CLK
5
8
VREF
VCC
+IN
2
3
EN
12IMON
PS0
DOUT LTC1197L
CS
3.3V
–IN
GND
4
2.2k
3
GND
8
TIMER
9
CT
0.1µF
422312 F01
3.3V
PRESENCE
ENABLE
RESET
10k
*OPTIONAL
Figure 1. Typical Advanced Mezzanine Card application for MicroTCA systems
Linear Technology Magazine • March 2008
27
L DESIGN FEATURES
Card Presence Detect
Ignores Contact Bounces
EN
5V/DIV
TIMER
2V/DIV
AUXOUT
5V/DIV
12VOUT
5V/DIV
EN
2V/DIV
TIMER
1V/DIV
AUXPGOOD
5V/DIV
12PGOOD
5V/DIV
5ms/DIV
a fast acting current limit to prevent
the input supply from collapsing when
the output is shorted or overloaded.
The LTC4223-1 features a latch-off
circuit breaker, while the LTC4223-2
provides automatic retry after a fault.
Both options are available in 16-pin
SSOP and space-saving 5mm × 4mm
DFN packages.
Typical AdvancedMC
Hot Swap Application
In a typical AdvancedMC application,
the LTC4223 resides on the carrier
board, delivering 12V and 3.3V auxiliary power to the modules, as shown
in Figure 1. It controls the 12V main
supply with an external N-channel
MOSFET and the 3.3V auxiliary supply
with an integrated 0.3Ω switch. The
current for the 12V supply is monitored
via sense resistor RS. The monitored
current is reproduced as a relative
voltage signal at the 12IMON pin. This
signal can be fed to a control system
using an LTC1197L ADC.
Resistor R3 prevents high frequency
MOSFET self-oscillation in Q1, and
RG/CG compensates the active current
limit loop. R2/C2 ilters the input power supply, VCC, from supply transients.
Several timers are conigured by the
capacitor, CT, including the debounce
cycle delay (CT • 741[ms/µF]), aux
current limit time-out during startup (CT • 123[ms/µF]) and 12V supply
overcurrent response (CT • 6[ms/µF]).
The two supplies can be independently
controlled by their respective ON pins,
and their power-good and fault status
are indicated using open-drain outputs with internal pull-ups.
28
Figure 3. Normal power-up sequence with CL1
= 2200µF and CL2 = 150µF after a debounce
timing cycle
In a typical AdvancedMC
application, the LTC4223
resides on the carrier board.
It controls the 12V main
supply with an external
N-channel MOSFET and
the 3.3V auxiliary supply
with an integrated 0.3Ω
switch. The current for the
12V supply is monitored
via sense resistor and is
reproduced at the 12IMON
pin as a relative voltage
signal, which can be fed to
a control system using an
LTC1197L ADC.
ILOAD
Power-Up Sequence
Figure 3 shows the 3.3V auxiliary
and 12V supplies powering up in
sequence after EN transitions low.
The pre-conditions for start-up are:
VCC and the input supplies exceed
VSENSE
– +
Q1
12V
12VSENSE
12VIN
RIN
5k
LOAD
12VGATE
–
Figure 2. TIMER blanking time prevents false
fault reset from EN pin contact bounces.
20ms/DIV
+
FAULT
2V/DIV
Contact bounces as connector pins are
mated can trigger unwanted system
resets or can cause supplies to turn
on unintentionally. To prevent this
behavior, the LTC4223 ignores these
contact bounces for one TIMER cycle
before turning on the supplies.
When the connector pin PS1 is
engaged low upon card insertion,
EN goes low and initiates a start-up
debounce cycle if the ON pin is high.
Any contact bounces on the EN pin
reset the TIMER and restart the ramp
up until it reaches 1.235V, at which
time the fault latches are cleared. If EN
remains low at the end of the debounce
cycle, the switches are allowed to turn
on. If EN toggles high indicating card
removal, all switches are turned off in
20µs, disconnecting the supplies to
the modules. Latched faults are not
cleared. However, because removing
the card could cause the EN pin voltage to bounce, the clearing of latched
faults is blanked internally by a TIMER
ramp-up time given by CT • 123[ms/
µF], as shown in Figure 2.
1µF
12IMON
ROUT
165k
VOUT
VREF
+IN
VCC
LTC1197L
–IN
LTC4223
3.3V
CLK
DOUT
TO SYSTEM
CONTROLLER
CS
GND
422312 F10
VOUT =
ROUT
RIN
• VSENSE = 33 • VSENSE
Figure 4. High side current sense monitor with the LTC1197L ADC
Linear Technology Magazine • March 2008
DESIGN FEATURES L
their undervoltage lockout thresholds,
TIMER is less than 0.2V and EN is
pulling low. If all of these conditions
are met, a debounce timing cycle is
initiated when the ON pin pulls high.
By default, the internal Aux switch
turns on irst if both ON pins are high
at the end of the debounce cycle. This
satisies the requirement to power up
the controller irst on the AdvancedMC
module before turning on the 12V
supply.
The charge current applied to the
output capacitor CL2 is limited to
240mA by an internal ACL (Active
Current Limit) ampliier, well below
the maximum 500mA allowed for AdvancedMC modules. When the current
limit is active, the TIMER pin ramps
up with a 10µA pull-up. AUXPGOOD
pulls low when AUXOUT exceeds its
power-good threshold of 2.901V unless the TIMER pin reaches 1.235V and
times out. When the TIMER pin falls
below 0.2V with a 2µA pull-down, the
12V supply external MOSFET turns
on by charging the GATE with a 10µA
current source. The GATE voltage rises
with a slope equal to 10µA/CG and the
inrush current lowing into the load
capacitor CL1 is limited to (CL1/CG) •
10µA. If the sense resistor voltage drop
becomes too large, the inrush current
is limited at 60mV/RS by the internal
current limit circuitry. 12PGOOD pulls
low when 12VOUT exceeds 10.36V.
Power Monitoring with
High Side Current Sense
The LTC4223 features a high side
current sense ampliier for the 12V
supply that translates the sense resistor voltage drop from the positive
rail to the negative rail. The voltage
at the 12IMON pin is equal to 33 •
VSENSE. This can drive the input of an
LTC1197L ADC, as shown in Figure
4 for data conversion, and allows
the system controller to monitor the
power consumed by the AdvancedMC
module. Full scale input to the current
sense ampliier is 82.5mV corresponding to an output of about 2.7V. If the
input exceeds 100mV, the 12IMON
output clamps at 3.2V.
FAULT
5V/DIV
∆VSENSE
200mV/DIV
12VOUT
5V/DIV
ILOAD
1A/DIV
12VGATE
5V/DIV
AUXOUT
5V/DIV
5µs/DIV
Figure 5. Fast acting current limit isolates
severe short-circuit fault on 12V output
Figure 6. Fast acting current limit isolates
short-circuit fault on 3.3V auxiliary output
Thermal Shutdown Protection
enters into active current limiting that
maintains 60mV across the sense resistor. When the sense voltage exceeds
the circuit breaker threshold (50mV
with 5% accuracy), the TIMER capacitor is pulled high with a 200µA current
until the TIMER reaches 1.235V, after
which it pulls down with 2µA. When
this occurs, FAULT pulls low and the
GATE is pulled down to ground with
1mA, turning off the MOSFET but not
the internal Aux switch. Similarly, an
internal ACL ampliier protects the
3.3V auxiliary supply from overcurrent
by pulling down the gate of the internal
pass transistor rapidly. Thereafter, the
gate recovers and servos the output
current to about 240mA for 25µs
before pulling down to ground gently,
turning the transistor off, as shown
in Figure 6. At this time, FAULT pulls
low and the 12V supply switch also
shuts off.
The internal 3.3V auxiliary supply
switch is protected by a thermal
shutdown circuit. If the switch’s
temperature reaches 150°C, the
Aux switch shuts off immediately
and FAULT pulls low. The external
12V supply switch also turns off.
The switches are allowed to turn on
again only after both the ON pins are
cycled low and then high after the
internal switch’s temperature falls
below 120°C.
Fast Acting Current Limit
Isolates Fault
The LTC4223 features an adjustable
current limit with circuit breaker
function that protects the external
MOSFET against excessive load current on the 12V supply. In the event of
a severe short-circuit fault as shown
in Figure 5, the LTC4223 brings the
surge current under control within
1µs by pulling the MOSFET’s GATE
down to the SOURCE pin. Thereafter,
the GATE recovers rapidly due to the
RG/CG compensation network and
Auto-Retry after a Fault
The LTC4223-1 latches off after an
overcurrent fault while the LTC4223-2
automatically restarts. Following an
TIMER
1V/DIV
TIMER
1V/DIV
FAULT
5V/DIV
FAULT
5V/DIV
12VGATE
5V/DIV
AUXOUT
5V/DIV
12IMON
2V/DIV
ILOAD
0.5A/DIV
20ms/DIV
Figure 7. Auto-retry with 0.5% duty
cycle during 12V output short
Linear Technology Magazine • March 2008
5µs/DIV
50ms/DIV
Figure 8. Auto-retry with 6.5% duty cycle
during 3.3V auxiliary output short
29
L DESIGN FEATURES
BACKPLANE
CARD
CONNECTOR CONNECTOR
RS
2.5m
Q1
HAT2160H
12V
18A
12V
R1
10
Z1
SMAJ13A
15V
R3
10
C1
100nF
12VIN
+
CG
RG 15nF
47
CL1
1000 F
12VSENSE 12VGATE
12VOUT
3.3V
AUXIN
Z2
SMAJ5.0A
7V
R7
2.7
C3
100nF
3.3VAUX
150mA
AUXOUT
R2
51
+
3.3V
R9
10k
AUXON
R8
10k
BD_SEL
CL2
150 F
VCC
C2
330nF
R5
10k
R4
10k
LTC4223-1
R6
10k
AUXPGOOD
12PGOOD
FAULT
12ON
EN
3.3V
1 F
12IMON
VREF
VCC
+IN
LTC1197L
TIMER
GND
CLK
DOUT
TO
CONTROLLER
CS
–IN
GND
CT
0.1 F
Figure 9. A 12V at 18A card-resident application
overcurrent fault, the LTC4223-1 does
not power up again until the fault is
cleared by either pulling the ON or
EN pin from high to low or by VCC
falling below its UVLO threshold. The
LTC4223-2 automatically clears faults
after a cool-off cycle and powers up
again, as shown in Figure 7 for a 12V
fault and Figure 8 for a 3.3V fault.
The latched fault is cleared during
the start-up cycle and FAULT pulls
high. If the output-short persists, the
device powers up into an output-short
with active current limiting until the
TIMER times out and FAULT pulls
low again. The device restarts after a
cool-off cycle and the process repeats
until the output-short is removed.
The cool-off cycle time is given by
CT • 1358[ms/µF] after a 12V supply
fault and CT • 1482 [ms/µF] after an
auxiliary supply fault. If the TIMER
capacitor CT = 0.1µF, the auto-retry
duty cycle is 0.5% for the 12V supply
and 6.5% for the 3.3V auxiliary.
30
12V at 18A
Card-Resident Application
In addition to the AdvancedMC application where the LTC4223 is on
the carrier board, it can also reside on
the plug-in card side of the connector.
Figure 9 shows a typical 12V at 18A
application. The 3.3V auxiliary supply
may be used to power up on-board logic
drawing not more than 150mA. The
12V supply inrush current is limited
to 0.7A by RG/CG compensation network when powering up into a large
load capacitor of 1000µF. When no
bulk capacitor is present on the card
supply, transient voltage suppressors
(Z1, Z2) are required to clamp supply
transients and the snubber (R1/C1,
R7/C3) eliminates ringing during an
output-short. For the LTC4223 to work
with a different load at 12V output,
choosing the correct sense resistor
and external MOSFET is crucial.
This ensures that the circuit breaker
threshold is not exceeded under the
maximum load condition, and that
the power dissipated in the MOSFET
is well within its safe operating area
(SOA) while in active current limit
during an output-short.
Conclusion
The LTC4223 provides Hot Swap
control for a 3.3V auxiliary and a
12V supply. It features board insertion and extraction detection, active
current limit into large load capacitors and sequenced supply turn-on
with power-good status, all critical
in Advanced Mezzanine Card applications. Its tight 5% circuit breaker
threshold accuracy and fast acting
current limit protect the supplies
against overcurrent faults. The current
monitor output allows measurement of
the 12V supply’s power consumption.
With these features, the LTC4223 offers a compact Hot Swap solution that
simpliies the Advanced Mezzanine
Card design. L
Linear Technology Magazine • March 2008
DESIGN IDEAS L
Tiny, Fast and Efficient Comparator
Regenerates Clock Signals
by Jim Sousae
up to 3MHz
What is it?
The LTC6702 is a tiny dual comparator that is designed to bridge the gap
between relatively slow ultralow power
comparators and very fast high power
comparators. The LTC6702 combines
speed, low voltage operation and
micropower operation, making it ideal
in battery powered circuits that require
high performance. Additional features
such as built-in hysteresis (to ensure
stable operation) and CMOS inputs
simplify designs and allow the use of
large source impedances. Offered in
the tiny 2mm × 2mm DFN package,
the LTC6702 is the smallest dual
comparator currently available, with
a footprint nearly 40% smaller than
that of a SOT-23.
The LTC6702 combines
speed, low voltage operation
and micropower operation,
making it ideal in battery
powered circuits that require
high performance.
with a supply voltage as low as 1.7V
optimizes battery life.
Excellent Output Swing
The LTC6702 uses patented breakbefore-make circuitry in its output
stage to minimize shoot-through cur800
VSTEP = 100mV
OVERDRIVE = 50mV
700 TA = 25 C
Guaranteed Speed
The two main benchmarks of a comparator are propagation delay and
supply current. Most comparators
only list a typical value for propagation delay. The LTC6702 goes one
step further, fully testing and guaranteeing a propagation delay of 500ns
maximum from –40°C to 125°C. It
manages to do this while drawing only
30µA maximum supply current per
comparator. Guaranteed operation
PROPAGATION DELAY (ns)
What’s So Special?
600
500
V+ = 1.7V
FALLING
400
300
200
RISING
+
V = 3V, 5V
10
FALLING
RISING
100
1000
LOAD CAPACITANCE (pF)
10000
Figure 1. Speed is maintained
with high capacitive loads
rent when the output changes states
(typically problematic in CMOS output
stages). The result is an output stage
with three times better swing than the
typical bipolar output stage and much
lower shoot-through current than the
typical CMOS output stage, allowing
the LTC6702 to maintain its eficient
operation, even at high toggle rates.
The push-pull output stage topology
provides rail-to-rail operation without
the need of a pull-up resistor.
Capacitive Load Handling
The LTC6702 has the ability to drive
large capacitive loads due to its high
output drive current, unusual in
such a small, low quiescent current
device. The output current is speciied
at ±15mA from –40°C to 125°C and
has a typical short circuit current of
±250mA. Competing products show
capacitive load handling to 400pF due
to the degradation of their rise/fall
times and propagation delay with
higher capacitive loads. Figure 1
shows the LTC6702’s ability to drive
up to 10,000pF with signiicantly less
degradation to these parameters. The
high output drive current also allows
the LTC6702 to drive low current
relays directly.
V += 3V
DESIGN IDEAS
Tiny, Fast and Efficient Comparator
Regenerates Clock Signals up to 3MHz
.........................................................31
IN
VR = 400mV
REFERENCE
Jim Sousae
9V to 80V Ideal Diode Reduces Heat
Dissipation by Order of Magnitude
over Schottky ....................................33
Meilissa Lum
CMOS Op Amp Outperforms Bipolar
Amps in Precision Applications .........35
Hengsheng Liu
0.1 F
LT6650
+
1/2 LTC6702
OUT
C1
0.1 F
–
–
1k
+
FB
RECOVERED
CLOCK
(UP TO 3MHz)
CLOCK INPUT > 100mVP-P
Figure 2. Clock recovery circuit efficiently recovers clock signals up to 3MHz
Linear Technology Magazine • March 2008
31
L DESIGN IDEAS
Inputs Operate Above V+
Most comparators protect their inputs
from ESD strikes by diode clamping
the inputs to supply. Instead, the
LTC6702 uses a ground referenced
ESD device on each input pin, thus
allowing the inputs to operate above
the positive supply without additional
input current or damage to the device.
As long as one input is within the allowed common-mode range, the other
input can go as high as the absolute
maximum rating of 6V, regardless of
the supply voltage.
1.8V
LT6650
OUT
IN
GND
0.1 F
500mV
–
24.9k
1 F
FB
1/2 LTC6702
6V
0V
OUT
5.5V
0V
5.5V
LT6650
OUT
IN
GND
0.1 F
500mV
–
24.9k
1 F
FB
1/2 LTC6702
2V
0V
+
100k
Level Translation
The LTC6702’s push-pull output stage
and its ability to operate with either
input above the positive supply rail
simpliies logic level translation. Many
comparators use an open collector or
open drain type output stage to enable
1.8V
0V
+
100k
What’s It Good For?
Clock Regeneration
The high toggle rate and eficiency of
the LTC6702 is ideal for clock regeneration in battery powered circuits.
It is no longer necessary to waste
milliamps of supply current powering
an ultrafast comparator when ultrafast
speeds are not required. The simple
circuit in Figure 2 can recover clock
signals with frequencies up to 3MHz
while burning only 225µA of supply
current.
OUT
Figure 3. Level translation without the need of an additional supply or pull-up resistor
level translation and require a pull-up
resistor and separate supply to set
the output logic level. The circuits in
Figure 3 show how the LTC6702 can
perform both high-to-low and low-tohigh level translation without the need
of an additional pull-up resistor, thus
reducing component count and saving
board space.
Current Sense Alarm
A typical swing of 300mV from either
rail with ±60mA output drive allows
the LTC6702 to directly drive an LED
or relay for alarm annunciation or load
protection switching. Figure 4 shows a
dual load current sense alarm circuit
that provides resistor programmable
thresholds, turns on an LED when an
overload condition is detected and has
a quiescent current of only 31µA.
Conclusion
The unique feature set of the LTC6702
makes it a very versatile dual comparator. Its tiny footprint and rail-to-rail
output capability allow the designer
to conserve board space, while it’s
high speed to power ratio and low
voltage operation enable eficient clock
regeneration and maximize battery
life. L
5.5V
V+
LTC6702
LED B
1k
–
LOAD B
ILOADB
RSB
0.1
LT6650
IN
0.1 F
GND
FB
OUT
COMP B
500mV
1 F
+
24.9k
LED A
1k
37.4k
+
61.9k
COMP A
–
LOAD A
ILOADA
RSA
0.1
GND
LEDA ON IF ILOADA > 2.5A
LEDB ON IF ILOADB > 5.0A
Figure 4. This micropower dual low side current sense alarm can drive an LED or relay
32
Linear Technology Magazine • March 2008
DESIGN IDEAS L
9V to 80V Ideal Diode Reduces Heat
Dissipation by Order of Magnitude
by Meilissa Lum
over Schottky
Introduction
High availability systems often employ
parallel-connected power supplies
or battery feeds to achieve redundancy and enhance system reliability.
Schottky ORing diodes have long been
used to connect these supplies at
the point of load. Unfortunately, the
forward voltage drop of these diodes
reduces the available supply voltage
and dissipates signiicant power at
high currents. Costly heat sinks and
elaborate layouts are needed to keep
the Schottky diode cool.
A better solution is to replace the
Schottky diode with a MOSFET-based
ideal diode. This reduces the voltage
drop and power dissipation, thereby
reducing the complexity, size and cost
of the thermal layout and increasing
system eficiency. The LTC4357 is an
ideal diode controller that drives an
N-channel MOSFET and operates over
a voltage range of 9V to 80V.
How It Works
The LTC4357’s basic operation is
straightforward. The external MOSFET
source is connected to the input supply and acts like the anode of a diode,
while the drain is the cathode. When
power is irst applied, the load current
initially lows through the body diode of
the MOSFET. The LTC4357 senses the
voltage drop and drives the MOSFET
on. The LTC4357’s internal ampliier
and charge pump try to maintain a
25mV drop across the MOSFET. If the
load current causes more than 25mV
FDB3632
VINA = 48V
IN
GATE
LTC4357
IN
VDD
0.1µF
IN
GATE
LTC4357
OUT
VDD
GND
Figure 1. Two load-sharing, redundant, 48V/10A power supplies using an ideal diode
of voltage drop, the MOSFET is driven
fully on, and the forward drop becomes
equal to RDS(ON) • ILOAD. If the load current reverses, as may occur during an
input short, the LTC4357 responds by
quickly pulling the MOSFET gate low
in less than 0.5µs.
Load Sharing
Redundant Supplies
Figure 1 shows a 48V/10A ideal diode-OR application. An MBR10100
Schottky diode would dissipate 6W
under these operating conditions.
In contrast, the FDB3632 7.5mΩ
MOSFET dissipates only 7.5mΩ •
(10A)2 = 0.75W. The reduced power loss
increases eficiency and saves space
required for heat sinking. If the power
supply voltages are nearly equal, the
GATE
LTC4357
OUT
+
12V
BATTERY
LOAD
GND
Figure 2. Solar panel charging 12V battery through ideal diode to prevent back feeding
Linear Technology Magazine • March 2008
VOUT TO LOAD
FDB3632
VINB = 48V
100Ω
14V
SHUNT
REGULATOR
VDD
GND
FDB3632
100W
SOLAR
PANEL
OUT
load current is shared between the two
supplies. Otherwise, the supply with
the highest output voltage provides
the load current.
Load sharing is accomplished using
a simple technique known as droop
sharing. Load current is irst taken
from the highest supply output. As this
output falls or droops with increased
loading, the lower supply begins to
contribute. Regulating the forward
voltage drop to 25mV ensures smooth
load sharing between outputs without
oscillation. The degree of sharing is a
function of MOSFET Rds(on), the output
impedance of the supplies and their
initial output voltages. Backfeeding of
one supply into the other is precluded
by the diode action of the LTC4357.
Solar Power Application
In solar power systems, Schottky diodes are used to prevent discharge of
the battery during hours of darkness.
Unfortunately, the voltage drop and
power dissipation of a Schottky diode
can be quite large when used with
high wattage solar panels, thus reducing the amount of power available to
charge the battery. Figure 2 uses the
33
L DESIGN IDEAS
LTC4357 with a FDB3632 MOSFET
to replace the Schottky diode.
When the solar panel is illuminated
by full sunlight, it charges the battery.
A shunt regulator absorbs any excess
charging current to prevent overcharging. If the forward current is greater
than 25mV/RDS(ON), the MOSFET is
fully enhanced and the voltage drop
rises according to RDS(ON) • (I BATTERY +
I LOAD). In darkness, or in the event of
a short circuit across the solar panel
or a component failure in the shunt
regulator, the output voltage of the
solar panel will be less than the battery voltage. In this case, the LTC4357
shuts off the MOSFET, so the battery
will not discharge. The current drawn
from the battery into the LTC4357’s
OUT pin is only 7µA at 12V.
Protecting Against
Reverse Inputs
In automotive applications, the
LTC4357 inputs can be reversed.
An additional component, shown in
Figure 3, prevents the MOSFET from
turning on and protects the LTC4357.
Si4874DY
VIN
12V
CLOAD
IN
GATE
LTC4357
VOUT
12V
10A
OUT
VDD
GND
MMBD1205
Figure 3. –12V Reverse input protection blocks reverse input voltage to the load
With a reverse input, the diode connected to system ground is reverse
biased. The GND pin is pulled by the
second diode to within 700mV of the
reverse input voltage. Any loading
or leakage current tends to hold the
output near system ground, biasing
the LTC4357 in the blocking condition.
If the output is held up at +12V by a
backup source or stored charge in
the output capacitor, roughly double
the input voltage appears across the
MOSFET. The MOSFET is off and held
in the blocking state.
Conclusion
the unregulated battery voltage, and
the COMP4 output alerts the system
to a low battery condition, allowing the
system to enter a standby or power
save mode.
The LTC2932 also provides a
mechanism to override a reset or fault
condition. This is accomplished by
pulling the RDIS pin low. With RDIS
pulled low, the RST output pulls up
to the V2 input voltage. Since V2 is
tied to V1, the reset high level is 5V.
The RDIS function allows the system
to have lexibility in controlling the
power sources without generating system faults. Additionally, the LTC2932
allows real time setting of the voltage
monitoring threshold. This could be
useful when changes in loading or
environment make for predictable
supply variances.
strained applications. The LTC2930 is
available in a 3mm × 3mm DFN, while
the LTC2931 and LTC2932 are available in 20-pin TSSOP packages.
All include design-time saving features for multi-voltage applications.
Voltage thresholds are accurate to
±1.5%, guaranteed over the entire
–40°C to 125°C temperature range.
This translates directly to simpliied
power supply design, as threshold
accuracy must be accounted for in
the entire power supply tolerance
budget.
Comparator glitch immunity eliminates false resets, with no effect on the
high accuracy of the monitor. These
devices support a variety of voltage
combinations, easily set with only a
few external components. The reset
timeout period is also adjustable with
a single capacitor.
Lastly, the features which differentiate the LTC2930, LTC2931 and
LTC2932 give users the lexibility to
choose one for any application. L
The LTC4357 ideal diode controller can
replace a Schottky diode in many applications. This simple solution reduces
both voltage drop and power dissipation, thereby shrinking the thermal
layout and reducing power loss. Its
wide 9V to 80V supply operating range
and 100V absolute maximum rating
accommodate a broad range of input
supply voltages and applications,
including automotive, telecom and industrial. A dual version, the LTC4355,
is available in 4mm × 3mm DFN-14
or SSOP-16 packages. L
LTC29x, continued from page 15
boost regulator and monitored by the
LTC2931. The LTC3780 is protected
from transients by the LT4356DE-1
and is capable of delivering full power
to the load with a supply voltage as low
as 6V. The LTC2931 is conigured to
monitor four ixed and two adjustable
voltages, including two independent
5V supplies. 1.5% voltage monitoring accuracy is guaranteed over the
entire operating temperature range.
Additionally, each voltage monitoring channel has its own comparator
output that can be used by the microprocessor to identify a fault condition.
The comparator outputs are pulled up
to the 5V bus that powers both voltage monitoring devices. The LTC2931
has an adjustable watchdog timer,
which allows the LTC2931 to report
a malfunctioning microprocessor to
the rest of the system.
The unregulated battery voltage
and power supplies delivered to the in
cabin electronics are monitored by the
LTC2932. This application monitors
34
Conclusion
The LTC2930, LTC2931 and LTC2932
can each monitor six supplies, saving
valuable board area in space con-
Linear Technology Magazine • March 2008
DESIGN IDEAS L
CMOS Op Amp Outperforms Bipolar
Amps in Precision Applications
by Hengsheng Liu
Introduction
Superior Precision
CMOS Op Amp
Bipolar ampliiers can have low offset
and low offset drift, but their nA level
input bias current make them inappropriate for high input impedance
applications such as photodiode
ampliiers. CMOS ampliiers usually
offer inferior offset drift, CMRR, and
PSRR speciications and therefore
are not suitable for precision applications. Chopper stabilized ampliiers,
also known as zero drift ampliiers,
can achieve superior offset and offset
drift by means of offset cancellation,
but have clock noise and fold-back
noise due to sampling. LTC6081 and
LTC6082, however, are continuous
time CMOS operational ampliiers,
which use a patented methodology
to improve their offset voltage, offset
voltage drift and CMRR. They combine
the features of low input bias current,
low offset drift and low noise.
30
NUMBER OF AMPLIFIERS (OUT OF 100)
The LTC6081 and LTC6082 are dual
and quad low offset, low drift, low noise
CMOS operational ampliiers with railto-rail input and output stages. Their
0.8µV/°C maximum offset drift, 1pA
input bias current, 1.3µVp-p of 0.1Hz to
10Hz noise, 120dB open loop gain and
110dB CMRR and PSRR make them
perfect for precision applications. The
LTC6081 and LTC6082 have a gain
bandwidth product of 3.6MHz, with
each ampliier only consuming about
330µA current for a supply voltage of
2.7 to 5.5V. The 10-lead DFN package of the LTC6081 offers a shutdown
function to reduce each ampliier’s
supply current to 2µA.
25
R4/R3 and R6/R5 is critical for CMRR.
Gain can be changed by simply changing R0 without affecting the resistor
matching.
The input referred offset of the
ampliier is
LTC6081MS8
TA = –40°C TO 125°C
VS = 3V
VCM = 0.5V
20
15
10
VOS = VOSB − VOSA +
5
0
–0.20
–0.10
0
0.10
VOSDRIFT (µV/°C)
0.20
≈ VOSB − VOSA
0.30
Figure 1. VOS drift histogram of LTC6081
In this two stage structure, the differential voltage passes through the
irst gain stage with gain of 1 + 2R1/R0
while the common mode voltage has
only unity gain at the irst stage, thus
improving CMRR. Ratio matching of
VIN1
+
LTC6081/2
A
–
VOUT

2R1  R5 
=  1+
V − VIN1
R0   R3  IN2

(
)
Linear Technology Magazine • March 2008
Statistically, the total VOS is √2
times the VOS of a single op amp.
Since a single LTC6081 op amp drifts
less than 0.8µV/°C, the ampliier in
Figure 2 will drift less than 1.1µV/°C.
One drawback of the circuit in Figure
2 is its common mode operating range
is no longer rail-to-rail. Assuming
R3
R4
R1
–
R0
LTC6081/2
C
VOUT
+
–
VIN2
LTC6081/2
B
+
R2
R5
R6
Figure 2. Typical three op amp structure of instrumentation amplifier
VIN1
+
LTC6081/2
A
–
R3
R4
–
LTC6081/2
C
Instrumentation Amplifier
Figure 2 shows a typical three op amp
instrumentation ampliier. If R1 = R2,
R3 = R5 and R4 = R6, then
VOSC
2R1
1+
R0
VOUT
+
–
VIN2
LTC6081/2
B
+
R5
R6
Figure 3. Instrumentation amplifier with unity gain buffers
35
L DESIGN IDEAS
the differential and common mode
input voltage are VIN(DM) and VIN(CM)
respectively, the output voltages of op
amp A and B are then VIN(CM) – (2R1/
R0)VIN(DM) and VIN(CM) + (2R1/R0)VIN(DM)
respectively. So
2R
V – < VIN(CM) ± 1 VIN(DM) < V +
R0
2R
V – + 1 VIN(DM) < VIN(CM) <
R0
2R
V + − 1 VIN(DM)
R0
where V+ and V– are the positive and
negative supply voltage respectively.
The larger the irst stage gain or input
differential signal is, the narrower
the input common mode range is. To
widen the input common mode range,
the irst stage gain can be reduced,
but this will compromise CMRR performance.
Figure 3 is a reduced circuit of Figure 2 with a unity gain buffer at the
front stage. This circuit can achieve
rail-to-rail input range. As mentioned
previously, it won’t have the high
CMRR of the circuit in Figure 2 since
we reduced the front stage gain to
unity. If the input resistance requirement can be eased, Figure 3 can be
reduced to Figure 4, a single stage
difference ampliier. The impedance of
the non-inverting and inverting inputs
are R3 and R5 + R6, respectively. An
obvious advantage of the LTC6081 is
its super low input bias current. Even
with a 1MΩ input resistor R3 or R5,
the less than 1pA input bias current
of LTC6081 will add less than 1µV
to VOS.
The above discussion assumes
a perfect matching of R4/R3 and
R6/R5. If
(
)
R6
R4
= 1+ ε
R5
R3
AV
ε
where AV is the differential gain of
the instrumentation ampliier. For
example, at gain of 10, to achieve 80dB
5V 0.1µF
+
–
1M
+
1µF
1M
1/2
LTC6081
VOUT = 10mV/°C
0°C TO 500°C
–
5V
LT1025
2.49M
K
R–
R4
–
LTC6081/2
VOUT
+
R5
R6
VIN2
Figure 4. Difference amplifier
with no input buffers
CMRR, mismatch of R4/R3 and R6/R5
should be less than 0.1%. This is true
for all the above three circuits. The
advantage of the circuit in Figure 2 is
that gain can be put at the front stage
to ease the matching requirements of
the second stage. Matching of R1 and
R2 in Figure 2 is not important.
Thermocouple Amplifier
then the CMRR degrades to
20log
R3
VIN1
10k
100pF
SENSOR: OMEGA 5TC-TT-K-30-36 K-TYPE THERMOCOUPLE
1M RESISTORS PROTECT CIRCUIT TO ±350V WITH NO PHASE REVERSAL OF AMPLIFIER OUTPUT
1pA MAX IBIAS TRANSLATES TO 0.05°C ERROR
90µV VOS → 2°C OFFSET
Figure 5. Thermocouple amplifier
Figure 5 shows the LTC6081 in a
thermocouple ampliier. The 1MΩ resistors protect the circuit up to ±350V
with no phase reversal to ampliier
output. The 1pA maximum IBIAS of
the LTC6081 translates to a miniscule
0.05°C temperature error with the
1MΩ input protection resistor. The
±90µV offset over the entire operating
temperature range ensures a less than
2°C temperature offset.
Conclusion
The LTC6081 and LTC6082 are high
performance dual and quad op amps
combining excellent noise, offset drift,
CMRR, PSRR and input bias current
speciications. They perform in a variety of topologies without compromising
performance. LTC6081 is available in
8-lead MSOP and 10-lead DFN packages. LTC6082 is available in 16-lead
SSOP and DFN packages. L
LTM4605/07, continued from page 19
Conclusion
The LTM4605 and LTM4607 µModule
regulators simplify the design of buckboost power supplies. Their low proile
15mm × 15mm × 2.8mm packages
and minimal component count help
free up valuable PCB area. High input
and high output ratings suit these
36
regulators to networking, industrial,
automotive systems and high power
battery-operated devices. Their optimized internal 4-switch architecture
provides high eficiency and high
performance. Overall, the LTM4605
and LTM4607 reduce product design
and test time with a mix of high per-
formance features, lexible settings
and ease-of-use. L
Notes
1 For more about layout with Kelvin sense resistors, see “Using Current Sensing Resistors with
Hot Swap Controllers and Current Mode Voltage
Regulators” by Eric Trelewicz in Linear Technology
Magazine, September 2003, page 34
Linear Technology Magazine • March 2008
NEW DEVICE CAMEOS L
New Device Cameos
Low Voltage Hot Swap
LTC4210-3 retries after circuit breaker
Second, the LTC6652 exhibits low
Controller Provides Additional timeout, whereas the LTC4210-4 con- noise, making it a good choice for
GPIO Capability
tinue latches off till system reset.
systems that require large dynamic
The LTC4215-1 is a low voltage Hot
Swap controller with an onboard
ADC and I2C compatible interface.
Functionally, the LTC4215-1 is similar to the LTC4215 with additional
General Purpose Input/Output (GPIO)
functionality for a total of three GPIO
pins.
The additional GPIO pins of the
LTC4215-1 may be used to light LEDs
to lag a card for service, turn on or
reset downstream circuits, or monitor digital signals from other blocks.
GPIO1 and GPIO2 default to a high
impedance state for an output high or
digital input, and GPIO3 defaults to
output low. This mix of states provides
lexibility in the application—a simple
circuit can be used to invert the state
of the pin or provide a higher current
if desired.
The GPIO1 pin may still be conigured to signal power good, as in the
LTC4215, and the GPIO2 pin may also
still be used to generate fault alerts in
addition to the new GPIO functionality.
The GPIO3 pin replaces the ADR2 pin
on the LTC4215, reducing the available
addresses from 27 to 9.
The LTC4215-1 works in applications from 12V (with transients to 24V)
down to 3.3V where the operating voltage can drop to 2.9V and is available
in a 4mm × 5mm QFN package.
Low Voltage Current Limiting
Hot Swap Controller
The LTC4210-3 and LTC4210-4 are
new members of the LTC4210 family
of tiny SOT-23 Hot Swap controllers.
These two products are ideal for low
voltage applications from 2.7V to 7V
where superior current limit responses
are absolutely essential to high performance systems. The LTC4210
rides through short duration of overload transients. Severe load faults
are isolated after a programmable
circuit breaker time-out to prevent
system and MOSFET damages. The
Linear Technology Magazine • March 2008
The gate driver maximum output
voltage is clamped to ground with a
12V Zener.
The LTC4210-3 and LTC4210-4
allow safe board insertion and removal with inrush current control.
The LTC4210-3 and LTC4210-4 also
can be utilized as high side gate driver
to control a small footprint logic level
MOSFET.
Precision Voltage Reference
with Wide Operating
Temperature Range Simplifies
High Temperature Industrial
and Automotive Design
The LTC6652 is a high precision, low
noise voltage reference with internal
output buffering that is fully speciied
over the –40°C to 125°C temperature
range. The wide operating temperature
range, combined with low noise and
low power consumption, allows system
designers to achieve high performance
in the most demanding applications.
First and foremost, the LTC6652
is a high precision voltage reference.
Speciications include a maximum
0.05% initial accuracy and 5ppm/°C
temperature drift. This precision satisies the high performance application
requirements of monitor and control
systems, instrumentation and test
equipment.
The part is designed using high
precision circuitry carefully tailored to
high temperature operation. In order
to reliably meet the speciications in
high volume manufacturing, the low
temperature drift is consistent from
part to part. These characteristics
make it easy to design systems using
the LTC6652. Perhaps most important,
the performance is measured at the
temperature extremes for every unit,
not just sample tested. This gives system designers additional conidence in
their product’s quality. The electrical
speciications are also guaranteed
across the entire temperature range.
range. At only 2.1ppmP–P (0.1Hz to
10Hz), the noise is well below the
drift, allowing full access to the high
performance accuracy and drift characteristics. The LTC6652 makes it easy
to achieve repeatability, stability and
wide dynamic range in just about any
high performance application.
Third, the LTC6652 simpliies system designs by eliminating the need for
a buffer ampliier. The output can both
sink and source 5mA, and is stable for
a wide range of load capacitances. Both
line and load regulation are exceptional, maintaining system performance
under a wide range of conditions while
reducing complexity.
Finally, the LTC6652 requires only
a small amount of power and board
space. It draws only 350µA supply
current, and is available in an 8-lead
MSOP package.
With its high precision, wide
temperature range, low noise and
small size, system designers should
be able to meet the most demanding
speciications. Using parts that are
individually and fully tested at three
temperatures will help them sleep at
night.
Dual/Triple Supply Monitor
Maintains 1.5% Accuracy
Over Temperature
The LTC2919 is a triple/dual input
monitor intended for a variety of system monitoring applications. The 0.5V
threshold of the two adjustable inputs
features a tight 1.5% accuracy over the
entire operating temperature range.
Additionally, an accurate threshold at
the VCC pin provides an undervoltage
monitor for a 2.5V, 3.3V or 5V supply.
Each input features glitch rejection,
which ensures that the outputs operate reliably without false triggering.
The polarity selection pin (SEL) and
buffered reference output (REF) allow
the LTC2919 to monitor positive and
negative supplies for undervoltage
37
L NEW DEVICE CAMEOS
(UV) and overvoltage (OV) conditions.
It can monitor two supplies for UV
or OV, or a single supply for UV and
OV simultaneously. The adjustable
trip thresholds are set with external
resistive divider networks, giving users
complete control over the trip voltage.
An open-drain RST output is held low
when any adjust supply is invalid
or VCC is in undervoltage. When all
the inputs are valid, the RST pin is
released after a timeout delay, which
can be set to 200ms, adjusted with
an external capacitor, or conigured
for no-delay.
When compared to the LTC2909,
the LTC2919 provides two additional
independent output pins to indicate
the status of each adjustable input.
When connected to the enable pins of
power supplies these outputs can be
used to implement start-up sequencing. In addition to providing a highly
versatile, precise solution for supply
monitoring, the low quiescent current
of 50µA and the tiny DFN package
makes the LTC2919 an ideal choice
in space limited applications. With
the addition of a single external current limiting resistor, the LTC2919’s
onboard 6.5V shunt regulator permits
operation from a high voltage supply.
The IC is offered in 10-pin plastic
MSOP and 3mm × 2mm DFN packages
and is speciied over the C, I, and H
temperature ranges. L
LT6105/6/7, continued from page 26
constraints of the absolute maximum
ratings, is that the negative supply to
the LT6105 be at least as negative as
the supply it is monitoring.
lexibility of external gain setting. The
–0.1V to 44V input range of the LT6105
enables the current in switched supplies to be monitored from initial
turn-on/turn-off to the steady state
value. The LT6106 provides a simple
but accurate solution for systems with
a single supply. The LT6107 extends
the temperature range of current
measurements to 150°C. L
noid to increase the solenoid current.
It lets go to decrease current, and the
solenoid voltage freewheels around the
Schottky diode. Current measurement
waveforms are shown in Figure 7. The
small glitches occur due to the action of
the solenoid plunger, and this provides
an opportunity for mechanical system
monitoring without an independent
sensor or limit switch.
Figure 8 shows another solenoid
driver circuit, a high side drive approach with one end of the solenoid
grounded and a P-Channel MOSFET
pulling up on the other end. In this
case, the inductor freewheels around
ground, imposing a negative input
common mode voltage of one Schottky
diode drop. This voltage may exceed
the input range of the LT6105. This
does not endanger the device, but it
degrades the accuracy. In order to
avoid exceeding the input range, pullup resistors may be used as shown.
Conclusion
The LT6105, LT6106, and LT6107
provide simple, lexible solutions to
high side (and low side) current sensing. Common to all the parts is the
CURRENT FLOW
+15V
POSITIVE
SUPPLY
+IN
38
5VDC
V+
–15V
V–
–
20mΩ
1%
LT6105
100Ω
1%
TO +15V
LOAD
–IN
VOUT
VOUT = 1V/A
4.99k
1%
LT6105
LT6105 Application:
Supply Monitor
The input common mode range of
the LT6105 also makes it suitable for
monitoring either positive or negative
supplies. Figure 9 shows one LT6105
applied as a simple positive supply
monitor, and another LT6105 as
a simple negative supply monitor.
Note that the schematics are practically identical, and both have outputs
conveniently referred to ground. The
only requirement for negative supply
monitoring, in addition to the usual
+
100Ω
1%
–15V
V–
5VDC
V+
–15V
NEGATIVE
SUPPLY
VOUT = 1V/A
4.99k
1%
–IN
100Ω
1%
VOUT
+IN
–
20mΩ
1% +
100Ω
1%
TO –15V
LOAD
CURRENT FLOW
Figure 9. The LT6105 can monitor the current of either positive or negative supplies, without
a schematic change. Just ensure that the current flow is in the correct direction.
Linear Technology Magazine • March 2008
DESIGN TOOLS L
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Linear Technology Magazine • March 2008
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