V19N1 - MARCH

LINEAR TECHNOLOGY
MARCH 2009
IN THIS ISSUE…
COVER ARTICLE
Battery Stack Monitor
Extends Life of Li-Ion Batteries
in Hybrid Electric Vehicles ..................1
Michael Kultgen and Jon Munson
Linear in the News… ...........................2
DESIGN FEATURES
DC/DC Converter, Capacitor Charger
Takes Inputs from 4.75V to 400V ........9
VOLUME XIX NUMBER 1
Battery Stack Monitor
Extends Life of Li-Ion
Batteries in Hybrid
Electric Vehicles
by Michael Kultgen and Jon Munson
Robert Milliken and Peter Liu
How to Choose a Voltage Reference ...14
Brendan Whelan
Introduction
1.2A Monolithic Buck Regulator
Shrinks Supply Size and Cost with
Programmable Output Current Limit
.........................................................20
The cost of running a car on electricity
is equivalent to paying $0.75/gallon
for gasoline, and if that electricity
comes from carbon neutral sources,
car owners are saving both money
and the environment (gasoline combustion produces 9kg of CO2 per US
gallon). Advancements in battery
technology (see sidebar), especially
with Lithium-based chemistries, hold
the greatest promise for converting
the worldwide leet of cars to hybrid
or fully electric.
Tom Sheehan
Boost Converters for Keep-Alive Circuits
Draw Only 8.5µA of Quiescent Current
.........................................................22
Xiaohua Su
Industrial/Automotive Step-Down
Regulator Accepts 3.6V to 36V and
Includes Power-On Reset and Watchdog
Timer in 3mm × 3mm QFN ................24
Ramanjot Singh
Complete APD Bias Solution in 60mm2
with On-the-Fly Adjustable Current
Limit and Adjustable VAPD ...................27
Xin (Shin) Qi
DESIGN IDEAS
Don’t Want to Hear It? Avoid the Audio
Band with PWM LED Dimming at
Frequencies Above 20kHz ..................30
+
Lithium battery packs offer the
highest energy density of any current battery technology, but high
performance is not guaranteed simply by design. In real world use, a
battery management system (BMS)
makes a signiicant difference in the
performance and lifetime of Li-Ion
batteries—arguably more so than
the design of the battery itself. The
LTC6802 multicell battery stack
monitor is central to any BMS for the
continued on page 12-CELL BATTERY
MODULE
12-CELL BATTERY
MODULE
12-CELL BATTERY
MODULE
12-CELL BATTERY
MODULE
+
+
+
+
CURRENT
SENSOR
Eric Young
–
–
–
BATTERY
MONITORING
& BALANCING
BATTERY
MONITORING
& BALANCING
BATTERY
MONITORING
& BALANCING
BATTERY
MONITORING
& BALANCING
DATA BUS
DATA BUS
DATA BUS
DATA BUS
DATA BUS
DATA BUS
DATA BUS
DATA BUS
BATTERY
MONITORING
& BALANCING
BATTERY
MONITORING
& BALANCING
BATTERY
MONITORING
& BALANCING
BATTERY
MONITORING
& BALANCING
SERVICE SWITCH
Eliminate EMI Worries with 2A,
15mm × 9mm × 2.82mm µModule™
Step-Down Regulator ........................33
–
David Ng
Diode Turn-On Time Induced Failures
in Switching Regulators ....................34
CAN
Jim Williams and David Beebe
µModule Regulator Fits a (Nearly)
Complete Buck-Boost Solution in
15mm × 15mm × 2.8mm for
4.5V–36V VIN to 0.8V–34V VOUT ..........39
Judy Sun, Sam Young and Henry Zhang
New Device Cameos ...........................41
–
HOST
CONTROLLER
SPI
–
+
12-CELL BATTERY
MODULE
–
+
12-CELL BATTERY
MODULE
–
+
12-CELL BATTERY
MODULE
–
+
12-CELL BATTERY
MODULE
Design Tools ......................................43
Sales Offices .....................................44
Figure 1. 96-cell battery pack
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. Adaptive Power, Bat-Track, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView,
µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational
Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, TimerBlox, True
Color PWM, UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks
of the companies that manufacture the products.
L LINEAR IN THE NEWS
Linear in the News…
EDN Highlights Linear for Innovation Awards
Linear CEO Comments on Growth Markets
EDN magazine in February chose several Linear Technology
products as inalists for their annual Innovation Awards, to
be announced later this month. And the nominees are:
Last month in EE Times, Linear Technology CEO Lothar
Maier discussed the challenging market conditions and
the bright spots on the horizon: “In these times our customers will continue to invest in new products and new
product development. Innovation will return growth to the
semiconductor market—speciically to analog. Now is the
time to get new products out, to be irst to market and to
have products that target emerging growth markets.” He
discussed several key markets:
q Automotive. “Automotive manufacturers are
forecasting automotive electronic content to grow 2–3
times over the next few years, so we will continue
to provide new products to the automotive area. In
addition, every major automotive manufacturer in the
world is now working on hybrid vehicles, which will
add even more electronic content in cars. We have
just introduced an innovative device, the LTC6802,
a highly integrated battery stack monitor that
signiicantly eases the design of battery monitoring
systems for hybrid/electric vehicles.”
q Green Growth Markets. “Products targeted toward
energy conservation or energy harvesting will
see growth opportunities and are insulated from
the current market conditions. Energy costs and
environmental concerns, as well as the need to
extend battery life for mobile devices, have led to a
focus on power optimization. Our energy-eficient
products enable customers to convert power more
eficiently, consume less power and extend battery
life. Our LED drivers enable a new generation of
low power lighting for a range of applications, from
cars and medical instruments to laptops and ofice
lighting. Our eficient analog solutions will help drive
innovative cleantech markets such as solar and wind
power systems.”
q Communications Infrastructure. “Wireless systems
continue to produce signiicant market opportunities
for products in wireless and network infrastructure.
Our high speed data converters and high frequency
products are designed into the next generation of
cellular basestations. And our Hot Swap™ and Power
over Ethernet products are proliferating in networks.”
q Industrial. “The broad industrial market continues
to provide a solid core of business and is somewhat
more insulated from market swings. Linear’s analog
products are used in a broad range of industrial
systems, including factory automation, industrial
process control, medical, instrumentation and
security.”
Lothar Maier concluded, “Finally, I believe that Linear’s
strategy of customer, market and geographic diversity will
be a hedge against the current market conditions and will
provide the conduit to future growth.” L
Best Contributed Article—“High Voltage,
Low-Noise DC/DC Converters” by Jim Williams
You can ind the article in its entirety on the EDN website
at www.edn.com/jimwilliams.
Battery ICs Category—LTC6802 Battery Stack Monitor
The LTC6802 is a highly integrated multicell battery
monitoring IC capable of precisely measuring the voltages
of up to 12 series-connected battery cells. Using a novel
stacking technique, multiple LTC6802s can be placed in
series without optocouplers or isolators. See the cover
article of this issue for an overview of this part.
Power ICs Category—LTC3642 50mA
Synchronous Step-Down Converter
The LTC3642 uses a unique high voltage synchronous
rectiication design, capable of continuous input voltages of 45V and offers transient protection up to 60V. Its
internal synchronous rectiication and its programmable
peak current mode control feature enable it to deliver up
to 93% eficiency, maximizing battery run time.
Power ICs: Modules—LTM4606 Ultralow EMI,
6A DC/DC µModule Regulator
The LTM4606 DC/DC µModule™ regulator signiicantly
reduces switching regulator noise by attenuating conducted
and radiated energy at the source. The µModule device
is a complete DC/DC system-in-a-package, including the
inductor, controller IC, MOSFETs, input and output capacitors and the compensation circuitry, housed in an enclosed
surface-mount plastic package resembling an IC.
The LTM4606 reduces switching regulator noise at the source.
2
Linear Technology Magazine • March 2009
DESIGN FEATURES L
LTC6802, continued from page large battery stacks common in electric vehicles (EVs) and hybrid electric
vehicles (HEVs). Its robust design and
high accuracy helps guarantee the
performance and lifetime of expensive
battery packs.
For instance, to meet a 15-year,
5000 charge cycle goal, only a portion
(say 40%) of the battery pack’s cellcapacity can be used. Of course, using
only 40% of the capacity essentially
lowers the energy density of the pack.
This is the problem: increasing battery
lifetime is traded against the need
to use as few kg of batteries as possible—the most expensive component
in any EV. Only a well-designed BMS
can maximize battery performance and
lifetime in the face 200A peak charge
and discharge currents.
Battery Management System
Optimizes Li-Ion Run Time
and Lifetime
In any battery stack, the more accurately you know state of charge (SOC)
of each cell, the more cell capacity you
can use while still maximizing cell life.
In a laptop computer, gas gauging
comes from monitoring cell voltage
and counting coulombs in and out of
the stack of four to eight cells. Voltage, current, time and temperature
are combined in a robust algorithm
to give an indication of the SOC. Unfortunately, it’s nearly impossible to
count coulombs in a car. The battery
drives an electric motor, not a motherboard, so it must handle current
spikes of 200A, followed by low level
idling. Furthermore, you have from 96
Li-ion Batteries in Electric Vehicles and Hybrids
So why aren’t all cars electric? One
regenerative braking means the gas
Table 1. Energy density comparison
reason is energy density. Gasoline
engine runs less often and runs at a
holds 80 times the energy per kg as
higher eficiency, effectively doubling
Medium
Wh/kg
Li-ion batteries (Table 1) and refuels
the mpg.
Diesel Fuel
12,700
in three minutes, essentially allowing
In the 1970s the only available high
Gasoline
12,200
indeinite driving. Even a big lithium
power battery chemistry was lead
pack only gives a passenger car
acid, too heavy to reasonably power
Li-Ion Battery
150
about a 100-miles after an 8-hour
anything larger than a golf cart. Then
NiMh Battery
100
charging cycle. To drive a passenger
came NiMh batteries, which improved
car further than 100 miles you still
energy density enough to enable the
Lead Acid Battery
25
need a gasoline engine, but even
irst commercially successful HEVs,
so, batteries improve gas mileage in
like the Toyota Prius and Ford Escape.
hybrid electric vehicles (HEVs). The peak eficiency of Li-ion batteries take energy density another step forward,
the Otto cycle engine is only 30% at high RPMs and the by offering another 50% improvement. The safety of Liaverage eficiency is about 12%. Using batteries to sup- ion was a concern, but new battery technologies like the
ply torque during acceleration and recover joules during A123 nanophosphate cell, the EnerDel Spinel-Titanate
chemistry, the GS Yuasa EH6 design and others are as
safe as NiMh, offer extremely high power (200A peak disMG1 INVERTER
BATTERY
MG2 INVERTER
charge rates), and last 10 to 15 years with proper charge
management. By model year 2012, the majority of hybrid
cars and trucks will use lithium battery technology.
Figure 1 shows a shows a block diagram of the batGASOLINE
SILENT
ENGINE
CHAIN
POWER SPLIT
tery
pack with a BMS, and Figure 2 shows a typical HEV
DEVICE
power train. The battery pack building block is a 2.5V
to 3.9V, 4Ahr to 40Ahr Li-ion cell. 100 to 200 cells are
connected in series to bring the battery pack voltage into
the hundreds of volts. This DC power source drives a
30kW to 70kW electric motor. The pack voltage is high
ELECTRIC MOTOR/
ELECTRIC MOTOR/
so that the average current is low for a given power level.
GENERATOR 1 (MG1)
GENERATOR 2 (MG2)
Lower current reduces I2R power losses, so cables can
be smaller, thus reducing weight and cost. The pack
REDUCTION
should be able to deliver 200A under peak conditions
GEARS
and be quickly rechargeable. In other words, the battery
needs to offer high energy density and high power denAXLES
sity, speciications that can be met by Li-ion batteries.
FRONT
WHEELS
Systems for busses and tractor-trailers use up to four
parallel packs of 640V each. L
DIFFERENTIAL
Figure 2. Toyota Prius “split power” hybrid drive train
Linear Technology Magazine • March 2009
3
4.5
4.5
4.0
4.0
CELL VOLTAGE (V)
to 200 cells in series, in groups of 10
or 12. The cells age at different rates,
were manufactured from multiple lots,
and vary in temperature. Their capacities diverge constantly. Different cells
with the same coulomb count can have
wildly different charge levels.
That’s why the BMS focuses on
cell voltage. If you can accurately
measure the voltage of every cell, you
can know the cell’s SOC with reasonable accuracy (Figure 3). The trick is
to improve the accuracy of the voltage
measurement by taking into account
temperature effects on battery ESR
and capacity. By constantly measuring
each cell’s voltage, you keep a running
estimation of each cell’s charge level.
If some cells are overcharged and
some under, they can be balanced by
bleeding off charge (passive balancing) or redistributing charge (active
balancing).
CELL VOLTAGE (V)
L DESIGN FEATURES
3.5
3.0
1C
2C
5C
10C
20C
50C
2.5
2.0
1.5
0
3.5
3.0
2.5
–20°C
0°C
30°C
60°C
2.0
1.5
10 20 30 40 50 60 70 80 90 100
DISCHARGE (%)
0
Figure 3. State of charge vs current and temperature for a typical Li-ion cell
NEXT 12-CELL
PACK ABOVE
LTC6802-1 SERIAL DATA
TO LTC6802-1
ABOVE
DIE TEMP
V+
REGISTERS
AND
CONTROL
12-CELL
BATTERY
STRING
MUX
12-BIT
∆∑ ADC
Accurate Monitoring is Key to
Raising Battery Performance
while Lowering Costs
4
V–
EXTERNAL
TEMP
NEXT 12-CELL
PACK BELOW
VOLTAGE
REFERENCE
SERIAL DATA
TO LTC6802-1
BELOW
100k
100k NTC
Figure 4. Simplified block diagram of the LTC6802
estimation of SOC is accurate to 3%.
The BMS must charge cells to no more
than 37% (40% – 3%) of their capacity
to guarantee the 15-year lifetime.
Now consider a monitor IC with
10mV error over similar conditions.
In this case, the BMS can only use
32% (40% – 10mV • 1%/1.25mV) of
the cells’ capacity and still guarantee a
15-year life. This seemingly negligible
increase in measurement error results
in a signiicant 14% reduction in the
usable capacity. That is, a vehicle
requires least 14% more batteries, or
9k
0.30
COST OF TYPICAL BATTERY PACK ($)
0.25
MEASUREMENT ERROR (%)
The LTC6802 (Figure 4) is a precision data acquisition IC optimized for
measuring the voltage of every cell in
a large string series-connected batteries. In the BMS, the LTC6802 does the
heavy lifting analog function, passing
digital voltage and temperature measurements to the host processor for
SOC computation. The LTC6802’s high
accuracy, excellent noise rejection,
high voltage tolerance, and extensive
self-diagnostics make it robust and
easy-to-use. The high level of integration means a substantial cost savings
for customers when compared to
discrete component data acquisition
designs.
Increasing measurement accuracy
reduces battery cost, as illustrated
by the following example. Figure 5
shows the typical performance of the
LTC6802, where 0.1% total error from
–20°C to 60°C translates to 4mV precision for a 3.7V cell. Suppose that to
achieve a 15-year battery lifetime, you
are limited to 40% of a cell’s capacity
per charge cycle, and assume the cell
voltage vs charge level of the battery
is very lat, e.g., 1.25mV/%SOC. A
measurement error of 4mV means the
10 20 30 40 50 60 70 80 90 100
DISCHARGE (%)
7 REPRESENTATIVE
UNITS
0.20
0.15
0.10
0.05
0
–0.05
–0.10
–0.15
–0.20
–0.25
–0.30
–50
8k
7k
6k
5k
4k
3k
–25
0
25
50
75
TEMPERATURE (°C)
100
125
Figure 5. Typical measurement accuracy
vs temperature of seven samples
0
5
20
25
10
15
MEASUREMENT ERROR (mV)
30
Figure 6. High BMS accuracy is important to
keeping battery costs in check, as shown in
this cost vs measurement error model.
Linear Technology Magazine • March 2009
DESIGN FEATURES L
TP0610K
CELL12
6µs
370V
1M
2.2M
LTC6802-1
270V
0 = REF_EN
GPIO2
10kHz
VSTACK12
0 = CELL1
GPIO1
Figure 8. Inverter noise example
WDTB
1M
1M
LT1461A-4
10M
1M
DNC DNC
DNC
VIN
SD VOUT
GND DNC
VREG
2N7002
1µF
90.9k
2N7002
V−
4.096V
2.2µF
C1
150Ω
TP0610K
+
TP0610K TP0610K
100Ω
VDD CH0 CH1 SEL
SD LT1636
100nF
TC4W53FU
–
CELL1
COM INH VEE VSS
1M
Figure 7. Improving accuracy with calibration
Linear Technology Magazine • March 2009
of 3.7V, which we need to measure
to 4mV. Breaking the battery stack
into 12-cell modules further reduces
The LTC6802’s 0.1% total
measurement error from
–20°C to 60°C translates to
4mV precision for a 3.7V cell.
Batteries are expensive. It
takes about $4000 worth
of batteries to drive 50
miles, so just increasing
measurement error to 10mV
means $560 in additional
cells. This is why BMS
designers scrutinize every
0.01% of measurement error.
Diagnostic Features of the
LTC6802 Improve Robustness
Automotive systems require that “no
bad cell reading be misinterpreted
as a good cell reading.” Two of the
more common faults that can cause
false readings are open circuits and
IC failures. If there is an open circuit
in the wiring harness and if there is
a ilter capacitor on the ADC input
(Figure 11), the capacitor will tend
to hold the input voltage at a point
midway between the adjacent cells.
Some type of open wire detection or
cell resistance measuring function
is necessary. The LTC6802 includes
100µA current sources to load the cell
inputs. The current source will cause
large changes in cell readings if there
is an open circuit in the harness.
0
0
VCM(IN) = 5VP-P
72dB REJECTION
–10 CORRESPONDS TO
LESS THAN 1 BIT
–20 AT ADC OUTPUT
–10
–20
REJECTION (dB)
REJECTION (db)
at least 14% more weight, cost and
electronics to travel an equivalent
distance as a vehicle with the more
accurate BMS. Batteries are expensive. It takes about $4000 worth of
batteries to drive 50 miles, so the
increased measurement error means
$560 in additional cells. This is why
BMS designers scrutinize every 0.01%
of measurement error. Figure 6 shows
a simple battery cost model as a function of BMS accuracy.
Adding a low drift reference, an initial factory calibration, and a periodic
self-calibration routine can improve
the measurement accuracy of the
LTC6802 to 0.03%. For example, in
Figure 7 the LT1461A-4 is periodically
applied to channel C1. The temperature stable LT1461 measurement is
used to correct temperature drift in
the LTC6802. The initial error of the
LTC6802 and LT1461A is corrected by
measuring and storing a calibration
reference after board assembly.
Inverter noise can seriously interfere with cell voltage measurements.
When a 100-cell stack is loaded by an
electric motor it can have a 370V open
circuit voltage and up to 100V switching transients (Figure 8). Spreading
the transient equally over the 100 cells
means the top cell has 370V of common mode voltage, 100V of common
mode transients, 1V of differential
transients and an average DC value
the common mode voltage. In a pack
like Figure 2, each LTC6802 (one
per module) sees up to 12V common
mode transients and 1V differential
transients per cell. The transients
are at the PWM frequency of 10kHz
to 20kHz. The LTC6802 has excellent
common mode rejection (Figure 9) to
eliminate this error term. The SINC2
ilter inherent in the delta-sigma ADC
attenuates the differential noise by
40dB (Figure 10). External iltering or
measurement averaging can be used to
further reduce the differential noise.
–30
–40
–30
–40
–50
–50
–60
–60
–70
10
100
1k
10k 100k
FREQUENCY (Hz)
1M
Figure 9. Cell measurement
common mode rejection
10M
–70
10
100
1k
10k
FREQUENCY (Hz)
100k
Figure 10. Cell measurement filtering
5
L DESIGN FEATURES
The host controller must be able
to run diagnostics on all the modules
during normal operation to detect IC
failures. If these periodic self-tests fail,
then the control algorithm is suspect
and the battery pack must be taken off
line. The LTC6802 includes a built-in
self-test in combination with external
support circuits to allow the BMS to
completely verify the data acquisition
system. See the LTC6802 data sheets
for more details.
LTC6802-1
C4
B4
CF4
B3
CF3
C2
MUX
C1
V–
The LTC6802 Isolates
Communications from
Swings in Ground Potential
Breaking a ~100 cell pack into modules makes it easier to integrate the
analog circuits. Unfortunately, we are
left with the task of getting the data
from measurement IC to the host controller when the difference in ground
potential exceeds 300V. The LTC6802
can solve this problem in a number of
ways, depending on the speciic needs
of the application.
The LTC6802 comes in two lavors,
depending on the desired data communication scheme. The LTC6802-1
offers a built-in stackable serial
peripheral interface (SPI) solution
designed for easy daisy chaining of the
interface. The addressable LTC6802-2
is designed for bus-oriented (parallel)
SPI communication, but it can also be
used in a parallel-addressable, daisy
chained interface for a robust and rela-
C3
100µA
Figure 11. Current sources help detect open circuits.
tively inexpensive solution. All three
schemes are described below.
SPI Bus Communication with
the Addressable LTC6802-2
and Digital Isolators
The most straightforward approach is
to use a bus communications scheme,
with a digital isolator between each
module and the host controller. Figure 12 shows a 96-cell pack using
eight multicell modules monitored
by the LTC6802. The physical layer
is a 4-wire SPI bus. An addressing
scheme allows the control module to
talk to the battery modules separately
or in unison. The data buses on the
modules are isolated from one another.
This is a robust scheme, but it has
one major drawback: digital isolators
are expensive and require an isolated
power supply so that the battery cells
don’t have to provide the power to the
cell side of the isolator.
Daisy Chaining the SPI Interface
with the LTC6802-1
The LTC6802-1 provides ixed 1mA
signaling between stacked devices to
enable easy implementation a daisy
chained SPI interface with inexpensive
support circuitry. The digital isolators
are eliminated as shown in Figure 13.
The interface exploits the fact that the
positive supply of module “N” is the
same voltage as the ground of module
“N+1.” A 1mA current is used to transmit data between adjacent modules.
Like the analog circuits, the modular
approach means the data bus has to
deal with a fraction of the total pack
voltage.
BATTERY MODULE 8
LTC6802
12 Li-Ion
SERIES
BATTERIES
BATTERY
MONITOR
DIGITAL
ISOLATOR
BATTERY MODULE 1
CONTROL MODULE
GALVANIC
ISOLATOR
LTC6802
µCONTROLLER
12 Li-Ion
SERIES
BATTERIES
BATTERY
MONITOR
DIGITAL
ISOLATOR
SPI
CAN
CAN
TRANSCEIVER
TO VEHICLE
CAN BUS
Figure 12. Using digital isolators to communicate to the LTC6802
6
Linear Technology Magazine • March 2009
DESIGN FEATURES L
BATTERY MODULE 8
LTC6802
12 Li-Ion
SERIES
BATTERIES
BATTERY
MONITOR
CONTROL MODULE
BATTERY MODULE 1
GALVANIC
ISOLATOR
LTC6802
µCONTROLLER
12 Li-Ion
SERIES
BATTERIES
BATTERY
MONITOR
SPI
CAN
CAN
TRANSCEIVER
TO VEHICLE
CAN BUS
Figure 13. Using the daisy chained SPI to eliminate digital isolators
The disadvantage of any pure daisy
chain is that a fault in one module
results in a loss of communications
with all the modules above it in the
stack. Also, since there is no galvanic
isolation between modules, the interface needs to handle large voltages
that occur during fault conditions.
For example if the “service switch” in
Figure 1 is open and there is a load
on the pack then the data bus connection between modules 4 and 5 will
see a reverse voltage equal to the total
pack voltage (–300V to –400V). The
LTC6802 interface relies on external
discrete diodes to block the reverse
voltage during fault conditions.
The Best of Both Worlds:
Daisy Chained, Addressable
Interface with the LTC6802-2
With inexpensive external circuitry,
the LTC6802-2 can also be used in
a stacked SPI coniguration like the
LTC6802-1, but with more lexibility
in the operating parameters.
The SPI port of the LTC6802-2 is
a 4-wire connection: chip select in
(CSBI), clock in (SCKI), data in (SDI),
and data out (SDO). The inputs are
conventional CMOS levels and the
output is an open-drain NMOS. The
SDO pin must have an external pull-up
current or added resistance suitable
for the intended data rate. The IC also
provides a versatile always-on 5V output (VREG), which can produce up to
Linear Technology Magazine • March 2009
4mA to energize low power auxiliary
circuitry.
Figure 14 shows a complete stacked
LTC6802-2 SPI interface for a 36cell application. The stack can be
increased in size by replicating the
circuit of the middle IC. In Figure 14,
the VREG and V– pins of each stacked
IC are used to bias common-base
connected transistors to form a signal
translation current for each SPI data
line. Each LTC6802 can monitor up
VBATT
LTC6802-2
IC #3
VREG
1M
1.8k
WDT
2.2k
2.2k
2.2k
NDC7002N
ALL NPN: CMPT8099
ALL PNP: CMPT8599
ALL PN: RS07J
ALL SCHOTTKY: CMD5H2-3
SDI
SCKI
CSBI
SDO
V−
LTC6802-2
IC #2
VREG
100Ω
2.2k
2.2k
2.2k
100Ω
2.2k
2.2k
2.2k
SDI
SCKI
CSBI
SDO
V−
LTC6802-2
IC #1
VREG
SDI
SCKI
CSBI
SDO
V−
R12
2.2k
CS
CK
DI
DO
HOST µP
500kbps MAX DATA RATE
Figure 14. Inexpensive SPI daisy chain for parallel-addressed LTC6802-2
7
L DESIGN FEATURES
to 12 cell-potentials, which could sum
to 60V in certain instances, so the
transistors selected for the SPI translation need to have a VCBO over 60V, but
they should be the highest available fT
to prevent undue slowing of the logic
signals. A suitable NPN candidate is
the CMPT8099, while the CMPT8599
is its PNP complement, both from
Central Semiconductor. These are fast
80V devices (fT > 150MHz).
Sending Signals Upwards
At the bottom-of-stack IC, the logic
signal is furnished by the host connection, be it a microprocessor or an
SPI isolation device. By simply pulling
down the emitter leg of an NPN having
a VREG base potential through a known
resistance, a speciic current is formed
for a logic low input signal. In the case
of the component values shown, the
current is about 2mA for a logic low,
and conversely, the transistor is essentially turned off with a logic high
(~0mA for 5V logic).
Since the collector current is nearly
identical to the emitter current, the
same current pulls on the next higher
cascode circuit. Since that next circuit
is the same as the irst, the voltage on
the upper emitter resistor reproduces
that of the bottom circuit logic level for
the upper IC. This continues up the
daisy chain, eventually terminating at
the top potential of the battery stack.
Since each IC is provided the same input waveforms, this structure forms a
parallel bus from a logical perspective,
even though each IC is operating at a
different potential in the stack.
The NPN transistors at the top IC
source the logic current directly from
the battery stack. Only small base
currents low from any VREG output.
The 600V collector diodes provide reverse-voltage protection in the event a
battery group interconnection is lost,
perhaps during service (these are not
required for functionality and could
be omitted in some situations).
Bringing Data Down the Stack
The SDO cascode chain is similar in
concept, except the current starts at
the top of the stack and lows downward. At the top IC, a PNP transistor
8
with its base connected to the local
V– pin has current injected into its
emitter by a pullup resistor. Here
again, the collector current is essentially identical to the emitter current,
and so current flows downward
through each successive PNP and terminates into a resistor at the bottom
of stack. In this case, the presence of
the current in the termination resistor,
about 2mA for the component values
shown, forms a logic high potential
for the host interface.
A Schottky diode is connected from
each SDO pin to the emitter of a local
PNP thereby allowing any LTC6802
on the stack to divert the pullup current to the local V– when outputting
a logic low. This effectively turns off
the emitter current to the local PNP
transistor and all points lower in the
stack, so the voltage on the bottom
termination resistor then drops to a
logic low level. Since each SDO pin
can force a low level, this forms a
wire-OR function that is equivalent
to paralleled connections as far as
the host interface is concerned. Note
the bottom of stack SDO diode is connected slightly differently; it forms a
direct wire-OR at the host interface.
Since the LTC6802-2 is designed to
use addressed readback commands,
this line is properly multiplexed and
no inter-IC contention occurs.
To eliminate the pull-up current
during standby, a general purpose
N-channel MOSFET is used to interrupt the top PNP emitter current when
the watchdog timer bit goes low. The
watchdog timeout will release when
clock activity is present, so the SDO
line will reactivate as needed. Here
again, an NPN is used at the top of
stack to ensure the pull-up current
comes directly from the battery, rather
than loading VREG.
Collector diodes are added here as
well to provide a high reverse voltage
protection capability, plus some added
series resistance is included to protect
the lower transistor emitters from
transient energy (once again, these
protection parts don’t add any other
functionality to the data transmission and could be omitted in some
circumstances).
External SPI Advantages
Since the LTC6802-2 uses a parallel
addressable SPI protocol, the conventional method of connecting multiple
devices in a stack is to provide isolation
for each SPI connection, then parallel
the signals on the host side. Isolators
are relatively expensive and often need
extra power circuitry, thus adding signiicantly to the total solution cost. The
transistor circuitry shown here is quite
inexpensive and offers the option to
make certain design tradeoffs as well.
With the propagation delays involved
and desire to keep power fairly low, this
circuit as shown still communicates
at over 500kbps. Lower SPI currents
could be chosen in applications that
don’t demand the high data rate by
simply raising the resistance values
accordingly.
The main feature of the transistorized SPI bus is the wide compliance
range that is afforded by the unconstrained collector-base operating
range of the transistors. In normal
operation the VCB ranges from just
less than the cells connected to the
LTC6802, to some ive volts below that,
depending on the logic level transmitted. This becomes important since
voltage luctuations on the battery,
due to load dynamics or switching
transients, affect the VCB of the transistors even though the V+ and ADC cell
inputs may be iltered. Some vehicle
manufacturers are requiring that a
BMS tolerate 1V steps with 200ns
rise/fall time per cell in the stack, so
this is a 12V waveform edge as seen by
the transistors in a typical application.
With the low collector capacitance and
2mA logic level of the transistor chain,
SPI transmissions remain error free
with even this high level of noise.
Conclusion
EVs and HEVs are here to stay. Inherently safe lithium batteries, which
combine energy density, power density, and cycle life, will continue to
evolve to improve the performance of
these vehicles. Battery management
systems using the LTC6802 extract
the most driving distance and lifetime
from the battery pack while lowering
system cost. L
Linear Technology Magazine • March 2009
DESIGN FEATURES L
DC/DC Converter, Capacitor Charger
Takes Inputs from 4.75V to 400V
by Robert Milliken and Peter Liu
Introduction
High voltage power supplies and capacitor chargers are readily found in
a number of applications, including
professional photolashes, security
control systems, pulsed radar systems,
satellite communication systems, and
explosive detonators. The LT3751
makes it possible for a designer to
meet the demanding requirements
of these applications, including high
reliability, relatively low cost, safe
operation, minimal board space and
high performance.
The LT3751 is a general purpose
lyback controller that can be used as
either a voltage regulator or as a capacitor charger. The LT3751 operates in
boundary-mode, between continuous
conduction mode and discontinuous
conduction mode. Boundary-mode
operation allows for a relatively small
transformer and an overall reduced
PCB footprint. Boundary-mode also
reduces large signal stability issues
that could arise from using voltagemode or PWM techniques. Regulation
is achieved with a new dual, overlapping modulation technique using both
damage. When used as a regulator, the
LT3751’s feedback loop is internally
compensated to ensure stability. The
LT3751 is available in two packages,
either a 20-pin exposed pad QFN or a
20-lead exposed pad TSSOP.
2V/DIV
GND
250ns/DIV
Figure 1. Gate driver waveform
in a typical application
peak primary current modulation and
duty-cycle modulation, drastically reducing audible transformer noise.
The LT3751 features many safety
and reliability functions, including
two sets of undervoltage lockouts
(UVLO), two sets of overvoltage
lockouts (OVLO), no-load operation,
over-temperature lockout (OTLO), internal Zener clamps on all high voltage
pins, and a selectable 5.6V or 10.5V
internal gate driver voltage clamp (no
external components needed). The
LT3751 also adds a start-up/shortcircuit protection circuit to protect
against transformer or external FET
New Gate Driver with Internal
Clamp Requires No External
Components
There are four main concerns when
using a gate driver: output current
drive capability, peak output voltage,
power consumption and propagation
delay. The LT3751 is equipped with a
1.5A push-pull main driver, enough to
drive +80nC gates. An auxiliary 0.5A
PMOS pull-up only driver is also integrated into the LT3751 and is used in
parallel with the main driver for VCC
voltages of 8V and below. This PMOS
driver allows for rail-to-rail operation.
Above 8V, the PMOS driver must be
deactivated by tying its drain to VCC.
Most discrete FETs have a VGS limit
of 20V. Driving the FET higher than
20V could cause a short in the internal gate oxide, causing permanent
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY
VTRANS
10V TO 24V
T1*
1:10
+
OFF ON
C1
10µF
TO µP
RVTRANS
CHARGE
CLAMP
RDCM
R7
18.2k
VCC
R8
40.2k
LT3751
RVOUT
•
C4
100µF
•
VOUT
50V TO 450V
+
C5
0.47µF
VOUT
100V/DIV
GND
HVGATE
LVGATE
CSP
FAULT
M1
VCC
R5
6mΩ
1W
UVLO1
R2, 475k
OVLO1
VCC
D2
DONE
R1, 154k
VTRANS
C2
2.2µF
s5
R6
40.2k
C3
680µF
D1
CSN
UVLO2
ALL RESISTORS ARE 0805,
1% RESISTORS UNLESS
OTHERWISE NOTED
D1,D2: VISHAY MURS260
M1: IRF3710Z
T1: WURTH 750310349
FB
OVLO2
GND RBG
R9
4.7nF
Y RATED
* LIMIT OUTPUT POWER TO
40W FOR 65°C T1 MAX
AMBIENT OPERATION
IIN(AVG)
2A/DIV
0
VIN = 24V
COUT = 100µF
20ms/DIV
Figure 3. Isolated high voltage capacitor
charger charging waveform
Figure 2. Isolated high voltage capacitor charger from 10V to 24V input
Linear Technology Magazine • March 2009
9
L DESIGN FEATURES
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY
VTRANS
10V TO 24V
T1**
1:10
+
R6
40.2k
C3
680µF
OFF ON
C1
10µF
TO µP
RVTRANS
CHARGE
CLAMP
RDCM
R7
18.2k
VCC
R8
40.2k
LT3751
RVOUT
HVGATE
LVGATE
CSP
R1, 154k
UVLO1
R2, 475k
OVLO1
VCC
D2
•
C4
100µF
•
VOUT
400V
ALL RESISTORS ARE 0805,
1% RESISTORS UNLESS
OTHERWISE NOTED
+
C5
0.47µF
C4: CDE 380LX101M500J042
C5: TDK CKG57NX7R2J474M
D1,D2: VISHAY MURS260
M1: IRF3710Z
T1: WURTH 750310349
DONE
FAULT
VTRANS
C2
2.2µF
s5
D1
CSN
M1
* USE TWO SERIES 1206,
1% RESISTORS FOR R10
R10: 249k s2
VCC
R5
6mΩ
1W
** LIMIT OUTPUT POWER TO
40W FOR 65°C T1 MAX
R10*
AMBIENT OPERATION
499k
UVLO2
OVLO2
FB
C6
10nF
GND RBG
R11
1.54k
R9
787Ω
Figure 4. A 10V to 24V input, 400V regulated power supply
damage. To alleviate this issue, the
LT3751 has an internal, selectable
5.6V or 10.5V gate driver clamp. No
external components are needed, not
even a capacitor. Simply tie the CLAMP
pin to ground for 10.5V operation or
tie to VCC for 5.6V operation. Figure
1 shows the gate driver clamping at
10.5V with a VCC voltage of 24V.
Not only does the internal clamp
protect the FET from damage, it also
reduces the amount of energy injected
into the gate. This increases overall
eficiency and reduces power consumption in the gate driver circuit. The
gate driver overshoot is very minimal,
as seen in Figure 1. Placing the external
FET closer to the LT3751 HVGATE pin
reduces overshoot.
High Voltage, Isolated
Capacitor Charger from
10V to 24V Input
The LT3751 can be conigured as
a fully isolated stand-alone capacitor charger using a new differential
discontinuous-conduction-mode
(DCM) comparator—used to sense
the boundary-mode condition—and
a new differential output voltage
(VOUT) comparator. The differential
operation of the DCM comparator and
VOUT comparator allow the LT3751 to
accurately operate from high voltage
input supplies of greater than 400V.
Likewise, the LT3751’s DCM comparator and VOUT comparator can work with
input supplies down to 4.75V. This
accommodates an unmatched range
of power sources.
Figure 2 shows a high voltage capacitor charger driven from an input
supply ranging from 10V to 24V. Only
ive resistors are needed to operate
the LT3751 as a capacitor charger.
The output voltage trip point can be
continuously adjusted from 50V to
450V by adjusting R9 given by:


0.98 • N
R9 = 
 • R8
 VOUT(TRIP) + VDIODE 
The LT3751 stops charging the
output capacitor once the programmed
output voltage trip point (VOUT(TRIP)) is
reached. The charge cycle is repeated
by toggling the CHARGE pin. The
maximum charge/discharge rate in
402
90
EFFICIENCY
VDRAIN
20V/DIV
GND
GND
IPRIMARY
5A/DIV
0
IPRIMARY
5A/DIV
0
10µs/DIV
a. Switching waveform for IOUT = 100mA
401
80
75
VOUT (V)
VDRAIN
20V/DIV
EFFICIENCY (%)
85
LOAD REGULATION
400
70
65
10µs/DIV
b. Switching waveform for IOUT = 10mA
60
0
20
40
60
80
LOAD CURRENT (mA)
399
100
c. Efficiency and load regulation
Figure 5. High voltage regulator performance
10
Linear Technology Magazine • March 2009
DESIGN FEATURES L
the output capacitor is limited by the
temperature rise in the transformer.
Limiting the transformer surface temperature in Figure 2 to 65°C with no
air low requires the average output
power to be ≤40W given by:
VOUT AC RIPPLE
10V/DIV
IIN(AVG)
20mA/DIV
0
PAVG =
1
C
• FREQUENCY •
2 OUT
2VOUT(TRIP) • VRIPPLE – VR2IPPLE
(
2s/DIV
)
Figure 6. The LT3751 protecting the
output during a no-load condition
≤ 40 W
where VOUT(TRIP) is the output trip
voltage, VRIPPLE is the ripple voltage
on the output node, and frequency is
the charge/discharge frequency. Two
techniques are used to increase the
available output power: increase the
airlow across the transformer, or increase the size of the transformer itself.
Figure 3 shows the charging waveform
and average input current for a 100µF
output capacitor charged to 400V in
less than 100ms (R9 = 976Ω).
For output voltages higher than
450V, the transformer in Figure 2 must
be replaced with one having higher
primary inductance and a higher
turns ratio. Consult the LT3751 data
sheet for proper transformer design
procedures.
High Voltage Regulated Power
Supply from 10V to 24V Input
The LT3751 can also be used to convert
a low voltage supply to a much higher
voltage. Placing a resistor divider from
the output node to the FB pin and
ground causes the LT3751 to operate as a voltage regulator. Figure 4
shows a 400V regulated power supply
operating from an input supply range
of 10V to 24V.
The LT3751 uses a regulation control scheme that drastically reduces
audible noise in the transformer and
the input and output ceramic bulk
capacitors. This is achieved by using
an internal 26kHz clock to synchronize
the primary winding switch cycles.
Within the clock period, the LT3751
modulates both the peak primary
current and the number of switching cycles. Figures 5a and 5b show
heavy-load and light-load waveforms,
respectively, while Figure 5c shows
eficiency over most of the operating
range for the application in Figure 4.
The clock forces at least one switch
cycle every period which would overcharge the output capacitor during a
no-load condition. The LT3751 handles no-load conditions and protects
against over-charging the output node.
Figure 6 shows the LT3751 protecting
during a no-load condition.
Resistors can be added to RVOUT and
RBG to add a second layer of protection, or they can be omitted to reduce
component count by tying RVOUT and
RBG to ground. The trip level for the
VOUT comparator is typically set 20%
higher than the nominal regulation
voltage. If the resistor divider were to
fail, the VOUT comparator would disable
switching when the output climbed to
20% above nominal.
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY
T1*** D1
1:3
F1, 1A
C3
100µF
450V
+
R7
88.7k + 7.5k
OFF ON
RVTRANS
CHARGE
RDCM
CLAMP
VCC
C1
10µF
LT3751
RVOUT
DONE
TO µP
FAULT
R1**
1.5M
C4
220µF
550V
•
R8
137k ×3
R11
14.7k +
17.4k
VCC
M1
FQB4N80
R2**, 9M
OVLO1
CSP
R3, 154k
VCC
UVLO2
R4, 475k
CSN
OVLO2
4.7nF
Y RATED
GND RBG
R5
1.11k
FB
C5
0.47µF
630V
ALL RESISTORS ARE 0805,
1% RESISTORS UNLESS
OTHERWISE NOTED
UVLO1
VTRANS
+
R9
66.5k
R10*
208k
R13,20Ω
HVGATE
LVGATE
•
VOUT
500V
R12
68mΩ
1/4W
C4: HITACHI PS22L221MSBPF
C5: TDK CKG57NX7R2J474M
T1: COILCRAFT HA4060-AL
D1,D2: VISHAY US1M
F1: BUSSMANN PCB-1-R
* USE THREE SERIES 1206, 0.1%
RESISTORS FOR R6 & R10
R6: 249k ×2 + 127k
R10: 66.5k ×2 + 75k
** USE TWO SERIES 1206, 1%
RESISTORS FOR R1 & R2
R1: 750k ×2
R2: 4.53M ×2
530
1000
520
850
VOUT,TRIP
700
510
CHARGE TIME
550
500
490
100
CHARGE TIME (ms)
VCC
10V TO 24V
C2
2.2µF
630V
s5
R6*
625k
D2
VOUT,TRIP (V)
VTRANS
100V TO 400V DC
200
300
INPUT VOLTAGE (V)
400
400
Figure 8. Isolated capacitor charger VOUT(TRIP)
and charge time with respect to input voltage
*** OUTPUT POWER LIMITED TO
20W FOR 65°C T1 AMBIENT
OPERATION
Figure 7. A 100V to 400V input, 500V output, isolated capacitor charger
Linear Technology Magazine • March 2009
11
L DESIGN FEATURES
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY
VTRANS
100V TO 400V DC
T1*** D1
1:3
F1, 1A
+
R7
95.3k
C3
100µF
OFF ON
VCC
10V TO 24V
C1
10µF
R6*
615k
C2
2.2µF
s5
RVTRANS
CHARGE
RDCM
CLAMP
VCC
LT3751
D2
•
C4
100µF
•
VOUT
400V
ALL RESISTORS ARE 0805,
1% RESISTORS UNLESS
OTHERWISE NOTED
+
R8*
411k
C5
0.47µF
C4: CDE 380LX101M500J042
C5: TDK CKG57NX7R2J474M
T1: COILCRAFT HA4060-AL
D1,D2: VISHAY US1M
F1: BUSSMANN PCB-1-R
R9
66.5k
RVOUT
* USE THREE SERIES 1206, 1%
RESISTORS FOR R6 & R8
R6: 205k ×3
R8: 137k ×3
DONE
TO µP
FAULT
R13,20Ω
HVGATE
LVGATE
VCC
** USE TWO SERIES 1206, 1%
RESISTORS FOR R1, R2 & R11
R1: 750k ×2
R2: 4.53M ×2
R11: 249k ×2
M1
FQB4N80
R1**, 1.5M
UVLO1
VTRANS
R2**, 9M
OVLO1
R3, 154k
CSP
UVLO2
VCC
R4, 475k
OVLO2
GND RBG
*** OUTPUT POWER LIMITED TO
20W FOR 65°C T1 AMBIENT
OPERATION
R10
68mΩ
¼W
CSN
R11**
499k
FB
R12
1.54k
C6
10nF
Figure 9. A 100V to 400V input, 400V output, capacitor charger and voltage regulator
Note that the FB pin of the LT3751
can also be used for a capacitor
charger. The LT3751 operates as a
capacitor charger until the FB pin
reaches 1.225V, after which the
LT3751 operates as a voltage regulator.
This keeps the capacitor topped-off
until the application needs to use its
energy. The output resistor divider
forms a leakage path from the output
capacitor to ground. When the output
voltage droops, the LT3751 feedback
circuit will keep the capacitor topped-
off with small, low current bursts of
charge as shown in Figure 6.
High Input Supply Voltage,
Isolated Capacitor Charger
As mentioned above, the LT3751 differential DCM and VOUT comparators
allow the part to accurately work from
very high input supply voltages. An
ofline capacitor charger, shown in
Figure 7, can operate with DC input
voltages from 100V to 400V. The transformer provides galvanic isolation from
90
398
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
80
70
60
VIN = 100V
VIN = 250V
VIN = 400V
50
40
0
25
50
397
396
IOUT = 10mA
IOUT = 25mA
IOUT = 50mA
75
395
100
OUTPUT CURRENT (mA)
200
300
INPUT VOLTAGE (V)
a. Overall efficiency
b. Line regulation
400
the input supply to output node—no
additional magnetics required.
Input voltages greater than 80V
require the use of resistor dividers
on the DCM and VOUT comparators
(charger mode only). The accuracy of
the VOUT trip threshold is heightened
by increasing current IQ through R10
and R11; however, the ratio of R6/R7
should closely match R10/R11 with
tolerances approaching 0.1%. A trick
is to use resistor arrays to yield the
desired ratio. Achieving 0.1% ratio accuracy is not dificult and can reduce
the overall cost compared to using
individual 0.1% surface mount resistors. Note that the absolute value of
the individual resistors is not critical,
only the ratio of R6/R7 and R10/R11.
The DCM comparator is less critical
and can tolerate resistance variations
greater than 1%.
The 100V to 400VDC input capacitor charger has an overall VOUT(TRIP)
accuracy of better than 6% over the
entire operating range using 0.1% resistor dividers. Figure 8 shows a typical
performance for VOUT(TRIP) and charge
time for the circuit in Figure 7.
Figure 10. High voltage input and output regulator performance
12
Linear Technology Magazine • March 2009
DESIGN FEATURES L
ISOLATION
BOUNDARY
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY
D2
R2, 10Ω
T1
•
Npb
VTRANS
100V TO 200V DC
F1, 2A
+
R1
49.9k
1/2W
M1
C2
1µF
D1
1
VCC
Ns
Np
•
R10, 4.3M
2
RVOUT
FAULT
HVGATE
LVGATE
CSP
R15
221k
R16, 1k
OVLO1
Nsb
•
M2
C5
0.01µF
630V
CSN
C9
3.3µF
R13
5.11Ω
1
R12, 442k
GND RBG
R8
2.49k
1
2
1
R7
475Ω
C8
22nF
LT4430
R17
3.16k
OC
FB
D7
U1
FB
OVLO2
VIN COMP
C10
0.47µF
2
UVLO2
2
U2
D6
VCC
R6
40mΩ
1/4W
R14
249k
VOUT
282V
225mA
2
2
D4
R11, 84.5k
4.7nF
Y RATED
C7
400µF
330V
RDCM
DONE
UVLO1
VCC
+
C6
0.1µF
630V
R4, 105k
LT3751
R9, 2.7M
VTRANS
D5
•
D3
R5, 210k
ALL RESISTORS ARE 0805,1% RESISTORS
UNLESS OTHERWISE NOTED
C7: 330FK400M22X38
D1: 12V ZENER
D2: MURS140
D3: P6kE200A
D4, D5: STTH112A
D6: BAT54
D7: BAS516
M1: IRF830
M2: STB11NM60FD
T1: TDK SRW24LQ-UxxH015
(Np:Ns:Npb:Nsb=1:2:0.08:0.08)
U1: PS2801-1
U2: LT4430
RVTRANS
TO µP
1
C4
1µF
250V
1 s2
R3
210k
CHARGE
CLAMP
OFF ON
1
C1
100pF
1
C3
22µF
350V
1 s2
OPTO GND
VCC
R18
274Ω
2
1
2
1
Figure 11. Fully isolated, high output voltage regulator
High Input Supply Voltage,
Non-Isolated Capacitor
Charger/Regulator
The FB pin of the LT3751 can also
be conigured for charging a capacitor from a high input supply voltage.
Simply tie a resistor divider from the
output node to the FB pin. The resistor dividers on the RVTRANS and RDCM
pins can tolerate 5% resistors, and all
the RV(OUT) and RBG pin resistors are
removed. This lowers the number and
the tolerance of required components,
reducing board real estate and overall
design costs. With the output voltage
resistor divider, the circuit in Figure
9 is also a fully functional, high-eficiency voltage regulator with load
and line regulation better than 1%.
Eficiency and line regulation for the
circuit in Figure 9 are shown in Figure
10a and Figure 10b, respectively.
Alternatively, a resistor can be tied
from VOUT to the OVLO1 pin or OVLO2
pin. This mimics the VOUT comparator, stopping charging once the target
voltage is reached. The FB pin is tied
to ground. The CHARGE pin must be
toggled to initiate another charge sequence, thus the LT3751 operates as
a capacitor charger only. Resistor R12
is omitted from Figure 9 and resistor
R11 is tied from VOUT directly to OVLO1
or OVLO2. R11 is calculated using the
following equation:
VDRAIN
100V/DIV
VDRAIN
100V/DIV
GND
IPRIMARY
2A/DIV
0
GND
IPRIMARY
2A/DIV
0
R11 =
VOUT(TRIP) − 1.225
50µA
Note that OVLO1 or OVLO2 will
cause the FAULT pin to indicate a
fault when the target outpaut voltage,
VOUT(TRIP) , is reached.
High Voltage Input/Output
Regulator with Isolation
Using a resistor divider from the output
node to the FB pin allows regulation
but does not provide galvanic isolation.
Two auxiliary windings are added to
the transformer in circuit shown in
Figure 11 to drive the FB pin, the
continued on page 42
20µs/DIV
20µs/DIV
a. IOUT = 225mA
b. IOUT = 7.1mA
Figure 12. Switching waveforms
Linear Technology Magazine • March 2009
13
L DESIGN FEATURES
How to Choose a Voltage Reference
by Brendan Whelan
Why Voltage References?
5V
It is an analog world. All electronic
devices must in some way interact with
the “real” world, whether they are in
an automobile, microwave oven or cell
phone. To do that, electronics must be
able to map real world measurements
(speed, pressure, length, temperature)
to a measurable quantity in the electronics world (voltage). Of course, to
measure voltage, you need a standard
to measure against. That standard is
a voltage reference. The question for
any system designer is not whether he
needs a voltage reference, but rather,
which one?
A voltage reference is simply that—a
circuit or circuit element that provides
a known potential for as long as the circuit requires it. This may be minutes,
hours or years. If a product requires
information about the world, such
VIN
18k
LTC1286
1
2
LT1634-4.096
0.1µF
3
4
VREF
VCC
+IN
CLK
–IN
DOUT
GND CS/SHDN
8
5V
7
6
5
0.1µF
µC/µP
SERIAL
INTERFACE
Figure 1. Typical use of a voltage reference for an ADC
Reference Specifications
as battery voltage or current, power
consumption, signal size or characteristics, or fault identiication, then the
signal in question must be compared
to a standard. Each comparator, ADC,
DAC, or detection circuit must have a
voltage reference in order to do its job
(Figure 1). By comparing the signal of
interest to a known value, any signal
may be quantiied accurately.
Voltage references come in many forms
and offer different features, but in
the end, accuracy and stability are
a voltage reference’s most important
features, as the main purpose of the
reference is to provide a known output
voltage. Variation from this known
value is an error. Voltage reference
specifications usually predict the
uncertainty of the reference under
Table 1. Specifications for high performance voltage references
Temperature Initial
Coefficient Accuracy
LT1031
5ppm/°C
0.05%
VOUT
Voltage
Noise*
Long-Term
Drift
Package
Buried Zener
10V
0.6ppm
15ppm/kHr
H
2.5ppm
IS
Architecture
1.2mA
LT1019
5ppm/°C
0.05%
650µA
Bandgap
2.5V, 4.5V,
5V, 10V
LT1027
5ppm/°C
0.05%
2.2mA
Buried Zener
5V
0.6ppm
LT1021
5ppm/°C
0.05%
800µA
Buried Zener
5V, 7V, 10V
0.6ppm
15ppm/kHr SO-8, PDIP, H
LTC6652
5ppm/°C
0.05%
350µA
Bandgap
1.25V, 2.048V,
2.5V, 3V, 3.3V,
4.096V, 5V
2.1ppm
60ppm/√kHr
MSOP
LT1236
5ppm/°C
0.05%
800µA
Buried Zener
5V, 10V
0.6ppm
20ppm/kHr
SO-8, PDIP
LT1461
3ppm/°C
0.04%
35µA
Bandgap
2.5V, 3V, 3.3V,
4.096V, 5V
8ppm
60ppm/√kHr
SO-8
LT1009
15ppm/°C
0.2%
1.2mA
Bandgap
2.5V
20ppm/kHr
MSOP-8,
SO-8, Z
LT1389
20ppm/°C
0.05%
700nA
Bandgap
1.25V, 2.5V,
4.096V, 5V
20ppm
SO-8
LT1634
10ppm/°C
0.05%
7µA
Bandgap
1.25V, 2.5V,
4.096V, 5V
6ppm
SO-8,
MSOP-8, Z
LT1029
20ppm/°C
0.20%
700µA
Bandgap
5V
LM399
1ppm/°C
2%
15mA
Buried Zener
7V
LTZ1000
0.05ppm/°C
4%
Buried Zener
7.2V
SO-8, PDIP
20ppm/
month
SO-8, PDIP
20ppm/kHr
Z
1ppm
8ppm/√kHr
H
0.17ppm
2µV/√kHr
H
*0.1Hz–10Hz, Peak-to-Peak
14
Linear Technology Magazine • March 2009
DESIGN FEATURES L
Initial Accuracy
The variance of output voltage as
measured at a given temperature,
usually 25°C. While the initial output
voltage may vary from unit to unit, if
it is constant for a given unit, then it
can be easily calibrated.
Temperature Drift
This speciication is the most widely
used to evaluate voltage reference
performance, as it shows the change
in output voltage over temperature.
Temperature drift is caused by imperfections and nonlinearities in the
circuit elements, and is often nonlinear
as a result.
For many parts, the temperature
drift, TC, speciied in ppm/°C, is the
dominant error source. For parts with
consistent drift, calibration is possible.
A common misconception regarding
temperature drift is that it is linear.
This leads to assumptions such as
“the part will drift a lesser amount
over a smaller temperature range.”
Often the opposite is true. TC is generally speciied with a “box method” in
order to give an understanding of the
likely error over the entire operating
temperature range. It is a calculated
value based only on minimum and
maximum values of voltage, and does
not take into account the temperatures
at which these extrema occur.
For voltage references that are very
linear over the speciied temperature
range, or for those that are not carefully tuned, the worst-case error can
be assumed to be proportional to the
temperature range. This is because
the maximum and minimum output
voltages are very likely to be found at
the maximum and minimum operating
temperatures. However, for very carefully tuned references, often identiied
by their very low temperature drift,
the nonlinear nature of the reference
may dominate.
For example, a reference speciied as 100ppm/°C tends to appear
quite linear over any temperature
range, as the drift due to component
mismatches completely obscures the
Linear Technology Magazine • March 2009
The best use of the temperature drift
speciication is to calculate maximum
total error over the speciied temperature range. It is generally inadvisable
to calculate errors over unspeciied
temperature ranges unless the temperature drift characteristics are well
understood.
1.003
OUTPUT VOLTAGE (NORMALIZED) (V)
certain conditions using the following
deinitions.
10ppm/oC
FULL TEMP RANGE “BOX”
1.002
1.001
LT1019
CURVE
1.000
0.999
5ppm/oC
0oC TO 70oC “BOX”
0.998
UNCOMPENSATED
“STANDARD” BANDGAP
DRIFT CURVE
0.997
–50 –25
50
25
75
0
TEMPERATURE (˚C)
100
125
Figure 2. Voltage reference
temperature characteristics
inherent nonlinearity. In contrast, the
temperature drift of a reference speciied as 5ppm/°C will be dominated by
the nonlinearities.
Voltage references come
in many forms and offer
different features, but in the
end, accuracy and stability
are a voltage reference’s
most important features,
as the main purpose of
the reference is to provide
a known output voltage.
Variation from this known
value is an error. Voltage
reference specifications
usually predict the
uncertainty of the reference
under certain conditions.
This can be easily seen in the output
voltage vs temperature characteristic
of Figure 2. Note that there are two
possible temperature characteristics
represented. An uncompensated
bandgap appears as a parabola, with
minima at the temperature extrema
and maximum in the middle. A
temperature compensated bandgap,
such as the LT1019, shown here,
appears as an “S” shaped curve, with
greatest slope near the center of the
temperature range. In the latter case,
nonlinearity is exacerbated so that the
aggregate uncertainty over temperature is reduced.
Long Term Stability
This is a measure of the tendency of a
reference voltage to change over time,
independent of other variables. Initial
shifts are largely caused by changes
in mechanical stress, usually from the
difference in expansion rates of the
lead frame, die and mold compound.
This stress effect tends to have a large
initial shift that reduces quickly with
time. Initial drift also includes changes
in electrical characteristics of the
circuit elements, including settling
of device characteristics at the atomic
level. Longer-term shifts are caused
by electrical changes in the circuit
elements, often referred to as “aging.”
This drift tends to occur at a reduced
rate as compared to initial drift, and to
further reduce over time. It is therefore
often speciied as drift/√khr. Voltage
references tend to age more quickly
at higher temperatures.
Thermal Hysteresis
This often-overlooked speciication
can also be a dominant source of error. It is mechanical in nature, and is
the result of changing die stress due
to thermal cycling. Hysteresis can be
observed as a change in output voltage
at a given temperature after a large
temperature cycle. It is independent
of temperature coeficient and time
drift, and reduces the effectiveness
of initial voltage calibration.
Most references tend to vary around
a nominal output voltage during
subsequent temperature cycles, so
thermal hysteresis is usually limited
to a predictable maximum value. Each
manufacturer has their own method
for specifying this parameter, so typical
values can be misleading. Distribution
data, as provided in data sheets such
as the LT1790 and LTC6652, is far
more useful when estimating output
voltage error.
15
L DESIGN FEATURES
5V
4.7M
VOUT
1.25V
2.6V b VIN b 18V
LT1389-1.25
Figure 3. Shunt voltage reference
Other Specifications
Additional speciications that may be
important, depending on application
requirements include:
q Voltage Noise
q Line Regulation/PSRR
q Load Regulation
q Dropout Voltage
q Supply Range
q Supply Current
Reference Types
The two main types of voltage reference
are shunt and series. See Table 2 for
a list of Linear Technology series and
shunt voltage references.
Shunt References
The shunt reference is a 2-terminal
type, usually designed to work over a
speciied range of currents. Though
most shunts are of the bandgap type
and come in a variety of voltages, they
can be thought of and are as simple
to use as a Zener diode.
The most common circuit ties one
terminal of the reference to ground and
the other terminal to a resistor. The
remaining terminal of the resistor is
VOUT = 2.5V
1MF
Figure 4. Series voltage reference
then tied to a supply. This becomes, in
essence, a three terminal circuit. The
shared reference/resistor terminal
is the output. The resistor must be
chosen such that the minimum and
maximum currents through the reference are within the speciied range
over the entire supply range and load
current range. These references are
quite easy to design with, provided
the supply voltage and load current do
not vary much. If either, or both, may
change substantially, then the resistor must be chosen to accommodate
this variance, often forcing the circuit
to dissipate signiicantly more power
than required for the nominal case. It
can be considered to function like a
class A ampliier, in that sense.
Advantages of shunt references
include simple design, small packages
and good stability over wide current
and load conditions. In addition, they
are easily designed as negative voltage
references and can be used with very
high supply voltages, as the external
resistor holds off most of the potential,
or very low supplies, as the output can
be as little as a few millivolts below
+
7.5k
LT1790-2.5
0.1MF
the supply. Linear Technology offers
shunt products including the LT1004,
LT1009, LT1389, LT1634, LM399 and
LTZ1000. A typical shunt circuit can
be seen in Figure 3.
Series References
Series references are three (or more)
terminal devices. They are more like
low dropout (LDO) regulators, so they
have many of the same advantages.
Most notably, they consume a relatively ixed amount of supply current over
a wide range of supply voltages, and
they only conduct load current when
the load demands it. This makes them
ideal for circuits with large changes in
supply voltage or load current. They
are especially useful in circuits with
very large load currents as there is no
series resistor between the reference
and supply.
Series products available from Linear Technology include the LT1460,
LT1790, LT1461, LT1021, LT1236,
LT1027, LTC6652, LT6660, and many
others. Products such as the LT1021
and LT1019 may be operated either as
a shunt or a series voltage reference.
A series reference circuit is illustrated
in Figure 4.
Q13
200k
Q12
Q4
240mV, +0.8mV/°C
Q3
20pF
20pF
–
+
Q11
Q1
50k
600k
–
+
Q10
300k
Q5
2.8k
60mV, +0.2mV/°C
Q9 Q14
–
+
VREF
1.235V, 0mV/°C
14.9k
Q1
360mV, +1.2mV/°C
Q2
135k
82.4k
Q8
Q6
575mV, –2.2mV/°C
500k
5007
2.5k
60k
–
Figure 5. A bandgap circuit is designed for a theoretically zero temperature coefficient.
16
2k
Figure 6. A 200mV reference circuit
Linear Technology Magazine • March 2009
DESIGN FEATURES L
Reference Circuits
Linear Technology Magazine • March 2009
Type
Series
Zener-Based References
The buried Zener type reference is a
relatively simple design. A Zener (or
avalanche) diode has a predictable
reverse voltage that is fairly constant
over temperature and very constant
over time. These diodes are often very
low noise and very stable over time if
held within a small temperature range,
making them useful in applications
where changes in the reference voltage
must be as small as possible.
This stability can be attributed to
the relatively small number of components and die area as compared
to other types of reference circuits,
as well as the careful construction
of the Zener element. However, relatively high variances in initial voltage
and temperature drift are common.
Additional circuitry may be added to
compensate these imperfections, or
to provide a range of output voltages.
Both shunt and series references use
Zener diodes.
Devices like the LT1021, LT1236
and LT1027 use internal current
sources and ampliiers to regulate the
Zener voltage and current to increase
stability, as well as to provide various
output voltages such as 5V, 7V and
10V. This additional circuitry makes
the Zener diode more compatible with
a wide variety of application circuits,
but requires some additional supply
headroom and may cause additional
error.
Alternatively, the LM399 and
LTZ1000 use internal heating elements and additional transistors to
stabilize the temperature drift of the
Zener diode, giving the best combination of temperature and time stability.
In addition, these Zener-based products have extraordinarily low noise,
giving the best possible performance.
The LTZ1000 exhibits 0.05ppm/°C
temperature drift, 2µV/√kHr long
term stability and 1.2µVP-P noise. To
give some perspective, in a laboratory
instrument, the total uncertainty in
the LTZ1000’s reference voltage due
Table 2. Voltage references available from Linear Technology
Shunt
There are many ways to design a voltage reference IC. Each has speciic
advantages and disadvantages.
Part
Description
LT1019
Precision Bandgap
LT1021
Precision Low Noise Buried Zener
LT1027
Precision 5V Buried Zener
LT1031
Precision Low Noise/Low Drift 10V Zener
LT1236
Precision Low Noise Buried Zener
LT1258
Micropower LDO Bandgap
LT1460
Micropower Precision Bandgap
LT1461
Micropower Ultra-Precision Bandgap
LT1790
Micropower Low Dropout Bandgap
LT1798
Micropower LDO Bandgap
LT6650
Micropower 400mV/Adjustable Bandgap
LTC6652
Precision Low Noise LDO Bandgap
LM129
Precision 6.9V Buried Zener
LM185
Micropower 1.2V/2.5V Zener
LM399
Precision 7V Heated Zener
LT1004
Micropower 1.2V/2.5V Bandgap
LT1009
Precision 2.5V Bandgap
LT1029
5V Bandgap
LT1034
Micropower Dual (1.2V Bandgap/7V Zener)
LT1389
Nanopower Precision Bandgap
LT1634
Micropower Precision Bandgap
LTZ1000
Ultra-Precision Heated Zener
to noise and temperature would be
only about 1.7ppm plus a fraction of
1ppm per month due to aging.
Bandgap References
While Zener diodes can be used to
make very high performance references, they lack lexibility. Speciically,
they require supply voltages above 7V
and they offer relatively few output
voltages. In contrast, bandgap references can produce a wide variety
of output voltages with little supply
headroom—often less than 100mV.
Bandgap references can be designed
to provide very precise initial output
voltages and low temperature drift,
eliminating the need for time-consuming in-application calibration.
Bandgap operation is based on a
basic characteristic of bipolar junction
transistors. Figure 5 shows a simpliied version of the LT1004 circuit, a
basic bandgap. It can be shown that
a mismatched pair of bipolar junction
transistors has a difference in VBE
that is proportional to temperature.
This difference can be used to create
a current that rises linearly with temperature. When this current is driven
through a resistor and a transistor, the
change over temperature of the baseemitter voltage of the transistor cancels
the change in the voltage across the
resistor if it is sized properly. While
this cancellation is not completely
linear, it can be compensated with
additional circuitry to yield very low
temperature drift.
The math behind the basic bandgap
voltage reference is interesting in that
it combines known temperature coef17
L DESIGN FEATURES
icients with unique resistor ratios to
produce a voltage reference with theoretically zero temperature drift. Figure
5 shows two transistors scaled so that
the emitter area of Q10 is 10-times that
of Q11, while Q12 and Q13 hold their
collector currents equal. This creates
a known voltage between the bases of
the two transistors of:
ΔVBE =
VS
LT6700-1
LT6700HV-1
–
–INB
OUTB
+
–
COMP A
OUTA
+
+INA
18
VS
400mV
REFERENCE
 AREA Q10 
kT
• ln 
q
 AREA Q11 
where k is the Boltzmann constant in
J/kelvin (1.38 × 10-23), T is temperature in kelvin (273 + T(°C)) and q is
the charge of an electron in coulombs
(1.6x10-19). At 25°C, kT/q has a value
of 25.7mV with a positive temperature
coeficient of 86µV/°C. ΔVBE is this
voltage times ln(10), or 2.3, for a 25°C
voltage of approximately 60mV with a
tempco of 0.2mV/°C.
Applying this voltage to the 50k
resistor tied between the bases creates
a current that is proportional to temperature. This current biases a diode,
Q14 with a 25°C voltage of 575mV with
a –2.2mV/°C temperature coeficient.
Resistors are used to create voltage
drops with positive tempcos, which
are added to the Q14 diode voltage,
thus producing a reference voltage
potential of approximately 1.235V
with theoretically 0mV/°C temperature coeficient. These voltage drops
are shown in Figure 5. The balance of
the circuit provides bias currents and
output drive.
Linear Technology produces a
wide variety of bandgap references,
including the LT1460, a small and
inexpensive precision series reference, the LT1389, an ultralow power
shunt reference, and the LT1461 and
LTC6652, which are very high precision, low drift references. Available
output voltages include 1.2V, 1.25V,
2.048V, 2.5V, 3.0V, 3.3V, 4.096V,
4.5V, 5V and 10V. These reference
voltages can be provided over a wide
range of supplies and load conditions
with minimal voltage and current overhead. Products may be very precise,
as with the LT1461, LT1019, LTC6652
and LT1790; very small, as with the
LT1790 and LT1460 (SOT23), or
COMP B
GND
Figure 7. The LT6700 allows comparisons
with thresholds as low as 400mV.
LT6660 in a 2mm × 2mm DFN package; or very low power, such as the
LT1389, which requires only 800nA.
While Zener references often have
better performance in terms of noise
and long term stability, new bandgap
references such as the LTC6652, with
2ppm peak-to-peak noise (0.1Hz to
10Hz) are narrowing the gap.
Fractional Bandgap References
These are references based on the
temperature characteristics of bipolar
transistors, but with output voltages
that may be as low as a few millivolts.
They are useful for very low voltage
circuits, especially in comparator applications where the threshold must
1M
be less than a conventional bandgap
voltage (approximately 1.2V).
Figure 6 shows the core circuit from
the LM10, which combines elements
that are proportional and inversely
proportional to temperature in a
similar fashion to the normal bandgap
reference to obtain a constant 200mV
reference. A fractional bandgap usually uses a ΔVBE to generate a current
that is proportional to temperature,
and a VBE to generate a current that
is inversely proportional. These are
combined in the proper ratio in a
resistor element to generate a temperature-invariant voltage. The size
of the resistor may be varied to alter
the reference voltage without affecting
the temperature characteristic. This
differs from a traditional bandgap
circuit in that the fractional bandgap
circuit combines currents, while the
traditional circuits tend to combine
voltages, usually a base-emitter voltage and an I•R with opposite TC.
Fractional bandgaps like the LM10
circuit are based in part on a subtraction as well. The LT6650 has a 400mV
reference of this type, combined with
an ampliier. This allows the reference voltage to be altered by changing
the gain of the ampliier, and gives a
buffered output. Any output voltage
from 0.4V to a few millivolts below
the supply voltage can be generated
with this simple circuit. In a more
integrated solution, the LT6700
(Figure 7) and LT6703 combine a
VBATT
1.4V (MIN)
3V (NOM)
LT6700-3
+
1M
0.1µF
ALKALINE
AA CELLS
+
+
1M
COMP B
VBATT > 1.6V
63.4k
–
VR = 400mV
REFERENCE
–
VS
COMP A
VBATT > 2V
+
261k
MONITOR CONSUMES ~10µA
HYSTERESIS IS APPROXIMATELY
2% OF TRIP VOLTAGE
Figure 8. Higher thresholds are set by dividing the input voltage.
Linear Technology Magazine • March 2009
DESIGN FEATURES L
400mV reference with comparators,
and can be used as voltage monitors
or window comparators. The 400mV
reference allows monitoring of small
input signals, which decreases the
complexity of monitor circuits and
enables monitoring of circuit elements
working on very low supplies as well.
For larger thresholds, a simple resistor
divider may be added (Figure 8). Each
of these products is available in a small
footprint package (SOT23), consumes
low power (less than 10µA) and works
on a wide supply range (1.4V to 18V).
In addition, the LT6700 is available in
a 2mm × 3mm DFN package and the
LT6703 is available in a 2mm × 2mm
DFN package.
Choosing a Reference
So, now, with all those options, how do
you choose the right reference for your
application? Here are a few hints that
can narrow the range of options:
q Is the supply voltage very
high? Choose a shunt.
q Does the supply voltage or
load current vary widely?
Choose a series.
q Require high power efficiency?
Choose a series.
q Figure your real-world
temperature range. Linear
Technology provides guaranteed
speciications and operation
over various temperature ranges
including 0°C to 70°C, –40°C to
85°C and –40°C to 125°C.
q Be realistic about required
accuracy. It is important to
understand the precision required
by the application. This will help
identify critical speciications.
With the requirement in mind,
multiply temperature drift by the
speciied temperature range. Add
initial accuracy error, thermal
hysteresis, and long term drift
over the intended product life.
Remove any terms that will be
factory calibrated or periodically
recalibrated. This gives an
idea of total accuracy. For the
most demanding applications,
noise, line regulation and load
regulation errors may also
be added. As an example, a
Linear Technology Magazine • March 2009
reference with 0.1% (1000ppm)
initial accuracy error, 25ppm/°C
temperature drift over –40°C
to 85°C, 200ppm thermal
hysteresis, 2ppm peak-to-peak
noise and 50ppm/√kHr time drift
would have a total uncertainty
of over 4300ppm at the time the
circuit is built. This uncertainty
increases by 50ppm in the
irst 1000 hours the circuit is
powered. The initial accuracy
may be calibrated, reducing the
error to 3300ppm + 50ppm •
√(t/1000hours).
Linear Technology offers
a wide variety of voltage
reference products. These
include both series and
shunt references—using
Zeners, bandgaps and
other schemes. References
are available in multiple
performance and
temperature grades, as
well as in nearly every
conceivable package type.
q What is the real supply range?
What is the maximum expected
supply voltage? Will there be
fault conditions such as battery
load dump or hot-swap inductive
supply spikes that the reference
IC must withstand? This may
signiicantly reduce the number
of viable choices.
q How much power can the
reference consume? References
tend to fall into a few categories:
more than 1mA, ~500µA,
<300µA, <50µA, <10µA, <1µA.
q How much load current?
Will the load draw substantial
current or produce current that
the reference must sink? Many
references can provide only small
currents to the load and few can
absorb substantial current. The
load regulation speciication is a
good guide.
q How much room do you have?
References come in a wide
variety of packages, including
metal cans, plastic packages
(DIP, SOIC, SOT) and very small
packages, including the LT6660
in a 2mm × 2mm DFN. There is a
widely held view that references
in larger package sizes have less
error due to mechanical stress
than smaller packages. While
it is true that some references
may give better performance in
larger packages, there is evidence
that suggests performance
difference has little to do directly
with the package size. It is more
likely that because smaller dice
are used for products that are
offered in smaller packages, some
performance tradeoffs must be
made to it the circuit on the die.
Usually, the package’s mounting
method makes a more signiicant
performance difference than
the actual package—careful
attention to mounting methods
and locations can maximize
performance. Also, devices with
smaller footprints can show
reduced stress when a PCB bends
compared to devices with larger
footprints. This is discussed in
detail in application note AN82,
“Understanding and Applying
Voltage References,” available
from Linear Technology.
Conclusion
Linear Technology offers a wide variety of voltage reference products.
These include both series and shunt
references designed with Zeners,
bandgaps and other types. References
are available in multiple performance
and temperature grades and nearly
every conceivable package type. Products range from the highest precision
available to small and inexpensive
alternatives. With a vast arsenal of
voltage reference products, Linear
Technology’s voltage references meet
the needs of almost any application.
See also Linear Technology’s application note AN82 “Understanding
and Applying Voltage References,”
available at www.linear.com. L
19
L DESIGN FEATURES
1.2A Monolithic Buck Regulator
Shrinks Supply Size and Cost with
Programmable Output Current Limit
by Tom Sheehan
Introduction
Power supplies are often overqualiied
for their job. This is because power
ICs often specify a current limit that
is more than twice the rated output
current of the device. The power supply
components are sized to handle the
maximum current that the IC can deliver, even if loads are unlikely to draw
that current during normal operation.
The components are bigger and more
expensive than they need to be.
There is, however, an alternative:
set an accurate maximum output current on the supply once the real world
load is known. Accurately setting the
maximum output current reduces the
required current rating of the regulator’s power path components, thus
replacing big, expensive components
with smaller, less expensive ones.
A limit on the regulator’s maximum
output limits the maximum power
dissipation of both the supply and
the load, thus reducing the potential
for localized heating. Monitoring and
controlling the output current also
makes for a robust solution, which is
able to withstand harsh overload and
short circuit conditions.
The LT3653 and LT3663 are monolithic step-down switching regulators
that have an accurate output current
limit programmable from 400mA to
1.2A. The LT3663 is a general purpose high voltage step-down regulator
while the LT3653 is designed for use
with Linear Technology Bat-Track™
enabled battery chargers and power
management ICs (PMICs). The maximum input voltages of 30V (LT3653)
or 36V (LT3663) with 60V transient
ride through capability are well suited
C3
0.1µF
10V
BOOST
HIGH VOLTAGE INPUT
7.5V TO 30V
TRANSIENT
TO 60V
VIN
C1
4.7µF
50V
R1
27.4k
USB
WALL
ADAPTER
C4
10µF
6.3V
R4
6.04k
TO µC
TO µC
C5
0.1µF
ILIM
VC
VBUS
SW
LT3653
GND
VC
OVGATE
D1
ISENSE
HVOK VOUT
WALL
LTC4098
R2
3.01k
SYSTEM
LOAD
C2
22µF
6.3V
L2
3.3µH
SW
VOUT
IDGATE
OVSENS
D0–D2
CHRG
NTC
CLPROG
L1
4.7µH
M1
(OPTIONAL)
BAT
PROG
BATSENS
GND
SINGLE-CELL
Li-Ion
R3
1k
L1 = TDK, VLCF5020T-4R7NIR7-1
L2 = COILCRAFT, LPS4018-332MLC
M1 = VISHAY, Si 2333DS
D1 = DIODES INC., DFLS240
SEE THE LTC4098 DATASHEET FOR MORE INFORMATION
ON CONFIGURING THE NTC BATTERY TEMPERATURE
QUALIFICATION OR REDUCED IDEAL DIODE IMPEDANCE.
Figure 1. Charging a single cell Li-ion battery from either a USB input or high voltage input. This
solution offers a seamless, highly efficient, low part count approach to dual input charging and
PowerPath™ control of a Li-ion battery-powered application. If additional integration is required
for more system supplies, the LT3653 can be used in a similar fashion with the LTC3576 PMIC.
20
to automotive, industrial, distributed
supply, and wall transformer applications.
Programmable
Output Current Limit
Monolithic switching regulators typically limit the peak switch current to
protect the internal switch from being
damaged during an overload or short
circuit event. The maximum switch
peak current limit is typically more
than two times the maximum output
current rating of the part. While the
peak switch current limit prevents
overstressing the IC, it does not keep
the entire application from overheating during an overload condition. For
example, a regulator with an output
current rating of 1A is typically capable
of providing over 2A at the output.
During an output overload condition,
the power dissipation of the regulator could more than double, making
thermal management more dificult.
The LT3653 and LT3663 reduce localized hot spots by controlling the total
power dissipation of the application
with a programmable, accurate current limit.
Conservative design principles call
for power path components that are
rated for worst-case currents. In the
above example, where a 1A part is
capable of delivering 2A, the power
path components must be sized for
greater than 2A, because during an
output short circuit or overload the
inductor and diode can conduct up
to 2A. In contrast, the PowerPath
components in LT3653 and LT3663
applications are sized based on the
programmed maximum output current limit. Therefore, an application
with a 750mA output current limit
requires only 750mA rated components. This allows for smaller, lower
Linear Technology Magazine • March 2009
DESIGN FEATURES L
The LT3653 Plays Well with
Bat-Track Battery Chargers
The LT3653 is a 1.5MHz constant
frequency, current mode control,
step-down regulator designed for use
with Linear Technology’s Bat-Trackenabled battery charger PowerPath
power managers. The LT3653 steps
down a high voltage input to power the
system load and charge a single-cell
Li-ion battery charger.
Minimizing the voltage across
a linear battery charger increases
eficiency. To accomplish this, a BatTrack battery charger controls the
LT3653’s VC Pin, overriding the error
ampliier. In this way, the output voltage of the LT3653 is regulated by the
battery charger to a potential slightly
above the battery voltage, typically
300mV.
VIN
VIN
2.2µF
BOOST
0.1µF
LT3663
SW
ON OFF
DIODES,
INC.
DFLS240
RUN
6.8µH
ISENSE
ILIM
VOUT
28.7k
VOUT
59k
22µF
FB
GND
11k
Figure 3. A LT3663 application producing
5V at 1.2A from an input of 7.5V to 36V. The
input is capable of handling 60V transients.
Linear Technology Magazine • March 2009
6
RILIM = 28.7k
5
OUTPUT VOLTAGE (V)
cost devices and a smaller overall
application footprint.
In early product development,
system designers usually don’t know
how much current their load will draw.
Once they choose a power supply, they
are committed. However, with the programmable current limit of the LT3653
and LT3663, once the load has been
fully characterized, they can change
the output current limit by changing
an inexpensive 1% resistor.
The output current limit is implemented by monitoring and controlling
the average inductor current. When an
overcurrent event occurs, the regulator
disables the power switch. This robust
solution withstands short circuit and
overload conditions throughout the
entire input voltage range.
4
3
2
1
0
0
0.2
0.4 0.6 0.8
1
OUTPUT CURRENT (A)
1.2
1.4
Figure 2. The LT3663 output
current limit at 1.2A
Input overvoltage protection allows
the LT3653 to handle 60V input transients. The HVOK pin indicates that
the internal bias supplies are present
and no faults have occurred (i.e., overtemperature and input overvoltage
and undervoltage). The LT3653 includes internal compensation, and
an internal boost diode to minimize
the number of external components.
The LT3653 is available in an 8-lead
2mm × 3mm DFN package with an
exposed pad.
Charging a Single Cell Li-Ion
Battery from Either a USB
or High Voltage Input
Figure 1 shows a LT3653 and LTC4098
application charging a single cell Liion battery from either a USB input
or high voltage input. This solution
offers a seamless, highly eficient, low
part count approach to dual input
charging and power path control of a
Li-ion battery-powered application. If
additional integration is required for
more system supplies, the LT3653 can
be used in a similar fashion with the
LTC3576 PMIC.
When a high voltage input is applied, the LT3653 HVOK pin signals
the LTC4098 that it is capable of
delivering power. The LTC4098 takes
control of the LT3653’s VC pin and
regulates the output voltage to just
above the battery voltage. This BatTrack function optimizes the battery
charger eficiency.
When present, the high voltage
input supplies the battery charge current and the system load current. If
the total current increases beyond the
LT3653 programmed current limit, the
regulator’s output voltage decreases to
reduce charge current as the battery
charger enters dropout. If the system
load continues to increase, the battery
charge current irst decreases to zero
and then reverses direction to deliver
power to the system load, supplementing the LT3653. The transitions
between these modes of operation are
seamless to the system load. The output current from the LT3653 regulator
never exceeds the programmed output
current limit.
The LT3663 Directly
Accepts 36V Inputs
The LT3663 is a 1.5MHz constant
frequency, current mode control,
general purpose, monolithic switching regulator suited for automotive
batteries, industrial power supplies, distributed supplies, and wall
transformers. The LT3663 includes
a low current shutdown mode, input
overvoltage and undervoltage lockout,
and thermal shutdown. The LT3663 is
available in 8-lead (2mm × 3mm) DFN
package with exposed pad. An 8-lead
MSOP package with exposed pad will
be available soon.
The LT3663 can also function as
a constant current, constant voltage
(CC/CV) source to charge a supercapacitor or other energy storage device.
The IC operates in constant current
mode at the programmed current
limit until the capacitor reaches the
programmed output voltage. It then
operates in a constant voltage mode
to maintain that voltage.
Figure 2 shows the LT3663 output current limit at 1.2A. For output
currents below 1.2A the regulator is
in constant voltage mode. When the
output current is increased to 1.2A it
goes into constant current mode. The
output current is maintained at 1.2A
from VOUT nominal down to 0V.
7.5V–36V to 5V Buck
Regulator with 1.2A
Output Current Limit
Figure 3 shows a LT3663 application
producing 5V at 1.2A from an input
of 7.5V to 36V. The input is capable
continued on page 29
21
L DESIGN FEATURES
Boost Converters for Keep-Alive
Circuits Draw Only 8.5µA of
by Xiaohua Su
Quiescent Current
Introduction
Industrial remote monitoring systems
and keep-alive circuits spend most of
their time idle. Many of these systems
use batteries, so to maximize run time
power losses,even during low power
idle modes, must be minimized. Even
at no load, power supplies draw some
current to produce a regulated voltage
for keep-alive circuits.
The LT8410/-1 DC/DC boost
converter features ultralow quiescent
current and integrated high value
feedback resistors to minimize the
draw on the battery when electronics
are idle.
An entire boost converter takes very
little space, as shown in Figure 1.
Ultralow Quiescent Current
Low Noise Boost Converter
with Output Disconnect
When a micropower boost converter
is in regulation with no load, the
input current depends mainly on
two things—the quiescent current
(required to keep regulation) and the
output feedback resistor value. When
the output voltage is high, the output
feedback resistor can easily dissipate
more power than the quiescent current
of the IC. The quiescent current of the
LT8410/-1 is a low 8.5µA, while the
integrated output feedback resistors
have very high values (12.4M/0.4M).
This enables the LT8410/-1 to dissipate very little power in regulation
at no load. In fact, the LT8410/-1 can
regulate a 16V output at no load from
3.6V input with about 30µA of average
input current. Figures 2, 3 and 4 show
the typical quiescent and input current
in regulation with no load.
The LT8410/-1 controls power
delivery by varying both the peak
inductor current and switch off time.
This control scheme results in low
output voltage ripple as well as high
eficiency over a wide load range. As
shown in Figure 5, even with a small
0.1µF output capacitor, the output
ripple is typically less than 10mV. The
part also features output disconnect,
which disconnects the output voltage
from the input during shutdown. This
output disconnect circuit also sets a
maximum output current limit, allowing the chip survive output shorts.
An Excellent Choice for
High Impedance Batteries
A power source with high internal
impedance, such as a coin cell battery,
may show normal output voltage on
a voltmeter, but its voltage can collapse under heavy current demands.
This makes it incompatible with high
Figure 1. The LT8410/-1 is designed
to facilitate compact board layout.
switch-current DC/DC converters.
The LT8410/-1 has an integrated
power switch and Schottky diode,
and the switch current limits are very
low (25mA for the LT8410 and 8mA
for the LT8410-1). This low switch
current limit enables the LT8410/-1
to operate very eficiently from high
impedance sources, such as coin cell
batteries, without causing inrush
current problems. Figure 6 shows
the LT8410-1 charging an electrolytic
capacitor. Without any additional external circuitry, the input current for
12
10
1000
6
4
2
AVERAGE INPUT CURRENT (µA)
8
QUIESCENT CURRENT (µA)
QUIESCENT CURRENT (µA)
VCC = 3.6V
10
8
6
4
2
100
VCC = 3.6V
0
– 40
0
40
80
TEMPERATURE (°C)
Figure 2. Quiescent current vs
temperature—not switching
22
120
0
0
4
8
12
VCC VOLTAGE (V)
Figure 3. Quiescent current
vs VCC voltage—not switching
16
10
0
10
20
30
OUTPUT VOLTAGE (V)
40
Figure 4. Average input current
in regulation with no load
Linear Technology Magazine • March 2009
DESIGN FEATURES L
100µH
VIN
2.5V to 16V
2.2µF
SW
CAP
VCC
VOUT
VOUT = 16V
0.1µF*
LT8410
VREF
SHDN
CHIP
ENABLE
604K
GND
0.1µF
FBP
412K
*HIGHER VALUE CAPACITOR IS REQUIRED
WHEN THE VIN IS HIGHER THAN 5V
10
100
VIN = 3.6V

R1 
1.30 •  1 + 
 R2 
VIN = 12V
90
8
EFFICIENCY (%)
VOUT PEAK-TO-PEAK RIPPLE (mV)
the SHDN pin below 0.3V shuts down
the part and reduces input current to
less than 1µA. When the part is on, and
the SHDN pin voltage is close to 1.3V,
0.1µA current lows out of the SHDN
pin. A programmable enable voltage
can be set up by connecting external
resistors as shown in Figure 7.
The turn-on voltage for the coniguration is:
0.1µF
6
4
2
VIN = 5V
80
and the turn-off voltage is:
VIN = 3.6V
70

R1 
(1.24 − R3 • 10 −7 ) •  1 +  − (R1• 10 −7 )
 R2 
60
50
0
0.01
0.1
1
LOAD CURRENT (mA)
40
0.01
10
0.1
1
10
LOAD CURRENT (mA)
100
Figure 5. General purpose bias with wide input voltage and low output voltage ripple
the entire charging cycle is less than
8mA.
Tiny Footprint with
Small Ceramic Capacitors
Available in a tiny 8-pin 2mm × 2mm
DFN package, the LT8410/-1 is internally compensated and stable for
a wide range of output capacitors. For
most applications, using 0.1µF output
capacitor and 1µF input capacitor is
suficient. An optional 0.1µF capacitor
at the VREF pin implements a soft-start
feature. The combination of small
package size and the ability to use
small ceramic capacitors enable the
VIN
2.5V to 16V
LT8410/-1 to it almost anywhere.
Figure 1 shows the size of a circuit
similar to that shown in Figure 4,
illustrating how little board space
is required to build a full featured
LT8410/-1 application.
where R1, R2 and R3 are resistance
in Ω. Programming the turn-on/turnoff voltage is particularly useful for
applications where high source impedance power sources are used, such as
energy harvesting applications.
By connecting an external capacitor (typically 47nF to 220nF) to the
VREF pin, a soft-start feature can be
implemented. When the part is brought
continued on page 29
ENABLE VOLTAGE
SHDN Pin Comparator and
Soft-Start Reset Feature
R1
An internal comparator compares the
SHDN pin voltage to an internal voltage reference of 1.3V, giving the part
a precise turn-on voltage level. The
SHDN pin has built-in programmable
hysteresis to reject noise and tolerate
slowly varying input voltages. Driving
R3
CONNECT TO
SHDN PIN
R2
Figure 7. Programming the enable
voltage by using external resistors
L1
220µH
C1
2.2µF
TURN ON/OFF
SW
CAP
VCC
VOUT
LT8410-1
VREF
SHDN
GND
FBP
C2
1.0µF
C3
10000µF
R1
604k
R2
412k
C1: 2.2μF, 16V, X5R, 0603
C2: 1.0μF, 25V, X5R, 0603*
C3: 10000μF, Electrolytic Capacitor
C4: 0.1μF, 16V, X7R, 0402
L1: COILCRAFT LPS3008-224ML
* HIGHER CAPACITANCE VALUE IS REQUIRED FOR
C2 WHEN THE VIN IS HIGHER THAN 12V
SHDN VOLTAGE
2V/DIV
C4
0.1µF
VOUT = 16V
VOUT VOLTAGE
10V/DIV
INPUT CURRENT
5mA/DIV
INDUCTOR
CURRENT
10mA/DIV
VIN = 3.6V
20s/DIV
Figure 6. Capacitor charger with the LT8410-1 and charging waveforms
Linear Technology Magazine • March 2009
23
L DESIGN FEATURES
Industrial/Automotive Step-Down
Regulator Accepts 3.6V to 36V
and Includes Power-On Reset and
Watchdog Timer in 3mm × 3mm QFN
by Ramanjot Singh
Introduction
As the number of microprocessors in
automotive and industrial applications continues to expand, so does the
need for rugged step-down regulators
that can operate over a wide input
voltage range and withstand high
voltage transients and output shorts.
Microprocessor-based applications
also require supervisory functions,
such as power-on reset (POR) and
watchdog timing, to ensure high system reliability. The regulator must
have high eficiency at light loads
to increase battery life. The LT3689
delivers all of these features in tiny
16-pin 3mm × 3mm QFN and 16-pin
MSOP packages.
Features of the LT3689
Step-Down Regulator
The LT3689 employs a constant frequency, current mode architecture to
provide 800mA of continuous output
current. The part operates from a
wide 3.6V to 36V input range and can
protect itself from input transients up
to 60V. It is internally compensated,
which helps to lower the external component count. The switching frequency
can be set anywhere between 350kHz
and 2.2MHz by tying a resistor from
VIN
4.5V TO 36V
TRANSIENT TO 60V
µP
2.2µF
I/O
Soft-Start and Output
Short Circuit Protection
The LT3689 includes a soft-start feature that limits the maximum inrush
current during start-up and recovery
from fault conditions. The soft-start
circuit ramps up the peak switch
current limit in approximately 150µs,
reducing the peak input current.
The DA pin is used to monitor the
current in the catch diode. If the catch
diode current at the end of switch
cycle is higher than the DA current
limit then the part delays the switch
turn-on until the catch diode current
drops below the DA current limit. This
protects the LT3689 in the face of
inductor current runaway situations,
EN/UVLO
I/O
WDO
LT3689
RST
CWDT
CPOR
68nF
tRST = 157ms
CPOR
OUT
BST
SW
WDI
RESET
CWDT
10nF
tWDU = 182ms
tWDL = 5.9ms
VIN
the R T pin to ground, allowing the designer to optimize component size and
eficiency. The switching frequency can
also be synchronized to an external
clock for noise sensitive applications.
An external resistor divider programs
the output voltage to any value above
the part’s 0.8V reference. Also, the
boost diode is integrated into the IC
to minimize solution size and cost.
Figure 1 shows a typical application
of LT3689.
GND
DA
FB
RT
SYNC
0.1µF
12µH
MBRM140 10pF
3.3V
800mA
316k
21k
100k
fSW = 700kHz
22µF
especially during output overload or
short at high switching frequencies
with high input voltages and small
inductor values. Other protection
features such as frequency foldback,
cycle-by-cycle current limit, and thermal shutdown together ensure that
the part is not damaged by excessive
switch currents during startup, overload or short circuit.
Pin Selectable Modes of
Operation: Low Ripple
Burst Mode Operation
and Pulse-Skipping Mode
Two modes of operation can be selected
through the SYNC pin. Applying a logic
low to the SYNC pin enables the low
ripple Burst Mode® operation, which
maintains high eficiency at light loads
while keeping output ripple low. In
Burst Mode operation, the LT3689
delivers single cycle bursts of current
to the output capacitor followed by
sleep periods. Between bursts, all circuitry associated with controlling the
output switch is shutdown, reducing
the VIN pin and OUT pin currents in
a typical application to a mere 50µA
and 75µA, respectively. As the load
current decreases to a no load condition, the percentage of sleep time
increases, thus decreasing average
input current.
A logic high on SYNC disables Burst
Mode operation, allowing the part to
skip pulses at light loads. The advantage of this pulse-skipping mode over
Burst Mode operation is that the part
continues to switch at the programmed
frequency (set by R T) down to very low
load currents, above 15mA at 12VIN
in a typical application.
Figure 1. LT3689 typical application circuit with reset time
set to 157ms and watchdog timeout period set to 182ms
24
Linear Technology Magazine • March 2009
DESIGN FEATURES L
The LT3689 can be shutdown by pulling the EN/UVLO pin below 0.3V. In
shutdown, quiescent current is less
than 0.5µA. The EN/UVLO pin can
also be used to perform an accurate
undervoltage lockout (UVLO) function.
A resistor divider from VIN pin can be
used to program the UVLO threshold
of the circuit using the 1.26V accurate
threshold of the EN/UVLO pin. A 4µA
current hysteresis on this pin is also
provided to allow the user to program
desired voltage hysteresis. The LT3689
also has an internal UVLO that prevents the part from switching if VIN
pin ever goes below 3.3V (typical). The
part only starts switching when VIN is
higher than 3.4V and EN/UVLO pin
is above the 1.26V threshold.
Low Dropout
The LT3689 features low dropout for
output voltages above 3V. The minimum operating voltage of the device
is determined either by the LT3689’s
internal undervoltage lockout or
by its maximum duty cycle. Unlike
many buck regulators, the LT3689
can extend its duty cycle by staying
on for multiple cycles, provided that
the boost capacitor is charged above
the minimum voltage of 2.5V. Eventually, after several switching cycles, the
boost capacitor discharges. Internal
circuitry detects this condition and
charges the boost capacitor only when
needed. Also, a bigger boost capacitor
allows even higher duty cycle, allowing extremely low dropout operation.
The dropout voltage for a 5V typical
application is about 400mV at 200mA
load and 900mV at 800mA load.
VOUT
2V/DIV
RST
2V/DIV
tRST = 165ms
CPOR = 71.3nF
CPOR
1V/DIV
50ms/DIV
Figure 2. Power-on reset feature of LT3689
Linear Technology Magazine • March 2009
Power-On Reset (POR)
Many microprocessor -based applications powered by the output
of a switching regulator must know
when the regulator output is ready
and stable before the microprocessor
starts operating. Likewise, once running, the electronic system must be
warned when the regulator output has
dropped below a minimum tolerable
threshold, such as during overload or
shutdown conditions. This is required
to prevent unreliable operation and to
allow the microprocessor to perform
housekeeping operations before power
is completely lost.
The LT3689’s accurate internal
voltage reference and glitch immune
precision POR comparator and timer
circuit feed these speciic needs of
microprocessor-based applications.
The switcher’s output voltage must
be above 90% of programmed value
for its RST pin to remain high (refer
to Figure 2). The LT3689 asserts
RST during power-up, power-down
and brown-out conditions. Once the
output voltage rises above the RST
threshold, the adjustable reset timer
is started and RST is released after the
reset timeout period. On power-down,
once output voltage drops below RST
threshold, RST is held at a logic low.
The reset timer is adjustable using an
external capacitor. The RST pin has a
weak pull up to the OUT pin.
The POR comparator is designed to
avoid false triggering. High frequency
noise on the FB pin can falsely trip
RST, particularly when the monitored
output is already near the reset threshold. This can cause oscillatory behavior
at the RST pin. The traditional way of
tackling this problem is to add some
DC hysteresis in the comparator input,
which changes the threshold point
once the output lips. The problem
is that the addition of DC hysteresis
makes the trip voltage less accurate,
since the trip point changes once the
output changes. The LT3689 does not
use hysteresis. Instead, it performs an
integration-like function on transient
events at the comparator. In this way
the magnitude and duration of the
event are both important to the comparator threshold. Figure 3 illustrates
the typical transient duration versus
comparator overdrive (as a percentage
of trip threshold) required to trip the
comparator.
Selecting the Reset
Timing Capacitor
The reset timeout period is adjustable
in order to accommodate a variety of
microprocessor applications. The reset
timeout period, tRST, is adjusted by
connecting a capacitor between the
CPOR pin and ground. The value of this
capacitor is determined by:
CPOR = tRST • 432 • 10–9
with CPOR in Farads and tRST in seconds. The CPOR value per millisecond
of delay can also be expressed as
CPOR/ms = 432 (pF/ms).
Leaving the CPOR pin unconnected
generates a minimum reset timeout
of approximately 25µs with 10kΩ
pull-up to 5V on RST pin. Maximum
reset timeout is limited by the largest
available low leakage capacitor. The
accuracy of the timeout period will
be affected by capacitor leakage (the
nominal charging current is 2µA) and
capacitor tolerance. A low leakage ceramic capacitor is recommended.
Watchdog Modes:
Timeout or Window
The LT3689 also includes an adjustable watchdog timer that monitors a
microprocessor’s activity. If a code
execution error occurs in a µP, the
watchdog detects the error and sets
the WDO pin low. This signal can
be used to interrupt a routine or to
reset a µP.
800
700
TRANSIENT DURATION (µs)
Programmable
Undervoltage Lockout
600
500
400
RESET OCCURS
ABOVE THE CURVE
300
200
100
0
0.10
1.00
10.00
100.00
POR COMPARATOR OVERDRIVE VOLTAGE AS PERCENTAGE
OF RESET THRESHOLD, VRST (%)
Figure 3. Typical transient duration
vs POR comparator overdrive
25
L DESIGN FEATURES
The watchdog is operated either
in timeout or window mode (refer to
Figure 4). In timeout mode, the microprocessor needs to toggle the WDI pin
before the watchdog timer expires to
keep the WDO pin high. If the voltage
on the WDI pin does not transition
during the programmed timeout period
then the circuitry pulls WDO low.
In window mode, the WDI pin’s
negative-going pulses must appear
inside a programmed time window to
prevent WDO from going low. If more
than two falling pulses are registered
in the lower boundary period (tWDL),
the WDO pin is forced low. The WDO
pin also goes low if no negative edge
is supplied to the WDI pin within the
upper boundary period (tWDU).
During a code execution error, the
microprocessor outputs WDI pulses
that are either too fast or too slow.
This condition asserts WDO low and
forces the microprocessor to reset the
program.
In window mode, the WDI signal is
bounded by an upper and lower boundary periods for normal operation. The
period of the WDI input signal should
be longer than the window mode’s lower boundary period and shorter than
the upper boundary period to keep
WDO high under normal conditions.
The window mode’s lower and upper
boundary periods have a ixed ratio
of 31. These times can be increased
or decreased by adjusting an external
capacitor on the CWDT pin.
In both watchdog modes, when
WDO is asserted, the reset timer is
enabled. Any WDI pulses that appear
while the reset timer is running are
ignored. When the reset timer expires,
the WDO is allowed to go back high
again. Therefore, if no input is applied
to the WDI pin then the watchdog
circuitry produces a train of pulses
on the WDO pin. The high time of
this pulse train is equal to the upper
boundary period and low time is equal
to the reset period. Also, WDO and
RST cannot be logic low simultaneously. If WDO is low and there is an
undervoltage lockout fault, RST goes
low and WDO will go high.
The WDE pin allows the user to turn
on or off the watchdog function. This
26
WDI
WDO
tWDU
tRST
WATCHDOG TIMING (W/T = HIGH), TIMEOUT MODE
t < t WDL
tRST
WDI
WDO
tRST
tWDU
WATCHDOG TIMING (W/ T = LOW), WINDOW MODE
tRST = PROGRAMMED RESET PERIOD
tWDU = WATCHDOG UPPER BOUNDARY PERIOD
tWDL = WATCHDOG WINDOW MODE LOWER BOUNDARY PERIOD
VUV = OUTPUT VOLTAGE RESET THRESHOLD
Figure 4. Watchdog timing diagram
feature can be used to reliably program
the connected microprocessor in the
factory. During factory programming
of the microprocessor, WDE pin can
be kept high to prevent WDO from
toggling and thus prevents WDO from
interfering with the microprocessor’s
programming procedure.
Tying the WDO and RST pins
together will generate a reset signal
when either the output voltage falls
10% below the regulation value or if
there is a watchdog error.
Leaving the CWDT pin unconnected
generates a minimum watchdog upper boundary period of approximately
200µs with 10kΩ pull-up to 5V on
WDO pin. Maximum timeout is limited
by the largest available low leakage
capacitor. The accuracy of the upper
and lower boundary periods is affected
by capacitor leakage (the nominal
charging current is 2µA) and capacitor tolerance. A low leakage ceramic
capacitor is recommended.
Selecting the Watchdog
Timing Capacitor
The wide input range, low quiescent
current, supervisory features, robustness and small size of the LT3689
makes it an ideal candidate to power
automotive and industrial applications. The part withstands 60V VIN
transients and normal operation is
guaranteed for max VIN of 36V, and
the part is robust against inrush and
short circuit conditions. The Burst
Mode circuitry provides high eficiencies at light loads. Programmable
switching frequency allows the designer to trade off between component
size and eficiency. The accurate POR
and Watchdog circuitry of LT3689
allows complete supervisory control
of a microprocessor connected to
the output of the LT3689 switching
regulator. L
The watchdog upper boundary period
is adjustable and can be optimized
for software execution. The watchdog
upper boundary period is adjusted by
connecting a capacitor between the
CWDT pin and ground. Given a speciied
watchdog upper boundary period, the
capacitor is determined by:
CWDT = tWDU • 55 (pF/ms)
The window mode lower boundary
period has a ixed relationship to upper
boundary period for a given capacitor.
The lower boundary period is related
to the upper boundary period by the
following:
tWDL = 1/31 • tWDU
Conclusion
Linear Technology Magazine • March 2009
DESIGN FEATURES L
Complete APD Bias Solution in 60mm2
with On-the-Fly Adjustable Current
By Xin (Shin) Qi
Limit and Adjustable VAPD
Introduction
The overriding factor limiting functionality in iber-optic communication
systems is available space. A compact
APD (avalanche photo diode) bias
solution with a high degree of feature
integration is the key to breaking
new ground in system size and performance. The LT3571 offers such a
solution in a tiny 3mm × 3mm QFN
package.
The LT3571 combines a current
mode step-up DC/DC converter and
a high side ixed voltage drop APD
current monitor with an integrated
75V power switch and Schottky diode.
The combination of a traditional voltage loop and a unique current loop
allows customers to set an accurate
APD current limit at any given bias
voltage. The integrated high side current monitor provides an 8% accurate
current that is proportional to the load
current, making it possible to adjust
the APD bias voltage via the CTRL
pin. This feature-rich device makes
it possible to produce a single stage
boost converter to bias high voltage
APDs in only 60mm2.
Low Noise APD Bias Supply
The gain of the APD is dependent on the
bias voltage, so the bias supply must
minimize the noise contamination
from switching regulators and other
sources. Figure 1 shows the LT3571
conigured to produce an ultralow
noise power supply for a 45V APD
with 2.5mA of load current capability.
The MONIN voltage is regulated by
the internal voltage reference and the
resistor divider made up of R1 and R2.
Resistor RSENSE is selected to set the
APD current limit at 200mV/1.2RSENSE
– 0.2mA.
The CTRL pin can override the
internal reference, making it possible
to optimize the APD bias on the ly
to maximize receiver performance.
Linear Technology Magazine • March 2009
L1
10µH
VIN
5V
OFF ON
VIN
SHDN
SW
VOUT
RSENSE
20Ω
VREF
C1
1µF
CTRL
LT3571
MONIN
50V
C3
10nF
RT
SYNC
R1
1M
FB
GND MON
R2
20.5k
APD
RT
12.1k
1MHz
R4
49.9Ω
C5
10nF
R3
10k
C2
0.1µF
45V
C4
0.1µF
L: TDK VLF3010AT – 100MR49
C1: TDK X7R C1608X7R1C105KT
C2, C4: MURATA X7R GRM188R72A104KA35
C3: AVX X7R 06031C103K
C5: MURATA X7R GRM155R71H103K
Figure 1. Low noise APD bias supply
When the CTRL pin is connected to a
supply above 1V, the output voltage is
regulated with feedback at 1V. When
driven below 1V, the feedback and the
output voltage follow accordingly.
The APD pin, the output of the current monitor, provides a voltage to the
APD load that is ixed 5V below the
MONIN pin. The LT3571 includes a
precise current mirror with a factorof-ive attenuation. The proportional
current output signal at the MON pin
can be used to accurately indicate the
APD signal strength. The voltage variance of APD pin voltage is only ±200mV
over the entire input current range and
the whole temperature range. Figure
2 shows the evaluation board for this
topology.
The topology uses several ilter
capacitors to achieve ultralow noise
performance. The capacitor at VOUT
pin and the 0.1µF capacitor at the
APD pin suppress switching noise. The
10nF feedforward capacitor across the
MONIN and FB pins ilters out high
frequency internal reference and error
ampliier noise. Figure 3 shows the
measured switching noise is less than
500µVP–P at 1mA load current. This
exceptionally low noise bias voltage
500µV/DIV
IAPD = 1mA
Figure 2. The LT3571 evaluation board
500ns/DIV
Figure 3. AC-coupled noise ripple at APD pin
27
L DESIGN FEATURES
gives the APD greater sensitivity and
dynamic range.
L1
10µH
VIN
3.3V
Fast APD Current Monitor
Transient Response
Design efforts in modern communications systems increasingly focus
on 10Gbits/s GPON systems, which
demand that the transient response of
the APD current monitor is less than
100ns for a two-decades-of-magnitude
input current step. To meet this challenging requirement, many designers
rely on a simple discrete current mirror
topology to reduce parasitic capacitance on the signal path, sacriicing
monitor accuracy and board space. In
contrast, the LT3571’s APD current
monitor is carefully designed to provide
not only a ixed voltage drop and high
accuracy, but also the required fast
transient response.
Figure 4 shows a compact circuit
that responds quickly to current
transients. Unlike the ultralow noise
topology shown in Figure 1, the ilter
capacitor at the APD pin is moved to
the MONIN pin. C2, C3 and RSENSE form
a π ilter to isolate the APD current
monitor from high frequency switching
noise. The capacitor at the MON pin is
also removed to reduce the transient
delay on the measurement path.
The transient speed is measured
using the same technique described in
the Linear Technology Design Note 447
“A Complete Compact APD Bias Solution for a 10GBit/s GPON System.”
Figures 5 and 6 show the measured
input signal falling transient response
and input signal rising transient response, respectively, where the input
current levels are 10µA and 1mA.
Note that there is an inversion and
DC offset present in the measurement.
The measurements show a transient
response time of less than 100ns, well
within the stringent speed demands of
the 10Gbits/s GPON system.
APD Bias Voltage
Temperature Compensation
Typically, the APD reverse bias voltage
is designed with a compensatory positive temperature coeficient. This can
be easily implemented via the CTRL
pin of the LT3571—a less complex
28
VIN
SHDN
OFF ON
SW
VOUT
RSENSE
20Ω
VREF
C1
1µF
CTRL
LT3571
55V
MONIN
R1
1M
RT
SYNC
GND
C2
0.1µF
FB
MON
R2
18.2k
APD
RT
26.1k
500kHz
C3
0.1µF
50V
0.5pF
FOR TEST PURPOSES,
REPLACE APD WITH
THIS SIMPLE TEST SETUP
APD PIN
4.99k
PMBT3904
–
2.5V
4.99k
LT1815
1k
+
MEASURE
HERE
L1: TDK VLF3010AT-100MR49
0.1µF
C1: MURATA X7R GRM21BR71C105KA01B
C2, C3: MURATA X7R GRM188R72A104KA35
PWM
–VLO
–VHI
Figure 4. APD bias supply with ultrafast current monitor transient speed
PWM GND
PWM GND
PWM
1V/DIV
IAPD = 10µA
IAPD = 10µA
PWM
1V/DIV
IAPD = 1mA
IAPD = 1mA
OUT
500mV/DIV
OUT
500mV/DIV
TFD < 100ns
TRD < 100ns
OUT GND
OUT GND
50ns/DIV
50ns/DIV
Figure 5. Transient response on input
signal falling edge (1mA to 10µA)
Figure 6. Transient response on input
signal rising edge (10µA to 1mA)
L1
15µH
VIN
5V
R5
30.1k
Q2
R7
49.9k
R8
36.5k
R9
20k
OFF ON
R6
100k
VIN
SHDN
VREF
SW
VOUT
RSENSE
49.9Ω
LT3571
MONIN
CTRL
RT
SYNC
C6
0.1µF
TEMPERATURE
COMPENSATION BLOCK
C1
1µF
Q1
C3
10nF
R1
1M
FB
GND MON
R2
15k
APD
RT
33.2k
400kHz
R4
49.9Ω
C5
10nF
L1: TDK VLF4012AT – 150MR63
C1: TDK X7R C1608X7R1C105KT
C2: MURATA X7R GRM21AR72A224KAC5L
C3: AVX X7R 06031C103K
C4: MURATA X7R GRM188R72A104KA35
C5: MURATA X7R GRM155R71H103K
C6: MURATA X7R GRM155R71A104KA01D
55V
R3
10k
C2
0.22µF
50V
C4
0.1µF
Q1, Q2 = PHILIPS PEMT1
Figure 7. Temperature-compensated APD power supply
Linear Technology Magazine • March 2009
DESIGN FEATURES L
and expensive solution than typical
microprocessor-controlled methods.
The simplest scheme uses a resistor
divider from the VREF pin to the CTRL
pin, where the top resistor in the divider is an NTC (negative temperature
coeficient) resistor. While simple,
this method suffers from nonlinear
temperature coeficient of the NTC
resistor. A more precise method uses
a transistor network as shown in Figure 7. The PTC (Positive Temperature
Coeficient) of the CTRL pin voltage is
realized by an emitter follower of Q1
and a VBE multiplier of Q2.
Assuming:
VBE(Q1) = VBE(Q2) = VBE
dT
=
dVBE(Q2)
dT
=
2mV
°C
R8
V
R7 BE
with
60
PTC =
58
56
VAPD (V)
dVCTRL R8 2mV
=
•
dT
R7 °C
52
50
48
R8
=
R7
46
44
42
40
–50
–25
0
50
25
75
TEMPERATURE (°C)
100
125
Figure 8. Temperature response
of the circuit shown in Figure 7
R1
=
R2
VREF
VOUT
2mV
VBE +
•
°C dVOUT dT
VBE •
dVBE(Q2)
dT
=
2mV
°C
Conclusion
Given VOUT at room and dVOUT/DT,
the R1/R2 and R8/R7 can be calculated as follows
54
=
Simulation using LTspice always
gives a good starting point. The circuit
shown in Figure 7 is designed to have
VAPD = 50V (VOUT = 55V) at room and
dVAPD/dT = 100mV/°C (dVOUT/dT =
100mV/°C). The measured temperature response is shown in Figure 8,
which is very close to the design
target.
then the CTRL pin voltage is
VCTRL = VREF −
dVBE(Q1)
dT
and
dVBE(Q1)
Resistors R5–R9 are selected to make
I(Q1) = I(Q2) ≈ 10µA, and
dVOUT 2mV
+
• VOUT
dT
°C
−1
2mV
• VREF
°C
The LT3571 is a highly integrated,
compact solution to APD bias supply
design. It provides a useful feature set
and the lexibility to meet a variety of
challenging requirements, such as low
noise, fast transient response speed,
and temperature compensation. With
a high level of integration and superior performance, the LT3571 is the
natural choice for APD bias supply
design. L
LT840, continued from page 2
out of shutdown, the VREF pin is irst
discharged for 70µs with a strong pull
down current, and then charged with
10µA to 1.235V. This achieves soft
start since the output is proportional
to VREF. Full soft-start cycles occur
even with short SHDN low pulses
since VREF is discharged when the
part is enabled.
In addition, the LT8410/-1 features
a 2.5V to 16V input voltage range, up
to 40V output voltage and overvoltage
protection for CAP and VOUT.
Conclusion
The LT8410/-1 is a smart choice
for applications which require low
quiescent current and low input current. The ultralow quiescent current,
combined with high value integrated
feedback resistors, keeps the average
input current very low, signiicantly
100
LT65/6, continued from page 2
Conclusion
90
EFFICIENCY (%)
of handling 60V transients. Figure 4
shows the circuit eficiency at multiple
input voltages.
The current limit of the application
is set to 1.2A, therefore, the power path
components are sized to handle 1.2A
maximum. To reduce the application
footprint, the LT3663 includes internal
compensation and a boost diode. The
RUN pin, when low, puts the LT3663
into a low current shutdown mode.
extending battery operating time.
Low current limit internal switches
(8mA for the LT8410-1, 25mA for the
LT8410) make the part ideal for high
impedance sources such as coin cell
batteries. The LT8410/-1 is packed
with features without compromising
performance or ease of use and is
available in a tiny 8-pin 2mm × 2mm
package. L
The accurate programmable output
current limit of the LT3653 and
LT3663 eliminates localized heating
from an output overload, reduces the
maximum current requirements on the
power components, and makes for a
robust power supply solutions. L
VIN = 8V
VIN = 15V
80
VIN = 30V
70
60
50
40
0.1
0.3
0.5
0.7
0.9
OUTPUT CURRENT (A)
1.1
1.3
Authors can be contacted
at (408) 432-1900
Figure 4. Efficiency of the circuit in Figure 3
Linear Technology Magazine • March 2009
29
L DESIGN IDEAS
Don’t Want to Hear It? Avoid the
Audio Band with PWM LED Dimming
at Frequencies Above 20kHz by Eric Young
Introduction
The requirements of LED drivers become more demanding as application
designers exploit the unique characteristics of LEDs. Linear Technology offers
a complete portfolio of LED drivers
with the performance levels required to
meet even the most challenging design
requirements. One area where these
LED drivers especially excel is in the
performance and lexibility of their
PWM dimming capabilities. LEDs can
be turned on and off rapidly—it takes
only nanoseconds to illuminate or
extinguish the source. PWM dimming
exploits this characteristic to achieve
orders of magnitude dimming, even
while maintaining a constant output
spectrum over the entire dynamic light
intensity range.
The broad ield of available LED
drivers narrows quite a bit when
one considers PWM dimming at frequencies above 20kHz. Why 20kHz?
Although most LED light designers
worry about perceptible licker at
PWM frequencies below about 100Hz,
in some applications the human eye
is not the limiting factor; it is the human ear. The human ear perceives
vibrations up to about 20kHz, which
in some applications can become the
important factor in determining PWM
frequency. The versatile LT3755 and
LT3756 are members of an elite group
VPWM
VGATE
IL1
5A/DIV
ILED
0.5A/DIV
5µs/DIV
Figure 2. DCM operation of the
boost LED driver in Figure 1
30
VIN
8V TO
18V
L1
1.5µH
C1
2.2µF
x2
25V
499k
D1
10µF
x2
35V
VIN
SHDN/UVLO
GATE
VREF
100k
M1
SENSE
255k
LT3755
0.01Ω
CTRL
499k
75k
FB
17.8k
PWM
ISP
INTVCC
100k
0.1Ω
OPENLED
ISN
D2
VC
0.1µF
10k
UP TO
8 LEDS
26V
PWMOUT
SS
C2
4.7µF
350mA
22k
470pF
GND
M2
RT
13k
800kHz
L1: COILTRONICS DR125-1R5
D1: ON SEMI MBRS360
M1: VISHAY SILICONIX Si7850DP
M2: VISHAY SILICONIX Si2306DS
D2: IN4448HWT
Figure 1. This 10W boost LED driver stays out of the audio band by
achieving 50:1 PWM dimming at 20kHz. Lower PWM frequencies can result
in an audible hum as ceramic capacitors vibrate.
of LED controllers that can support
very high PWM dimming ratios, as
much as 50:1, at 20kHz. These controllers support a variety of topologies,
including buck mode, boost and buckboost at various power levels.
High Performance
PWM Dimming
The PWM dimming method is straightforward; the LED is driven by a
tightly regulated current for a ixed
interval in every PWM period. During
the off-phase, the current in the LED is
zero. During the on-phase, the current
is carefully regulated. It is important
that the “on” current is consistent,
since an LED’s output spectrum is a
function of forward current. The duty
cycle of the PWM signal corresponds
to the dimming value.
Although the concept is simple,
designing a controller that can achieve
this at a high PWM frequency is any-
thing but simple. The rise and fall
times of the pulsed current should
be fast, less than 100ns. Generating
a suitable PWM current pulse from
an arbitrary input voltage can prove
a challenge. This usually requires a
high bandwidth DC/DC converter to
regulate the current, a storage/ilter
capacitor across the LED to provide
current during PWM on/off transitions, and a disconnect switch to
ensure that the current waveform has
sharp turn-on and off edges.
Hysteretic converters, while simple
to use from the standpoint of closed
loop stability, have problems. The
slow LED current rise and fall times
are one consequence of using a large
value inductor to smooth the current
through the LED because there is
no output capacitor. And since the
average current in the LED is related
to the ripple current in the inductor,
which is in turn sensitive to input voltLinear Technology Magazine • March 2009
DESIGN IDEAS L
1M
trollers can be conigured into several
different converter circuits to provide a
high bandwidth, well regulated output
current that can be pulsed at intervals
as short as 1µs.
96
92
Discontinuous Conduction
Mode Is the Secret
to Maximizing PWM
Performance
88
84
80
0.0
0.2
1.0
an annoying buzz or hum next to a
handheld device containing one of
these circuits, then you have observed
this effect.
The use of a disconnect switch in
series with the LED greatly reduces
the voltage transient and therefore the
hum from the output capacitor. While
good design techniques can greatly
minimize audible noise for lower PWM
frequencies, the elimination of audible
emission is not assured so long as PWM
frequency is below 20kHz. Many application designers don’t want to tinker
with acoustics, preferring instead quiet
running circuits that do a reasonable
job of PWM dimming. The LT3755 and
LT3756 current-mode switching con-
The key to short on/off times is for the
switching regulator to operate in discontinuous conduction mode (DCM).
In this mode, the inductor current
always starts from zero at the beginning of each switching period and the
peak inductor current is determined
by the load and adjusted through
the switch duty cycle. In contrast,
continuous conduction mode (CCM)
maintains a relatively constant switch
duty cycle and adjusts the average
inductor current to meet the demands
of the load.
DCM is superior for high performance PWM dimming because it
delivers the required energy to the
output in a single switching period.
This allows the controller to bypass the
typical minimum PWM period of 3-4
switching cycles to reach steady state,
a familiar requirement of CCM. Operation in DCM places greater demands
ISP
VIN
0.2Ω
VREF
500mA
ISN
CTRL
UP TO
5 LEDS
16V
0.22µF
D2
6.2V
INTVCC
2200pF
OPENLED
L1
3.3µH
D1
M1
GATE
VLED = 16V
ILED = 0.5A
96
4.7µF 2x
25V
1M
LT3755
4.7µF
100
M2
PWMOUT
PWM
0.8
Figure 3. The efficiency of the boost LED
driver in Figure 1 is greater than 90%.
SHDN/UVLO
68.1k
0.4
0.6
LED CURRENT(A)
EFFICIENCY (%)
VIN
22V TO
36V
100
EFFICIENCY (%)
age transients, the LED light output
changes with input supply. In most
cases, this method cannot provide
acceptable PWM performance.
What determines PWM performance? The PWM interval or frequency
is determined by the application, and
there are several considerations to
bear in mind. First, the human eye
generally does not perceive licker if the
PWM frequency is greater than 120Hz,
thus a lower bound on the interval is
typically taken to be 8ms.
The achievable dimming ratio is
a function of the minimum on- and
off-times of the current pulse provided by the driver circuit. So an 8µs
minimum pulse yields a 1000:1 dimming capability at 120Hz. The 20kHz
audible requirement comes about
because audible physical vibrations
can be introduced to the PC board
by the ceramic capacitors, and these
caps are ubiquitous in high bandwidth
converter circuits because of their low
ESR, ruggedness, and long-term reliability. Ceramic capacitors physically
change dimension (as well as value)
with a change in applied voltage, and
rapid voltage transients during the
PWM transients cause rapid changes
in dimensions that couple vibrations
into the boards. If you ever noticed
92
88
84
SENSE
SS
VC
GND
FB
22k
0.1µF
470pF
RT
13k
800kHz
0.033Ω
2.2µF 2x
50V
80
15
20
25
30
VIN (V)
35
40
L1: TOKO 962BS_3R3M
M1: VISHAY SILICONIX Si7850DP
M2: VISHAY SILICONIX Si2306DS
D1: DIODES, INC SBM540
Figure 4. An 8W buck-mode LED driver with 50:1 PWM dimming at 20kHz and 90% efficiency
Linear Technology Magazine • March 2009
31
L DESIGN IDEAS
on switching components because the
switching components see higher peak
currents for a given load. Because
of this, a controller is easier to use
than a monolithic converter because
its maximum switching current can
be programmed to the needs of the
application, without having to change
the application’s features.
Operating in DCM does come at
a price when compared to CCM:
eficiency, input supply range and
analog dimming range all suffer some
reduction. The ratio of maximumto-minimum input supply range is
slightly less than the ratio of the
minimum PWM pulse width to the
minimum switch on-time. Likewise,
provided the input supply is ixed,
the maximum analog dimming ratio
is the same ratio of minimum PWM
pulse to minimum switch on-time.
Nevertheless, the beneit of this technique is that minimum PWM period
is four to ive times shorter compared
with continuous conduction mode. If
the application calls for high PWM
dimming ratio, DCM mode provides
a sure path to achieve that objective.
Three application circuits built with
LT3755 and shown here demonstrate
this technique.
ILED
500mA/DIV
3% PWM DUTY
50% PWM DUTY
97% PWM DUTY
5µs/DIV
Figure 5. Three PWM dimming settings
for the buck mode driver in Figure 4.
Even at 33kHz there is no perceptible
change in the LED current from
minimum to maximum duty cycle.
Figure 1 shows a 9W boost converter that regulates 26V of LEDs at a
steady 350mA from a supply ranging
between 8V and 18V. If the supply is
ixed at 12V, the regulator operates at
constant switching frequency for LED
currents programmed by the CTRL
pin between 125mA and 1A (2.4W to
27W). The minimum on-time is 1µs, as
is the minimum off-time. The switching waveforms in Figure 2 show the
operation at 50% duty cycle, 27V/1A
load and 12V supply. Notice the fast
rise and fall times of the LED current
signal, even at 1A. At maximum load,
the GATE pin is 7V for almost 1µs
(same as the minimum pulse width)
and the inductor current reaches zero
before the start of the each GATE pulse,
a characteristic of DCM operation.
Figure 3 shows the eficiency versus
LED current at 12V input, which peaks
at just over 90%.
Figure 4 shows a buck-mode converter that regulates a 16V LED string
at 500mA from a 22V to 36V supply.
This circuit has an external chargepump and level shift to drive the gate
of an LED disconnect NMOS. This level
shift provides much faster rise and fall
times than the familiar resistor level
shift driving a PMOS, and uses much
less current. The scope trace in Figure
5 shows PWM dimming at several duty
cycles—it is clear that the output LED
current has no perceptible variation
as pulse width is smoothly adjusted
between the minimum on-time and
the minimum off-time. The eficiency
of this 8W circuit exceeds 90%.
Figure 6 shows a SEPIC converter
driving a 1A, 20V LED string from
a 12V-to-36V supply. In addition to
providing step-up and step-down capability, this circuit is handy because
it provides input-output isolation and
built in protection from a short to GND
on the output. The eficiency of this
circuit exceeds 87%. The minimum
continued on page 40
L1
1.5µH
VIN
10V TO
36V
D1
1:1
1M
C1
4.7µF
50V
1µF x 2
100V
SHDN/UVLO
200k
10µF
x2
25V
VIN
GATE
VREF
M1
L1B
SENSE
0.01Ω
GATE
CTRL
169k
PWM
FB
LT3755
ISP
10k
0.1Ω
INTVCC
100k
1A
ISN
OPENLED
20W
LED
STRING
D2
PWMOUT
SS
VC
0.1µF
4.7µF
10k
22k
470pF
SWITCH CURRENT
(2A/DIV) PEAKS
AT CURRENT
LIMIT OF 10A
GND
OUTPUT SHORT
CIRCUIT CURRENT
(2A/DIV)
5µs/DIV
RT
13k
800kHz
Figure 7. The SEPIC converter in Figure 6
maintains control during an output fault to GND.
M2
L1: COILTRONICS DRQ125-1R5 COUPLED INDUCTOR
D1: ON SEMI MBRS360
M1: VISHAY SILICONIX Si7850DP
M2: VISHAY SILICONIX Si2306DS
Figure 6. A 20W SEPIC LED Driver with 50:1 PWM
dimming at 20kHz and output fault protection
32
Linear Technology Magazine • March 2009
DESIGN IDEAS L
Eliminate EMI Worries with 2A,
15mm × 9mm × 2.82mm
µModule Step-Down Regulator
by David Ng
Introduction
“We failed EMI.” Those three dreaded
words strike fear into the hearts and
minds of electronics design engineers.
There are four words that are even
worse: “We failed EMI again.” The
psyche of many a seasoned engineer
is scarred with dark memories of long
days and nights in an EMI lab, struggling with aluminum foil, copper tape,
clamp-on ilter beads and inger cuts to
ix a design that just won’t keep quiet. A
big part of the problem is the necessary
profusion of switching power supplies,
which contribute signiicantly to the
radiated system EMI.
The LTM8032 is a DC/DC switching
step-down µModule regulator built
speciically for low EMI. It is rated for
up to 36VIN, 10VOUT at 2A, and features
adjustable frequency, synchronization, a power good status lag and
soft-start. It is small, measuring only
15mm × 9mm × 2.82mm, integrating
the inductor, power stage and controller in a ROHS e3-compliant molded
LGA package.
10V/2A Supply Is
EN55022 and CSIPR 22
Class B Compliant
Like most other µModule regulators, the LTM8032 is easy to use. As
shown in Figure 1, all that is needed
for a complete power design are the
resistors to set the output voltage and
operating frequency, and the input
and output caps.
The LTM8032 is test-proven
EN55022 and CSIPR 22 class B
complaint, tested in an NRTL 5-meter
chamber, set up as shown in the photo
given in Figure 2. The LTM8032 is
mounted on a circuit board with no
bulk capacitance installed. The input
and output capacitance are the minimum ceramic values speciied in the
data sheet for proper operation.
Linear Technology Magazine • March 2009
VIN
5.5VDC TO 36VDC
OUT
VIN
22µF
FIN
RUN/SS
2.2µF
VOUT
3.3V
2A
AUX
LTM8032 BIAS
SHARE
PGOOD
RT SYNC GND ADJ
54.9k
78.7k
Figure 1. Just two resistors, input and output caps are needed
to complete a power supply design with the LTM8032.
The LTM8032 is a
DC/DC switching step-down
µModule regulator built
for low EMI. It is rated for
up to 36VIN, 10VOUT at 2A,
integrating the inductor,
power stage and controller
in a ROHS e3-compliant
molded LGA package.
The assembled unit is placed
atop an all-wood table. The all-wood
construction ensures that the test
setup does not shield or shadow noise
emanating from the device under test
(DUT). The power source, a linear lab
grade power supply, is on the loor. The
load for the LTM8032, with its heat
sink, is also on the table top.
Before measuring the emissions
from the LTM8032, a baseline measurement is taken to establish the
continued on page 8
Figure 2. For EMI testing, the DUT is mounted on a circuit board
and placed on a wooden table. The power source is on the floor.
33
L DESIGN IDEAS
Diode Turn-On Time Induced
Failures in Switching Regulators
Never Has So Much Trouble Been Had
by So Many with So Few Terminals
by Jim Williams and David Beebe
+V
This article is excerpted from the Linear
Technology Application Note AN22
with the same title.
+V
VIN
IC REGULATOR
VIN
Introduction
Most circuit designers are familiar with
diode dynamic characteristics such
as charge storage, voltage dependent
capacitance and reverse recovery
time. Less commonly acknowledged
and manufacturer speciied is diode
forward turn-on time. This parameter
describes the time required for a diode
to turn on and clamp at its forward voltage drop. Historically, this extremely
short time, units of nanoseconds, has
been so small that user and vendor
alike have essentially ignored it. It
is rarely discussed and almost never
speciied. Recently, switching regulator clock rate and transition time have
become faster, making diode turn-on
time a critical issue. Increased clock
rates are mandated to achieve smaller
magnetics size; decreased transition
times somewhat aid overall eficiency
but are principally needed to minimize
IC heat rise. At clock speeds beyond
about 1MHz, transition time losses are
the primary source of die heating.
A potential dificulty due to diode
turn-on time is that the resultant
OUTPUT
VREG
SWITCH PIN
IC REGULATOR
CONTROL
SWITCH
SWITCH
PIN
OUTPUT
VREG
CONTROL
SWITCH
REF
REF
GND
GND
FEEDBACK NODE
FEEDBACK NODE
STEP-UP
STEP-DOWN
Figure 1. Typical voltage step-up/step-down converters. Assumption
is diode clamps switch pin voltage excursion to safe limits.
transitory “overshoot” voltage across
the diode, even when restricted to
nanoseconds, can induce overvoltage
stress, causing switching regulator
IC failure. As such, careful testing is
required to qualify a given diode for a
particular application to insure reliability. This testing, which assumes
low loss surrounding components
and layout in the inal application,
measures turn-on overshoot voltage
due to diode parasitics only. Improper
IC BREAKDOWN LIMIT
associated component selection and
layout will contribute additional overstress terms.
Diode Turn-On Time
Perspectives
Figure 1 shows typical step-up and
step-down voltage converters. In both
cases, the assumption is that the diode
clamps switch pin voltage excursions
to safe limits. In the step-up case, this
limit is deined by the switch pins
PULSE IN
tRISE ≤ 2ns
AMPLITUDE = 5V + VFWD
DIODE ON VOLTAGE
5Ω
MEASUREMENT POINT
DIODE
UNDER
TEST
DIODE TURN-ON TIME
AN122 F02
AN122 F03
Figure 2. Diode forward turn-on time permits transient excursion above
nominal diode clamp voltage, potentially exceeding IC breakdown limit.
34
Figure 3. Conceptual method tests diode turn-on time at 1A. Input
step must have exceptionally fast, high fidelity transition.
Linear Technology Magazine • March 2009
DESIGN IDEAS L
PULSE CURRENT
AMPLIFIER
tRISE = 2ns
PULSE GENERATOR
tRISE < 1ns
OSCILLOSCOPE
1GHz BANDWIDTH
tRISE = 350ps
TYPICALLY
5V TO 6V, 30ns
WIDE
5Ω
Z0 PROBE
≈1A
DIODE
UNDER
TEST
AN122 F04
Figure 4. Detailed measurement scheme indicates necessary performance parameters for various elements. Subnanosecond rise time pulse
generator, 1A, 2ns rise time amplifier and 1GHz oscilloscope are required.
VIN = 20V
+V
+
LT1086
22µF
+
120Ω
+V TYPICAL 17V
22µF
*
1k
Q1
Q4
1Ω
Q5
1Ω
+V ADJUST (RISE TIME TRIM)
1k
+V
PULSE
INPUT
*
EDGE PURITY
100Ω
MINIMIZE INDUCTANCE IN ALL PATHS
Q2
50Ω
62Ω
2pF TO 12pF
EDGE
PURITY
= 2N3866
OUTPUT
+V
5Ω**
= 2N3375
*
** = TEN PARALLELED 50Ω RESISTORS
* = BYPASS EVERY TRANSISTOR WITH
22µF SANYO OSCON PARALLELED WITH
2.2µF MYLAR
Q3
Q6
1Ω
AN122 F05
Figure 5. Pulse amplifier includes paralleled, darlington driven RF transistor output stage. Collector voltage adjustment
(“rise time trim”) peaks Q4 to Q6 FT, input RC network optimizes output pulse purity. Low inductance layout is mandatory.
maximum allowable forward voltage.
The step-down case limit is set by
the switch pins maximum allowable
reverse voltage.
Figure 2 indicates the diode requires
a inite length of time to clamp at its
forward voltage. This forward turnon time permits transient excursions
above the nominal diode clamp voltage, potentially exceeding the IC’s
breakdown limit. The turn-on time is
typically measured in nanoseconds,
making observation dificult. A further
complication is that the turn-on overshoot occurs at the amplitude extreme
of a pulse waveform, precluding high
resolution amplitude measurement.
These factors must be considered
when designing a diode turn-on test
method.
Linear Technology Magazine • March 2009
Figure 3 shows a conceptual method
for testing diode turn-on time. Here,
the test is performed at 1A although
other currents could be used. A pulse
steps 1A into the diode under test via
the 5Ω resistor. Turn-on time voltage excursion is measured directly
at the diode under test. The igure
1V/DIV
2ns/DIV
AN122 F06
Figure 6. Pulse amplifier output into 5Ω. Rise time is 2ns with minimal pulse-top aberrations.
35
36
50Ω
2pF TO 12pF
EDGE
PURITY
62Ω
+
22µF
Q3
+V
Q2
+V
Q1
+V
LT1086
*
*
*
Q6
Q5
Q4
1k
1k
22µF
+V, TYPICAL 17V
1Ω
1Ω
1Ω
DIODE
UNDER
TEST
Z0 PROBE = TEKTRONIX
P-6056, 500Ω
5Ω**
≈
5.5V
+V ADJUST (RISETIME TRIM)
120Ω
+
Figure 7. Complete diode forward turn-on time measurement arrangement includes subnanosecond rise time
pulse generator, pulse amplifier, Z0 probe and 1GHz oscilloscope.
ADJUST PULSE GENERATOR AMPLITUDE FOR 5.5V AMPLITUDE AT 5Ω RESISTOR
** = TEN PARALLELED 50Ω RESISTORS
* = BYPASS EVERY TRANSISTOR WITH
22µF SANYO OSCON PARALLELED WITH
2.2µF MYLAR
= 2N3375
= 2N3866
MINIMIZE INDUCTANCE IN ALL PATHS
HP-215A
PULSE GENERATOR
tRISE = 800ps
PWIDTH = 30ns
215A
EDGE PURITY
100Ω
≈
6.7V
VIN = 20V
7A29
7B15
TEKTRONIX
7104/7A29/7B10/7B15
1GHz (tRISE = 350ps)
OSCILLOSCOPE
7A29
7B10
7104
AN122 F05
L DESIGN IDEAS
Linear Technology Magazine • March 2009
DESIGN IDEAS L
200mV/DIV
200mV/DIV
AN122 F08
2ns/DIV
2ns/DIV
Figure 9. “Diode Number 2” peaks ≈750mV before settling
in 6ns... > 2x steady state forward voltage.
Figure 8. “Diode Number 1” overshoots steady state
forward voltage for ≈3.6ns, peaking 200mV.
200mV/DIV
AN122 F09
200mV/DIV
AN122 F10
2ns/DIV
Figure 10. “Diode Number 3” peaks 1V above nominal
400mV VFWD, a 2.5x error.
5ns/DIV
AN122 F11
Figure 11. “Diode Number 4” peaks ≈750mV with lengthy
(note horizontal 2.5x scale change) tailing towards VFWD value.
200mV/DIV
5ns/DIV
AN122 F12
Figure 12. “Diode Number 5” peaks offscale with extended tailing (note
horizontal slower scale compared to Figures 8 thru 10).
Linear Technology Magazine • March 2009
37
L DESIGN IDEAS
is deceptively simple in appearance.
In particular, the current step must
have an exceptionally fast, high-idelity
transition and faithful turn-on time
determination requires substantial
measurement bandwidth.
Detailed Measurement
Scheme
A more detailed measurement scheme
appears in Figure 4. Necessary performance parameters for various
elements are called out. A subnanosecond rise time pulse generator, 1A,
2ns rise time ampliier and a 1GHz
oscilloscope are required. These speciications represent realistic operating
conditions; other currents and rise
times can be selected by altering appropriate parameters.
The pulse ampliier necessitates
careful attention to circuit coniguration and layout. Figure 5 shows the
ampliier includes a paralleled, Darlington driven RF transistor output
stage. The collector voltage adjustment
(“rise time trim”) peaks Q4 to Q6 FT;
an input RC network optimizes output
pulse purity by slightly retarding input
pulse rise time to within ampliier
passband. Paralleling allows Q4 to Q6
LTM802, continued from page Diode Testing and
Interpreting Results
The measurement test ixture, properly equipped and constructed,
permits diode turn-on time testing
with excellent time and amplitude
resolution.5 Figures 8 through 12
show results for ive different diodes
from various manufacturers. Figure 8
(Diode Number 1) overshoots steady
state forward voltage for 3.6ns, peaking
200mV. This is the best performance
of the ive. Figures 9 through 12 show
2A, from the maximum input voltage,
36V. There are two traces in the plot,
one for the vertical and horizontal
orientations of the test lab’s receiver
antenna. As shown in the igure, the
LTM8032 easily meets the CISPR 22
class B limits, with 20db of margin for
most of the frequency spectrum, with
either antenna orientation.
90
90
80
80
70
70
60
50
EN55022
CLASS B
LIMIT
40
30
20
10
Conclusion
The LTM8032 switching step-down
regulator is both easy to use and quiet,
meeting the radiated emissions requirements of CISPR22 and EN55022
class B by a wide margin. L
Authors can be contacted
at (408) 432-1900
EN55022
CLASS B
LIMIT
40
30
20
10
0
–10
100 200 300 400 500 600 700 800 900 1000
FREQUENCY (MHz)
Notes
1 An alternate pulse generation approach appears in
Linear Technology Application Note 22, Appendix
F, “Another Way to Do It.”
2 Z0 probes are described in Linear Technology Application Note 22 Appendix C, “About Z0 Probes.”
See also References 27 thru 34.
3 The subnanosecond pulse generator requirement
is not trivial. See Linear Technology Application
Note 22 Appendix B, “Subnanosecond Rise Time
Pulse Generators For The Rich and Poor.”
4 See Linear Linear Technology Application Note
22 Appendix E, “Connections, Cables, Adapters,
Attenuators, Probes and Picoseconds” for relevant
commentary.
5 See Linear Technology Application Note 22 Appendix A, “How Much Bandwidth is Enough?” for
discussion on determining necessary measurement
bandwidth.
50
–10
0
increasing turn-on amplitude and
time which are detailed in the igure
captions. In the worst cases, turn-on
amplitudes exceed nominal clamp
voltage by more than 1V while turn-on
times extend for tens of nanoseconds.
Figure 12 culminates this unfortunate
parade with huge time and amplitude
errors. Such errant excursions can and
will cause IC regulator breakdown and
failure. The lesson here is clear. Diode
turn-on time must be characterized
and measured in any given application
to insure reliability. L
60
0
Figure 3. The baseline measurement of ambient
noise in the 5-meter chamber (no devices operating)
38
EMISSIONS LEVEL (dBµV/m)
EMISSIONS LEVEL (dBµV/m)
amount of ambient noise in the room.
Figure 3 shows the noise spectrum in
the chamber without any devices running. This can be used to determine the
actual noise produced by the DUT.
Figure 4 shows the worst case
LTM8032 emissions plot, which occurs at maximum power out, 10V at
to operate at favorable individual currents, maintaining bandwidth. When
the (mildly interactive) edge purity and
rise time trims are optimized, Figure
6 indicates the ampliier produces a
transcendently clean 2ns rise time
output pulse devoid of ringing, alien
components or post-transition excursions. Such performance makes diode
turn-on time testing practical.1
Figure 7 depicts the complete diode
forward turn-on time measurement arrangement. The pulse ampliier, driven
by a sub-nanosecond pulse generator,
drives the diode under test. A Z0 probe
monitors the measurement point and
feeds a 1GHz oscilloscope.2, 3, 4
0
100 200 300 400 500 600 700 800 900 1000
FREQUENCY (MHz)
Figure 4. The LTM8032 emissions for 20W out, 36VIN
Linear Technology Magazine • March 2009
DESIGN IDEAS L
µModule Regulator Fits a (Nearly)
Complete Buck-Boost Solution
in 15mm × 15mm × 2.8mm for
4.5V–36V VIN to 0.8V–34V VOUT
Linear Technology offers a number of
high eficiency synchronous 4-switch
buck-boost DC/DC converter solutions for applications where VOUT falls
within the range of VIN. The LTM4605,
LTM4607 and LTM4609 µModule
regulators are nearly self-contained
buck-boost solutions that share pincompatible 15mm × 15mm × 2.8mm
packages. The package includes the
controller, four power FETs and a
number of other discrete components.
Only an external inductor, a sensing
resistor, a voltage setting resistor and
a few input and output capacitors are
needed to complete a high eficiency
buck-boost converter.
Table 1 shows the input voltage,
output voltage and current speciications of these three buck-boost
µModule regulators. The LTM4609
is the latest addition to this family.
It satisies the needs of high output
voltage applications with an output
range of 0.8V–34V.
by Judy Sun, Sam Young and Henry Zhang
VIN
10V TO 36V
OPTIONAL
CLOCK SYNC
4.7µF
×2
50V
ON/OFF
VIN
PLLIN V
OUT
FCB
RUN
LTM4609
+
150µF
×2
35V
VOUT
30V
3A
SW1
SW2
RSENSE
L1: SUMIDA CDEP147
SENSE+
10nF
7mΩ
SS
SGND
SENSE–
PGND
VFB
2.74k
Figure 1. Just a few components form a complete 10V to 36V input, 30V/3A output converter
using the LTM4609.
As with all Linear Technology µModule
regulators, the LTM4609 requires only
a few external components to complete
a wide input range buck-boost converter. Figure 1 shows a 10V to 36V
input, 30V output design. The output
current capability is 3A at 10V VIN,
and 8A with 36V input.
Figure 2 shows the eficiency of
this converter, up to 98% in buck
mode and 95% in boost mode. The
low proile LGA package features low
thermal resistance from junction to
pin, thus maintaining an acceptable
junction temperature even at high
output power. The LTM4609’s high
VIN = 36V, VOUT = 30V, IOUT = 8A
VIN = 24V, VOUT = 30V, IOUT = 6A
High Performance with
Minimum Component Count
10µF
×2
35V
L1
3.4µH
100
95
EFFICIENCY (%)
Introduction
36VIN
24VIN
10VIN
90
85
CONTINUOUS CURRENT MODE
VOUT = 30V
fSW = 275kHz
80
0
2
4
6
LOAD CURRENT (A)
8
10
Figure 2. Efficiency of the 30V
buck-boost converter
VIN = 12V, VOUT = 30V, IOUT = 3A
Figure 3. Thermal-graph taken with the LTM4609 running at different input voltages. The LTM4609 is on the left, the inductor (Sumida CDEP147)
is on the right. No heat sink or forced air flow. Ambient temperature = 25°C.
Linear Technology Magazine • March 2009
39
L DESIGN IDEAS
eficiency combined with its excellent
thermal management capability enables it to deliver up to 240W output
power without a heat sink or forced
airlow. Figure 3 shows the thermalgraphs taken with three different input
voltages and loads at 25°C ambient
temperature. With 240W output and
36V input, the maximum temperature
rise of the LTM4609 is only 52.8°C.
L1
L1,L2: FAIR-RITE 2518065007Y6
L2
VIN
CBULK
100µF
VIN
+
CIN1
10µF
CIN2
10µF
LTM4609
GND
Figure 4. The LTM4609 µModule regulator with an input π filter.
Input Ripple Reduction
One way to improve eficiency in a
switching DC/DC converter is to minimize the turn-on and turn-off times
of the MOSFET—shorter transitions
correspond to lower switch losses.
However, fast transitions also lead
to high frequency switching noise,
which can pollute the input power
source. For the applications where the
input voltage ripple must be limited,
a simple LC π ilter can be inserted at
the input side to attenuate the high
frequency input voltage noise. Figure
4 shows the LTM4609 with an input
π ilter. The ilter includes two 10µF
low ESR ceramic capacitors and two
very small magnetic beads. For lower
output power applications, only one
magnetic bead is necessary.
Figure 5 shows the input ripple
reduction with the π ilter. Figure 5a
shows the input ripple with 100µF
aluminum electrolytic plus 2 × 4.7µF
VIN
200mV/DIV
VIN
200mV/DIV
VIN = 10V
VOUT = 30V
IOUT = 3A
10µs/DIV
CBULK = 100µF
CIN1, CIN2 = 4.7µF
5a. Input voltage waveform without
the input π filter shown in Figure 4
5b. Input voltage waveform with
input π filter as shown in Figure 4
Figure 5. The input π filter shown in Figure 4 effectively reduces
the input voltage spike caused by switching action of the MOSFETs.
ceramic input capacitors. Figure 5b
shows the input ripple with the ilter
shown in Figure 4. Both waveforms
are measured across the 100µF aluminum capacitor. A 67% reduction in
input ripple is obtained with the input
π ilter, which requires only two small
additional magnetic beads.
Table 1. Specification comparison of the LTM4605, LTM4607 and LTM4609
LTM4605
LTM4607
LTM4609
VIN
4.5V ~ 20V
4.5V ~ 36V
4.5V ~ 36V
VOUT
0.8V ~ 16V
0.8V ~ 24V
0.8V ~ 34V
IOUT
5A
(12A in buck mode)
5A
(10A in buck mode)
4A
(10A in buck mode)
Package
10µs/DIV
VIN = 10V
CBULK = 100µF
VOUT = 30V
CIN1, CIN2 = 10µF
IOUT = 3A
L1, L2: FAIR-RITE 2518065007Y6
15mm × 15mm × 2.8mm LGA
Conclusion
Buck-boost µModule regulators
are easy-to-use, high performance
solutions for applications where a
regulated output voltage sits within the
range of the input voltage. The 15mm
× 15mm × 2.8mm LTM4609 widens the
input/output voltage range of the pin
compatible LTM4605 and LTM4607.
The advanced package technology, as
well as the high eficiency design of
the LTM4609, allows it to deliver up
to 240W of output power without heat
sinks or forced airlow. For applications that require low input voltage
ripple, a simple π ilter can be added
by inserting one or two small magnetic
beads to signiicantly reduce the high
frequency input noise. L
LT755/56, continued from page 2
PWM on- and off-times are 1µs as
with the other circuits. Figure 7 shows
the waveforms during a short circuit
fault on the output. The input current remains in control as the switch
current ramps up to the set limit of
10A, then skips the next few cycles
while the current sensed by the LED
40
resistor ramps down to 1.5A. This
faulted mode of circuit operation can
continue indeinitely without damage
to the components.
Conclusion
The LT3755 and LT3756 offer unparalleled performance for an LED
controller generating PWM pulse
widths as narrow as 1µs, which
enables 50:1 PWM dimming at frequencies above the audible range.
Other features include open LED
protection, an open LED status indicator, and programmability of the LED
current via an analog input. L
Linear Technology Magazine • March 2009
NEW DEVICE CAMEOS L
New Device Cameos
Micropower Low Noise Boost
Converter with Output
Disconnect
Dual 550mA, 1MHz
Synchronous Boost Regulator
with Output Disconnect in a
The LT3495/LT3495B/LT3495-1/ 3mm × 3mm DFN
LT3495B-1 is a low noise boost converter with integrated power switch,
feedback resistor and output disconnect circuitry. The part controls power
delivery by varying both the peak
inductor current and switch off-time.
This new control scheme results in low
output voltage ripple as well as high
eficiency over a wide load range.
For the LT3495/LT3495-1, the
off-time of the switch is not allowed
to exceed a ixed level, guaranteeing
the switching frequency stays above
the audio band for the entire load
range. This feature is disabled for the
LT3495B/LT3495B-1, which leads to
higher eficiency at light load. The difference between the LT3495/LT3495B
and LT3495-1/LT3495B-1 is the
level of the switch current limit. The
LT3495/LT3495B has a typical peak
current limit of 650mA while the
LT3495-1/LT3495B-1 has a typical
peak current limit of 350mA.
The LT3495 series has an output
disconnect PMOS that blocks the
load from the input during shutdown.
During normal operation, the maximum current through this PMOS is
limited by circuitry inside the chip,
which helps the chip survive output
shorts.
The input voltage range of the
LT3495 series is a wide 2.5V to 16V
and the output voltage can be up
to 40V. In addition, the part is well
compensated internally, and can be
stable with very small ceramic output
capacitors.
Other features include low quiescent current (60µA in active mode and
0.1µA in shutdown mode), integrated
output dimming, maximum switching
on-time and undervoltage lockout.
Combining the small ceramic capacitors and space saving 10-pin 3mm ×
2mm DFN packages, the LT3495
enables compact solutions for many
applications.
Linear Technology Magazine • March 2009
The LTC3535 is a dual-channel 1MHz,
current mode synchronous boost
DC/DC converter with integrated
output disconnect and soft-start. The
LTC3535’s internal 550mA switches
deliver output voltages as high as
5.25V from an input voltage range of
0.7V at start-up/0.5V when running
to 5V, making it ideal for single- or
multicell alkaline/NiMH as well as Liion/polymer applications. Each of the
LTC3535’s channels has it own power
input and is completely independent,
offering maximum design lexibility.
For example, one channel can deliver
up to 50mA of continuous output current at 3.3V while the other channel
delivers up to 100mA at 1.8V to power
a microcontroller from a single alkaline
cell. The 1MHz switching frequency
minimizes external component sizes
while providing up to 94% eficiency.
Combined with a compact 3mm × 3mm
DFN-12 package, the LTC3535 dual
channel boost provides the tiny and
eficient solution footprint required in
handheld applications.
Burst Mode ® operation lowers
quiescent current to only 18µA (both
channels), providing extended battery
run time in handheld applications. The
LTC3535 is an ideal part for handheld
dual boost applications where small
solution size and maximum battery
run time are deining factors.
Ideal Diode Equipped High
Power Battery Charger
Handles All Chemistries
The LTC4012, LTC4012-1, LTC4012-2
and LTC4012-3 are a family of high
power buck battery chargers all in a
20-lead 4mm × 4mm QFN package.
Compared to the LTC4009 family of
chargers, the 4012 family adds Ideal
Diode™ input reverse current input
protection and extends the high
eficiency to higher current levels.
Combined with just a few external
components and external termination
control, the LTC4012 family facilitates
construction of chargers capable of
delivering up to 4A to batteries with
output power levels approaching 66W
in a very small footprint.
The LTC4012 family builds upon
the proven quasi-constant frequency,
constant off-time PWM buck control architecture as found in Linear
Technology’s LTC4008. This unique
buck topology provides continuous
switching with synchronous rectiication even with no load current,
critical to preventing audible noise
in constant voltage charge termination applications. However, LTC4012
family uses switching NFETs along
with an adaptive gate drive to avoid
overlap conduction losses. The higher
550kHz switching frequency reduces
both the inductor size and output
capacitance requirements while offering eficiencies up to 95% or more.
If the duty cycle goes below 20% or
above 80%, the LTC4012 lowers the
switching frequency to avoid pulse
skipping that might otherwise begin to
occur at 550kHz. Under low dropout
conditions requiring high duty cycle
operation, the internal watchdog timer
prevents the LTC4012 from switching
below 25kHz, achieving a maximum
duty cycle of 98% without producing
audible noise. There is also an input
current monitor function that prevents
input power overload when the input
power is shared with a load.
There are four versions of the
LTC4012. The LTC4012 and LTC40123 offer a user programmable voltage set
point using an external resistor divider
allowing for multi-chemistry support.
The Li-ion optimized LTC4012-1 and
LTC4012-2 support one to four series
cells via pin selection. The LTC40121 provides 4.1V/cell charging while
the LTC4012-2 produces 4.2V/cell.
Output voltage accuracy is typically
±0.5% and a maximum of ±0.8% over
temperature. These ICs contain a
switch that in shutdown removes voltage divider current drained from the
41
L NEW DEVICE CAMEOS
battery whether external or internal.
Programming the charge current only
requires a single external resistor.
The fault management system of the
LTC4012 family suspends charging
immediately for various conditions.
First is battery overvoltage protection,
which can occur with the sudden loss
of battery load during bulk charge.
Second, each IC features internal
over-temperature protection to prevent silicon damage during elevated
thermal operation.
The LTC4012 family has a logic-level
shutdown control input and three
open-drain status outputs. First is an
input current limit (ICL) status lag to
tell the system when VIN is running at
over 95% of its current capacity. The
input current limit accuracy is typically ±3% and a maximum of ±4% over
the full operating temperature range.
Next is the AC present status, which
indicates when VIN is within a valid
range for charging under all modes of
operation. The last is a charge status
output can indicate bulk or C/10
charge states. The control input and
status outputs of the LTC4012, along
with the analog current monitor output, can be used by the host system
to perform necessary preconditioning,
charge termination and safety timing
functions.
4MHz Synchronous StepDown DC/DC Converter
Delivers up to 1.25A from a
3mm × 3mm DFN
The LTC3565 is a high eficiency synchronous step-down regulator that
can deliver up to 1.25A of continuous
output current from a 3mm × 3mm
DFN (or MSOP-10E) package. Using
a constant frequency of (up to 4MHz)
and current mode architecture, the
LTC3565 operates from an input voltage range of 2.5V to 5.5V making it
ideal for single cell Li-Ion, or multicell
Alkaline/NiCad/NiMH applications.
It can generate output voltages as
low as 0.6V, enabling it to power the
latest generation of low voltage DSPs
and microcontrollers. An independent
RUN pin enables simple turn-on and
shutdown. Its switching frequency
is user programmable from 400kHz
to 4MHz, enabling the designer to
optimize eficiency while avoiding critical noise-sensitive frequency bands.
The combination of its 3mm × 3mm
DFN-10 (or MSOP-10) package and
high switching frequency keeps external inductors and capacitors small,
providing a very compact, thermally
eficient footprint.
The LTC3565 uses internal switches
with an RDS(ON) of only 0.13Ω (N-Channel lower FET) and 0.15Ω (P-Channel
upper FET) to deliver eficiencies
as high as 95%. It also utilizes low
dropout 100% duty cycle operation
to allow output voltages equal to VIN,
further extending battery run time.
The LTC3565 utilizes Automatic Low
Ripple ( < 25mVP–P) Burst Mode®
operation to offer only 40µA no load
quiescent current. If the application is
noise sensitive, Burst Mode operation
can be disabled using a lower noise
pulse-skipping mode, which still offers
only 330µA of quiescent current. The
LTC3565 can be synchronized to an
external clock throughout its entire
frequency range. Other features include ±2% output voltage accuracy and
over-temperature protection. L
LT75, continued from page Conclusion
The ability to run from any input
supply voltage ranging from 4.75V
to greater than 400V and the abundance of safety features make the
LT3751 an excellent choice for high
voltage capacitor chargers or high
voltage regulated power supplies. In
fact, the LT3751 is, for now, the only
42
100
0.5
OUTPUT VOLTAGE ERROR (V)
95
EFFICIENCY (%)
LT3751 controller, and the optocoupler
on the feedback resistor divider. The
auxiliary windings provide the desired
galvanic isolation boundary while
maintaining an isolated feedback path
from the output node to the LT3751
FB pin. Figures 12 and 13 show the
regulator’s performance.
The fully isolated, high voltage input/output regulator yields over 90%
eficiency. Load regulation is excellent
as shown in Figure 13b, due mainly
to the added gain of the optocoupler
circuit.
90
85
80
POUT = 63W
POUT = 48W
POUT = 25W
75
70
100
120
140
160
180
200
INPUT DC VOLTAGE (V)
a. Efficiency
0.25
0
–0.25
–0.5
0
50
100
150
200
250
IOUT (mA)
b. Load regulation
Figure 13. Fully isolated, high voltage regulator performance
boundary-mode capacitor charger
controller that can accurately operate
from extremely high input voltages.
The LT3751 simpliies design by integrating many functions that—due
to cost and board real-estate—would
otherwise not be realizable. Although
several designs are shown here, the
LT3751 includes many more features
than we can show in one article. We
recommended consulting the data
sheet or calling the Linear Technology
applications engineering department
for more in-depth coverage of all available features. L
Linear Technology Magazine • March 2009
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Linear Technology Magazine • March 2009