Combined Voltage and Current Control Loops Simplify LED Drivers, High Capacity Battery/Supercap Chargers & MPPT* Solar Applications

July 2012
I N
T H I S
I S S U E
pushbutton controller 10
15V, 2.5A monolithic buckboost with 95% efficiency
and low noise 17
easy 2-supply current
sharing 20
Volume 22 Number 2
Combined Voltage and Current
Control Loops Simplify LED Drivers,
High Capacity Battery/Supercap
Chargers & MPPT* Solar Applications
Xin (Shin) Qi
* Maximum Power Point Tracking
sub-1mm height 24V, 15A
monolithic regulator 26
deliver 25A at 12V from
inputs to 60V 28
The rapid expansion of constant-current/constant-voltage (CC-CV)
applications, especially in LED lighting and high capacity battery
and supercapacitor chargers challenges power supply designers to
keep pace with the increasingly complicated interplay of current and
voltage control loops. A switch-mode converter designed specifically
for CC-CV offers a clear advantage,
especially when the supply has limited
power, or its power is allocated
among several competing loads.
The LTC4155 is a monolithic switching battery charger that efficiently delivers
3.5A charge current in a compact PCB footprint. See page 13.
Caption
w w w. li n ea r.com
Consider, for instance, the challenge of charging a
supercapacitor in a minimum amount of time from
a power-limited supply. To maintain constant input
power, the controlled charging current must decrease
as the output (supercapacitor) voltage increases. The
LT®3796 solves the problem of power limited or constant current/constant voltage regulation by seamlessly
combining a current regulation loop and two voltage regulation loops to control an external N-channel
power switch. The inherent wired-OR behavior of its
three transconductance error amplifiers summed into
the compensation pin, VC , ensures that the correct loop
(that is, the one closest to regulation) dominates.
(continued on page 4)
The LT3796’s wide VIN range (6V to 100V) and rail-to-rail
(0V to 100V) output current monitoring and regulation allow
it to be used in a wide variety of applications from solar
battery chargers to high power LED lighting systems.
(LT3796, continued from page 1)
HIGH POWER LED DRIVER WITH
ROBUST OUTPUT SHORT CIRCUIT
PROTECTION
The additional, standalone current sense
amplifier can be configured for any
number of functions, including input current limit and input voltage regulation.
Figure 1 shows the LT3796 configured as a
boost converter to drive a 34W LED string
from a wide input range. The LED current is derated at low input voltages
to prevent external power components
from overheating. The front-end current
sense amplifier monitors the input current by converting the input current to
a voltage signal at the CSOUT pin with
The LT3796’s wide VIN range (6V to
100V) and rail-to-rail (0V to 100V) output current monitoring and regulation
allow it to be used in a wide variety of
applications from solar battery chargers to high power LED lighting systems. The fixed switching frequency,
current-mode architecture results in
stable operation over a wide range of
supply and output voltages. The LT3796
incorporates a high side current sense,
enabling its use in boost, buck, buckboost or SEPIC and flyback topologies.
Figure 1. A 34W LED driver with robust
output short-circuit protection.
VCSOUT = IIN • RSNS1 •
The LT3796 includes short-circuit protection independent of the LED current sense. The short-circuit protection
feature prevents the development of
excessive switching currents and protects the power components. The
protection threshold (375mV, typ) is
designed to be 50% higher than the
default LED current sense threshold.
R6
R5
The resistor network at the FB1 pin
provides OPENLED protection, which
limits the output voltage and prevents
the ISP pin, ISN pin and several external
VIN
9V TO 60V
100V (TRANSIENT)
R1
1M
R2
118k
R3
499k
OPTIONAL INPUT
CURRENT REPORTING
R4
97.6k
VS
CSN
CSP
EN/UVLO
GATE
R6
40.2k
PWM
SYNC
LED CURRENT REPORTING
INTVCC
R10
100k
R9
100k
FAULT
VMODE
RSNS
15mΩ
FB1
LT3796
SYNC
ISP
ISMON
Q1
M2
TG
FAULT
INTVCC
VMODE
R11 402k (OPT)
RLED
620mΩ
ISN
C4
0.1µF
R11 OPTIONAL
FOR FAULT LATCHOFF
UP TO
400mA
GND
PWM
VREF
M1: INFINEON BCS160N10NS3-G
M2: VISHAY SILICONIX Si7113DN
L1: COILTRONICS DR127-220
D1: DIODES INC PDS5100
D2: VISHAY ES1C
Q1: ZETEX FMMT589
LED: CREE XLAMP XR-E
R8
13.7k
M1
SENSE
CSOUT
C3
10nF
C2
2.2µF
×4
100V
R7
1M
CTRL
CSOUT
D1
R5
2k
VIN
circuits.linear.com/558
L1 22µH
RSNS1 50mΩ
IIN
C1
2.2µF
×3
LTspice IV
4 | July 2012 : LT Journal of Analog Innovation
components from exceeding their maximum rating. If an LED fails open or if
the LED string is removed from the high
power driver, the FB constant voltage
loop takes over and regulates the output to 92.5V. The VMODE flag is also
asserted to indicate an OPENLED event.
SS
FB2
RC
10k
C6
0.1µF
CC
10nF
D2
C5
4.7µF
RT
VC
INTVCC
85V LED
RT
31.6k
250kHz
design features
The LT3796 solves the problem of power limited,
or constant-current/constant-voltage regulation
by seamlessly combining a current regulation
loop and two voltage regulation loops to control
an external N-channel power switch.
SS
2V/DIV
SS
2V/DIV
LED+
50V/DIV
LED+
50V/DIV
FAULT
10V/DIV
FAULT
10V/DIV
IM2
1A/DIV
IM2
1A/DIV
If there is no resistor between the SS pin
and VREF pin, the converter enters hiccup
mode and periodically retries as shown
in the Figure 2. If a resistor is placed
between VREF and SS pin to hold SS pin
higher than 0.2V during LED short, then
the LT3796 enters latchoff mode with
GATE pin low and TG pin high, as shown
in Figure 3. To exit latchoff mode, the
EN/UVLO pin must be toggled low to high.
5ms/DIV
5ms/DIV
Figure 2. Short LED protection: hiccup mode
(without R11 in Figure 1)
Figure 3. Short LED protection: latchoff mode
(with R11 in Figure 1)
LED DRIVER WITH HIGH PWM
DIMMING RATIO
Once the LED overcurrent is detected, the
GATE pin drives to GND to stop switching, the TG pin is pulled high to disconnect the LED array from the power
path and the FAULT pin is asserted. The
Schottky diode D2 is added to protect
the drain of PMOS M2 from swinging well below ground when shorting
to ground through a long cable. The
PNP helper Q1 is included to further limit
the transient short-circuit current.
Using an input referred LED string
allows the LT3796 to act as a buck
mode controller as shown in Figure 4.
The 1MHz operating frequency enables
a high PWM dimming ratio. The
OPENLED regulation voltage is set to
1.25V •
VIN
16V TO 36V
RLED 250mΩ
1A
Figure 4. A buck
mode LED driver
with 3000:1 PWM
dimming ratio
R3
100k
R1
1M
VIN
R2
100k
VS
VREF
CSN
LT3796
R4
100k
8V
LED
TG
CSP
PWM
PWM
ISN
ISP
EN/UVLO
CTRL
CSOUT
FB1
LED CURRENT REPORTING
C5
0.1µF
R6
124k
L1
10µH
SENSE
INTVCC
RSNS
33mΩ
R9
100k
FAULT
VMODE
GND
FAULT
VMODE
VC
M1: VISHAY SILICONIX Si73430DV
M2: VISHAY SILICONIX Si7113DN
D1: ZETEX ZLLS2000TA
L1: WÜRTH 744066100
LED: CREE XLAMP XM-L
M1
GATE
ISMON
INTVCC
SYNC
RT
RC
10k
CC
4.7nF
RT
6.65k
1MHz
SS
C4
0.1µF
C2
10µF
×4
25V
R5 1M
FB2
R8
100k
through the independent current sense
amplifier at CSP, CSN and CSOUT pins.
During the PWM off phase, the LT3796
disables all internal loads to the VC pin
LED+
M2
C3
4.7µF
R3  R5 
•  + 1
R6  R4 
Figure 5. 3000:1 PWM dimming ratio of the circuit in
Figure 4 at VIN = 24V and PWM = 100Hz
D1
C1
2.2µF
×2
50V
PWM
5V/DIV
IL
1A/DIV
INTVCC
IL
1A/DIV
1µs/DIV
July 2012 : LT Journal of Analog Innovation | 5
Voltage drops in wiring and cables can cause load regulation errors. These errors can
be corrected by adding remote sensing wires, but adding wires is not an option in
some applications. As an alternative, the LT3796 can adjust for wiring drops, regardless
of load current, provided that the parasitic wiring or cable impedance is known.
OUT
•
1:1
C1
10µF
M1
C3
10µF
L1B
+
C4
100µF
25V
RWIRE
VLOAD
12V, 1A CURRENT LIMIT
RSNS
33mΩ
VIN
C8
0.1µF
VREF
GATE
SENSE
GND
EN/UVLO
ISP
ISMON
ISN
PWM
LT3796
SYNC
C5
0.1µF
R7
100k
SS
VS
CTRL
FAULT
FAULT
R4
287k
FB1
VMODE
TG
VC
RT
19.6k
400kHz
and preserves the charge state. It also
turns off the PMOS switch M2 to disconnect the LED string from the power path
and prevent the output capacitor from
discharging. These features combine to
greatly improve the LED current recovery
time when PWM signal goes high. Even
with a 100Hz PWM input signal, this buck
mode LED driver can achieve a 3000:1
dimming ratio as illustrated in Figure 5.
OUT
CSOUT
RT
L1: WÜRTH 744871220
D1: ZETEX ZLLS2000TA
M1: VISHAY SILICONIX Si4840DY
C7
1µF
CSP
R6
100k
VMODE
R2
38.3k
R3
154k
VREF
INTVCC
R1
38.3k
CSN
FB2
INTVCC
RC
24.9k
CC
10nF
R5
12.4k
INTVCC
C6
4.7µF
SEPIC CONVERTER WITH R WIRE
COMPENSATION
Voltage drops in wiring and cables can
cause load regulation errors. These errors
can be corrected by adding remote sensing
wires, but adding wires is not an option in
some applications. As an alternative, the
LT3796 can adjust for wiring drops, regardless of load current, provided that the parasitic wiring or cable impedance is known.
Figure 6 shows a 12V SEPIC converter that
uses the RWIRE compensation feature.
RSNS1 is selected to have 1A load current
6 | July 2012 : LT Journal of Analog Innovation
RSNS1
250mΩ
D1
•
Figure 6. This SEPIC converter
compensates for voltage drops
in the wire between the controller
and the load (RWIRE)
C2
10µF
L1A 22µH
VIN
12V
limit controlled by the ISP, ISN pins. The
resistor network R1–R5, along with the
LT3796’s integrated current sense amplifier
(CSAMP in Figure 7), adjusts the OUT node
voltage (VOUT) to account for voltage
drops with respect to the load current.
This ensures that VLOAD remains constant at 12V throughout the load range.
Figure 7 shows how the LT3796’s internal
CSAMP circuit plays into the operation. The
LT3796’s voltage loop regulates the FB1 pin
at 1.25V so that I3 stays fixed at 100µ A for
R5 = 12.4k. In Figure 7, VOUT changes
design features
The LT3796 in a 28-lead TSSOP package performs
tasks that would otherwise require a number of control
ICs and systems. It offers a reliable power system with
simplicity, reduced cost and small solution size.
ILOAD
R3 2 • (R WIRE )
=
R1
R SNS1
RWIRE
CSN
CSP
–
I1
CSAMP
R4
RWIRE = 0.5Ω
VOUT
800mA
12.5
VLOAD
LT3796
+
R3
12V
R2
R1
13.0
I2
VOUT/VLOAD (V)
VOUT
RSNS1
IOUT
500mA/DIV
12.0
11.5
11.0
VOUT
500mV/DIV
(AC-COUPLED)
10.5
10.0
CSOUT
FB1 = 1.25V
R5
200mA
0
200
400
600
800
1000
1200
500µs/DIV
ILOAD (mA)
I3
Figure 7. RWIRE voltage drops are compensated for
via the LT3796’s CSAMP circuit
Figure 8. Measured VLOAD and VOUT with respect to
ILOAD
Figure 9. Load step response of the circuit in
Figure 6
with current I2 as VOUT = 1.25V + I2 • R4.
If the change of I2 • R4 can offset
the change of ILOAD • (RSNS1 + RWIRE),
then VLOAD will stay constant.
output current is what gives VOUT the
positive load regulation characteristic.
The positive load regulation is just what is
needed to compensate for the cable drop.
SOLAR PANEL BATTERY CHARGER
Referring to Figure 7, the divider
R1/R3 from VOUT sets the voltage
regulated at CSP by the current I1 flowing in R2. I1 is conveyed to the FB1
node where it sums with I2 .
The measured VLOAD and VOUT with respect
to ILOAD are shown in Figure 8. Clearly,
VLOAD is independent of ILOAD when
ILOAD is less than the 1A current limit.
When ILOAD approaches 1A, the current
loop at ISP and ISN pins begins to interfere
with the voltage loop and drags the output
voltage down correspondingly. The load
transient response is shown in Figure 9.
As the output current increases, I1
decreases due to the increasing voltage
drop across RSNS ; its decrease must be
compensated by a matching increase in
the current I2 to maintain the constant
100µ A into FB2. This increase in I2 with
Solar powered devices rely on a highly
variable energy source, so for a device to
be useful at all times, energy from solar
cells must be stored in a rechargeable
battery. Solar panels have a maximum
power point, a relatively fixed voltage at
which the panel can produce the most
power. Maximum power point tracking
(MPPT) is usually achieved by limiting a
converter’s output current to keep the
panel voltage from straying from this
value. The LT3796’s unique combination
of current and voltage loops make it an
ideal MPPT battery charger solution.
July 2012 : LT Journal of Analog Innovation | 7
WÜRTH SOLAR PANEL
VOC = 37.5V
VMPP = 28V
C6
2.2µF 100V
OUT
BAT
RSNS1
250mΩ
•
D1
D2
15V
1:1
C1
4.7µF
50V
•
VIN
L1A 33µH
M2
R4
301k
INTVCC
R1
10k
C2
10µF
L1B
R10
30.1k
R9
10k
NTC
VCHARGE = 14.6V
VFLOAT = 13.5V
AT 25°C
+
BAT
R5
137k
R2
475k
VIN
VS
CSN
CSP
M1
GATE
EN/UVLO
R3
20k
SENSE
RSNS
15mΩ
CTRL
CSOUT
C3
0.1µF
GND
CSOUT
R6
100k
R11
93.1k
FB1
FB2
PWM
LT3796
VREF
ISP
OUT
ISN
BAT
R8
113k
M3
ISMON
C6
0.1µF
SS
C4
0.1µF
VMODE
SYNC
TG
M1: VISHAY SILICONIX Si7456DP
M2: VISHAY SUD19P06-60-E3
M3: ZETEX ZXM61N03F
L1: COILCRAFT MSD1260-333
D1: ON SEMI MBRS260T3G
D2: CENTRAL SEMI CMDZ15L
R9: MURATA NCP18XH103F03RB
R12
10.2k
INTVCC
VC
RT
RC
499Ω
CC
22nF
FAULT
RT
19.6k
400kHz
VMODE
R7
49.9k
C5
4.7µF
INTVCC
R13
49.9k
FAULT
Figure 10. A solar panel battery charger maximum power point tracking (MPPT)
Figure 10 shows a solar panel to sealed
lead acid (SLA) battery charger driven by
the LT3796. The charger uses a three-stage
charging scheme. The first stage is a constant current charge. Once the battery is
charged up to 14.35V, the charging current
The charging current is programmed
by the resistor network at the CSP and
CSOUT (CTRL) pins as follows,
1.2
1.0
V −V

V
VCTRL = R6 •  IN INTVCC − INTVCC  ,

R4
R5 
 R4 
FOR VIN ≥ VINTVCC 1+ 
 R5 
VCTRL = 0 V,
ICHARGE (A)
0.8
0.6
0.4
 R4 
FOR VIN < VINTVCC 1+ 
 R5 
0.2
0
begins to decrease. Finally, when the
required battery charge current falls below
100m A, the built-in C/10 termination
disables the charge circuit by pulling down
VMODE, and the charger enters float charge
stage with VFLOAT = 13.5V to compensate
for the loss caused by self-discharge.
20
25
30
35
VIN (V)
Figure 11. ICHARGE vs VIN for the solar
charger in Figure 10
8 | July 2012 : LT Journal of Analog Innovation
40
Maximum power point tracking is
implemented by controlling the maximum output charge current. Charge
current is reduced as the voltage on
the solar panel output falls toward
28V, which corresponds to 1.1V on the
CTRL pin and full charging current, as
shown in Figure 11. This servo loop thus
acts to dynamically reduce the power
requirements of the charger system to
the maximum power that the panel
can provide, maintaining solar panel
power utilization close to 100%.
SUPERCAPACITOR CHARGER WITH
INPUT CURRENT LIMIT
Supercapacitors are rapidly replacing
batteries in a number of applications
from rapid-charge power cells for cordless tools to short term backup systems
for microprocessors. Supercapacitors are
longer lasting, greener, higher performance
and less expensive over the long run, but
charging supercapacitors requires precise
control of charging current and voltage
design features
RSNS1 150mΩ
1.33A MAX
VS
L1B
CSP
CSN
EN/UVLO
OUTPUT CURRENT REPORTING
GATE
ISMON
PWM
VREF
INPUT CURRENT
REPORTING AND LIMIT
CSOUT
SYNC
R2
124k
C4
0.1µF
VOUT = 0V TO 28V
D1
R1
20k
VIN
C3
0.1µF
C6
10µF
•
C1
10µF
C7
0.1µF
L1A 33µH
•
VIN
28V
C2
4.7µF
×2
50V
R8
536k
R9
24.9k
M1
SENSE
RSNS
33mΩ
LT3796
1.67A
MAX
GND
CSOUT
FB1
FB2
ISP
RSNS2
150mΩ
SS
ISN
SUPERCAP
TG
INTVCC
R7
100k
R6
100k
INTVCC
FAULT
FAULT
CHGDONE
C6
4.7µF
VMODE
CTRL VREF
RT
VC
RC
499Ω
VOUT
L1: COILCRAFT MSD1260-333
D1: ON SEMI MBRS260T3G
M1: VISHAY SILICONIX Si7850
Q1: ZETEX FMMT591A
R3
499k
C5
0.1µF
R10
499k
R5
1M
R4
30.1k
CC
22nF
Q1
RT
19.6k
400kHz
Figure 12. A 28V/1.67A supercapacitor charger with input current limit
regulation until the input current moves
close to the 1.33A input current limit.
Some applications require that the
input current is limited to prevent the
input supply from crashing. Figure 12
shows a 1.67A supercapacitor charger
with 28V regulated output voltage and
1.33A input current limit. The input
current is sensed by RSNS1, converted
to a voltage signal and fed to the FB2
pin to provide input current limit.
CONCLUSION
In each charging cycle, the supercapacitor is charged from 0V. The feedback loop
from VOUT to the RT pin through R3, C5, R5,
R10, R4, and Q1 to RT works as frequency
foldback to keep regulation under control.
In Figure 13, the input current and output
charging current are plotted against
output voltage for this charger, showing
the LT3796 maintaining the output current
The LT3796 is a versatile step-up
DC/DC controller that combines accurate
current and voltage regulation loops. Its
unique combination of a single current
loop and two voltage loops makes it easy
to solve the problems posed by applications that require multiple control loops,
such as LED drivers, battery or supercapacitor chargers, MPPT solar battery
chargers, and step-up or SEPIC converters
with input and output current limit. It
also includes a number of fault protection and reporting functions, a top gate
driver and current loop reporting.
The LT3796 in a 28-lead TSSOP package performs tasks that would other­
wise require a number of control ICs
and systems. It offers a reliable power
system with simplicity, reduced
cost and small solution size. n
1800
IOUT
1600
INPUT/OUTPUT CURRENT (mA)
limiting to prevent any system-wide damage or damage to the supercapacitor.
1400
1200
IIN
1000
800
600
400
200
0
0
5
10
15
20
25
30
VOUT (V)
Figure 13. Input/output current vs output voltage for
28V/1.67A supercapacitor charger in Figure 12
July 2012 : LT Journal of Analog Innovation | 9
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