January 2013 I N T H I S I S S U E 2.7V to 40V monolithic buck-boost regulates through automotive cold-crank and load-dump transients 9 precision monolithic Volume 22 Number 4 High Accuracy (±1°C) Temperature Sensors Improve System Performance and Reliability Christoph Schwoerer and Gerd Trampitsch op amp works from ±4.75V to ±70V 14 bus buffers simplify design of large, noisy I2C systems 17 The march toward increasingly dense computing power has amplified the challenges related to heat. In many systems, the capabilities of the cooling system are a significant limitation to overall performance. Standard cooling components—bulky heat sinks and power-hungry noisy fans (or expensive quiet ones)—impose size limitations on tightly packed electronics. The only way to maximize performance, minimize cooling requirements, and ensure the health of the electronics is with accurate, precise and comprehensive temperature monitoring throughout the system. With this in mind, Linear Technology has developed a family of highly accurate temperature monitors that can be easily distributed throughout a system. Included in this family: •The LTC®2997 accurately measures either its own temperature or the temperature of an external diode. •The LTC2996 adds monitoring functionality by comparing the measured temperature with a high and a low temperature threshold and communicating any temperature excess via open drain alert outputs. •The LTC2995 combines the LTC2996 with a dual supply voltage monitor, allowing it to measure temperature, compare temperature to configurable thresholds, and supervise two supply voltages. (continued on page 2) Analog CaptionCircuit Design, Volume 2 now available. See page 3. w w w. li n ea r.com In this issue... COVER STORY (LTC299x continued from page 1) High Accuracy (±1°C) Temperature Sensors Improve System Performance and Reliability Christoph Schwoerer and Gerd Trampitsch 1 The LTC2997 in a 2mm × 3mm 6-pin DFN package is perfectly suited to measure temperature of an FPGA or microprocessor as shown in Figure 1. DESIGN FEATURES 2.7V to 40V Monolithic Buck-Boost DC/DC Expands Input Capabilities, Regulates Seamlessly through Automotive Cold-Crank and Load-Dump Transients John Canfield 9 Monolithic Operational Amplifier Works from ±4.75V to ±70V and Features Rail-to-Rail Output Swing and Low Input Bias Current Michael B. Anderson 14 Bus Buffers Simplify Design of Large, Noisy I2C Systems Rajesh Venugopal 17 Ideal Diode and Hot Swap™ Controller Enables Supply Redundancy and Isolates Faults Chew Lye Huat 24 DESIGN IDEAS What’s New with LTspice IV? Gabino Alonso 30 20V, 2.5A Synchronous Monolithic Buck with Current and Temperature Monitoring K. Bassett THE LTC2997 IS A TINY HIGH PRECISION TEMPERATURE SENSOR To this end, the LTC2997 sends measurement currents to the temperature monitoring diode of the FPGA or microprocessor and generates a voltage proportional to the temperature of the diode on its VPTAT output. LTC2997 also provides a 1.8V reference voltage at the VREF output, which can be used as reference voltage for the onboard ADC in the FPGA or microprocessor. The measurement error in this configuration with external sensor element is guaranteed to ±1°C over the wide temperature range from 0°C to 100°C and to ±1.5°C from –40°C to 125°C; typical temperature measurement error is far better, as shown in Figure 2. Tying the D+ pin to VCC configures the LTC2997 to use its own internal temperature sensor. The VPTAT voltage has a slope of 4mV/K and is updated every 3.5ms. OPERATING PRINCIPLES The LTC2997 achieves impressive accuracy by measuring the diode voltage at multiple test currents and using the measurements to remove any process-dependent errors and series resistance errors. The diode equation can be solved for T, where T is temperature in Kelvin, IS is a process dependent factor on the order of 10–13A, η is the diode ideality factor, k is the Boltzmann constant and q is the electron charge: 32 T= Sub-Milliohm DCR Current Sensing with Accurate Multiphase Current Sharing for High Current Power Supplies Muthu Subramanian, Tuan Nguyen and Theo Phillips 34 High Performance Single Phase DC/DC Controller with Digital Power Management q VD • η • k ln ID I S This equation has a relationship between temperature and voltage, dependent on the process-dependent variable IS . Measuring the same diode (with the same value IS) at two different currents yields an expression that is independent of (continued on page 4) Yi Sun 37 back page circuits 40 Figure 1. Remote CPU temperature sensor 2.5V TO 5.5V 0.1µF D+ CPU/ FPGA/ ASIC 2 | January 2013 : LT Journal of Analog Innovation VCC VREF 1.8V LTC2997 470pF D– GND VPTAT 4mV/K Linear in the news Linear in the News ANALOG CIRCUIT DESIGN BOOK SEQUEL PUBLISHED The much anticipated Analog Circuit Design, Volume 2 has just been published by the Newnes Press imprint of Elsevier Science & Technology Books. Edited by industry gurus, Bob Dobkin and the late Jim Williams, the new volume, Analog Circuit Design, Volume 2, Immersion in the Black Art of Analog Design, extends the reach of the first volume, at 1250 pages, covering a broad range of analog circuit design techniques. The book includes an extensive power management section, covering such topics as power management tutorials, switching regulator design, linear regulator design, powering illumination devices and automotive and industrial power design. Other sections of the book span a wide array of topics in data conversion, signal conditioning and high frequency/RF. This volume also features an extensive section of circuit collections with numerous hands-on examples across a variety of application areas. Readers of Analog Circuit Design, Volume 2 will be treated to the insight, technique and fascinating design approaches of Bob Dobkin, Jim Williams, Carl Nelson, Bob Widlar and many others. For more information, go to www.linear. com/designtools/acd_book.php To purchase Analog Circuit Design, Volume 2, click on the Elsevier link for a 30 percent discount on the cover price or go to the Amazon link at the bottom of the page. NEXT GENERATION BATTERY STACK MONITOR FOR HYBRID/ELECTRICS Linear has just announced its nextgeneration battery stack monitor for hybrid/ electric vehicles, the LTC6804, at press conferences in Europe, Asia and the US. This device can measure up to 12 seriesconnected battery cells at voltages up to 4.2V with 16-bit resolution and better than 0.04% accuracy. This high precision is maintained over time, temperature and operating conditions by a subsurface Zener voltage reference similar to references used in precision instrumentation. When stacked in series, the LTC6804 enables measurement of every battery cell voltage in large high voltage systems. “The LTC6804 combines 30 years of analog experience with hard-earned lessons in automotive battery management,” stated Mike Kultgen, design manager for Linear Technology. Multiple LTC6804s can be interconnected over long distances and operated simultaneously using Linear’s proprietary 2-wire isoSPI™ interface. The LTC6804 operates with a companion device, the LTC6820 isoSPI transceiver, which enables bidirectional transmission of the serial peripheral interface (SPI) bus across an isolated barrier up to 100 meters. For more information, visit www.linear.com/product/LTC6804 and www.linear.com/product/LTC6820. CONFERENCES & EVENTS Car-Ele Japan 2013, 5th Annual Automotive Electronics Technology EXPO, Tokyo Bigsight, Japan, January 16-18, 2013, East Hall 2, Booth E11-49:—Focus on automotive applications solutions, including LTC6804 battery management system. More info at www.car-ele.jp/en Advanced Automotive Battery Conference, Pasadena Convention Center, California, February 4-8, 2013, Booths 300-301—Presenting Linear’s battery management solutions. Erik Soule will present “Measuring the EV Battery Stack.” More info at www.advancedautobat.com/conferences/automotivebattery-conference-2013/index.html APEC 2013, Applied Power Electronics Conference, Long Beach Convention Center, California, March 17-21, 2013, Booths 1111 & 1113—Presenting Linear’s broad line of high performance power solutions. Brian Shaffer will present “Advancements in Energy Harvesting Transducers & the Challenges They Present for Power Management Solutions.” More info at www.apec-conf.org/ Electronica China 2013, Shanghai New International Expo Centre, China, March 19-21, 2013, Hall E1, Booth 1332—Linear will showcase its high performance analog and power management portfolio. More info at www.electronicachina.com/en/home n January 2013 : LT Journal of Analog Innovation | 3 The LTC2997 in a 2mm × 3mm 6-Pin DFN package is perfectly suited to measure temperature of an FPGA or microprocessor via the processor’s temperature measuring diode. The measurement error in this configuration is guaranteed to ±1°C over the temperature range from 0°C to 100°C and to ±1.5°C from –40°C to 125°C. (LTC299x continued from page 2) Figure 2. Temperature error vs temperature (LTC2997 at same temperature as remote diode) IS . The value in the natural logarithm term becomes the ratio of the two currents, which is process independent: 3 q V – VD1 T= • D2 η •k ID2 ln ID1 TRMT ERROR (°C) 2 Resistance in series with the remote diode causes a positive temperature error by increasing the measured voltage at each test current. The composite voltage equals: VD + VERROR = η TINTERNAL = TREMOTE 1 Capacitances larger than 1nF start to impact the settling of the sensor voltage at the various sense currents and therefore introduce additional temperature reading errors. For example, a 10m long CAT 6 cable has about 500pF of capacitance. resistance and the sensor temperature can be determined using currents I1 and I2 . Unlike many remote diode sensors, the LTC2997 accurately tracks fast changing temperatures due to its short update time (3.5ms) and its robust temperature measurement algorithm in the face of temperature variations, even during a measurement interval. Figure 4 shows the step response of the LTC2997’s internal sensor when the entire device is dipped into boiling water immediately after sitting in ice water. Series resistance up to 1k typically causes less than 1°C of temperature error as indicated in Figure 3b, which makes LTC2997 the ideal device to read out diode sensors that are several meters away from the temperature management system. Indeed, the maximum distance is limited more by the line capacitance than by the line resistance. The LTC2997 has many advantages over its digital counterparts when applied in temperature regulation loops. Its fast response time and analog output temperature eliminate much of the complexity required by digital systems. For example, Figure 5 shows the LTC2997 in a heater that regulates at 75°C. In this application, the 0 –1 –2 kT I • ln D + RS • ID I S q –3 –50 where RS is the series resistance. The LTC2997 removes this error term from the sensor signal by subtracting a cancellation voltage (see Figure 3a). A resistance extraction circuit uses one additional measurement current (I3) to determine the series resistance in the measurement path. Once the correct value of the resistor is determined VCANCEL equals VERROR . Now the temperature to voltage converter’s input signal is free from errors due to series Figure 3. Series resistance cancellation –25 0 25 50 TA (°C) 75 125 100 a. Simplified block diagram b. Temperature error vs series resistance 6 LTC2997 I1, I2 I3 4 ERROR (°C) 2 D+ RSERIES RESISTANCE EXTRACTION CIRCUIT VERROR VBE D– + – VCANCEL = VERROR 0 –2 VBE TEMPERATURE TO VOLT CONVERTER VPTAT –4 –6 4 | January 2013 : LT Journal of Analog Innovation 0 200 400 600 800 1000 SERIES RESISTANCE (Ω) 1200 design features The LTC2997 has many advantages over its digital counterparts when applied in temperature regulation loops. Its fast response time and analog output temperature eliminate much of the complexity required by digital systems. Figure 4. LTC2997 internal sensor thermal step response MEASURE TEMPERATURE AND SET TARGET TEMPERATURE WITH RESISTIVE DIVIDER INTEGRATE TEMPERATURE ERROR PWM OSCILLATOR 5V 125 100µF LTC2997 CONNECTED VIA 5 INCH 30AWG WRAPPING WIRES 100 0.1µF VPTAT (°C) 75 VCC VPTAT 1k D– GND AIR –50 0 1 3 2 TIME (s) 4 + 22k BOILING WATER ZXM64PO35 – 100k LTC6079 VREF 200k 5V + 0 ICE –25 WATER – LTC2997 470pF 25 100pF 10M D+ 50 10M VTARGET 75k VREF 100k LTC6079 CET 3904 1M 10Ω RHEATER 5 reference voltage is used to generate—by means of a resistive divider—a target voltage of 1.392V (= [75 + 273.15]K • 4mV/K). The first micropower rail-to-rail amplifier, the LTC6079, integrates the difference between the VPTAT output of the LTC2997 and the target voltage. The integrated error signal is converted to a pulse width modulated signal by the PWM oscillator, which in turn drives the switch of the PMOS, controlling the current through the heating resistor. The LTC2997 can also be used to build a Celsius thermometer (Figure 6), a Fahrenheit thermometer (Figure 7), a thermocouple thermometer with cold junction compensation (Figure 8), or in countless other applications where accurate and fast temperature measurements are required. Figure 5. 75°C analog PWM heater controller 0.1µF 150k 2.5V TO 5.5V 0.1µF VCC D+ 5V VREF 1.8V LTC2997 D– 1.8k VPTAT 4mV/K 100k 1k GND – 62k 143k 7 LTC1150 1 + 4 10mV/°C 0V AT 0°C 1µF –5V Figure 6. Celsius thermometer 0.1µF 255k 2.5V TO 5.5V 0.1µF VCC D+ VREF 1.8V LTC2997 D– VPTAT GND 4mV/K 100k 5V 270k – 7 LTC1150 1 + 4 62k 10mV/°F 0V AT 0°F 1µF –5V Figure 7. Fahrenheit thermometer January 2013 : LT Journal of Analog Innovation | 5 Unlike many remote diode sensors, the LTC2997 accurately tracks fast changing temperatures due to its short update time (3.5ms) and its robust temperature measurement algorithm in the face of temperature variations, even during a measurement interval. THE LTC2996 TEMPERATURE MONITOR 5V + OUT = 4mV/K LTC6078 TYPE K THERMOCOUPLE – 1.3k 127k 5V 10k 5.6pF 0.1µF VCC D+ VPTAT LTC2997 D– GND VREF Figure 8. Thermocouple thermometer with cold junction compensation 2.25V TO 5.5V 0.1µF 1.8V VCC VREF 43k OT LTC2996 UT VTH 36k VPTAT VTL 102k OT T > 70°C UT T < –20°C TEMPERATURE CONTROL SYSTEM 4mV/K D+ 470pF Figure 9. Remote temperature monitor with overtemperature and undertemperature thresholds GND MMBT3904 D– 5V 1.8V 30.9k 40.2k VREF 1.09V 1.49V VCC 0.1µF 10Ω RHEATER VPTAT LTC2996 VTH OT VTL D+ 110k HIGH IF T < 0°C MMBT3904 B6015L12F IRF3708 470pF D– GND UT HIGH IF T < 100°C Figure 10. Bang-bang controller maintains temperature between 0°C and 100°C 6 | January 2013 : LT Journal of Analog Innovation 2N7000 The LTC2996 adds threshold inputs VTH and VTL to the LTC2997 and continuously compares VPTAT to these thresholds to detect overtemperature (OT) or undertemperature (UT) conditions. The threshold input voltages can be conveniently set by resistive dividers from the built-in reference voltage, as depicted in Figure 9. If the temperature of the remote diode in Figure 9 increases above 70°C, the VPTAT voltage exceeds the high temperature threshold at VTH. The LTC2996 detects this overtemperature condition and alerts the temperature control system by pulling the OT pin low. In the same way, a temperature falling below –20°C is communicated via the UT pin. Note that the LTC2996 pulls on the open drain alert outputs only if the temperature exceeds the corresponding threshold for five consecutive update intervals of 3.5ms each. The OT and the UT pin have internal weak 400k pull-up resistors to VCC —no external resistors are required in many applications. The LTC2996 can be used to implement a bang-bang controller, keeping the temperature of a sensitive device (e.g., a battery) in a certain desirable temperature range, as shown in Figure 10. In this application, the undertemperature input threshold is set to 100°C, whereas the overtemperature input threshold input is set to 0°C. This seemingly upside down arrangement is linked to the fact that OT und UT are pulled low when a threshold is exceeded. Therefore, in this design features The LTC2996 adds threshold inputs VTH and VTL to the LTC2997 and continuously compares VPTAT to these thresholds to detect overtemperature (OT) or undertemperature (UT) conditions. of each cell individually with minimal additional wiring, as shown in Figure 11. 2.25V TO 5.5V 0.1µF VCC VREF LTC2996 D+ OT 43.2k VTH UT VTL VPTAT BATTERY SUPERVISOR 10k TALERT INT 28k 110k GND D– LOW IF TEMPERATURE OF ANY CELL TCELL > 70°C OR TCELL < 0°C 0.1µF VCC VREF LTC2996 D+ UT VTL VPTAT 28k 110k GND D– Figure 11. Supervising temperature of cells in a battery stack configuration, UT and OT both pull the gates of the NMOS transistors low while the temperature remains within the desired range (over the overtemp and under the undertemp), and the heating resistor and the cooling fan are turned off. If the temperature rises above 100°C, the undertemperature open drain output UT is released high and the fan THE LTC2995 COMBINES A TEMPERATURE AND A DUAL VOLTAGE MONITOR / SUPERVISOR In addition to temperature monitoring, nearly every electronic system requires multisupply voltage supervision. To serve this need, the LTC2995 combines the LTC2996 with a dual voltage supervisor, monitoring two supply lines for overvoltage and undervoltage conditions as shown in Figure 12. OT 43.2k VTH In fact, if the cells are connected in series (battery stack) only three additional lines—VCC , GND and an alert output—are required to monitor whether the temperature of any cell leaves the desired operating range. If the cells are connected in parallel, and a battery with a terminal voltage between 2.25V and 5.5V (e.g., Li-ion) is monitored, even a single additional line—the alert output—is sufficient to supervise the temperature of each cell. is switched on. Similarly, a temperature below 0°C turns on the heater. In the context of batteries, the LTC2996 can also be used to supervise the temperature of a large battery composed of several different cells. A damaged, shorted or worn out cell typically heats up, and can, in worst case, catch fire. The LTC2996 supervises the temperature The LTC2995 adds two additional high and low voltage inputs per channel, which are continuously compared to an internal 500mV reference. As soon as the voltage at either VH1 or VH2 falls below 500mV, the LTC2995 flags an undervoltage condition by pulling the UV output pin low. Similarly, an overvoltage condition is indicated by pulling the OV pin low if either VL1 or VL2 rise above 500mV. To prevent spurious resets due to noise on the monitored supply voltages, the LTC2995’s lowpass filter causes the January 2013 : LT Journal of Analog Innovation | 7 To prevent spurious resets due to noise on the monitored supply voltages, the LTC2995’s lowpass filter causes the output of the comparator to be integrated before asserting UV or OV. Any transient at the input of the comparator must be of sufficient magnitude and duration before the comparator triggers the output logic. output of the comparator to be integrated before asserting UV or OV. Any transient at the input of the comparator must be of sufficient magnitude and duration before the comparator triggers the output logic. Furthermore, the LTC2995 has an adjustable timeout period (tUOTO) that holds UV and OV asserted after any faults have cleared. This delay minimizes the effect of input noise with a frequency above 1/tUOTO. The timeout period (tUOTO) is adjustable by connecting a capacitor, CTMR, between the TMR pin and ground in order to accommodate a variety of applications. The LTC2995 includes temperature measuring and monitoring features that provide more flexibility than the LTC2997 and LTC2996. While the latter devices always switch to external mode if an external diode is connected, requiring D+ to be connected to VCC to measure the internal diode, the LTC2995 provides an additional diode select (DS) pin, allowing switching between the internal and an external diode on the fly. If the DS pin is left floating, the LTC2995 goes into “pingpong” mode, where it alternates between CONCLUSION internal and external diode measurement with a period of about 20ms. Finally, the LTC2995 can configure its two temperature thresholds both as overtemperature or both as undertemperature limits using the polarity select (PS) pin. This feature allows systems to react in levels to changes in temperature. As an example you might want to get a warning if the temperature rises above 75°C (e.g., to switch on a fan) and an alert if it increases above 125°C (e.g., to switch off the system) as depicted in Figure 12. Figure 12. Dual OV/UV ±10% supply and 75°C/125°C OT/OT remote temperature monitor ASIC/ CPU/ FPGA 2.5V 1.2V D+ 470pF VCC 0.1µF D– PS DS 64.4k VH1 LTC2995 10.2k If You Need Digital Output The LTC2990 and the LTC2991 feature digital I2C output and control as well as voltage and current monitoring functions. For more information, go to www.linear.com/2990 or www.linear.com/2991. 8 | January 2013 : LT Journal of Analog Innovation Linear Technology’s new family of accurate temperature sensors/monitors can use an internal or external diode as a sensor and produce analog outputs proportional to measured temperature. The family ranges from a tiny temperature sensor to a combined temperature and dual voltage supervisor that can signal out-ofrange conditions. These devices make it easy to build analog temperature control loops or to monitor temperatures (and voltages) with minimum complexity. n VPTAT VL1 45.3k TO2 194k TO1 VH2 OV 10.2k UV VL2 45.3k TMR GND 5nF 140k VT1 VT2 20k VREF 20k A/D OT T > 125°C OT T > 75°C +10% –10% design features 2.7V to 40V Monolithic Buck-Boost DC/DC Expands Input Capabilities, Regulates Seamlessly through Automotive Cold-Crank and Load-Dump Transients John Canfield Handheld devices, industrial instruments and automotive electronics all demand power supply solutions that can support an expansive range of input voltages resulting from automotive input voltage transients, resistive line drops and a wide variety of power sources. As a further design challenge, applications often require a variety of regulated voltage rails, including some that fall within the input voltage range. The LTC3115-1 buck-boost DC/DC converter, with its wide 2.7V to 40V input and output voltage capability, high efficiency, small footprint and seamless transition between step-up and step-down modes of operation, easily meets the requirements of such applications. For automotive electronics, the LTC3115-1 provides uninterrupted operation through load dump transients and even the harshest cold-crank conditions. Its programmable switching frequency optimizes efficiency and supports operation at 2MHz to ensure that switching noise and harmonics are located above the AM broadcast band. The LTC3115-1 employs a proprietary low noise PWM control algorithm that minimizes electromagnetic emissions over all operating conditions even during transitions between the step-up and step-down modes of operation and over the full range of load current. An internal phase-locked loop allows switching edges to be synchronized with an external clock for further control of EMI in noise-sensitive applications. An accurate RUN pin provides a programmable input undervoltage lockout threshold with independent control of hysteresis. By consuming only 30µ A of quiescent current in Burst Mode® operation and 3µ A in shutdown, the LTC3115-1 reduces standby current drain on automobile batteries to negligible levels. Figure 1. 5V regulator with wide 2.7V to 40V input range The LTC3115-1 is also well suited for handheld devices, which are required to interface to an expanding array of power sources. While it was once common for portable devices to be powered by a dedicated AC adapter or a single power source, many must now be compatible with a variety of inputs including automotive, USB, Firewire and unregulated wall adapters. Next generation military radios and support electronics are an extreme example, requiring the capability to operate from all available power sources for emergency use and to minimize the number of battery varieties carried in the field. Additionally, in an effort to reduce design overhead, many product families utilize a single power supply design that is shared across multiple versions of a product. This requires that the common power supply support the widest range of possible input voltages that will be seen by any device within the family. With its wide 2.7V to 40V input and output voltage ranges, internal power switches and high efficiency the LTC3115-1 has the features and flexibility required for these demanding applications. 5V, 2MHz MINIATURE SIZE AUTOMOTIVE SUPPLY The proliferation of electronic subsystems in automobiles has created demand for small size, high reliability power supplies that can operate under the stringent conditions presented by the automotive environment. The LTC3115-1 is well suited for such applications given its ability to provide a stable well-regulated voltage over automotive operating conditions even when the battery voltage falls below the required output rail due to battery state of charge, line transients induced by switched high current loads and cold-cranking events. January 2013 : LT Journal of Analog Innovation | 9 Of commonly utilized power sources, the automotive supply rail presents one of the most challenging inputs to a power supply. Its nominal voltage varies from 10.6V to 15V depending on the state of charge of the battery, the ambient temperature and whether the alternator is charging or idle. Cold-crank conditions can push the rail below 4V and line transients can produce 40V spikes. Figure 2 shows a 5V automotive supply ideal for use in engine control units and other critical functions including safety, fuel system and drive train subsystems where processors must remain powered without glitch during even the most severe input voltage transients. This application uses a 2MHz switching frequency to minimize its footprint and eliminate interference with the AM broadcast band. regulator. Figure 3 shows the efficiency of this application circuit with a 500m A load for input voltages from 3.3V to 40V. RIDING THROUGH AUTOMOTIVE LOAD-DUMP AND INDUCTIVE LINE TRANSIENTS Of commonly utilized power sources, the automotive supply rail presents one of the most challenging inputs to a power supply. Its nominal voltage varies from 10.6V to 15V depending on the state of charge of the battery, the ambient temperature and whether the alternator is charging or idle. In addition to the variability in its nominal voltage, the automotive power rail is also subject to a wide range of dynamic disturbances induced by changes in engine RPM, transitioning loads such as power windows, wipers and air conditioning, and inductive transients in the wiring harness. The VCC rail provides power to the internal circuitry of the LTC3115-1 including the power device gate drivers and is ordinarily powered from the input rail via an internal linear regulator. In this application, diode D1 bypasses the internal linear regulator and delivers power to the VCC rail directly from the regulated output to improve efficiency and output current capability. This is particularly advantageous in applications with higher switching frequencies, given that the increased gate drive current is provided more efficiently from the converter’s output rail than through the internal linear However, the most extreme conditions occur during a load-dump transient which can produce voltages in excess of 120V for a duration of hundreds of milliseconds. Figure 2. A 5V, 2MHz automotive supply with cold-crank capability CBST2 0.1µF SW2 BST2 PVOUT PVIN VIN LTC3115-1 PWM/SYNC BURST PWM VC RUN OFF ON 90 CO 47µF CFB RFB 1000pF 237k FB RT 17.8k 10 | January 2013 : LT Journal of Analog Innovation PVCC VCC GND PGND CFF 10pF RFF 15k RBOT 249k D1: PANASONIC MA785 L1: COILCRAFT LPS6225 RT RTOP 1M ILOAD = 500mA 85 5V 0.5A EFFICIENCY (%) BST1 SW1 CIN 4.7µF Automotive electronics must also be designed to survive a double-battery jump start, where they are subjected to 24V for extended durations as the vehicle is jump started using a series-connected second battery or from a commercial vehicle with a dual battery electrical system. An additional overvoltage condition on the automotive bus is caused by alternator voltage regulator failure and is often Figure 3. 5V, 2MHz automotive supply efficiency versus VIN L1 3.3µH CBST1 0.1µF AUTOMOTIVE 3.3V TO 40V A load-dump transient occurs when the alternator is charging the vehicle’s battery and an electrical open-circuit causes a momentary disconnection of the battery from the alternator. Until the voltage regulator can respond, the full alternator charging current is applied directly to the automotive power bus, raising its voltage to potentially dangerous levels. Such a transient could be caused through a physical disconnection of the battery by a mechanic working on the vehicle, but could also result from a faulty connection in the battery cable or corrosion at the battery terminals. 80 75 70 D1 65 C1 4.7µF 60 0 5 10 15 20 25 30 INPUT VOLTAGE (V) 35 40 design features Typically, automotive electronics located downstream from passive protection networks must survive up to a 40V transient without damage. Critical systems must survive high level transients, and function seamlessly through such transients without interruption. The LTC3115-1 can maintain uninterrupted regulation of a 5V supply rail through a 13.8V-to-40V momentary line transient with 1ms rise and fall times. included in the battery of tests conducted by automotive electronics OEMs. Such a malfunction can result in full application of the alternator charge current to the battery and an overvoltage of approximately 18V for extended durations. The automotive power rail is also polluted with short duration overvoltage transients due to rapid load changes produced by switching high power loads such as power doors, fans and cooling fan motors interacting with the significant inductance in the vehicle’s wiring harness. In most vehicles a passive protection network consisting of a lowpass LC filter and transient voltage suppression (TVS) array is used as a first line of defense to clamp the peak excursions of the power bus. Typically, automotive electronics located downstream from the protection network must survive up to a 40V transient without damage. Critical systems must not only survive, but must also function seamlessly through such transients Figure 4. A 13.8V to 40V load-dump line transient without interruption. Figure 4 illustrates the ability of the LTC3115-1 to maintain uninterrupted regulation of a 5V supply rail through a 13.8V-to-40V momentary line transient with 1ms rise and fall times. SEAMLESS OPERATION THROUGH AUTOMOTIVE COLD-CRANK TRANSIENTS High voltage transients are a problem on the automotive power bus, but perhaps the more challenging problem is undervoltage transients. The most severe of these is known as cold crank, which occurs when the engine is initially started. A typical cold-crank voltage waveform is shown in Figure 5. The initial low voltage plateau is the most extreme and is caused when the starter motor begins turning over the engine from a dead stop. During this phase, the vehicle’s bus voltage can fall below 4V. Colder temperatures exacerbate the situation since the higher viscosity of the engine oil results in a higher required torque from the starter Figure 5. A 12V to 4.5V cold-crank line transient 12V 40V VIN 10V/DIV VIN 2V/DIV 13.8V 6V 4.5V VOUT 200mV/DIV motor. The first plateau is followed by a second somewhat higher voltage plateau, typically near half the nominal battery voltage, as the starter maintains the engine rotation. Once the engine starts, the battery recovers to its nominal voltage. Safety devices and engine critical components such as the engine control unit and fuel injection system are required to remain operational throughout a coldcrank transient. As shown in Figure 5, the LTC3115-1’s buck-boost architecture enables it to maintain output regulation through even the most severe cold-crank transients by automatically and seamlessly switching to boost mode operation during the undervoltage event. Cold-crank capability for automotive electronics has expanded in importance as cars now include automated fuel-saving, on-demand engine start/stop, whereby the vehicle’s engine is turned off during momentary vehicle stops at stoplights or in traffic. Vehicles equipped with ondemand starting are subjected to frequent cranking undervoltage events. As a result, auxiliary electrical components that previously had no need to function through the occasional cold-crank event in a traditional vehicle must now operate through such transients to eliminate any disturbance to infotainment, navigation, dashboard electronics and lighting systems. VOUT 200mV/DIV 1ms/DIV 5ms/DIV January 2013 : LT Journal of Analog Innovation | 11 The LTC3115-1’s buck-boost architecture enables it to maintain output regulation through even the most severe cold-crank transients by automatically and seamlessly switching to boost mode operation during the undervoltage event. LOW EMI AND NO EMISSIONS IN THE AM BAND within the AM broadcast band is free from any significant spectral emission. The LTC3115-1 supports switching frequencies up to 2MHz so that the fundamental switching frequency component, and all of its harmonics, can be located above the AM frequency band to minimize interference with radio reception. Figure 6 shows the spectral emission of the LTC3115-1 over the AM band for the automotive application circuit of Figure 2 operating at no load and with a 500m A load. In both cases the entire range of frequencies HANDLING MULTIPLE POWER SOURCES – UNREGULATED WALL ADAPTER, AUTOMOTIVE INPUT, USB, USB-PD AND FIREWIRE To increase flexibility and enhance the user’s experience, many portable electronic devices are being designed to work from various power sources. These power sources can vary widely in voltage, especially when accounting for connector and cable drops. 2MHz FUNDAMENTAL 10 0 SW AMPLITUDE (dBV) The LTC3115-1 features a low noise forced PWM mode where both switch pins operate at constant frequency for all loads, producing a low noise spectrum, independent of operating conditions. The predictable spectrum and minimal subharmonic emissions help reduce interference and aid in compliance with strict automotive EMI standards. 20 –10 –20 AM BAND –30 –40 NO LOAD –50 500mA LOAD –60 –70 0 0.5 1 1.5 FREQUENCY (MHz) 2 2.5 Figure 6. Fixed frequency low noise PWM minimizes emissions across the AM band Under USB 3.0, the nominal supplied voltage is 5V ±5%, but a fully compliant powered device must be able to operate down to 4V when accounting for allowable cable and connector voltage drops. In addition, a downstream USB power rail is permitted to drop as low as 3.67V under transient conditions such as when additional devices are plugged into the host or powered hub. Figure 7. For high efficiency, this dual input 5V supply uses a LTC4412 low loss PowerPath™ controller and a P-channel MOSFET in the battery path instead of a Schottky diode. An inexpensive Schottky diode is used on the higher voltage input where its voltage drop is insignificant. The newly approved USB PD (power delivery) specification allows for higher power delivery over USB with support for supply voltages up to 20V. Firewire ports deliver an unregulated power rail with a voltage that varies over a wide range, typically 9V to 26V depending on the class of the power provider. Figure 8. Overall efficiency of the PowerPath and LTC3115-1 L1 10µH CBST2 0.1µF D1 UNREGULATED WALL ADAPTER 8V TO 28V LITHIUM CELL 3V–4.2V BST1 SW1 CIN 4.7µF M1 SW2 BST2 PVIN VIN PVOUT LTC3115-1 VC RUN + 100 COUT 47µF ×2 CFB RFB 4700pF 100k FB PVIN GATE SENSE LTC4412HV GND RT 47.5k 12 | January 2013 : LT Journal of Analog Innovation PWM/SYNC PVCC VCC RT GND PGND RTOP 1M CFF 47pF RFF 51k RBOT 249k C1 4.7µF VIN = 4.2V 5V 500mA COUT: GRM43ER60J476 D1: B360A-13-F L1: COILCRAFT LPS6225 M1: Si8487DB 80 EFFICIENCY (%) CBST1 0.1µF 60 VIN = 13.8V 40 20 0.01 0.1 ILOAD (A) 1 2 design features The LTC3115-1 features a low noise forced PWM mode where both switch pins operate at constant frequency for all loads, producing a low noise spectrum independent of operating conditions. The predictable spectrum and minimal subharmonic emissions help reduce interference and aid in compliance with strict automotive EMI standards. L1 22µH CBST1 0.1µF 20V TO 30V BST1 SW1 VCAP CIN 4.7µF + CBST2 0.1µF CBULK 1000µF ×2 35V ALUMINUM ELECTROLYTIC SW2 BST2 PVIN VIN PVOUT 1µF RUN LTC3115-1 VC + RFB 25k CO 82µF CFB 3300pF FB COUT: OS-CON 35SVPF82M L1: TOKO 892NBS-220M RT PWM/SYNC PVCC VCC GND PGND RT 47.5k RTOP 1M CFF 47pF RFF 51k The LTC3115-1 operates directly from all of these portable power sources as well 2V/DIV CAPACITOR BANK VOLTAGE (VCAP) RBOT 43.2k C1 4.7µF Figure 9. 24V industrial rail restorer with brownout ride-through The ubiquitous wall adapter remains perhaps the most common source of power for portable devices. A typical wall adapter is simply a transformer followed by a bridge rectifier, offering no active regulation. That task is left to the end device to avoid the effects of cable drop. Unregulated wall adapters are designed to provide rated current at the specified typical output voltage. Being unregulated, the output voltage is a load line function, increasing substantially at lighter loads and decreasing under heavy load. In addition, the AC line voltage is permitted to vary between 105V and 125V, adding an additional 10% variability in the unregulated wall adapter’s output. It is not uncommon for a 12V unregulated wall adapter to produce an output voltage of 17V or greater at light load. NOISY 24V INPUT RESTORED 24V RAIL RAIL WITH DROPOUT (LTC3115-1 OUTPUT) 24V 1.5A as from a variety of battery chemistries including lithium (single cell or series connected), sealed lead acid, three or more series alkaline cells and even a bank of supercapacitors for backup applications. Multiple power sources can be combined through a Schottky diode-OR circuit. For higher efficiency, the LTC3115-1 can be combined with an ideal diode PowerPath controller to provide automatic switchover between multiple power sources using the low voltage drop of a power P-channel MOSFET to replace the Schottky diode. Figure 7 shows how the LTC3115-1 can be combined with the LTC4412HV to obtain a dual input—single lithium and unregulated wall adapter—5V supply. In this case, a series PMOS is used on the lower voltage lithium input while an inexpensive Schottky diode is used on the higher voltage input where its ILOAD = 1A 10ms/DIV Figure 10. The LTC3115-1 regulates the output rail through input brownouts voltage drop is insignificant. The overall efficiency of this supply including the converter and PowerPath is given in Figure 8 for each power input. 24V INDUSTRIAL RAIL RESTORER AND BACKUP Industrial control and monitoring systems commonly utilize a 24V bus to power DIN mounted instrumentation such as programmable logic controllers, actuators and sensors. Being subject to high power switching loads and possible fault conditions, this bus can become corrupted with transients and momentary undervoltage transients. In severe cases there may even be momentary interruptions in bus power. Critical rail-powered systems are required to remain powered throughout such events to ensure control and monitoring of critical functions. (continued on page 16) January 2013 : LT Journal of Analog Innovation | 13 Monolithic Operational Amplifier Works from ±4.75V to ±70V and Features Rail-to-Rail Output Swing and Low Input Bias Current Michael B. Anderson Monolithic operational amplifiers have been around since the 1960s, but this ubiquitous device still sees steady improvements in performance. The LTC6090 precision monolithic operational amplifier takes a big step forward by extending the supply voltage to ±70V without compromising the features that are expected in a precision op amp. The LTC6090 is available in a small 8-lead SO package and a 16-lead TSSOP package. Both packages feature exposed pads to reduce thermal resistance, eliminating the need for a heat sink. An easy interface to low voltage control lines and built-in thermal safety features simplify the task of high voltage analog design. Operational amplifiers are expected to have low input bias current, low offset, and low noise. The LTC6090 is no exception. Designed with a MOS input stage the input bias current is typically 3pA at 25°C and less than 100pA at 85°C. This makes it well suited for high impedance applications such as a photodiode amplifier Figure 2. LTC6090 output voltage 140VP–P 10kHz sine wave OUTPUT VOLTAGE SWING (V) V+ – 0.8 V+ – 1.0 V+ – 1.2 V – + 0.8 V – + 0.6 14 | January 2013 : LT Journal of Analog Innovation V– 0.001 0.01 SINK 0.1 1 10 LOAD CURRENT (mA) + 8 LTC6090 1 5 200k 1% 100mW 6 4 VOUT 22.1k 1% –3V fast slew rate and rail-to-rail output stage rated for ±10m A that can drive up to 200pF. An example shown in Figure 2 is a 140VP–P 10k Hz sine wave. Figure 3 shows the output swing is well maintained as load current is increased. And the fidelity of the output voltage at 100VP–P extends out to 8kHz as shown in Figure 4. Figure 4. LTC6090 total harmonic distortion plus noise vs. frequency SOURCE TA = 125°C TA = 25°C TA = –40°C 3 7 Figure 1. Extended dynamic range 1M transimpedance photodiode amplifier V+ – 0.6 V – + 0.2 – VOUT = IPD • 1M OUTPUT NOISE = 21µVRMS (1kHz – 40kHz) OUTPUT OFFSET = 150µV MAXIMUM BANDWIDTH = 40kHz (–3dB) OUTPUT SWING = 0V TO 12V VS = ±70V V – + 0.4 2 –3V Figure 3. LTC6090 output voltage swing vs load current V+ – 0.4 125V PHOTODIODE SFH213 On the output side, precision op amps are expected to maintain precision when driving loads. Again, the LTC6090 does not disappoint. The unity gain stable output drive capability includes a 10MHz GBW product, V+ 25µs/DIV IPD shown in Figure 1. The low input offset voltage is less than 1.6mV, and the noise is 11nV/√Hz at 10kHz. The input common mode range is to 3V of either rail or a range of 134V across a 140V supply. V+ – 0.2 VOUT 20V/DIV 10M 1% TOTAL HARMONIC DISTORTION + NOISE (%) HIGH VOLTAGE AND HIGH PERFORMANCE 0.3pF 100 10 VS = ±70V AV = 5 RL = 10k CF = 30pF 1 0.1 VOUT = 100VP-P 0.01 0.001 VOUT = 50VP-P VOUT = 10VP-P 10 100 1000 10000 FREQUENCY (Hz) 100000 design features Operational amplifiers are expected to have low input bias current, low offset and low noise. The LTC6090 is no exception. Designed with a MOS input stage, the input bias current is typically 3pA at 25°C and less than 100pA at 85°C. This makes it well suited for high impedance applications. HIGH IMPEDANCE APPLICATIONS REQUIRE LOW LEAKAGE CIRCUITS The low input bias current of the LTC6090 make it an excellent choice for high impedance applications that require high voltage. As shown in Figure 5, input bias current is logarithmically dependent on temperature, doubling for every 10°C increase. In addition, input protection devices sit in an isolated pocket where leakage increases as the voltage on the input pin increases with respect to V–. In Figure 5 the input pin is held at mid-supply. In order to maintain low input bias current, care should be taken during PCB layout. Special low leakage board material can be considered. In critical applications, consider using guard rings. The TSSOP package with exposed pad has guard ring pins that can be used to protect the input pins from leakage currents. An example PCB layout of an inverting amplifier is shown in Figure 6. Note that the solder mask should be pulled back over the guard ring to expose the PCB metal. It is important that the PCB be clean and moisture free. Consider cleaning it with a solvent and rinsing any residue with tap water, then baking the board to remove any moisture. We have also found that thoroughly washing the board using soap and tap water (without solvent) yields good results. INTERFACING LOW VOLTAGE CONTROL LINES TO A HIGH VOLTAGE OP AMP The low voltage control lines on the LTC6090 can be interfaced as low as the negative supply rail, or as high as 5V below the positive supply rail. The COM pin acts as a common to interface to the low voltage control lines, and can be connected to the low voltage system ground or left to float. The output disable, OD, and overtemperature, TFLAG, pins are now referred to the low voltage system ground. COM, OD and TFLAG pins are protected with diodes and resistors as shown in Figure 7. If left floating the COM pin will be pulled above mid-supply by the OD pin internal pull-up resistor to 21V when the supplies are ±70V. THERMAL PROTECTION: USE OD AND TFLAG At 140V total supply voltage and 2.7m A typical quiescent current, the LTC6090 consumes 378mW of power. Add a load and the power can exceed a watt, making good thermal design a priority. Both packages, the SO and TSSOP, feature an exposed pad on the bottom of the Figure 7. The low voltage interface configured to automatically disable the output stage when the junction temperature of the die reaches 145°C LTC6090 V+ 2M 10k OD Figure 6. PCB guard ring example layout Figure 5. LTC6090 input bias current versus junction temperature V+ OUT 2M 10k COM 1000 2M INPUT BIAS CURRENT (pA) ±70V 100 V– 10k 500Ω ±5V 10 +IN TFLAG 10k 1 R2 –IN 0.1 0 20 40 60 80 100 120 JUNCTION TEMPERATURE (°C) 140 C1 R1 30k V– January 2013 : LT Journal of Analog Innovation | 15 An important feature designed to protect the LTC6090 from exceeding 150°C junction temperature shuts down the output stage when the junction temperature gets too high. This is accomplished by connecting the overtemperature pin to the output disable pin. The overtemperature pin, or TFLAG pin, is an open drain pin that pulls low when the junction temperature of the die reaches 145°C. The 5°C built-in hysteresis releases the TFLAG pin when the junction temperature reaches 140°C. The output disable pin, or OD pin, is an active low pin that turns off the output stage and lowers the quiescent current of the device to 670µ A when pulled low with respect to the COM pin. When these two pins are tied together, the LTC6090 is disabled if the junction temperature of the die reaches 145°C. Note that these pins can float and be tied together. An additional thermal safety feature shuts off the output stage when the junction temperature of the die reaches approximately 175°C. The 7°C of hysteresis enables the output stage when it returns to approximately 168°C as shown Figure 8. LTC6090 thermal shutdown hysteresis plot 3.0 2.5 SUPPLY CURRENT (mA) package, which is internally connected to the negative supply rail, V–, and must be connected to the negative power plane. Connect as much PCB metal as practical to the exposed pad—the thermal resistance of the package is proportional to the amount of metal soldered to the exposed pad. In a best case scenario the thermal resistance, qJA , of the SO package is 33°C/W. For 1W of power, the junction temperature of the die increases 33°C above ambient temperature. 2.0 1.5 1.0 0.5 0 162 164 166 168 170 172 174 176 178 JUNCTION TEMPERATURE (°C) in Figure 8. Note that Figure 8 shows the junction temperature. This feature is intended to prevent the device from thermal catastrophic failure. Operating the LTC6090 above its absolute maximum junction temperature of 150°C can reduce reliability and is discouraged. CONCLUSION The LTC6090 features the high performance specs of a low voltage precision amplifier, but with the ability to work with ±70V for high voltage applications. These features include high gain, low input bias current, low offset and low noise for a precision front end. A rail-to-rail output stage can drive a 200pF load capacitor and ±10m A of load current, making this part suitable for precision high voltage applications such as high impedance amplifiers. Easily interfaced control lines for disabling the output and a thermal shutdown function are simple to implement. Small 8-lead SO and 16-lead TSSOP packages both have exposed pads to reduce thermal resistance, eliminating the need for a heat sink. n (LTC3115-1 continued from page 13) In addition, many devices must remain operating for a period of time after bus failure in order to initiate a controlled shutdown. The LTC3115-1 application shown in Figure 9 is a 24V rail restorer application that maintains a clean and well-regulated 24V output rail from a noisy input supply rail, which can fluctuate above and below the regulation target. In addition, as shown in the waveforms of Figure 10, this supply is able to maintain regulation of its 24V output through momentary interruptions in bus power. CONCLUSION The flexibility and high efficiency of the LTC3115-1 make it perfectly suited to meet the demanding needs of the next generation of automotive electronics and 16 | January 2013 : LT Journal of Analog Innovation portable devices, especially those operated from multiple power sources. Its internal power switches and programmable switching frequency minimize the power solution footprint, supporting the increasing demand for miniaturization of electronic devices in the portable and automotive arenas. Low Burst Mode operation and shutdown quiescent currents prolong battery life and facilitate use in always-active automotive applications. The LTC3115-1 is ideal for noise-sensitive applications, given its low noise, fixed frequency PWM mode, which produces a predictable and well controlled EMI spectrum with switching edges that can be synchronized to a system clock. Internal soft-start minimizes inrush current during start-up and an internal divider in the control path reduces the impact of input voltage variations, and makes the loop easier to compensate in applications with widely varying input voltages. A programmable input undervoltage lockout allows the input voltage at which the part is enabled to be set by the user, and provides for independent control of the hysteresis. The LTC3115-1 also features complete disconnect of the output from the input in shutdown, and is fully protected with output short-circuit protection and overtemperature shutdown. n design features Bus Buffers Simplify Design of Large, Noisy I2C Systems Rajesh Venugopal The original I2C specification limited the maximum bus operating frequency to 100k Hz; it is now 400k Hz . As systems grew larger, bus buffers were introduced to buffer bus capacitance and solve several other common I2C issues. Early bus buffers degraded certain I2C specifications in a manner that can be unacceptable in large noisy systems. The LTC4313 and LTC4315 family of bus buffers offers the benefits of traditional bus buffers while maintaining compliance to all I2C voltage DRIVER HIGH 0.9 • VCC 0.7 • VCC VOLTAGE The I2C bus and its derivatives—such as SMBus, PMBus, the DDC bus of HDMI and IPMB bus of ATCA—are used in a variety of large systems to transfer vital system information. These bus specifications have gained wide acceptance due to ease of use. The I2C bus is a digital serial 2-wire bus consisting of a single clock (SCL) and single data (SDA) line. The I2C protocol employs open drain pull-downs to drive the bus low, and resistors or current sources to pull the bus high. The maximum allowed pull-up current and bus capacitance are 4mA and 400pF, respectively. LOGIC HIGH NOISE MARGIN VOH VIH RECEIVER THRESHOLD BAND 0.3 • VCC 0.4V LOGIC LOW NOISE MARGIN VIL VOL DRIVER LOW Figure 1. I2C bus voltage specifications and resulting noise margins specifications. This makes it the preferred choice for use in large noisy systems. Figure 1 shows the I2C specification requirements for logic high and logic Table 1. A list of LTC4313 and LTC4315 features and benefits FEATURE BENEFITS I 2 C Buffers • Break up bus capacitance, which allows large I 2 C compliant systems to be built, by keeping the capacitance of each section < 400pF High V IL • High logic-low noise margin up to 0.3 • V CC • Operation with noncompliant I 2 C devices Automatic Buffer Turn-Off Voltage Adjustment • Compatible with devices whose RTA turn-on voltage is lower than 0.3 • V CC • Interoperable with other LTC buffers Level Translation • Provides I 2 C communication between buses with voltages from 1.4V to 5.5V • Reduce rise time Rise Time Accelerators (RTAs) • Allow larger bus pull-up resistors for better logic low noise margin • Selectable RTA pull-up current strength Disconnection and Recovery from Stuck Bus Fall Time Control Hot Swapping • Free masters to resume upstream communications • Generates up to 16 clock pulses and a stop bit on the stuck buses to get the bus to release high • Minimizes transmission line effects in systems • Waits for bus idle or stop bit before making a connection • Precharges bus to minimize disturbance January 2013 : LT Journal of Analog Innovation | 17 The LTC4315 and LTC4313 are high noise margin bus buffers that solve a number of problems associated with large I2C systems. They provide capacitance buffering, level translation for bus supplies ranging from 1.4V to 5.5V, high logic-low noise margins up to 0.3 • VCC and reject noise above 0.3 • VCC when the bus is a logic high. low voltages on the bus. For I2C compliance, driven logic low signals must be below an output low level (VOL) of 0.4V. Logic high signals require the bus to be pulled up above an output high level (VOH) of 0.9 • VCC, where VCC is the bus supply voltage. I2C compliant receivers must interpret any voltage below an input low level (VIL) of 0.3 • VCC as a logic low and any voltage above an input high level (VIH) of 0.7 • VCC as a logic high. These requirements yield a logic low noise margin of 0.3 • VCC – 0.4V and a logic high noise margin of 0.2 • VCC. Over time, as systems grew larger, bus capacitances increased well beyond 400pF. Bus buffers were introduced to break the large I2C bus into smaller segments and to drive the capacitance associated with each segment. A higher operating frequency coupled with increasing bus capacitance also required a decrease in signal rise times. Rise time accelerators (RTAs) were incorporated into the bus buffers to reduce bus rise times—by sourcing strong pull-up currents into the bus during these transitions. In addition, bus buffer products offered by Linear Technology also incorporated several additional features like SDA, SCL Hot Swap, precharge and stuck bus recovery to improve robustness of I2C systems and voltage level translation to ease communication across voltage domains. deviations from the I2C specification. There are three reasons for this: •First, buffers require a scheme to differentiate an externally driven logic low from their own driven low. This is required to prevent locking the bus into a permanent low state. As a result, some buffers drive VOLs above the 0.4V I2C specification and require all other devices to drive below 0.4V. Others drive an output VOL that is a small offset higher than the driven input VOL. As systems grew, the compressed logic low noise margin of existing buffers increased the bus’ susceptibility to noise. Typically larger systems require a bus buffer that restores logic low noise margin to the I2C specification, namely a fast buffer that is active until the bus voltage crosses the VIL value of 0.3 • VCC and does not load the bus. •Second, to maximize RTA operating range, Linear Technology bus buffers turn off their pull-down devices and turn on their RTAs at voltages slightly higher than the I2C VOL. An additional requirement in large systems is backward compatibility with buffer products whose RTAs turn on below 0.3 • VCC or with products that drive a noncompliant VOL of 0.6V. An adjustable RTA current is also advantageous, especially in large systems where multiple RTAs can be activated simultaneously. Large RTA currents result in sharp edges and raise concerns about unwanted effects like inductive ringing and EMI. •Third, all buffers capacitively load the bus when they are active and need to be turned off at as low a voltage as possible in order to reduce bus rise time. As a result, most existing bus buffers detect a logic low only if the bus voltage is < 0.6V. Most buffers turn on their RTAs at 0.8V. Some buffers drive a noncompliant VOL > 0.4V. All these result in reducing the logic low Figure 2. The LTC4315 driving the parasitic backplane capacitance in a large system. Only the SCL pathway is shown for simplicity. The LTC4315 (12-pin) and the LTC4313 (8-pin) parts specifically solve these problems while retaining the beneficial features of other Linear Technology bus buffer CONTROLLER CARD 5V 0.01µF SCL1 The downside of buffer and RTA insertion into a bidirectional I2C bus is the introduction of noise margin from (0.3 • VCC – 0.4V) to 0.2V or even lower, and slowing the bus rising edge by the capactive load of the buffers when they are active. 5.1k VCC VCC2 LTC4315 SCLIN SCLOUT ACC GND BACKPLANE 5.1k SCL2 *CBP 690pF *LARGE PARASITIC BACKPLANE CAPACITANCE 18 | January 2013 : LT Journal of Analog Innovation design features In extended I2C systems, long PCB traces and large backplanes with long cables generate large parasitic bus capacitances. The LTC4315’s high noise margin buffers can drive these capacitances without degrading signal integrity or reducing operating frequency. products. Table 1 lists the key features of these products. This document references the LTC4315, but all text applies to the LTC4313 as well, unless otherwise noted. The LTC4315 has a high 0.3 • VCC guaranteed minimum VIL , ensuring a high logic-low noise margin. The LTC4315 is interoperable with devices that drive a high VOL > 0.4V and with products whose RTAs turn on at voltages below 0.3 • VCC. The LTC4315 allows user selection of the RTA current level in order to control bus rise rates. The LTC4315 retains capacitance buffering, Hot Swap, precharge, stuck bus recovery and level translation features of other Linear Technology bus buffers. Since its buffers do not load the bus, the LTC4315 is capable of operation up to 1MHz and is compatible with the I2C standard mode and fast mode, SMBus and PMBus specifications. In summary, LTC4315 provides all the benefits of the existing buffers without compromising any I2C specification. CAPACITANCE BUFFERING AND NOISE REJECTION IN LARGE SYSTEMS In large I2C systems, long PCB traces and large backplanes with long cables cause large parasitic bus capacitances. As shown in Figure 2, the LTC4315’s high noise margin buffers can drive these large capacitances without degrading signal integrity or reducing operating frequency. Another issue of large I2C systems, like the one in Figure 2, is noise susceptibility. Noise and signal coupling in the cable and between PCB traces can disrupt input and output clock and data signals, causing system level failures. A particularly VOH – 0.33 • VMIN, where VMIN is the lower of the VCC and VCC2 voltages. For all versions of the LTC4313, VMIN defaults to VCC. In Figure 3, when the SCLIN voltage drops below 0.33 • VMIN, SCLOUT tracks SCLIN. No output glitches occur as the input crosses the VIL level of 0.33 • VMIN . Assuming a worst-case DC VOL of 0.4V on the bus, the LTC4315’s logic low noise margin is 0.33 • VMIN – 0.4V = 1.25V. These noise suppression features make the LTC4315 a solid choice for large, noisy I2C systems. Ideally, system designers of large, noisy I2C systems should use LTC4315s on all boards for maximum noise immunity. SCLIN 2V/DIV SCLOUT 2V/DIV 500ns/DIV Figure 3. The LTC4315 transmits a clean logic high at SCLOUT even when a noisy 400kHz I2C signal is applied to SCLIN. extreme example of a noisy SCL waveform is shown in Figure 3 to illustrate the robust noise rejection the LTC4315 features. OPERATION WITH NON-COMPLIANT I 2C DEVICES Figure 4 shows the LTC4315’s compatibility with devices that drive non-compliant VOL s —in this case 0.6V. The LTC4315 passes the 0.6V to the microprocessor where it is interpreted as a logic low. The high buffer turn-off voltage of the LTC4315—1.089V in this circuit—yields a logic low noise margin of 489mV. Figure 3 shows the LTC4315’s handling of sinusoidal noise superimposed on a 400kHz square wave at its input. The noise applied to the logic high state is not propagated to the other side as long as that bus voltage does not drop below 0.33 • VMIN . The logic high state of SCLOUT is not affected by noise on SCLIN. For the LTC4315, logic high noise margin is 3.3V Figure 4. The LTC4315 communicating with a noncompliant I2C device. 0.01µF 10k 10k 10k 10k VCC 5V VCC2 10k 10k LTC4315 DISCEN ENABLE FAULT µP READY SCLIN SCLOUT SDAIN SDAOUT ACC GND NON-COMPLIANT I2C DEVICE VOL = 0.6V January 2013 : LT Journal of Analog Innovation | 19 The LTC4315 detects RTA current from other devices and turns off its buffers to prevent contention between its buffers and other RTAs. This permits the LTC4315 to be interoperable with any combination of all older Linear Technology bus buffers, whose RTAs turn on at voltages < 0.3 • VCC. INTEROPERABILITY WITH OTHER LINEAR TECHNOLOGY BUFFERS In large systems older Linear Technology buffers might be present on the same bus with the LTC4315. These older buffers may have RTAs that turn on at voltages below the LTC4315 buffer turn-off voltage of 0.3 • VCC . Glitch-free operation under these circumstances is critical for system integrity. The LTC4315 detects RTA current from other devices at bus voltages below 0.3 • VCC and turns off its buffers to prevent contention between its buffers and other RTAs, to facilitate interoperability. Figure 5 shows the LTC4315 operating in a dynamic system that changes as cards are plugged into or out of the backplane. For simplicity, a single 3.3V supply is chosen and only the SCL pathway is shown. Cards have buffers at their edges in order to shield the I2C devices on the card from the large backplane capacitance and to keep the card capacitances isolated from each other and to aid in hot swapping. The cards in the I/O CARD #1 LTC4300A & LTC4307 RTAs TURN ON SCL3 SCL2 SCL1 3.3V 3.3V 3.3V 0.01µF 5.1k VCC VCC2 LTC4315 SCLIN SCLOUT SCL1 ACC GND Figure 6. SCL waveforms of one LTC4315 operating with three LTC4300As and one LTC4307. 20 | January 2013 : LT Journal of Analog Innovation SCL2 CB2* 690pF LTC4300A SCLIN SCLOUT GND 2.7k SCL3 I/O CARD #5 3.3V *PARASITIC BACKPLANE CAPACITANCE VCC LTC4307 SCLIN SCLOUT GND 5.1k SCL4 BACKPLANE Figure 5. The LTC4315 operating with multiple LTC4300As and LTC4307s in a cascaded application. application shown have LTC4300A or LTC4307 buffers on their edges. The RTAs of these products turn on at 0.6V and 0.8V, respectively, while the LTC4315’s buffers turn off at 0.3 • VCC (~1V). Figures 6–9 track backplane and card SCL waveforms in this system as its configuration changes. Figure 6 shows the SCL waveforms for the system configuration shown in Figure 5, where three LTC4300As and one LTC4307 operate with one LTC4315. In Figure 7, the LTC4307 is LTC4315 BUFFER TURNS OFF 0.5V/DIV LTC4300A RTAs TURN ON SCL3 SCL2 SCL1 500ns/DIV VCC 2.7k CB1 100pF LTC4315 BUFFER TURNS OFF 0.5V/DIV I/O CARD #2 TO #4 LTC4315 BUFFER TURNS OFF 0.5V/DIV LTC4300A RTA TURNS ON SCL3 SCL2 SCL1 500ns/DIV Figure 7. SCL waveforms of one LTC4315 operating with three LTC4300As. 500ns/DIV Figure 8. SCL waveforms of one LTC4315 operating with one LTC4300A. design features BACKPLANE CARD CONNECTOR CONNECTOR I/O PERIPHERAL CARD 1 5V C1 0.01µF 3.3V VCC R1 10k R2 10k R3 10k R4 10k VCC2 R5 10k DISCEN C2 0.01µF R6 10k ACC LTC4315 READY READY FAULT FAULT SCLOUT CARD 1_SCL SCL SCLIN SDAOUT CARD 1_SDA SDA SDAIN ENABLE ENABLE 1 R7 10k GND ••• I/O PERIPHERAL CARD N ••• C3 0.01µF VCC VCC2 R8 10k DISCEN C4 0.01µF R9 10k ACC LTC4315 READY FAULT SCLOUT CARD N_SCL SCLIN SDAOUT CARD N_SDA SDAIN ENABLE ENABLE N R10 10k GND Figure 10. The LTC4315 in an I2C Hot Swap application with staggered pin lengths in the connector. swapped out, leaving three LTC4300As and one LTC4315. In Figure 8, two more LTC4300As are swapped out, leaving one LTC4315 BUFFER TURNS OFF 0.5V/DIV LTC4300A & LTC4307 RTAs TURN ON SCL3 SCL2 SCL1 500ns/DIV Figure 9. SCL waveforms of one LTC4315 operating with one LTC4300A and one LTC4307. LTC4315 and one LTC4300A. Finally in Figure 9, the LTC4307 is reconnected, making the system one LTC4307, one LTC4300A and one LTC4315. The SCL waveforms remain monotonic during the entire sequence of events due to the automatic adjustment of the LTC4315 buffer turn-off voltage in response to varying amounts of LTC4300A and LTC4307 RTA current. Figures 6–9 illustrate the interoperability of the LTC4315 with various combinations of LTC4300As and LTC4307s in a moderately complex system. As a general rule, the LTC4315 is interoperable with any number or combination of older Linear Technology buffers. Nevertheless, given the varying number and variety of buffers that can interact with each other, interoperability cannot be tested and hence guaranteed under all circumstances. Useful guidelines on card capacitances, bus pullup resistances and buffer combinations to ensure interoperability in large systems are provided in the LTC4315 data sheet. HOT SWAP AND CAPACITANCE BUFFERING I/O cards with LTC4315s on their edges can be hot swapped into a live backplane as shown in Figure 10. The corresponding waveforms are shown in Figure 11. Communication at the backplane end is not disrupted during hot plug because January 2013 : LT Journal of Analog Innovation | 21 Circuits on a card that has an LTC4315 on its edge drive only the < 10pF input capacitance of the LTC4315. The LTC4315 drives the large combined capacitance of backplane and all the cards that plug into it. The LTC4315 can drive up to 1.2nF of capacitance on its SDA and SCL pins. This capacitance buffering feature, combined with RTAs, permits 400kHz operation in large systems. the LTC4315’s small input capacitance causes minimal disturbance during connection to the backplane. Furthermore the LTC4315 precharges its clock and data lines to 1V before they contact the backplane, minimizing the voltage step on the backplane bus. The LTC4315 waits for a stop bit or bus idle condition to enable its buffers, ensuring that a partial message is not transmitted across its buffers. When hot plugging into a live backplane, a staggered connector should be used. Make ENABLE the shortest pin with a pull-down resistor to GND on the card, VCC and GND the longest pins and SCL and SDA medium length pins. This ensures that the part is powered up and SDA and SCL pins are precharged to 1V, before they connect to the backplane. Holding ENABLE low during this period ensures correct operation of the stop bit and bus idle circuitry and allows any transients associated with card insertion to settle before the LTC4315 is activated. Figure 11 shows waveforms when the LTC4315 is hot plugged into a live backplane using a staggered connector. VCC and ENABLE SCLIN 2V/DIV PRE-CHARGE SDAIN INVALID STOP BIT(IGNORED) SDAOUT 2.7k 2.7k VALID STOP BIT 500µs/DIV Figure 11. Waveforms during an LTC4315 Hot Swap event into a live backplane using a staggered connector. VCC2, as the longest pins, have already contacted the backplane and are powering the LTC4315 and the output buses. At this time SDAIN and SCLIN are precharged to 1V by the LTC4315. Once SDAIN and SCLIN contact the backplane, they are driven by backplane circuitry. Stop bits at the input are ignored by the LTC4315 as ENABLE is low. The outputs of the LTC4315 idle high (SCLOUT not shown), until a stop bit is detected at the input after ENABLE has been asserted high and is stable. The LTC4315 3.3V 0.01µF buffers turn on at this time and establish a connection between the input and output. Partial messages are not propagated across the LTC4315. If a staggered connector is not used, ENABLE should be held low until all transients associated with card insertion into a live system die out. CONNECTOR BOUNCE 5V VCC VCC2 10k 1.3k 1.3k 10k DISCEN ENABLE LTC4315 READY READY SCL1 SCLIN SCLOUT SCL2 SDA1 SDAIN SDAOUT SDA2 FAULT ACC Figure 12. The LTC4315 in a level translating application. 22 | January 2013 : LT Journal of Analog Innovation GND FAULT Circuits on a card that has an LTC4315 on its edge drive only the < 10pF input capacitance of the LTC4315. The LTC4315 drives the large combined capacitance of the backplane and all the cards that plug into it. The LTC4315 can drive up to 1.2nF of capacitance on its SDA and SCL pins. This capacitance buffering feature, combined with RTAs, permits 400kHz operation in large systems. RISE TIME ACCELERATORS The RTAs of the LTC4315 can be configured either in the current source mode (ACC open), slew limited switch mode (ACC grounded) or disabled (ACC high). In the current source mode the RTAs source a constant 2.5m A current into the bus. In the slew controlled switch mode, the RTAs turn on in a controlled manner and source current into the buses, making them rise at a typical rate of 40V/µs. To selectively disable RTAs only on the outputs, ground VCC2 and either ground ACC or leave ACC open. The LTC4313 comes with 3 different versions of RTAs. The LTC4313-1 RTAs are slew controlled switches, the LTC4313-2 RTAs are 2.5m A current sources and the LTC4313-3 has no RTAs. design features The LTC4315 and LTC4313 disconnect stuck buses and allow I/O cards to be hot swapped into and out of live systems. They level translate signals down to 1.4V and provide user-selectable RTA current that permits operation at frequencies up to 1MHz. LEVEL TRANSLATION The circuit shown in Figure 12 illustrates the level translation feature of the LTC4315. The operating ranges for the LTC4315 supplies are VCC from 2.9V–5.5V and VCC2 from 2.25V–5.5V. Tying the input bus to VCC and the output bus to VCC2 permits level translation between 2.9V–5.5V inputs and 2.25V–5.5V outputs. The example shown in Figure 12 translates a 3.3V input to a 5V output. Level translation to voltages lower than the minimum allowed VCC and VCC2 values imposes other constraints. Level translation to output voltages less than 2.25V requires VCC2 to be tied low to disable output RTAs. Level translation to input voltages less than 2.9V requires all RTAs to be disabled by tying ACC high for the LTC4315 or using the LTC4313-3. This prevents overdriving of the input bus by the RTA. Under these conditions, level translation to a bus voltage of 1.4V is possible. The buffer turn-off voltage in both cases is 0.3 • VCC and a high logic-low noise margin is maintained. STUCK BUS DETECTION AND RECOVERY Occasionally, slave devices get confused and get stuck in a low state. The LTC4315 monitors the output I2C bus to see if clock and data have been simultaneously high at least once in 45ms. If this condition is not detected, the LTC4315 asserts the FAULT flag low. If DISCEN is tied high, the LTC4315 also disconnects the input and output sides and generates clock pulses on SCLOUT in an attempt to free the stuck bus. Clocking READY 5V/DIV FAULT 5V/DIV AUTOMATIC CLOCKING SCLOUT 5V/DIV SDAIN 5V/DIV SDAOUT 5V/DIV STOP BIT GENERATED DISCONNECT AT TIMEOUT RECOVERS HIGH STUCK LOW > 45ms If automatic stuck bus disconnection is not desired, this feature can be disabled in the LTC4315 by tying DISCEN low. In this case, during a stuck bus event, the FAULT flag is asserted low, but no stop bit or clock generation occurs and the input and output sides stay connected. Stuck bus disconnection and output clocking cannot be disabled in the LTC4313. DRIVEN LOW 1ms/DIV Figure 13. Bus waveforms during an SDAOUT stuck low and recovery event. is stopped when data releases high or 16 clocks have been generated. After the final clock pulse, a stop bit is generated to reset the bus for further communication. When a stuck bus releases high, connection is reestablished when a stop bit or bus idle condition is detected on both buses. No user intervention is required. Figure 13 shows the waveforms during an SDAOUT stuck low and recovery event with DISCEN tied high. In Figure 13, the FAULT flag is asserted low after the 45ms timeout period and the input and output sides are disconnected. This causes SDAIN to release high. Clock pulses are generated on SCLOUT. SDAOUT releases high before 16 clock pulses have been generated. Clock pulsing is stopped and a stop bit is generated. As SDAOUT recovers and a stop bit is detected, connection is reestablished and signals propagate from the input to the output. If SDAOUT stays low, an input to output connection can be forced by toggling ENABLE low, then high. CONCLUSION The LTC4315 and LTC4313 are high noise margin bus buffers that solve a number of problems associated with large I2C systems. They provide capacitance buffering, level translation for bus supplies ranging from 1.4V to 5.5V, high logic-low noise margins up to 0.3 • VCC and reject noise above 0.3 • VCC when the bus is a logic high. Their high bandwidth buffers and integrated RTAs enable operation at frequencies up to 1MHz. The buffers can drive noncompliant buses with parasitic capacitance as large as 1.2nF. They disconnect stuck buses and allow I/O cards to be hot swapped into and out of live systems. These buffers are interoperable with noncompliant I2C devices that drive a high VOL and with legacy buffers whose RTAs turn on at low voltages. The LTC4315 and LTC4313 ease practical design issues associated with large I2C bus systems. n January 2013 : LT Journal of Analog Innovation | 23 Ideal Diode and Hot Swap Controller Enables Supply Redundancy and Isolates Faults Chew Lye Huat Schottky diodes are used in a variety of ways to implement multisource power systems. For instance, high availability electronic systems—such as µTCA network and storage servers—employ power Schottky diode-OR circuits in redundant power systems. Diode ORing is also used in systems with alternate power sources, such as an AC wall adapter and a backup battery feed. The problem is that the Schottky diodes consume power due to the forward voltage drop—the resulting heat must be dissipated with dedicated copper area on the PCB, or by heat sinks bolted to the diode, both of which require significant space. The family of products comprising the LTC4225, LTC4227 and LTC4228 minimize power loss by using external N-channel MOSFETs for pass elements, minimizing the voltage drop from the supply to the load when the MOSFETs are turned on. When an input source voltage drops below the output common supply voltage, the appropriate MOSFET is turned off, thereby matching the function and performance of an ideal diode. As shown in Figure 1, by adding a current sense resistor and configuring two MOSFETs back-to-back with separate gate control, the LTC4225 enhances the ideal diode performance with inrush current limiting and overcurrent protection. This allows the boards to be safely inserted and removed from a live backplane without damaging the connector. The LTC4227 can be used with the current sense resistor and the Hot Swap MOSFET added 24 | January 2013 : LT Journal of Analog Innovation Figure 1. An overview of different configurations with sense resistor and external N-channel MOSFETs for the LTC4225, LTC4227 and LTC4228 VOUT1 VIN1 IN1 SENSE1 DGATE1 HGATE1 OUT1 LTC4225* IN2 SENSE2 DGATE2 HGATE2 OUT2 VIN2 VOUT2 VIN1 VOUT VIN2 IN1 DGATE1 IN2 DGATE2 SENSE+ SENSE– HGATE OUT LTC4227* VIN1 VOUT1 IN1 DGATE1 SENSE1+ SENSE1– HGATE1 OUT1 IN2 DGATE2 SENSE2+ SENSE2– HGATE2 OUT2 LTC4228* VIN2 VOUT2 *ADDITIONAL DETAILS OMITTED FOR CLARITY after the parallel-connected ideal diode MOSFET to save one MOSFET. By configuring the sense resistor between the ideal diode and Hot Swap MOSFET, the LTC4228 improves on the LTC4225 by recovering more quickly from input brownouts to preserve the output voltage. The LTC4225-1, LTC4227-1 and LTC4228-1 feature a latchoff circuit breaker, while the LTC4225-2, LTC4227-2 and LTC4228-2 provide automatic retry after a fault. Both options are available in 24-pin, 20-pin and 28-pin 4mm × 5mm QFN and SSOP packages for LTC4225, LTC4227 and LTC4228, respectively. IDEAL DIODE CONTROL The LTC4225 and LTC4228 function as an ideal diode by monitoring the voltage between IN and OUT pins (IN and SENSE+ pins for LTC4227) with an internal gate drive amplifier, which drives the DGATE pin. The amplifier quickly pulls up the DGATE pin, turning on the MOSFET for ideal diode control, when it senses a large forward voltage drop (Figure 2). An external capacitor connected between the CPO and IN pins provides the charge needed to quickly turn on the ideal diode MOSFET. An internal charge pump charges up this capacitor at device power-up. design features The LTC4225, LTC4227 and LTC4228 minimize power loss by using external N-channel MOSFETs for pass elements, minimizing the voltage drop from the supply to the load when the MOSFETs are turned on. When an input source voltage drops below the output common supply voltage, the appropriate MOSFET is turned off, thereby matching the function and performance of an ideal diode. ON 5V/DIV CPO 10V/DIV HGATE 10V/DIV OUT 10V/DIV DGATE 10V/DIV OUT 10V/DIV PWRGD 10V/DIV 20ms/DIV HOT SWAP CONTROL 50ms/DIV Figure 2. Ideal diode controller CPO and DGATE pull up when IN supply turns on Figure 3. Hot Swap controller HGATE starts up and PWRGD pulls low after 100ms delay when ON toggles high The DGATE pin sources current from the CPO pin and sinks current into the IN and GND pins. The gate drive amplifier controls DGATE to servo the forward voltage drop across the sense resistor and the two external N-channel MOSFETs to 25mV. Figure 4. The LTC4225 in a µTCA application to supply 12V power to two µTCA slots VIN1 12V If the load current causes more than 25mV of voltage drop, the gate voltage RS1 0.004Ω BULK SUPPLY BYPASS CAPACITOR R2 137k R1 20k CF1 10nF R4 137k CF2 10nF CCP1 0.1µF IN1 SENSE1 DGATE1 LTC4225 ON2 CPO2 IN2 SENSE2 DGATE2 CCP2 0.1µF VIN2 12V HGATE1 INTVCC GND BULK SUPPLY BYPASS CAPACITOR HGATE2 RH2 10Ω RS2 0.004Ω PLUG-IN CARD 1 MH1 MD1 Si7336ADP Si7336ADP RH1 10Ω CPO1 ON1 C1 0.1µF R3 20k Pulling the ON pin high and the EN pin low initiates a 100ms debounce timing cycle. After this timing cycle, a 10µ A current from the charge pump ramps up the HGATE pin. When the Hot Swap MOSFET turns on, the inrush current is limited at a level set by an external sense resistor connected between the IN and SENSE pins for LTC4225 (SENSE+ and MH2 MD2 Si7336ADP Si7336ADP 12V 7.6A RHG1 47Ω CHG1 15nF OUT1 FAULT1 PWRGD1 EN1 TMR1 TMR2 EN2 PWRGD2 FAULT2 OUT2 RHG2 47Ω CHG2 15nF + CL1 1600µF VIN1 R5 100k R6 100k CT1 47nF CT2 47nF R7 100k PLUG-IN CARD 2 R8 100k VIN2 + IN 10V/DIV rises to enhance the MOSFET used for ideal diode control. In the case of an input supply short-circuit when the MOSFETs are conducting, a large reverse current starts flowing from the load toward the input. The gate drive amplifier detects this failure condition as soon as it appears and turns off the ideal diode MOSFET by pulling down the DGATE. CL2 1600µF 12V 7.6A BACKPLANE January 2013 : LT Journal of Analog Innovation | 25 If the main supply loses power, the controller reacts quickly to turn off the ideal diode MOSFET in the main supply path and turn on the MOSFET in the redundant supply path, providing a smooth supply switchover to the output load. The Hot Swap MOSFETs remain on so they do not affect the supply switchover. SENSE– pins for LTC4227 and LTC4228). An active current limit amplifier servos the gate of the MOSFET so that 65mV appears across the current sense resistor. If the sense voltage exceeds 50mV for more than a fault-filter delay configured at the TMR pin, a circuit breaker trips and pulls HGATE low. Inrush current can be further reduced, if desired, by adding a capacitor from HGATE to GND. When the MOSFET’s gate overdrive (HGATE to OUT voltage) exceeds 4.2V, the PWRGD pin pulls low (Figure 3). powering down the system. The LTC4225 and LTC4228, which both include dual ideal diode and Hot Swap controllers, are ideal for these applications—they provide smooth supply switchover between two supplies and overcurrent protection. If the main supply loses power, the controller reacts quickly to turn off the ideal diode MOSFET in the main supply path and turn on the MOSFET in the redundant supply path, providing a smooth supply switchover to the output load. The Hot Swap MOSFETs remain on so they do not affect the supply switchover. The controller turns off a Hot Swap MOSFET when the respective ON pin is pulled low or EN pin is pulled high. When an overcurrent fault is detected at the output, the gate of the Hot Swap MOSFET is pulled down quickly, COMBINING THE IDEAL DIODE AND HOT SWAP CONTROL In a typical µTCA application with redundant supplies (Figures 4 and 9), the outputs are diode-ORed at the backplane, so cards can be removed or inserted without Figure 5. LTC4225 for 2-channel power prioritizer with IN1 as the prioritizing input 5V PRIMARY SUPPLY RS1 0.006Ω INPUT 1 R1 20k C1 R4 0.1µF 41.2k In a traditional diode-ORed multisupply system, the input supply with the higher voltage is passed to the output, while the lower voltage supply is shut out. This simple solution satisfies the needs of applications where the priority of the supplies is not simply a matter of the higher voltage supply winning. Figure 5 shows a backup supply system where the 5V primary supply (INPUT1) is passed to the output whenever it is available, while the 12V backup MH1 SiR466DP RH1 10Ω CCP1 0.1µF OUT1 FAULT1 ON1 PWRGD1 IN1 SENSE1 DGATE1 HGATE1 INTVCC TMR1 TMR2 LTC4225 GND ON2 PWRGD2 FAULT2 EN2 CPO2 + Z2 SMAJ13A BV=14.4V R3 3.92k 26 | January 2013 : LT Journal of Analog Innovation IN2 SENSE2 DGATE2 HGATE2 CCP2 0.1µF INPUT 2 12V BACKUP SUPPLY RHG1 47Ω CHG1 33nF EN1 CPO1 CF1 0.1µF PRIORITIZING A POWER SUPPLY + Z1 SMAJ13A BV=14.4V R2 49.9k MD1 SiR466DP after which the output is regulated in current limit until the fault filter delay set by the TMR pin capacitor times out. The Hot Swap MOSFET is turned off and the FAULT pin is latched-low to indicate a fault. The electronic circuit breaker is reset by pulling the ON pin below 0.6V. RS2 0.006Ω D1 LS4148 MD2 SiR466DP MH2 SiR466DP OUT2 CT2 47nF CL 470µF CT1 47nF VOUT 5A design features In a typical µTCA application with redundant supplies, the outputs are diodeORed at the backplane, so cards can be removed or inserted without powering down the system. The LTC4225 and LTC4228, which both include dual ideal diode and Hot Swap controllers, are ideal for these applications—they provide smooth supply switchover between two supplies and overcurrent protection. RS1 0.006Ω BULK SUPPLY BYPASS CAPACITOR RH1 10Ω CPO1 PWREN2 IN1 LTC4225 GND IN2 SENSE2 HGATE2 supply (INPUT2) is called on only when the primary supply fails to deliver. As long as INPUT1 is above the 4.3V UV threshold set by the R1-R2 divider at the ON1 pin, MH1 is turned on, connecting INPUT1 to the output. When MH1 is on, PWRGD1 goes low, which in turn pulls ON2 low and disables the IN2 path by turning MH2 off. If the primary supply fails and INPUT1 drops below 4.3V, ON1 turns off MH1 and PWRGD1 goes high, allowing ON2 to turn on MH2 and connect the INPUT2 to the output. The ideal diode MOSFETs MD1 and MD2 prevent backfeeding of one input to the other under any condition. RH2 10Ω OUT1 FAULT1 PWRGD1 EN1 TMR1 TMR2 CT2 47nF CT1 47nF PLUG-IN CARD 2 RHG2 47Ω CHG2 15nF ZH2 ZD2 BULK SUPPLY BYPASS CAPACITOR RS2 0.006Ω CL1 1000µF EN2 PWRGD2 FAULT2 DGATE2 OUT2 ON2 CCP2 0.1µF VIN2 12V DGATE1 SENSE1 HGATE1 INTVCC CPO2 Figure 6. LTC4225 for application with the Hot Swap MOSFET on the supply side and the ideal diode MOSFET on the load side + RHG1 47Ω CHG1 15nF ON1 C1 0.1µF 12V 5A ZH1 ZD1 CCP1 0.1µF PWREN1 PLUG-IN CARD 1 MD1 SiR466DP + VIN1 12V MH1 SiR466DP MH2 SiR466DP MD2 SiR466DP CL2 1000µF 12V 5A BACKPLANE ZH1, ZD1, ZH2, ZD2: CMHZ4706, BV=19V SWAPPING THE DIODE AND HOT SWAP FET ON SUPPLY AND LOAD SIDE The LTC4225 allows applications with back-to-back MOSFETs to be configured with the MOSFET on the supply side as the ideal diode and the MOSFET on the load side as the Hot Swap control (Figure 4) or vice versa (Figure 6). In Figure 6, an external Zener diode clamp may be required between the GATE and SOURCE pins of the MOSFET to prevent it from breaking down if the MOSFET’s gate-to-source voltage is rated for less than 20V. In either arrangement, LTC4225 smoothly switches between supplies with its ideal diode ORing between the IN and OUT pins. DUAL IDEAL DIODE AND SINGLE HOT SWAP CONTROL Figure 7 shows a LTC4227 application where the sense resistor is placed after dual supply ideal diode MOSFETs connected in parallel, which is then followed by a single Hot Swap MOSFET. Here, the LTC4227 regulates an overloaded output at 1× the current limit before fault timeout, instead of 2×, as in the LTC4225 diode-OR application. As a result, power dissipation is reduced during an overload condition. The LTC4227 also features the D2ON pin, which allows the IN1 supply to be easily prioritized. For example, Figure 8 shows a simple resistive divider connecting IN1 to the D2ON pin, so that the January 2013 : LT Journal of Analog Innovation | 27 Tight 5% circuit breaker threshold accuracy and fast acting current limit protect the supplies against overcurrent faults. The LTC4228’s fast recovery from input brownouts preserves the output voltage in the face of such events. Figure 7. LTC4227 for card-resident diode-OR application with Hot Swap control MD1 SiR462DP VIN1 12V Z1 SMAJ13A BV=14.4V VIN2 12V CCP1 0.1µF MD2 SiR462DP RS 0.006Ω MH Si7336ADP + Z2 SMAJ13A BV=14.4V RH 10Ω CCP2 0.1µF CL 680µF 12V 7.6A RHG 47Ω CHG 15nF R2 137k R1 20k CF 10nF CPO1 ON INTVCC OUT D2ON R3 100k R4 100k FAULT PWRGD TMR GND CT 0.1µF C1 0.1µF CARD CONNECTOR FASTER OUTPUT RECOVERY FROM INPUT COLLAPSE IN1 supply is prioritized until IN1 falls below 2.8V, wherein MD2 is turned on and the diode-OR output is switched from the main 3.3V supply at IN1 to the auxiliary 3.3V supply at IN2. SENSE– HGATE LTC4227 EN BACKPLANE CONNECTOR IN2 DGATE2 SENSE+ IN1 DGATE1 CPO2 while the other supply is not available, HGATE is pulled low to turn off the Hot Swap MOSFET as the IN supply drops below the undervoltage lockout threshold. When the input supply recovers, HGATE is allowed to start up to In the LTC4225 µTCA application shown in Figure 4, if one of the input supplies collapses to ground momentarily Figure 8. Plug-in card IN1 supply controls the IN2 supply turn-on via D20N of LTC4227 MD1 SiR462DP VMAIN 3.3V Z1 SMAJ7A BV=7.78V VAUX 3.3V CCP1 0.1µF Z2 SMAJ7A BV=7.78V R2 22.1k R1 20k CPO1 CF1 0.1µF CARD CONNECTOR 28 | January 2013 : LT Journal of Analog Innovation R6 28.7k R5 20k MH Si7336ADP + IN1 DGATE1 CPO2 IN2 DGATE2 SENSE+ SENSE– HGATE OUT FAULT PWRGD ON LTC4227 D2ON INTVCC C1 0.1µF CF2 10nF RS 0.008Ω CCP2 0.1µF EN BACKPLANE CONNECTOR MD2 SiR462DP GND TMR CT 0.1µF R3 10k R4 10k CL 100µF 3.3V 5A design features The LTC4225, LTC4227 and LTC4228 enable ideal diode and Hot Swap functions for two power rails by controlling external N-channel MOSFETs. They feature fast reverse turn-off, smooth supply switchover, active current limit and status and fault reporting. turn on the MOSFET. As it takes a while to charge up HGATE and the depleted output capacitance, the output voltage may brown out during this period. This prevents the SENSE+ voltage from entering into undervoltage lockout and turning off the Hot Swap MOSFET. As the input supply recovers, it charges up the depleted load capacitance and instantly provides power to the downstream load, since the Hot Swap MOSFET remains on. In this situation, the LTC4228 offers an advantage over the LTC4225 by recovering more quickly to preserve the output voltage. As shown in Figure 9, the sense resistor is placed in between the ideal diode and Hot Swap MOSFET, allowing the SENSE+ pin voltage to be held up by the output load capacitance temporarily when the input supply collapses. supply switchover, active current limit and status and fault reporting. Their tight 5% circuit breaker threshold accuracy and fast acting current limit protect the supplies against overcurrent faults. The LTC4228’s fast recovery from input brownouts preserves the output voltage in the face of such events. n CONCLUSION The LTC4225, LTC4227 and LTC4228 enable ideal diode and Hot Swap functions for two power rails by controlling external N-channel MOSFETs. They feature fast reverse turn-off, smooth Figure 9. LTC4228 for µTCA application to supply 12V power to two µTCA slots MD1 Si7336ADP BULK SUPPLY BYPASS CAPACITOR RH1 10Ω CCP1 0.1µF CPO1 IN1 DGATE1 R1 20k R3 20k R4 137k CF2 10nF INTVCC LTC4228 GND OUT1 VSENSE1+ R5 100k R6 100k CL1 1600µF R7 100k CT1 47nF CT2 47nF + PLUG-IN CARD 2 EN2 PWRGD2 FAULT2 STATUS2 ON2 CPO2 IN2 DGATE2 SENSE2+ SENSE2– HGATE2 RH2 10Ω CCP2 0.1µF VIN2 12V RHG1 47Ω CHG1 15nF STATUS1 FAULT1 PWRGD1 EN1 TMR1 TMR2 ON1 C1 0.1µF 12V 7.6A SENSE1+ SENSE1– HGATE1 R2 137k CF1 10nF PLUG-IN CARD 1 MH1 Si7336ADP BULK SUPPLY BYPASS CAPACITOR MD2 Si7336ADP RS2 0.004Ω MH2 Si7336ADP OUT2 RHG2 47Ω CHG2 15nF R8 100k R9 100k R10 100k + VIN1 12V RS1 0.004Ω VSENSE2+ CL2 1600µF 12V 7.6A BACKPLANE January 2013 : LT Journal of Analog Innovation | 29 What’s New with LTspice IV? Gabino Alonso Follow @LTspice on Twitter for up-to-date information on models, demo circuits, events and user tips: www.twitter.com/LTspice DEMO CIRCUITS Step-Down Regulators • LT®3641: Dual high voltage buck with POR and WDT (7V–42V to 5V at 1A & 1.8V at 0.8A) www.linear.com/LT3641 • LT3976: 3.3V Step-down converter (4.3V–42V to 3.3V at 5A) www.linear.com/LT3976 • LT8300: 100V µ Power isolated flyback converter (22V–75V to 5V at 0.25A) www.linear.com/LT8300 LED Drivers, Battery Chargers and Negative & Inverting Regulators • LT3791: 98% efficient 100W buck-boost LED driver (15V–58V to 33V LED at 3A) www.linear.com/LT3791 • LT3959: Wide input voltage range boost converter (2V–10V to 12V at 0.5V–2A) www.linear.com/LT3959 • LT8611: Negative converter with 1A output current limit (3.8V–42V to –3.3V at 1A) www.linear.com/LT8611 • LT8300: 100V input µ Power isolated flyback converter with 150V/260m A switch www.linear.com/LT8300 • LTC3122: 15V, 2.5A synchronous step-up DC/DC converter with output disconnect www.linear.com/LTC3122 • LTC3633A: Dual channel 3A, 20V monolithic synchronous step-down regulator www.linear.com/LTC3633A • LT8611: CCCV Li-ion battery charger (3.8V–42V to 4.1V at 1A) www.linear.com/LT8611 • LTC3861-1: Dual, multiphase step-down voltage mode DC/DC controller with accurate current sharing www.linear.com/LTC3861-1 NEW MODELS Linear Regulators • LTC3839: Fast transient step-down converter with differential RSENSE sensing (4.5V–14V to 1.5V at 40A) www.linear.com/LTC3839 Switching Regulators • LTC3026-1: 1.5A low input voltage VLDO™ linear regulator www.linear.com/LTC3026-1 • LTC3861: High current, dual output synchronous buck converter with DCR current sensing (4V–14V to 1.2V at 25A & 1.8V at 25A) www.linear.com/LTC3861 • LT3791-1: 60V input 4-switch synchronous buck-boost controller www.linear.com/LT3791-1 • LTC3626: 2.5V, 1MHz step-down converter with average input current limit & monitor (3.6V–20V to 2.5V at 2.5A) www.linear.com/LTC3626 • LTM®8029: µ Power high voltage buck converter (5.6V–36V to 5V at 600m A) www.linear.com/LTM8029 • LT3761: 60V input LED controller with internal PWM generator www.linear.com/LT3761 • LT3959: Wide input voltage range boost/ SEPIC/inverting converter with 6A, 40V switch www.linear.com/LT3959 What is LTspice IV? LTspice® IV is a high performance SPICE simulator, schematic capture and waveform viewer designed to speed the process of power supply design. LTspice IV adds enhancements and models to SPICE, significantly reducing simulation time compared to typical SPICE simulators, allowing one to view waveforms for most switching regulators in minutes compared to hours for other SPICE simulators. LTspice IV is available free from Linear Technology at www.linear.com/LTspice. Included in the download is a complete working version of LTspice IV, macro models for Linear Technology’s power products, over 200 op amp models, as well as models for resistors, transistors and MOSFETs. 30 | January 2013 : LT Journal of Analog Innovation Overvoltage & Overcurrent Protection and Timing • LT4363-1: High voltage surge stopper with current limit www.linear.com/LT4363 • LTC4359: Ideal diode controller with reverse input protection www.linear.com/LTC4359 • LTC6905: 17MHz to 170MHz resistor set SOTV–23 oscillator www.linear.com/LTC6905 • LT3957A: Boost, flyback, SEPIC and inverting converter with 5A, 40V switch www.linear.com/LT3957A n design ideas Power User Tip IMPORTING AND EXPORTING DATA IN LTSPICE IV The LTspice IV waveform viewer is a handy way to perform basic measurements, but there are times when you need to export data from, or import data into LTspice to further evaluate a circuit. To export waveform data to an ACSII text file: The imported file must contain a list of two-dimensional points that represent time and value data pairs in a tab or comma delimited format—with no header information. The PWL function connects the dots in the data, constructing a waveform based on straight-line segments between the points defined in the text file. 1.Click to select the waveform viewer 2.Choose Export from the File menu. 3.Select the traces you want exported 4.Click browse to specifiy the file location and name to save the text file. Once this file is created you can analyze it further in applications like Microsoft Excel or MATLAB. Note that some applications like MATLAB expect imported files to contain only data, with no header information. If you need to remove the header, open the text file in a text editor or Excel and delete the header information. To import waveform data into LTspice IV you must attach a text file as a piecewise linear (PWL) function in a voltage or current source. To add a text file as a PWL function to a voltage or current source: 1.Right-click the symbol in the schematic editor 2.Choose advance 3.Select PWL FILE: and click Browse to choose the text file. The PWL statement is discussed in more depth in the previous issue of this magazine at cds.linear.com/docs/LT%20Journal/LTJournal-V22N3-2012-10.pdf. Happy simulations! January 2013 : LT Journal of Analog Innovation | 31 20V, 2.5A Synchronous Monolithic Buck with Current and Temperature Monitoring K. Bassett The LTC3626 integrates a number of easyto-use, but powerful, features that would normally require additional ICs and design time to implement. Specifically, with the addition of just a couple of passive components, the LTC3626 can be configured to provide accurate measures of its output current, input current, and on-die temperature. It can be just as easily programmed to limit each measured parameter. These built-in features expand the designer’s insight into the performance of the system and increase the level of control with remarkably little extra design investment. Additionally, optional internal loop compensation is available to minimize the design effort. 32 | January 2013 : LT Journal of Analog Innovation The LTC3626 also includes userselectable Burst Mode operation or forced-continuous mode, resistorprogrammable switching frequencies from 500kHz to 3MHz, power good status output, output tracking capability, and external clock synchronization. CURRENT MONITOR AND LIMIT One way to measure the overall performance of a system is to is to monitor the current at the output of the power supply. Supply current monitoring also informs designers if downsteam ICs are operating as expected—useful in design and debug, and during normal operation. The LTC3626 makes it easy to monitor the supply current by producing a fraction of its average output current at its IMONOUT pin, specifically, the current at the IMONOUT pin is equal to the average output current divided by 16,000. Figure 1 shows the typical performance of the output current measurement for an ambient temperature range of –40°C to 85°C. Figure 2 shows the error between the actual average output CALCULATED OUTPUT CURRENT, IMONOUT • 16000 (A) 156 VIN = 12V VOUT = 1.8V fO = 1MHz 2.25 2.00 125 1.75 1.50 94 1.25 1.00 TA = 85°C TA = 25°C TA = –40°C 0.75 63 0.50 31 0.5 0.75 1.0 1.25 1.5 1.75 2.0 2.25 2.5 OUTPUT CURRENT (A) Figure 1. Output current monitor vs output current current and the average output current as measured by the LTC3626. The current at the IMONOUT pin can be measured directly or converted to a voltage by placing a resistor from the IMONOUT pin to ground. Converting the output of the IMONOUT pin to a voltage makes it easy to scale the output for digitization via a microcontroller or standalone ADC. Figure 3 shows the LTC3626 configured to run with Figure 2. Output current monitor error vs output current MEASURED OUTPUT CURRENT ERROR (%) The LTC3626 is capable of supplying 2.5A of output current over an input voltage range of 3.6V to 20V from a tiny, 3mm × 4mm, 20-pin QFN package. Its patented controlled on-time architecture yields outstanding transient response and enables high step-down ratios at high switching frequencies, minimizing board footprint. 2.50 5 VIN = 12V VOUT = 1.8V fO = 1MHz 4 3 2 1 0 –1 –2 –3 TA = 85°C TA = 25°C TA = –40°C –4 –5 1 1.25 1.75 2 2.25 1.5 OUTPUT CURRENT (A) 2.5 IMONOUT CURRENT (µA) Increases in digital IC integration, coupled with advances in printed circuit board layout and assembly techniques, continue to push system performance and power density higher. Many of these systems, powered from a 12V rail or battery stack, utilize point-of-load regulators to maximize power chain efficiency while maintaining a small form factor. The LTC3626 synchronous, monolithic step-down regulator is ideally suited for these operating environments, given its ability to provide a flexible, highly efficient DC/DC conversion while occupying a very small footprint. design ideas VIN 12V C1 47µF 0.1µF C4 2.2µF 0.1µF RPGD 200k 0.1µF REFOUT COMP VCC IN CIOUT 1µF LTC2460 GND RIOUT 5.1k BOOST CBST 0.1µF L1 1.5µH LTC3626 INTVCC SW ITH TRACK/SS VON TSET TMON FB IMONIN RT PGOOD IMONOUT MODE/SYNC SGND R1 40.2k RT 324k PGND CF 22pF VOUT 1.8V COUT 2.5A 47µF R2 20k REF– Figure 3. 12V input to 1.8V output, 2.5A regulator with digital output current monitoring is useful for applications that must limit the average current drawn from the input supply. Figure 4 shows the LTC3626 configured to limit the average input current to 475m A while producing an output voltage of 2.5V from a 5V input voltage. the output current monitor activated while the LTC2460, 16-bit ADC, digitizes the result for digital processing. The LTC3626 also features an easily programmed average output current limit. Specifically, the LTC3626 contains an onchip current limit amplifier with a reference of approximately 1.2V. To program an average output current, simply size the resistor from IMONOUT to ground such that the resultant voltage is 1.2V for the current at which the limit should be activated. TEMPERATURE MONITOR AND LIMIT The LTC3626 produces an estimate of the on-die temperature at the TMON pin. This feature can be used to determine the quality of the ground connection to the QFN exposed pad made during assembly. The exposed pad for the QFN is intended to provide a low impedance electrical connection to the board as well as good thermal contact. Visual inspection of this critical connection can be difficult, and a poor exposed pad connection may not be apparent by simple observation of the regulated output voltage even though the on-die temperature may be far Similar to the average output current, the LTC3626 produces an estimate of the average input current at the IMONIN pin. That is, the current at the IMONIN pin is an estimate of the average input current divided by 16,000. Just like the average output current, the LTC3626 offers a simple mechanism to program a limit for the average input current. This feature too high for reliable, long-term part operation. Measurement of the TMON pin however gives the user insight into the exposed pad connection and hence the internal part operating environment. As an example, Figure 5 shows data taken on two parts, one with a good exposed pad connection to the PCB, the other with a poor exposed pad connection. Though both parts regulate to the expected output voltage, it is clear from the internal temperature measurement that the internal operating environment is very different between the two parts. If placed in a system with an ambient operating temperature of say 70°C, the device with the poor exposed pad connection will clearly exceed the maximum allowed junction temperature of 125°C and will thus have compromised long-term reliability. CONCLUSION The continuous push for higher performance and power density faced by today’s system designers require small, flexible, and efficient point-of-load converters to maximize overall power chain efficiency. The LTC3626’s combination of wide input voltage range, output current capability, flexible feature set, and very small form factor make it ideal for many of today’s point-of-load regulator applications. n Figure 5. It is easy to determine the quality of the exposed pad connection by examining temperature measurements made by the LTC3626. Figure 4. 5V Input to 2.5V output at 1MHz synchronized frequency with input current monitor and 475mA input current limit 80 VIN 5V C4 2.2µF RCOMP 13k CCOMP 220pF RIIN 40.2k C1 47µF RPGD 100k CIIN 1µF PVIN SVIN RUN BOOST LTC3626 TRACK/SS INTVCC RT SW TSET IMONOUT VON TMON PGOOD FB ITH IMONIN MODE/SYNC SGND PGND CBST 0.1µF L1 2.2µH R1 127k R2 40.2k EXTERNAL CLOCK CF 22pF COUT 47µF VOUT 2.5V 2.5A MEASURED ON-DIE TEMPERATURE (°C) SCK SDO CS PVIN SVIN RUN C2 1µF POOR EXPOSED CONNECTION 60 40 20 0 NORMAL EXPOSED CONNECTION 0 5 10 15 ILOAD (A) 20 25 January 2013 : LT Journal of Analog Innovation | 33 Sub-Milliohm DCR Current Sensing with Accurate Multiphase Current Sharing for High Current Power Supplies Muthu Subramanian, Tuan Nguyen and Theo Phillips The increasing functional complexity of electronic devices, combined with the desire for higher microprocessor computational speed and the quest for eco-friendly electronics, places stringent requirements on power supplies. High current supplies are expected to operate at top efficiency. In order to minimize conduction losses, power supplies are placed closer to the load, and multiple power stages are used on the same board. Individual power stages have had to shrink in size to fit the available board area. To achieve the best performance per board area, controllers must work with external power stages such as power blocks, DrMOS or external gate drivers with MOSFETs. SS VIN 7V TO 14V 20k 4.22k 40.2k VOUT LTC3861 VSNSOUT1 COMP2 FB2 SS CLKIN 500kHz EXTERNAL SYNC INPUT 1Ω 2.2µF 16V 53.6k RUN1 ILIM1 SGND ISNS1P ISNS1N ISNS2N ISNS2P SGND ILIM2 RUN2 VCC VIN RUN VIN VIN SS VCC RUN1 ILIM1 SGND ISNS1P ISNS1N ISNS2N ISNS2P SGND ILIM2 RUN2 34k 34 | January 2013 : LT Journal of Analog Innovation VIN RUN SS COUT2 : SANYO 2R5TPE330M9 COUT1 : MURATA GRM32ER60J107ME20 L1, L2, L3, L4 : COILCRAFT XAL1010-221ME VCC BOOT PHASE V FDMF6707B IN DISB VSWH PWM VDRV PGND VCIN SMOD CGND 0.22µF 2.87k L2 0.22µH 10k 0.22µF CIN4 22µF × 2 10k 16V 1Ω 2.2µF 16V 53.6k SS2 FREQ CLKIN CLKOUT PHSMD PGOOD2 PWMEN2 PWM2 VCC FB1 COMP1 VSNSP1,2 VSNSN1,2 VSNSOUT1,2 LTC3861 COMP2 FB2 2.87k 0.22µF 2.2µF 16V VCC RUN VCC SS1 VINSNS CONFIG IAVG PGOOD1 PWMEN1 PWM1 Figure 1. 4-phase, VIN =12V, VOUT = 0.9V/120A, step-down converter with DrMOS, fSW = 500kHz L1 0.22µH 10k 0.22µF VCC 34k 100pF 1µF IN VSWH DISB PWM PGND VDRV VCIN SMOD CGND 2.2µF 16V CIN3 22µF × 2 10k 16V 1Ω 2.2µF 16V VCC 5V BOOT PHASE V FDMF6707B VOUT 0.9V/ 120A SS2 FREQ CLKIN CLKOUT PHSMD PGOOD2 PWMEN2 PWM2 VCC VCC RUN FB1 COMP1 VSNSP1 VSNSN1 470pF 0.22µF CIN2 22µF × 2 10k 16V 100k 1µF 3.3nF VIN VCC VCC SS1 VINSNS CONFIG IAVG PGOOD1 PWMEN1 PWM1 374Ω 100pF 0.1µF CIN1 180µF VCC 5V 4.7nF IAVG BOOT PHASE FDMF6707B VIN VSWH DISB PWM PGND VDRV VCIN SMOD CGND 0.22µF 0.22µF VCC 1Ω 2.2µF 16V 2.2µF 16V 2.87k 10k 2.2µF 16V CIN5 22µF × 2 10k 16V L3 0.22µH BOOT PHASE V FDMF6707B IN VSWH DISB PWM PGND VDRV VCIN SMOD CGND 10k 0.22µF 2.87k L4 0.22µH COUT1 100µF × 8 6.3V COUT2 330µF × 12 2.5V design ideas The LTC3861 uses a constant-frequency voltage mode architecture, combined with a very low offset, high bandwidth error amplifier and a remote output sense differential amplifier per channel for excellent transient response and output regulation. 35 CURRENT IN EACH PHASE (A) 30 25 20 15 10 CHANNEL 4 CHANNEL 3 CHANNEL 2 CHANNEL 1 5 0 0 20 80 100 40 60 TOTAL LOAD CURRENT (A) Figure 3. Thermal image at 0.9V/120A, 400 FPM, fSW = 500kHz 120 Figure 2. Current sharing between the four phases with varying load current independent of any offsets between power ground and the controller’s ground. The LTC3861 is a multiphase dual output synchronous step-down DC/DC controller that can operate with power blocks, DrMOS and external gate drivers. It is flexible enough to operate as a dual output, 3+1 output, or up to a 12-phase single output step-down converter. In a voltage mode control loop, the error amplifier output is compared to a sawtooth ramp, which directly controls the converter duty cycle. The output voltage of the error amplifier depends on the magnitude of the error signal between the differentially sensed output voltage and the amplifier reference voltage. The 600mV reference has an accuracy of ±0.75% over a 0°C to 85°C temperature 100 90 EFFICIENCY (%) The LTC3861 uses a constant-frequency voltage mode architecture, combined with a very low offset, high bandwidth error amplifier and a remote output sense differential amplifier per channel for excellent transient response and output regulation. The error and differential amplifiers have a gain bandwidth of 40MHz, high enough not to affect the main loop compensation and transient behavior, especially when all ceramic low ESR output capacitors are used to minimize output ripple. The differential amplifiers sense the resistively divided feedback voltage differentially over the full output range from 0.6V to VCC – 0.5V, ensuring that the LTC3861 sees the actual output voltage, 80 range. This, combined with the low offset of the amplifiers, guarantees a total output regulation accuracy of ±1.3% over a –40°C to 125°C temperature range. The LTC3861 achieves outstanding line transient response using a feedforward correction scheme, which instantaneously adjusts the duty cycle to compensate for changes in input voltage, significantly reducing output overshoot and undershoot. This scheme makes the DC loop gain independent of the input voltage. The converter has a minimum on-time of 20ns, which is suitable for high stepdown ratio converters operating at high frequencies. The operating frequency is resistor programmable from 250kHz to 2.25MHz, or can be synchronized to an external clock through an onboard PLL. MULTIPHASE CURRENT SHARING 70 60 VIN = 12V VOUT = 0.9V fSW = 500kHz 0 20 40 80 60 ILOAD (A) 100 120 Figure 4. 4-phase, 0.9V/120A converter efficiency The controller allows the use of sense resistors or lossless inductor DCR current sensing to maintain current balance between phases and to provide overcurrent protection. In multiphase operation, the LTC3861 incorporates an auxiliary current January 2013 : LT Journal of Analog Innovation | 35 In multiphase operation, the LTC3861 incorporates an auxiliary current share loop, which is activated by configuring the FB pin and by adding an external capacitor on the IAVG pin. The maximum current sense mismatch between phases is ±1.25mV over the –40°C to 125°C temperature range. The current sharing accuracy between the four phases at full 120A load current is ±2.15%. 5mV (±0.28%) VOUT 2mV/DIV VOUT 20mV/DIV 60mV (±3.3%) 120A IOUT 20A/DIV 500ns/DIV 10µs/DIV Figure 5. Steady state voltage ripple share loop, which is activated by configuring the FB pin and by adding an external capacitor on the IAVG pin. The voltage on the IAVG pin corresponds to the instantaneous average inductor current of the master phase. Each slave phase integrates the difference between its inductor current and the master’s. A resistor connected to the ILIM pin sets the threshold for the positive and negative overcurrent fault protection comparator. The maximum current sense mismatch between phases is ±1.25mV over –40°C to 125°C temperature range. CIRCUIT PERFORMANCE Figure 1 shows a high efficiency 12V to 0.9V/120A 4-phase step-down converter with low DCR sensing. An inductor with DCR = 0.45mΩ is used in the design. The current sharing accuracy between the four phases at full 120A load current 36 | January 2013 : LT Journal of Analog Innovation 90A Figure 6. 30A Load step transient response from 90A to 120A is ±2.15%. Figure 2 shows the current sharing between phases as a function of varying load current. Figure 3 shows the thermal image at 120A load, and the hottest spot occurs on the MOSFETs of channels 2 and 3. The efficiency at full 120A load is close to 86%, as illustrated in Figure 4. Figure 5 shows the steady state voltage ripple as approximately ±0.3% of output voltage. Load step transient analysis was performed by stepping the load from 75% to 100% of full load. This resulted in a 30A load step from 90A to 120A. The peak to peak voltage overshoot and undershoot during a load step was 60mV, which is about ±3.3% of output voltage. CONCLUSION The LTC3861 is a voltage mode controller with accurate current sharing of up to 12 phases in parallel. Since it has a 3-state PWM output instead of a builtin gate driver output, the controller can be placed further from high current paths. Because output voltage is differentially sensed, offsets between power ground and the LTC3861’s ground do not affect load regulation. The LTC3861 works with DrMOS, power blocks, and external MOSFETs with an LTC4449 gate driver. It is used in high current distributed power systems, DSP, FPGA, and ASIC supplies, datacom and telecom systems, and industrial power supplies. The LTC3861 is available in a 36-pin 5mm × 6mm QFN package. In addition, the LTC3861-1 is a pin-compatible dropin replacement for the LTC3860, available in a 32-pin 5mm × 5mm QFN package. n design ideas High Performance Single Phase DC/DC Controller with Digital Power Management Yi Sun LTC3883 is a single phase synchronous step-down DC/DC controller featuring a PMBus interface for digital control and monitoring, and integrated MOSFET gate drivers. It can function either standalone or in a digitally managed system with other Linear Technology PMBus enabled parts. 5mΩ VIN 6V TO 24V 10µF 100Ω 1µF 100Ω 10nF 3Ω 10nF VIN 10k 10k PMBus INTERFACE 10k 10k 10k 10k •±0.5% output voltage accuracy over the operation temperature range of –40°C to 125°C. • PMBus, which provides programmable voltage, current limits, sequencing, margining, OV/UV thresholds, frequency synchronization and fault logging. •Telemetry read back including VIN, IIN, VOUT, IOUT, temperature and faults. •External voltage divider to set the chip address, switching frequency and the output voltage. •Input current sensing and inductor DCR auto calibration. 1.8V/30A SINGLE PHASE DIGITAL POWER SUPPLY WITH I IN SENSE Figure 1 shows a 7V to 14V input, 1.8V/30A output application that features inductor DCR current sensing. To improve the accuracy of DCR current sense, the LTC3883 senses inductor temperature and 10k 5k VDD33 PGOOD SDA M2 ALERT RUN 1.4k 1µF VDD25 20k 24.9k 10k 20k 12.7k 9.09k 23.2k 17.8k FREQ_CFG SCL VOUT_CFG 1.4k VTRIM_CFG 0.22µF SHARE_CLK ASEL GPIO SYNC WP VDD25 VDD33 1.0µF 0.56µH BG PGND 10µF M1 0.1µF SW VIN_SNS The LTC3883 features: •4.5V to 24V input voltage range and 0.5V to 5.5V output voltage range. TG LTC3883 BOOST IIN_SNS 22µF 50V 1µF D1 INTVCC ISENSE+ ISENSE– VSENSE+ VSENSE– + TSNS GND ITH 2200pF 1.0µF 100pF D1: CENTRAL CMDSH-3TR L: COILCRAFT XAL7070-551ME VOUT 1.8V 30A COUT 1520µF MMBT3906 4.99k M1: INFINEON BSC050N03LSG COUT: 4× 330µF SANYO 2R5TPE330M9, 2× 100µF AVX 12106D107KAT2A M2: INFINEON BSC011N03LSI Figure 1. 1.8V/30A single phase digital power supply with IIN sense compensates for the TC of the DCR. This method ensures the accuracy of the readback current and overcurrent limit. The LTC3883’s control loop uses peak current mode control, which offers fast transient response. Figure 2 shows the typical waveforms of a 10A load step transient. Figure 2. Transient performance of a 10A load step VOUT 100mV/DIV (AC-COUPLED) IOUT 10A/DIV (AC-COUPLED) 120mV 30A 20A IL 10A/DIV (AC-COUPLED) 20µs/DIV January 2013 : LT Journal of Analog Innovation | 37 The LTC3883 uses a proprietary inductor DCR autocalibration function, which enables output current read back accuracy within 3%, regardless of inductor DCR tolerance. LTPOWERPLAY DEVELOPMENT IIN VIN + CIN VIN VIN_SNS IIN_SNS TG LTC3883 Figure 3. DCR auto calibration INDUCTOR DCR AUTO CALIBRATION The problem with the conventional inductor DCR current sensing is that the tolerance of the DCR can be as large as ±10%, greatly limiting the current read back accuracy. To solve this problem, the LTC3883 uses a proprietary inductor DCR autocalibration function. Figure 3 shows the simplified diagram of this circuit. The LTC3883 accurately measures the input current, IIN, the duty cycle, D and calibrates the real DCR value based on the relation: LTpowerPlay LTpowerPlay™ software is available for free at www.llinear.com/ltpowerplay SW BG ISENSE+ ISENSE– The LTC3883 features input current sensing via a resistor in series with the input side of the buck converter—a 5mΩ sense resistor as shown in Figure 1. The sense voltage is translated into a power stage input current by the LTC3883’s 16-bit internal ADC. An internal sense resistor senses chip’s supply current at VIN, so it can provide both the chip and the power stage’s input current measurements. M1 M2 IOUT DCR + + – COUT VCS DCRCALIBRATED = VCS • + D IIN With this auto-calibration method, the output current read back accuracy can be within 3%, regardless of inductor DCR tolerance. 1.2V/60A 3-PHASE DIGITAL POWER SUPPLY VOUT – All digital power management functions can be controlled by LTpowerPlay, PC-based software compatible with all of Lintear Technology’s digital power products. With LTpowerPlay, designers can easily program and control the entire power system without writing a line of code. It is easy to configure any chip on the bus, verify the system’s status, read the telemetry, check fault status, control supply sequencing. CONCLUSION The LTC3883 combines a best-in-class analog DC/DC controller with complete digital power management functions and precision data converters for unprecedented performance and control. Multiple LTC3883s can be used with other Linear Technology PMBus products to optimize multirail digital power systems. Powerful LTpowerPlay software simplifies the development of complex power systems. The LTC3883 can be used for telecom, computing, data storage, and other applications. n The LTC3883 has an analog current control loop, which makes it ideal for PolyPhase® operation. Figure 4 shows an example of a 3-phase single output circuit, with one LTC3883 and one LTC3880 for a 7V to 14V input, 1.2V/60A output application. The LTC3880 is a 2-phase synchronous buck controller with digital Figure 5. Transient performance of a 30A load power system management. The interstep for 3-phase power supply connection between these two chips is straightforward and easy. Note how VOUT 50mV/DIV the input current sense resistor of (AC-COUPLED) the LTC3883 is used to sense the total input current for all three phases. Figure 5 shows the dynamic current sharing for a load step transient. All the three phases can share the current evenly. IL0 10A/DIV IL0 10A/DIV IL0 10A/DIV 20µs/DIV 38 | January 2013 : LT Journal of Analog Innovation 60mV design ideas 5mΩ VIN 6V TO 14V 10µF INTVCC 100Ω 1µF TG LTC3883 BOOST IIN_SNS 100Ω Figure 4. 1.0V/60A 3-phase digital power supply with IIN sense VIN_SNS 10nF 22µF 50V 1µF D1 M1A 0.1µF M1 L0 0.56µH SW 10nF M1B BG PGND 3Ω 10k 10µF 1k PGOOD 10k PMBus INTERFACE VDD25 VIN 10k 10k SDA VOUT_CFG SCL VTRIM_CFG ALERT 10k GPIO 5k SYNC WP ASEL 0.22µF ISENSE– VSENSE+ TSNS ITH GND D2 VIN TG0 TG1 LTC3880 22µF 1µF M2A 0.1µF M2 L2 0.56µH SW1 BG0 M2B BG1 PGND VDD33 1µF COUT1 530µF 10nF BOOST1 SW0 M3B VOUT 1V 60A MMBT3906 2200pF D3 INTVCC BOOST0 1µF + VSENSE 4.99k 0.1µF 1k 1k ISENSE+ 1.0µF M3A 17.8k – VDD25 VDD33 1.0µF M3 L1 0.56µH 11.3k SHARE_CLK 10k 10µF 20k FREQ_CFG RUN 10k 1µF 24.9k 1k SYNC 1µF SDA SCL VDD25 ALERT VOUT1_CFG GPIO1 SHARE_CLK VTRIM0_CFG RUN0 RUN1 VTRIM1_CFG 0.22µF TSNS0 ISENSE0+ TSNS1 ISENSE1+ ISENSE0– ISENSE1– ITH0 MMBT3906 11.3k 15.8k 17.8k 0.22µF 1k ITH1 GND 100pF 10µF D1-D3: CENTRAL CMDSH-3TR L0-L2: COILCRAFT XAL7070-301ME 1µF VSENSE1 VSENSE0– COUT2 530µF 20k FREQ_CFG VSENSE0+ + 10k ASEL WP 1k 24.9k VOUT0_CFG GPIO0 MMBT3906 10nF + COUT3 530µF M1, M2, M3: FAIRCHILD FDMS3620S COUT1, COUT2, COUT3: 330μH SANYO 4TPF330ML, 2× 100µF AVX 12106D107KAT2A January 2013 : LT Journal of Analog Innovation | 39 highlights from circuits.linear.com PRECISION HIGH VOLTAGE HIGH SIDE LOAD CURRENT MONITOR The LT6016/LT6017 are dual and quad rail-to-rail input operational amplifiers with input offset voltage trimmed to less than 50μV. These amplifiers operate on single and split supplies with a total voltage of 3V to 50V and draw only 315μA per amplifier. The Over-The-Top® input stage of the LT6016/LT6017 is designed for added protection in demanding environments. The input common mode range extends to inputs up to 76V above V– independent of V+. circuits.linear.com/609 5V VBAT = 1.5V TO 76V 200Ω 0.1Ω 10W 0.1µF + LT6016 200Ω 100Ω 1% BSP89 – 1V/A 0V TO 1V OUT LOAD 2k LTspice IV circuits.linear.com/609 PANEL VOLTAGE UP TO 60V 37V VIN REG POINT RSENSE_IN 10mΩ D1 RFILTA 1k D2 CFILT 1µF IVINP ENABLE Dn 70W, SOLAR ENERGY HARVESTER WITH MAXIMUM POWER POINT REGULATION The LT3763 is a fixed frequency, synchronous, step-down DC/DC controller designed to accurately regulate output currents up to 20A. The average current mode controller will maintain inductor current regulation over a wide output voltage range from 0V to 55V. Output current is set by analog voltages on the CTRL pins and an external sense resistor. Voltage regulation and overvoltage protection are set with a voltage divider from the output to the FB pin. circuits.linear.com/608 CREF 2.2µF RFILTB 1k IVINN VIN EN/UVLO RHOT 45.3k TG VREF CBOOST 100nF BOOST LT3763 FBIN SYNC RT RFB1 121k FAULT FB SS VC CSS 10nF RC 26.1k CC 4.7nF 2.5V VIN + – circuits.linear.com/607 RFB2 12.1k L1: COILCRAFT MSS1278-123 M1, M2: INFINEON BSC100N06LS3 M3: VISHAY VN2222LL RS: VISHAY WSL2512R0100FEA RFB3 182k M3 1.8pF 1VP-P 75Ω 150Ω 665Ω 3V A 133MHz DIFFERENTIAL AMPLIFIER WITH EXTERNAL GAIN SET, IMPEDANCE MATCHING TO A 75Ω SOURCE AND LEVEL SHIFTING Complete single-ended 75Ω input impedance to differential out, level shifting 2.5V input to 1.25V differential common mode, single-ended to differential gain of 2 using external resistors circuit example. circuits.linear.com/607 LTspice IV RSB 10Ω PWMOUT ISMON PWM RT 82.5k RSA 10Ω M2 3.6V CS 33nF SENSE– IVINMON circuits.linear.com/608 VOUT RS 10mΩ 14V MAXIMUM GND SENSE+ INTVCC LTspice IV CTRL1 CVCC 22µF RFAULT 47.5kΩ BG RFBIN2 12.1k L1 12µH + INTVCC RNTC 470k VREF M1 SW CTRL2 RFBIN1 348k CIN2 100µF CIN1 4.7µF 0.1µF 102Ω 1.25V VOCM – + + – 3V LT6660-2.5 IN OUT GND 0.1µF 2.5V 0.1µF 43.2Ω 150Ω 10µF 1.25V 1VP-P 1.25V 1VP-P LTC6406 665Ω GAIN = 2 1.8pF L, LT, LTC, LTM, Linear Technology, the Linear logo, LTspice, Burst Mode, Over-The-Top and PolyPhase are registered trademarks, and Hot Swap, isoSPI, PowerPath, LTpowerPlay and VLDO are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. © 2013 Linear Technology Corporation/Printed in U.S.A./58K Linear Technology Corporation 1630 McCarthy Boulevard, Milpitas, CA 95035 (408) 432-1900 www.linear.com Cert no. SW-COC-001530