Aug 1998 Low Noise LT1614 DC/DC Converter Delivers –5V at 200mA from 5V Input

DESIGN IDEAS
Low Noise LT1614 DC/DC Converter
Delivers –5V at 200mA from 5V Input
by Steve Pietkiewicz
The inverting DC/DC converter
function is traditionally realized with
a capacitor-based charge pump.
Although simple, the output impedances of the best charge pump
solutions are in the 5Ω to 10Ω range,
resulting in significant regulation
issues when the load current
increases beyond a few tens of milliamperes. The LT1614 inductor-based
inverting DC/DC converter uses
closed-loop regulation to obtain an
output impedance of 0.1Ω, eliminating output voltage droop under load.
Figure 1 details the 5V to –5V converter circuit. The LT1614 contains
an internal 0.6Ω switch rated at 30V,
C3
1µF
L1
22µH
VIN
5V
allowing up to 28V differential between
input and output. Quiescent current
is 1mA and the device contains a lowbattery detector with a 200mV
reference voltage. The device switches
at 600kHz, allowing the use of small,
inexpensive external inductors and
capacitors. In fact, the total cost of the
components specified in Figure 1
(excluding the LT1614) is approximately $0.70 in 10,000-piece
quantities.
The LT1614 operates by driving its
NFB pin to a voltage of –1.24V, allowing direct regulation of the negative
output. This converter topology, which
consists of inductors in series with
both input and output, results in low
output noise and also in low reflected
noise on the 5V input supply. The
output and switch nodes are shown
in Figure 2. Output ripple voltage of
40mV is due to the ESR of the tantalum output capacitor C2. Ripple
voltage can be reduced substantially
by replacing output capacitor C2 with
a 10µ F ceramic unit, as pictured in
Figure 3.
In layout, be sure to tie D1’s cathode directly to the LT1614’s GND pin,
as shown in Figure 1. This keeps the
switching current loops tight and prevents the introduction of high
frequency spikes on the output. The
L2
22µH
D1
SW
VIN
+
SHDN
VC
VOUT
–5V/200mA
R1
69.8k
LT1614
C1
33µF
V OUT
100mV/DIV
AC COUPLED
NFB
GND
R2
24.9k
+
100k
C2
33µF
1nF
D1: MBR0520
L1, L2: MURATA LQH3C220
C1, C2: AVX TAJB336M010
C3: AVX1206YC105KAT (CERAMIC, X7R)
Figure 1. 5V to –5V DC/DC converter uses an inverting topology and
delivers 200mA.
V OUT
100mV/DIV
AC COUPLED
V SW
5V/DIV
500ns/DIV
Figure 2. LT1614 output and switch node with a 33µF tantalum
capacitor and 200mA load current
V OUT
100mV/DIV
AC COUPLED
V SW
5V/DIV
V SW
5V/DIV
500ns/DIV
500ns/DIV
Figure 3. LT1614 output and switch node with a 10µF
ceramic output capacitor and 200mA load current
Figure 4. Improper placement of D1’s cathode results in 60mV
switching spikes at output, even with a 10µ F ceramic output
capacitor.
Linear Technology Magazine • August 1998
35
DESIGN IDEAS
ing spikes ruin an otherwise clean
output.
Efficiency of the circuit is detailed
in Figure 5. Efficiency reaches 73%
at a 50mA load, and is above 70% at
a 200mA load. Larger inductors with
less copper resistance can be used to
increase efficiency, although such
inductors are more expensive than
the Murata units specified.
90
80
EFFICIENCY (%)
low noise that can be achieved with a
ceramic capacitor may be corrupted
by noise spikes if proper layout practice is not followed. To illustrate this
point, output and switch waveforms
from Figure 1’s circuit, with a 10µ F
ceramic output capacitor and 200mA
load, but with D1’s cathode arbitrarily connected to the ground plane,
are shown in Figure 4. 60mV switch-
70
60
50
40
10
30
100
LOAD CURRENT (mA)
3
300
1610 TA02
Figure 5. 5V to –5V converter efficiency
reaches 73%.
4.5ns Dual-Comparator-Based Crystal
Oscillator has 50% Duty Cycle
and Complementary Outputs
by Joseph Petrofsky and Jim Williams
Figure 1’s circuit uses the LT1720
dual comparator in a 50% duty cycle
crystal oscillator. Output frequencies
of up to 10MHz are practical.
Resistors at C1’s positive input set
a DC bias point. The 2k–0.068µ F
path furnishes phase-shifted feed-
back and C1 acts like a wideband,
unity-gain follower at DC. The crystal’s
path provides resonant positive feedback and stable oscillation occurs.
C2, sensing C1’s input, provides a
low skew, complementary output. A1
compares band-limited versions of
2.7V–6V
2k
1MHz–10MHz
CRYSTAL (AT-CUT)
220Ω
620Ω
+
GROUND
CASE
C1
1/2 LT1720
the outputs and biases C1’s negative
input. C1’s only degree of freedom to
respond is variation of pulse width;
hence, the outputs are forced to 50%
duty cycle.
The circuit operates with AT-cut
fundamental crystals from 1MHz to
10MHz, over a 2.7V–6V power supply
range. 50% duty cycle is maintained
at all supply voltages, with output
skew below 800 picoseconds. Figure
2 plots skew, which is seen to vary by
about 800ps over a 2.7V–6V supply
excursion.
OUTPUT
–
1000
100k
2k
0.1µF
–
680Ω
0.1µF
100k
OUTPUT SKEW (ps)
0.068µF
800
+
A1
LT1636
600
400
200
+
C2
1/2 LT1720
OUTPUT
–
0
2.5
3.0
3.5 4.0 4.5 5.0
SUPPLY VOLTAGE (V)
5.5
6.0
AN70 F52
Figure 1. Crystal oscillator has complementary outputs and 50% duty cycle. A1’s feedback
maintains output duty cycle despite supply variations.
36
Figure 2. Output skew varies only 800ps over
a 2.7V–6V supply excursion.
Linear Technology Magazine • August 1998
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