A pp l ic a ti on No te , V 1.0 , M ay 2 00 1 AP2428.01 A/D Converter C500 and C166 Microcontroller Families Analog Aspects Microcontrollers N e v e r s t o p t h i n k i n g . A/D Converter Revision History: 2001-05 Previous Version: - Page V 1.0 Subjects (major changes since last revision) Controller Area Network (CAN): License of Robert Bosch GmbH We Listen to Your Comments Any information within this document that you feel is wrong, unclear or missing at all? Your feedback will help us to continuously improve the quality of this document. Please send your proposal (including a reference to this document) to: [email protected] Edition 2001-05 Published by Infineon Technologies AG 81726 München, Germany © Infineon Technologies AG 2006. All Rights Reserved. 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AP2428.01 C500 / C166 Microcontroller Families 1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2 2.1 2.2 2.3 2.4 2.5 Transfer Characteristic and Error Definition . . . . . . . . . . . . . . . . . . . . . . 6 Ideal Transfer Characteristic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Offset Error . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Gain Error . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 Differential Nonlinearity Error (DNLE) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 Integral Nonlinearity Error (INLE) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 3 3.1 3.2 3.3 3.4 Principle of Conversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Sample Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Charge-Redistribution Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Calibration Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Write Back Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 12 13 13 13 4 4.1 4.2 4.3 4.4 Calibration Mechanism (Error Correction) . . . . . . . . . . . . . . . . . . . . . . Calibration Principle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Reset Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Normal Calibration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Disturbance Filtering . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 15 15 16 16 5 5.1 5.2 5.3 5.3.1 5.3.2 5.4 5.4.1 5.4.2 5.4.3 5.4.3.1 5.4.3.2 5.5 5.5.1 5.5.2 5.5.3 5.5.4 5.6 5.6.1 5.6.2 5.6.3 Analog Input AN0 ... ANy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Electrical Model of the A/D Converter Input . . . . . . . . . . . . . . . . . . . . . . . . Accuracy at Sample Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Charge Flow during Sample Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Charge Balance between CAIN and CEXT . . . . . . . . . . . . . . . . . . . . . . . Charge of CAIN and CEXT via RASRC . . . . . . . . . . . . . . . . . . . . . . . . . . . RASRC Calculation with (0 pF < CEXT < (2r - 1) * CAIN) . . . . . . . . . . . . . . . Charge-Redistribution Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Cycle Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Calculation Example with (0 pF < CEXT < (2r -1) * CAIN) . . . . . . . . . . . . Resistance of the Analog Source RASRC . . . . . . . . . . . . . . . . . . . . . . Cycle Time tCYCLEn . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . RASRC Calculation with (CEXT > (2r -1) * CAIN) . . . . . . . . . . . . . . . . . . . . . External Capacitance CEXT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Cycle Time tCYCLEn . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Cutoff Frequency fC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Calculation Example with (CEXT > (2r - 1) * CAIN) . . . . . . . . . . . . . . . . . RASRC Calculation with (CEXT = 0pF) . . . . . . . . . . . . . . . . . . . . . . . . . . . . Resistance of the Analog Source RASRC . . . . . . . . . . . . . . . . . . . . . . . . Calculation Example with (CEXT = 0pF) . . . . . . . . . . . . . . . . . . . . . . . . . Calculation Example with the Formula in the Data Sheet . . . . . . . . . . . 17 17 19 20 20 22 23 25 26 27 27 27 28 28 28 30 30 32 32 33 34 Application Note 3 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families 6 6.1 6.1.1 6.1.2 6.2 6.2.1 6.3 6.3.1 6.4 Reference Voltage VAREF and VAGND . . . . . . . . . . . . . . . . . . . . . . . . . . . Sources for the Voltage Reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Supply Voltage of the Microcontroller . . . . . . . . . . . . . . . . . . . . . . . . . . External Voltage Reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . RAREF Calculation Including an External Capacitance . . . . . . . . . . . . . . . Calculation Example: . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . RAREF Calculation based on the Formula in the Data Sheet . . . . . . . . . . . Calculation Example: . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Ratiometric Configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 36 36 37 38 40 41 43 44 7 7.1 7.1.1 7.2 7.2.1 7.2.1.1 7.2.2 7.2.2.1 Overload and Leakage Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Leakage Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Calculation Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Overload Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Overload Current and Absolute Maximum Ratings . . . . . . . . . . . . . . . . Calculation Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Overload Current and Operating Conditions . . . . . . . . . . . . . . . . . . . . . Calculation Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45 45 46 47 47 47 48 48 8 8.1 8.2 8.3 8.4 8.5 PCB and Design Considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Component Placing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Power Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Ground Planes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Signal Lines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Clock Generation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49 49 49 49 50 50 9 Used Short Cuts . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51 Application Note 4 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Introduction 1 Introduction For analog signal measurement on most members of the C500 and C166 microcontroller families, an A/D (Analog/Digital) converter with multiplexed input channels and a sample and hold circuit has been integrated on-chip. Depending on the device type of the C500/C166 Family, an 8-bit or 10-bit A/D converter with 4, 8, 10, 12, 15, 16 or 24 multiplexed input channels, is integrated. The A/D converter uses the method of successive approximation. In principle, the A/D converter can be divided in two parts, the analog interface (including the converter with sample and hold circuit) and the digital part, which contains different registers and the digital control unit. This Application Note provides basic information and recommendations concerning the analog part of the A/D converter. Please refer to the corresponding User’s manual for the description concerning the digital part of the A/D converter. Based on the history and evolution of the microcontrollers, there are different implementations of the A/D converter available. This Application Note is referred to the actual status of A/D converters, which are implemented in the C500/C166 Family. The differences of the analog part concern mainly the values in the A/D converter characteristics specified in the Data Sheet. For details, please use the corresponding Data Sheet. The resolution (r) of the A/D converter refers to the number of quantization levels, an analog input voltage can be determined to. This number of smallest levels is given in bits and one of them is an LSB. Figure 1 shows an example of an A/D converter with 1024 quantization levels. This A/D converter has a 10-bit resolution. An input voltage of 5 V is quantized with a step size of 5 V / 2 10 = 4,88 mV. This theoretical accuracy of an A/D converter is degraded by inaccuracies of the A/D converter itself (total unadjusted error). Further the accuracy of the total A/D conversion system is degraded by the involved external elements which are connected to the analog input ANx and to the reference voltage VAREF. It is the task of the system designer to keep the inaccuracies caused by the external circuits as low as possible. This application note provides the necessary basic information to optimize the external circuits of the A/D converter. Application Note 5 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Transfer Characteristic and Error Definition 2 Transfer Characteristic and Error Definition The following diagrams show the ideal transfer characteristic of an A/D converter and the error definition for the different kind of errors: • • • • Offset error Gain error Differential nonlinearity error (DNLE) Integral nonlinearity error (INLE) The total unadjusted error (TUE) is specified in the Data Sheets of the C500 and C166 microcontrollers. 2.1 Ideal Transfer Characteristic Figure 1 defines the ideal transfer characteristic for an A/D converter. The Ideal Transfer Curve (1) transfers each input to an output. The Ideal ADC Transfer Curve (2) includes a quantization error, since all analog input values are presumed to exist, they must be quantized by partitioning the continuum into discrete digital values. All analog values within a given range (quantization step) are represented by the same digital value, which corresponds to the nominal mid- range value. That is the reason for the quantization uncertainty of +/- 0.5 LSB, which is a natural error and inherent to each A/D converter. The quantization step size is 1 LSB = VAREF / 2r. According to the Ideal Transfer Curve (1) the first digital transition, from 0 to 1, occurs at the analog value of 0.5 LSB. That is why the first step width of the Ideal ADC Transfer Curve (2) is 0.5 LSB and the last step width is 1.5 LSB. The inherent quantization error in relation to the analog input voltage is shown in Figure 2 The total unadjusted error includes all A/D converter related inaccuracies like production process deviations and internal noise. The TUE consists of offset error, gain error, DNLE and INLE but it is not simply the sum of individually measured errors. Since some errors of the ADC, like offset and gain error, can compensate each other, the TUE can be far less than the absolute sum of all individual errors. Figure 1 shows the definition of the TUE in relation to the Ideal ADC Transfer Curve (1). The real result of the A/D converter is in the range of Ideal ADC Transfer Curve (2) +/TUE. This area is shaded in Figure 1 and is between both TUE related to ideal ADC Transfer Curves (3) and (4). Application Note 6 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Transfer Characteristic and Error Definition Digital Output Ideal Transfer Curve (1) TUE related to ideal Transfer Curve 3FF (2) Ideal ADC Transfer Curve (3) TUE related to ideal ADC Transfer Curve 3FE (4) 10 Bit Resolution 5 4 3 TUE related to ideal Transfer Curve +|TUE| -|TUE| TUE related to ideal ADC Transfer Curve 2 1 0 0 0.5 1 2 3 4 5 1023 1024 Analog Input Voltage [LSB] 0.5 LSB Inherent Quantization Error Figure 1 1022 Ideal Transfer Characteristic Quantization Error [LSB] 0.5 0 1 2 3 4 5 -0.5 1022 1023 1024 Analog Input Voltage [LSB] -1.0 Figure 2 Quantization Error Application Note 7 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Transfer Characteristic and Error Definition 2.2 Offset Error The offset error is the deviation from the Ideal ADC Transfer Curve at the lowest transition level on the Real ADC Transfer Curve. It is the input voltage required to bring the digital output to zero and can be measured by determining the first digital transition, from 0 to 1, of the A/D converter. The offset error affects all codes by the same amount.. For the consideration in the figure below, all other kinds of errors (gain, DNLE, INLE) are excluded. Digital Output Ideal Transfer Curve 3FF 3FE Ideal ADC Transfer Curve 10 Bit Resolution 5 Real ADC Transfer Curve including the Offset Error 4 3 2 1 0 0 1 2 3 4 5 1023 1024 Analog Input Voltage [LSB] Offset Error Figure 3 1022 Offset Error Application Note 8 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Transfer Characteristic and Error Definition 2.3 Gain Error The gain error is the difference between the slopes of the real ADC Transfer Curve and the Ideal ADC Transfer Curve at the maximum digital out value. For the consideration in the figure below, all other kinds of errors (offset, DNLE, INLE) are excluded. Digital Output Gain Error 3FF Ideal Transfer Curve 3FE Ideal ADC Transfer Curve 10 Bit Resolution 5 Real ADC Transfer Curve 4 3 2 1 0 0 1 2 3 4 5 1022 1023 1024 Analog Input Voltage [LSB] Figure 4 Gain Error Application Note 9 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Transfer Characteristic and Error Definition 2.4 Differential Nonlinearity Error (DNLE) The differential nonlinearity error describes variations in the analog value between adjacent pairs of digital numbers, over the full range of the digital output. If each transition step width is exactly 1 LSB, the differential nonlinearity error is zero. If the transitions are 1 LSB +/- 1 LSB, then there is the possibility of a missing codes. If a missing code occurs then one value of the digital output is missing, e.g. the digital output might jump from 0011 to 0101 and missing out 0100; See figure below. If the differential nonlinearity error is less than 1 LSB, then a missing code is automatically excluded. For the consideration in the figure below, all other kinds of errors (offset, gain, INLE) are excluded. Ideal Transfer Curve Digital Output 3FF Ideal Transfer Curve ADC Transfer Curve 3FE Real ADC Transfer Curve 10 Bit Resolution 5 Ideal ADC Transfer Curve Missing Code 4 3 2 1 0 0 0.5 1 2 3 4 5 1022 1023 1024 Analog Input Voltage [LSB] DNLE = 1 LSB Figure 5 Differential Nonlinearity Error Application Note 10 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Transfer Characteristic and Error Definition 2.5 Integral Nonlinearity Error (INLE) The integral nonlinearity error is the maximum difference between the Ideal ADC Transfer Curve and the adjusted Real ADC Transfer Curve (without offset- and gain error). For the consideration in the figure below, DNLE is also excluded. Digital Output Ideal Transfer Curve 3FF 3FE Ideal ADC Transfer Curve 10 Bit Resolution 5 Real ADC Transfer Curve 4 3 2 1 0 0 0.5 1 2 3 4 5 1022 1023 1024 Analog Input Voltage [LSB] INLE = 1 LSB Figure 6 Integral Nonlinearity Error Application Note 11 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Principle of Conversion 3 Principle of Conversion The A/D converter is based on the principle of successive approximation. It uses a capacitor network in order to compare the analog input voltage with the actual digital approximation of this voltage. The capacitor network is also used for the sample and hold function. The conversion is performed in several steps. A total conversion consists of: • • • • Sample phase Charge-redistribution phase (conversion phase) Calibration phase Write back phase The sequence of the different phases is shown in Figure 7. The total ADC conversion time can be controlled via register ADCON (C166 Family). The block diagram in Figure 8 is related to an A/D converter with 10 bit resolution and represents the principle connections between the analog input ANx, conversion C-net, comparator and the result register ADDAT. Start of Conversion End of Conversion MSB Sample Phase LSB Charge-Redistribution Phase Calibration Phase Write back Phase ADC Conversion Time Figure 7 3.1 A/D Converter Timing Sample Phase During the sample phase, the conversion control unit connects the capacitors of the conversion C-net to one of the analog input channels via a multiplexer. The capacitor network is thus charged/discharged to the voltage level of the connected analog input channel. The hold capacitor CHOLD at the comparator holds the analog input voltage after sample phase. Application Note 12 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Principle of Conversion 3.2 Charge-Redistribution Phase At the end of the sample phase and with the start of the charge-redistribution phase, the conversion C-net is disconnected from the analog input. The goal now is to reconstruct the voltage level stored in the hold capacitor CHOLD by connecting the capacitors C9 to C0 individually to VAREF or VAGND. As the capacitor network (conversion C-net) is binary weighted (i.e. Cn = 2*Cn-1), the charge of the capacitors C9 to C0, corresponds directly to the voltage level of the connected analog input channel. The digital value is found successively starting from the most significant bit down to the least significant bit. The comparator is used to decide whether the actual voltage of the capacitor Cn is below or above the voltage stored in the hold capacitance. The charge-redistribution phase is finished after 10 steps of successive approximation. The conversion C-net for a 12-bit A/D converter consists of C11 to C0 and 12 steps are required. The conversion C-net for a 8-bit A/D converter consists of C7 to C0 and 12 steps are required. 3.3 Calibration Phase The conversion accuracy depends on the precision of the conversion C-net and the offset voltage of the comparator. In order to correct the errors that are introduced through process variations and offset voltage, an additional C-net (the calibration C-net) is used together with a calibration control logic. A detailed description of the calibration phase is shown in the chapter 4, Calibration Mechanism. 3.4 Write Back Phase During the write back phase, the result of the successive approximation is copied to the result register ADDAT. The duration of the write back phase is 4 TCL. During the write back phase, the conversion C-net is precharged with approximately VAREF / 2. Note: Because of parasitic capacitances caused by the pads and the analog multiplexer, the precharge voltage at the pins can differ from VAREF / 2. Application Note 13 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Principle of Conversion Comparator ADDAT CHOLD Calibration C-Net C7’ C6’ C5’ C4’ C3’ C2’ RAM C1’ C0’ Calibration Control Conversion C-Net C9 C8 C7 C6 C5 C4 C3 C2 C1 C0 Conversion Control ANx VAREF VAGND Figure 8 Block Diagram for the analog Part of a 10-bit A/D Converter with Calibration and Conversion C-Nets Application Note 14 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Calibration Mechanism (Error Correction) 4 Calibration Mechanism (Error Correction) An automatic self-calibration mechanism is implemented in the A/D-converter in order to compensate the offset error and to balance differences in the capacitive network. This is due to production variations, which can cause linearity deviations of the A/D conversion. The self-calibration mechanism consists of the calibration capacitor-net, the calibration RAM and the calibration control unit; See Figure 8. The self-calibration includes two kinds of calibrations: • Offset Calibration is the adjustment of the offset error. • Linearity Calibration is the binary weight adjustment between the capacitors of the conversion capacitor-net. 4.1 Calibration Principle The additional correction capacitor-net (calibration C-net) is used to add/subtract a capacitive charge to the comparator input of the A/D converter. This correction C-net allows an adjustment in the range of ± 4 LSB with a resolution of 1/32 LSB within ± 128 steps. The same calibration C-net is used for both the offset and the linearity calibrations. During offset calibration, the corrective charge, in order to zero-adjust the comparator, is determined. During linearity calibration, for each of the binary weighted capacitors of the conversion C-net, a correction value (with respect to the sum of the remaining capacitors) is determined. The results of the calibration are stored in the calibration RAM. During normal conversion, the stored values are used to correct the measurement. For this purpose, the calibration control unit is used to calculate the appropriate combination of the calibration capacitors. 4.2 Reset Calibration After a reset, the contents of the calibration RAM is cleared and the A/D converter automatically starts an initial full calibration sequence (power-up calibration). Both the offset and the linearity deviations are adjusted. This calibration sequence has a duration of 3328*tBC (0.66 msec @ fCPU = 20 MHz with the reset values of register ADCON). During the first quarter of this calibration sequence, a coarse adjustment with steps from 0.5 LSB down to 0.1 LSB is performed, which becomes more precise during the following three quarters of the sequence with calibration steps of 0.03 LSB. This scheme, guarantees a very fast reduction of the offset and linearity error. Application Note 15 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Calibration Mechanism (Error Correction) Note: After reset the positive and negative analog reference voltages (VAREF and VAGND) have to be stable and within the specified range, in order to perform a correct reset calibration. Note: The reset calibration can be interrupted by any conversion. In this case, the reset calibration is lengthened by the conversion time. The calibration sequence is performed with the actual values of register ADCON. A change of bit field ADCTC (A/D Conversion Time Control) also changes the duration of the calibration sequence. During the reset calibration sequence the specified maximum TUE can be exceeded. Note: When entering IDLE or Slow Down Mode, before reset calibration is finished, the reset calibration continues until it is finished. In this case, the Power Down current increases. It is recommended to wait until reset calibration is finished, before entering IDLE or Slow Down Mode. 4.3 Normal Calibration During A/D converter operation, a re-calibration is performed after each conversion, in order to perform an adaptation to changing operation conditions, e.g. temperature. This re-calibration is performed in single steps, where a maximum change of ± 1/32 LSB of the calibration value is possible. 4.4 Disturbance Filtering Due to the way the calibration operation is implemented, a filtering of disturbances during the calibration is achieved. For example noise on VAREF or VAGND can disturb calibration, but instead of performing a full correction of a detected deviation (either offset or linearity) in one cycle, the calibration circuit performs a step-by-step reduction of the deviation. Thus, if during one calibration cycle a deviation caused by a disturbance is detected, the last correction value will only be incremented or decremented by one (1/32 LSB). As an example, if the disturbance would cause an offset deviation of 1 LSB, then 32 calibration steps would be necessary to correct for this error. If, however, a deviation occurs during one calibration cycle, but has vanished during the next calibration cycle, the previous change of the correction value will be cancelled again. In other words, a wrong calibration caused by disturbances can only occur if the disturbance lasts for a long time. Also, disturbances occurring during the reset calibration will be eliminated due to the long calibration sequence and the re-calibration after each conversion. Application Note 16 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5 Analog Input AN0 ... ANy Each application, where an analog voltage has to be measured, needs an accurate calculation of the involved external elements. This is fundamental to ensure the sufficiently charging of the A/D converter input capacitance CAIN to the same potential of the analog source, during the sample time. An insufficient charging of CAIN causes an additional inaccuracy (ErrorANx) to the TUE of the A/D converter. This chapter shows how to calculate the external circuits for the analog inputs. The derivation of the necessary formulas is followed by calculation examples. Because of the different phases of a total conversion (sample- and charge-redistribution time) the calculation examples are shared into different electrical models which fit best to the values of the used external circuits. The basis for the way of proceeding is the voltage waveform of analog input ANx which can be observed during a conversion. Note: A detailed solution of the calculation without a simplified electrical model leads to a 2nd order differential equation and will not be discussed in the ApNote. 5.1 Electrical Model of the A/D Converter Input Figure 9 is a strongly simplified block diagram of the A/D converter. The block diagram includes only the relevant elements necessary for a calculation of the external circuits. The A/D converter input capacitance CAIN contains the capacitors of the conversion Cnet and all parasitic capacitors which have to be considered for the calculation. The A/D converter input capacitance CAIN is specified in the A/D converter characteristics in the Data Sheet. The value of the actual design steps is CAIN_max = 33 pF. Please refer to the Data Sheet for the exact value of the used microcontroller. RAIN is the internal series resistance of the A/D converter. The value is RAIN = 250 Ω. This value is not explicitly in the Data Sheet, but implicitly in the formula for the calculation of the internal resistance of the analog source RASRC. The sample switch represents an analog switch closed only during sample time. The multiplexer connects the selected analog input ANx with the internal conversion C-net. The external capacitance CEXT can be a real external capacitor for noise reduction or only the parasitic capacitance caused by the signal line between analog source and A/D converter input. The analog voltage source is represented by an ideal voltage source V0 and a series resistance RASRC. Application Note 17 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy A/D Converter Analog Source AN0 ANx VAINx V0 RAIN Comparator MUX RASRC Sample ANy CAIN CEXT VAGND Central Analog Ground Figure 9 Block Diagram of A/D Converter and Analog Source Application Note 18 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.2 Accuracy at Sample Time As already described in chapter "Principle of Conversion", a total conversion is divided in two phases: the sample phase and the charge-redistribution phase. The total accuracy of the A/D converter result depends on the TUE, the accuracy of VAREF and the voltage level difference between analog source V0 and VCAIN (ErrorANx) at the end of the sample phase. A detailed consideration of the voltage level at CAIN (or ANx, respectively) is the condition to determine the correct values for RASRC, CEXT, sample time and cycle time of a system. The worst case voltage deviation for the system is the maximum voltage difference between the precharge voltage of CAIN (approximately VAREF / 2) and V0 at the beginning of the sample phase. This case is given for V0 = VAREF or V0 = VAGND. Figure 10 shows the absolute voltage difference between V0 and CAIN (|V0 - VAREF / 2|) at the beginning of the sample phase. The formulas in this ApNote are all related to the possible absolute maximum V0 = VAREF. The result can also be transformed to V0 = VAGND. Voltages used in the calculations are all referred to VAGND. |V0 - VAREF/2| VAREF/2 VAREF/2 VAGND Figure 10 V0 VAREF Voltage Difference between V0 and CAIN (|V0 - VAREF / 2|) at the Start of the Sample Time Note: The assumed error (ErrorANx) used in this chapter (“Analog Input AN0 ... ANy”) for the calculation examples is referred to the allowed maximum input voltage at ANx (VAINx = VAREF). For input voltages at ANx smaller than VAREF the additional inaccuracy at VAINx is proportional less than the value of ErrorANx used in the example calculations. The real additional inaccuracy at VAINx is: ErrorANx_real = (VAINx / VAREF) * ErrorANx with the condition VAGND ≤ VAINx ≤ VAREF Application Note 19 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.3 Charge Flow during Sample Time The input impedance of the A/D converter is mainly capacitive (CAIN) with a small resistive (RAIN) part. This capacitance, however, applies only to the selected analog input pin ANx during the sample time. During the remaining time, the inputs are extremely high impendance (e.g. typical leakage currents are in the range of some 10 nA). See specification in the Data Sheet: Input leakage current of the ADC. During the sample phase, two different sequential processes are running. First, CAIN is charged from CEXT and the voltages at CAIN and CEXT get the same value. Secondly, the common voltage at CAIN and CEXT is adjusted to V0 via the resistance of the analog source RASRC. Depending on the performed phases of the A/D converter different time constants τ=have to be considered: τ1 : Time constant at the beginning of the sample time. It contains CAIN, CEXT and RAIN • τ2 : Time constant during sample time. It contains CAIN, CEXT and RASRC • τ3 : Time constant during and after charge-redistribution phase. It contains CEXT and RASRC • 5.3.1 Charge Balance between CAIN and CEXT The electrical model for τ1 is shown in Figure 11. The voltage at CAIN before switch Sample is closed, is approximately VAREF/2 because of precharging CAIN at the end of conversion. The voltage at CEXT is nearly V0 depending on the cycle time of the conversion. Sample VAINx Figure 11 CEXT RAIN VCAIN ~ VAREF/2 CAIN Electrical Model of the A/D Converter during τ1 When switch Sample is closed, then a charge balance between CAIN and CEXT is done with the time constant τ1; See Figure 11. Figure 13 presents the corresponding waveform at ANx. Application Note 20 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy τ1 C AIN ⋅ C EXT = R AIN ⋅ ------------------------------ C AIN + C EXT The possible maximum value is τ1 = 8.25 ns @ CAIN = 33 pF and CEXT = infinite, because RAIN and CAIN are fixed values of the A/D converter. For the calculation of the sample time, which is in the range of some µs, the duration of time constant τ1 is in most cases negligible (after 7.6*τ=the voltage error is less than 0.5 LSB). For typical values of 7.6*τ1 see Table 1. Table 1 CEXT 7.6*τ1 Values for 7.6*τ1 @ CAIN = 33 pF and RAIN = 250 Ω 1 pF 10 pF 100 pF 1 nF 10 nF 100 nF 1 µF 1.84 ns 14.58 ns 47.14 ns 60.70 ns 62.49 ns 62.68 ns 62.70ns The charge balance between CAIN and CEXT causes a voltage jump V∆= at the analog input ANx. Depending on the voltage on ANx when the sample phase starts, the voltage can be increased or decreased. The example of Figure 13 uses the worst case V0 = VAREF. At the end of 7.6*τ1 the voltage at ANx is reduced (or increased) by the value V∆ with an accuracy of 0.5 LSB. The charge balance between CEXT and CAIN results in the formula for V∆: C AIN ⋅ ( V 0 – V CAIN ) V ∆ = ------------------------------------------------C AIN + C EXT Table 2 Typical Values for the Voltage Jump V∆=@ CAIN = 33 pF, Precharge: V0 - VCAIN = 2.5 V and V0 = VAREF CEXT 1 pF 10 pF 100 pF 1 nF 10 nF 100 nF 1 µF V∆ 2.4 V 1.9 V 0.6 V 80 mV 8.2 mV 0.8 mV 0.08 mV Application Note 21 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.3.2 Charge of CAIN and CEXT via RASRC The electrical model during sample time with τ2 is shown in Figure 12. In this electrical model RAIN is neglected because in typical systems RASRC >> RAIN. The voltage at CAIN and CEXT is defined by V0 and V∆ at the beginning of the second phase (’startvoltage’ = V0 - V∆). RASRC V0 CEXT Figure 12 CAIN Electrical Model of the A/D Converter during τ2 After V∆= has reached the absolute maximum value, CEXT and CAIN are charged via RASRC from V0 with the time constant τ2. τ2 Application Note = R ASRC ⋅ ( C AIN + C EXT ) 22 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.4 RASRC Calculation with (0 pF < CEXT < (2r - 1) * CAIN) For reliable results it must be assured that during the sample time the input capacitance CAIN is completely charged to the desired value, which is then digitized by the converter. Under worst case conditions this capacity must be charged or discharged by the half input voltage when V0 = VAREF or V0 = VAGND. The input capacitance CAIN of the A/D converter, the external capacitance CEXT and the resistance of the analog source RASRC form an RC lowpass filter, which has the charging function VS(t). In normal systems, the sample time tS >> τ1, therefore τ1 is neglected in the formula for VS(t). The waveform is shown in Figure 13. V S (t) = V AREF – V ∆ ⋅ e -t ------τ2 The voltage on ANx at the end of the sample time can also be described with the formula VS(tS). The ErrorANx describes the maximum allowed deviation between the voltage on ANx and V0 when the sample time is finished. An assumed ErrorANx of 0.5 LSB is equivalent to 9.76 mV / 2.44 mV / 0.61 mV @ VAREF = 5 V and 8-bit / 10-bit / 12-bit A/D converter resolution. V S (t s ) = V AREF – Error ANx Now it is possible to calculate the maximum value of the analog source resistance RASRC. The formula for RASRC assumes that RAIN = 0 Ω. ts R ASRC = --------------------------------------------------------------------V∆ ( C AIN + C EXT ) ⋅ ln ---------------------Error ANx The formula is only valid for: V∆ / ErrorANx > 1 An assumed maximum ErrorANx = LSB / 2 leads to CEXT < (2r - 1) * CAIN Depending on the A/D converter resolution the relations between CEXT and CAIN for the calculation of RASRC are: 8-bit resolution: 10-bit resolution: 12-bit resolution: Application Note 0 pF <ΙCEXT < 255 * CAIN 0 pF <ΙCEXT < 1023 * CAIN 0 pF <ΙCEXT < 4095 * CAIN 23 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy VAIN τ2 V0 = VAREF τ3 ErrorANx VS(t) VS(tS) Sampled Voltage V∆ V0 - V∆ τ1 τ2 t tS tCR tC tCYCLEn tS : Sample time tC : Conversion time tCYCLEn : Cycle time of channel n τ1,=τ2,=τ3 : Time constants V0 : Voltage of the analog source tCR : Charge-redistribution time (conversion phase) V∆============: Voltage jump at the start of the sample phase ErrorANx : Voltage deviation between sampled voltage and voltage of the analog source Figure 13 Voltage Waveform at ANx Application Note 24 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.4.1 Charge-Redistribution Time During the charge-redistribution time, the Sample switch is open and the external capacitance CEXT is charged via the resistor of the analog source RASRC. RASRC V0 CEXT Figure 14 Electrical Model of the A/D Converter during τ3 The time constant during and after charge-redistribution time is τ3. τ3 = R ASRC ⋅ C EXT While the external capacitance CEXT is charged via RASRC, the A/D converter performs the successive approximation (charge-redistribution). This is the transformation of the analog voltage into a digital value. The reference for the transformation is the reference voltage at pin VAREF referred to VAGND. It is very important for an exact conversion result to hold the reference voltage and the reference ground on a constant level during the charge-redistribution time. More details can be found in the chapter "Reference Voltage VAREF and VAGND". Application Note 25 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.4.2 Cycle Time The cycle time tCYCLEn is the duration from the start of a conversion to the next conversion start of the same analog channel. The figure below shows the relation between the conversion time of an analog channel and the cycle time. chx chy chz chx tCx tCy tCz tCx tCYCLEn tCn : Conversion time of analog channel n tCYCLEn : Cycle time of analog channel n chn Figure 15 : analog channel n Cycle Time For continuous conversion mode of a channel, the conversion time tC can be equal to the cycle time tCYCLEn. The cycle time of consecutive conversions is important for the calculation of the voltage on CEXT at the start of next conversion. The voltage difference between the analog source V0 and the analog input ANx at the start of a conversion should be 0 V or negligible. The recommendation is: t CYCLE ≥ 7.6 ⋅ τ 3 + t s Note: After 7.6 * τ3 the remaining deviation from V0 is 0.049% of the assumed ErrorANx for VS(tS). Application Note 26 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.4.3 Calculation Example with (0 pF < CEXT < (2r -1) * CAIN) The assumed values used in the example are: CAIN RAIN CEXT ErrorANx = 33 pF, tS = 1.28 µs, = 250 Ω, tC = 7.8 µs, = 200 pF, = 0.5 LSB10 = VAREF / 2048 = 2.44 mV VAREF = V0 = 5 V, r = 10 (10-bit resolution), The calculation results in the values for RASRC and tCYCLEn. 5.4.3.1 Resistance of the Analog Source RASRC First the voltage jump V∆ during the sample phase is calculated: V∆ = (CAIN * (VAREF - VAREF/2)) / (CAIN + CEXT) V∆ = (33 pF * (5 V - 2.5 V)) / (33 pF + 200 pF) V∆ = 354 mV The allowed maximum resistance of analog source RASRC is: RASRC = tS / ((CAIN + CEXT) * ln(V∆ / ErrorANx)) RASRC = 1.28 µs/ ((33 pF + 200 pF) * ln(354 mV / 2.44 mV)) RASRC = 1103 Ω The table below shows the different results of RASRC with the assumed values used in the example. Maximum Values for RASRC and different CEXT Table 3 CEXT[pF] RASRC [kΩ] 1 20 40 60 80 100 150 200 250 500 1000 5.4 3.7 2.9 2.3 2.0 1.7 1.3 1.1 0.9 0.6 0.35 10000 0.1 Note: The capacitive load at the analog inputs ANx should be as small as possible because it reduces the allowed resistance of the analog source RASRC; See Table 3. The only exception is the use of a very high external capacitance, which supplies the A/D converter with the necessary charge during the sample phase. 5.4.3.2 Cycle Time tCYCLEn The recommended minimum value of the cycle time is tCYCLEn = 7.6 * RASRC * CEXT + tS tCYCLEn = 7.6 * 1103 Ω * 200 pF + 1.28 µs tCYCLEn = 2.95 µs The calculated cycle time is smaller than the conversion time and, in that case, continuous conversion of this analog channel is possible without inserting a waiting period to charge the external capacitance CEXT. Application Note 27 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.5 RASRC Calculation with (CEXT > (2r -1) * CAIN) The selected external capacitance has to be high enough so the total charge, which is necessary to load the internal C-net (CAIN) of the A/D converter, is supplied by the external capacitor CEXT. The considerations below include the value of the external capacitance CEXT with respect to the assumed maximal ErrorANx caused by the CEXT and the necessary time tCYCLEn to reload the external capacitor. 5.5.1 External Capacitance CEXT The calculation of the external capacitance CEXT is based on the assumption that VAREF - V∆ is the sampled voltage and V∆= is the maximum allowed ErrorANx; See Figure 16. After the charge balance (voltage jump, V∆= ) the voltage change at the capacitors during the sample phase is extremely small because of the high time constant τ3 of the external capacitance and the resistance of the analog source. The example is calculated with the assumption of a maximum allowed error, ErrorANx = LSBr / 2. Error = LSBr / 2 Error = VAREF / (2r * 2) Error > V∆ = (CAIN * (VAREF - VAREF/2)) / (CAIN + CEXT) CEXT > (2r - 1) * CAIN Depending on the A/D converter resolution the relations between CEXT and CAIN for the calculation of RASRC are: 8-bit resolution: 10-bit resolution: 12-bit resolution: CEXT > 255 * CAIN CEXT > 1023 * CAIN CEXT > 4095 * CAIN The condition CEXT > (2r - 1) * CAIN allows a free choice of the sample time tS without consideration of the resistance of the analog source RASRC but RASRC has a direct influence on the cycle time tCYCLEn of the conversion. 5.5.2 Cycle Time tCYCLEn The calculation of the cycle time takes into account that the external capacitor is not totally charged to the voltage of the analog source V0 (worst case V0 = VAREF or V0 = VAGND) but a small voltage rest VR is missing. See Figure 16. With the condition CAIN << CEXT the formula for V∆=can be simplified: V∆ = (CAIN * (VAREF - VAREF/2)) / (CAIN + CEXT) V∆ ~ CAIN * VAREF / 2 * CEXT Application Note 28 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy V∆ + VR ≤ ErrorANx The condition with VR = VAREF - VC(tCYCLEn) is based on Figure 16. The charge curve VC(t) of the capacitor CEXT via the resistance of the analog source RASRC is: V C (t) = V AREF – Error ANx ⋅ e -t ------τ3 With an assumed maximum error of LSBr / 2 (ErrorANx = (VAREF / 2r) / 2) and with τ3 = RASRC * CEXT the formulas result in the relation: C EXT t CYCLE ≥ R ASRC ⋅ C EXT ⋅ ln ----------------------------------------r C EXT – 2 ⋅ C AIN This formula is only valid for CEXT > 2r * CAIN VAIN tCYCLEn V0 = VAREF tCYCLEn VR ErrorANx VC(tCYCLEn) VC(t) V∆ Sampled Voltage t tCYCLEn : Cycle time of channel n V0 : Voltage of the analog source VC(t)=========: Charge curve for CEXT VR : Voltage rest at the end of tCYCLEn V∆============: Voltage jump at the start of the sample phase ErrorANx : Voltage deviation between sampled voltage and voltage of the analog source Figure 16 Voltage at CEXT with High Capacitance for Periodical Conversions Application Note 29 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.5.3 Cutoff Frequency fC The resistance of the analog source and the external capacitance CEXT act as a lowpass filter with the cutoff frequency fC. A check is necessary whether the cutoff frequency fits to the frequency of the analog source. If the relation between A/D converter cycle frequency (fCYCLE = 1 / tCYCLEn) and the cutoff frequency is fCYCLE / fC = 0.1 then the analog signal is damped with 5 o/oo (~1 LSB). 1 f C = ----------------------------------------------2 π ⋅ R ASRC ⋅ C EXT Note: If the external circuit reaches the cutoff frequency then the voltage of the analog source V0 is damped with the factor -3 dB (VAIN ~ 0.7 * V0 @ cutoff frequency fC). 5.5.4 Calculation Example with (CEXT > (2r - 1) * CAIN) The assumed values used in the example are: CAIN RAIN RASRC ErrorANx = 33 pF, tS = 1.28 µs, = 250 Ω, tC = 7.8 µs, = 20 kΩ, CEXT = 100 nF = 0.5 LSB10 = VAREF / 2048 = 2.44 mV, VAREF = V0 = 5 V, r = 10 (10-bit resolution), (CEXT = 3030 * CAIN) The calculation results in the value of cycle time tCYCLEn and cutoff frequency fC. The values of the external capacitance CEXT and resistance of the analog source RASRC are in a fixed relation with the cycle time tCYCLEn: tCYCLEn ≥ RASRC * CEXT * ln(CEXT / (CEXT - 2r * CEXT)) tCYCLEn ≥=20 kΩ=* 100 nF * ln(100 nF / (100 nF - 210 * 100 nF)) tCYCLEn ≥=0.82 ms The cutoff frequency is calculated via: fC = 1 / (2 * π=* RASRC * CEXT) fC = 1 / (2 * π=* 20 kΩ=* 100 nF) fC = 80 Hz Application Note 30 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy Table 4 includes calculation results of the cycle time in [ms] for different values of CEXT and RASRC with the assumed values of the example (ErrorANx = 0.5 LSB10). Table 4 Cycle Time tCYCLEn for Different Values of CEXT and RASRC RASRC [kΩ] tCYCLEn [ms] CEXT [nF] 34 40 50 75 100 500 1000 10000 1 0.17 0.07 0.06 0.04 0.04 0.03 0.03 0.03 5 0.87 0.37 0.28 0.22 0.21 0.17 0.17 0.17 10 1.73 0.75 0.56 0.45 0.41 0.35 0.34 0.34 15 2.60 1.12 0.84 0.67 0.62 0.52 0.52 0.52 20 3.47 1.49 1.13 0.90 0.82 0.70 0.69 0.68 35 4.33 1.86 1.41 1.12 1.03 0.87 0.86 0.85 30 5.20 2.24 1.69 1.35 1.24 1.05 1.03 1.02 40 6.93 2.98 2.25 1.80 1.65 1.40 1.38 1.35 50 8.66 3.73 2.82 2.25 2.06 1.75 1.72 1.69 100 17.33 7.45 5.63 4.49 4.12 3.50 3.44 3.38 Table 5 includes calculation results of the cutoff frequency in [Hz] for different values of CEXT and RASRC with the assumed values of the example (ErrorANx = 0.5 LSB10). Table 5 Cutoff Frequency fC for Different Values of CEXT and RASRC RASRC [kΩ] fC [Hz] CEXT [nF] 34 40 50 75 100 500 1000 10000 1 4681 3979 3183 2122 1592 318 159 16 5 936 796 637 424 318 64 32 3.2 10 468 398 318 212 159 32 16 1,6 15 312 265 212 141 106 21 11 1,1 20 234 199 159 106 80 16 8.0 0.8 35 187 159 127 85 64 13 6.4 0.6 30 156 133 106 71 53 11 5.3 0.5 40 117 99 80 53 40 8.0 4.0 0.4 50 94 80 64 42 32 6.4 3.2 0.3 100 47 40 32 21 16 3.2 1.6 0.2 Application Note 31 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.6 RASRC Calculation with (CEXT = 0pF) In this case, which is in real systems hard to realize, the external capacitance is neglected. The electrical model is shown in Figure 17. It can be used for a rough estimation of the external components if the value of CEXT is nearly zero pF. The internal C-net capacitance of the A/D converter is directly charged via RASRC and RAIN. RASRC RAIN CAIN V0 Figure 17 5.6.1 Electrical Model of the A/D Converter during τ2 with CEXT = 0 pF Resistance of the Analog Source RASRC When the external capacitance is CEXT = 0 pF then time constant τ1 = 0 s and the maximum voltage jump V∆=at the beginning of the sample time is approximately VAREF/2, equal to the precharge value of the internal C-net. V∆ = (CAIN * (VAREF - VAREF / 2)) / CAIN V∆ = VAREF / 2 The resistance of the analog source RASRC, is calculated via the formula for systems with a small external capacitance but without CEXT and with RAIN. ts R ASRC = -------------------------------------------- – R AIN V∆ C AIN ⋅ ln ---------------------Error ANx The calculation of the cycle time is not necessary because only during sample time, is the internal C-net connected to the analog source. In the other phases of the cycle time, the internal C-net is disconnected from the analog source. Therefore, no capacitance has to be charged via RASRC until the start of the next sample time. Application Note 32 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.6.2 Calculation Example with (CEXT = 0pF) The calculation example gives a rough estimation for the allowed maximum of RASRC if CEXT is nearly zero pF. The assumed values used in the example are: CAIN RAIN CEXT ErrorANx = 33 pF, tS =1.28 µs, = 250 Ω, tC = 7.8 µs, = 0 pF, = 0.5 LSB10 = VAREF / 2048 = 2.44 mV, VAREF = V0 = 5 V, r = 10 (10-bit resolution), The calculation results in the value for RASRC with V∆ = VAREF / 2. RASRC = tS / (CAIN * ln(V∆ / ErrorANx)) - RAIN RASRC = 1.28 µs / (33 pF * ln(2.5 V / 2.44 mV)) - 250 Ω RASRC = 5345 Ω The table below includes the maximum values for RASRC and different sample times with the assumed values of the example: Table 6 Maximum Values for RASRC and sample Times tS @ CEXT = 0 pF tS [µs] 1 2 3 4 5 6 7 8 9 10 RASRC [kΩ] 4.1 8.5 12.9 17.2 21.6 26.0 30.4 34.7 39.1 43.5 Note: The leakage current specified in the Data Sheet can have an influence to the accuracy of the analog input voltage, when the values of RASRC exceeds a certain limit. This limit depends on the allowed inaccuracy referred to VAINx which is given by the system demands. See chapter “Overload and Leakage Current”. Application Note 33 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Analog Input AN0 ... ANy 5.6.3 Calculation Example with the Formula in the Data Sheet The A/D converter Characteristics in the Data Sheet (example for C166 Family) include the formula for the calculation of the maximum ’Internal resistance of analog source’. With tS in ns and RASRC in kΩ the formula is: RASRC ≤ tS / 450 - 0.25 This formula in the C166 Family Data Sheets is based on the assumption, that the analog input ANx is only loaded with a small external parasitic capacitance: CEXT < 65 pF. For systems with an external capacitance, which exceeds this value, the external components have to be calculated as shown in the previous chapters. The table below includes the maximum values for RASRC calculated with the formula in the Data Sheets of the C166 Family. Table 7 Maximum Values for RASRC and sample Times tS @ CEXT = 65 pF tS [µs] 1 2 3 4 5 6 7 8 9 10 RASRC [kΩ] 2.0 4.2 6.4 8.6 10.9 13.1 15.3 17.5 19.8 22.0 Note: The leakage current specified in the Data Sheet can have an influence to the accuracy, see note at Table 6. Application Note 34 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND 6 Reference Voltage VAREF and VAGND During the charge-redistribution phase, and also during the calibration phase, each group of capacitors from the C-net is individually switched to either VAREF or VAGND. Because of this switching and the according charge transfers in the C-net, the A/D converter requires a dynamic current at pin VAREF. Thus, the resistance of the voltage reference source has to be low enough to supply the current for the charge-redistribution- and calibration phase. The external circuit at VAREF has a direct influence to the required resistance of the voltage reference. If an external capacitance CAREF, between VAREF and VAGND, is used then the voltage reference has to supply a small continuous current to charge the external capacitor. The necessary peak current during the charge-redistribution phase is supplied by the external capacitance CEXT. The continuous current and the charge duration (tCYCLE) have to be high enough to fill the external capacitance to a sufficient voltage level before the next charge-redistribution phase starts. If there is no external capacitance between VAREF and VAGND then the voltage reference has to supply the peak current directly. The maximum allowed resistance RAREF between the voltage reference VRF and VAREF using no external capacitance CAREF is specified in the Data Sheets of the C500/C166 Family. The specified value for RAREF in the Data Sheet is the worst case for the calculation of the minimum sourcing peak current, which has to be supplied by the voltage reference. V RF I AREF ≥ ---------------R AREF Application Note 35 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND 6.1 Sources for the Voltage Reference Depending on the system demands, several different kinds of voltage references can be used in a system. The supply voltage of the microcontroller can be selected for the reference voltage, but the accuracy is in percentage range. The accuracy of an external high precision voltage reference is in the per mille range. 6.1.1 Supply Voltage of the Microcontroller In most systems, the voltage reference used for the A/D converter is the supply voltage VDD of the microcontroller. The typical accuracy of voltage regulators is 2 %; See power semiconductor family TLE42xx from Infineon Technologies. When using the digital supply voltage of the microcontroller, it is recommended to insert a low pass filter between VDD and VAREF for the voltage reference; See figure below. The low pass filter suppresses noise on pin VAREF to improve the accuracy of the A/D converter results. RAREF VAREF CAREF Power Supply Tantalum Ceramic Central Analog Ground GND VAGND Microcontroller VDD 5V VSS Central Digital Ground Figure 18 Supply Voltage used for Voltage Reference The values of the capacitance CAREF and the resistor RAREF depend on the characteristics of the system. Typical values are RAREF = 47 Ohm and CAREF = 100 nF @ r = 10 and CAIN = 33 pF. The cutoff frequency of this low pass filter is fC = 34 kHz. If there is noise on the system supply voltage with a very low frequency, then the cutoff frequency can be reduced via an appropriate tantalum capacitance in parallel to CAREF, which stabilizes the voltage reference. Note: The impedance and the noise caused by the connection between Central Analog Ground and Central Digital Ground should be as low as possible. Application Note 36 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND 6.1.2 External Voltage Reference The source for an external voltage reference can be a standard supply voltage with increased accuracy or with less noise. For systems where a high accuracy is demanded, high precision voltage reference can be used with a typical accuracy in the range of 2.5 mV ... 20 mV. VDD RAREF 5.000 V VAREF CAREF Tantalum Ceramic GND VAGND Central Analog Ground Power Supply 5V VDD GND VSS Microcontroller Voltage Reference Central Digital Ground Figure 19 External Voltage Reference Note: If the supply voltage of the microcontroller and the voltage reference of the A/D converter are switched on and off at different times, then it is very important that the voltage reference is switched on or off only when the supply voltage of the microcontroller is on otherwise the voltage reference supplies the system with current via the ESD clamp diode. In that case, it is necessary to reduce the overload current to the specified absolute maximum ratings; See chapter overload and leakage current. The overload current can be reduced via a resistor or a diode. If the additional external clamp resistor causes an unacceptable additional error at VAREF then an external clamp diode should be used. Application Note 37 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND 6.2 RAREF Calculation Including an External Capacitance The calculation is based on the assumption that there is an external capacitance CAREF between VAREF and VAGND. The selected external capacitance has to be high enough that the total charge, which is necessary to load the internal C-net (CAIN) for a total conversion phase, is supplied by the external capacitor CAREF. Voltage Reference IAREF RAREF_INT A/D Converter VERROR VAREF RAREF VRF CAREF RAIN VAREF tbit_conversion CAIN VAGND Central Analog Ground Figure 20 A/D Converter during Conversion Phase with CAREF The following considerations include the value of the external capacitance CAREF with respect to the assumed maximum voltage error at VAREF (ErrorAREF) caused by CAREF and the necessary time tCYCLE to reload the external capacitor. The relation between the external capacitance CAREF, the internal C-net CAIN and the assumed maximum error caused by CAREF is: C AREF ≥ 2 with: r = 8: 8-bit resolution r = 10: 10-bit resolution r = 12: 12-bit resolution Application Note r+E ⋅ ( C AIN ) ⁄ 2 E = 0: ErrorAREF = 1 LSBr E = 1: ErrorAREF = LSBr / 2 E = 2: ErrorAREF = LSBr / 4 38 ErrorAREF = 1 / 2E LSBr LSBr = VAREF / 2r V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND Note: The maximum voltage error (ErrorAREF) at VAREF caused by CAREF is referred to the allowed maximum input voltage at ANx (VAINx = VAREF). For input voltages at ANx smaller than VAREF the additional inaccuracy at VAINx is proportional less than the value of ErrorAREF used in the example calculations. The real additional inaccuracy at VAINx is: ErrorAREF_real = (VAINx / VAREF) * ErrorAREF with the condition VAGND ≤ VAINx ≤ VAREF The condition (CAREF ≥ 2r+E CAIN / 2) allows a free choice of the A/D converter basic clock tBC but the cycle time tCYCLE has a direct influence on the accuracy of the conversion. The cycle time has to be long enough to recharge the external capacitance CAREF before the next charge-redistribution phase is started. The external capacitance CAREF has to be charged from the voltage reference. The minimum current, which is drawn from the voltage reference, is based on the charge that is necessary for a complete conversion. The charge Q for a complete chargeredistribution phase and a calibration phase is: Q = C AIN ⋅ V AREF The current for the voltage reference depends on the minimum cycle time for a total conversion: Q I AREF = ----------------t CYCLE The external resistance RAREF between the voltage reference and the input VAREF of the A/D converter has an enormous influence on the accuracy. This resistor should be chosen as small as possible! Because the continuous current IAREF causes a voltage difference VERROR between the voltage reference VRF and the reference voltage input VAREF of the A/D converter; See Figure 20. V ERROR = R AREF ⋅ I AREF Application Note 39 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND 6.2.1 Calculation Example: The assumed values used in the example are: CAIN = 33 pF, tBC = 160 ns, RAIN = 250 Ω, r = 10, (10-bit resolution) VRF = 5 V, tCYCLE = 7.8 µs (C166 Family: minimum time @ fCPU = 20 MHz), E = 2, ErrorAREF = 0.25 LSB, VERROR = VRF / 4096 = 1.22 mV The value for the external capacitance between VAREF and VAGND is: CAREF ≥ 2 r+E * CAIN / 2 CAREF ≥ 210+2 * 33pF / 2 CAREF ≥ 68 nF Note: A typical recommendation for the value of the external capacitance is CAREF = 100 nF With the assumption VAREF = VRF, the minimum continuous current which has to be supplied by the voltage reference is: IAREF ≥=CAIN * VAREF / tCYCLE IAREF ≥=33 pF * 5 V / 7.8 µs IAREF ≥=21µA The allowed maximum value for the resistor RAREF between voltage reference VRF and input VAREF of the A/D converter is: RAREF ≤=VERROR=/=IAREF RAREF ≤ 1.22 mV / 21µA RAREF ≤=58 Ω Note: In case of an overload condition, it is possible that RAREF has to be increased, to limit the overload current to the specified values. If that value of RAREF exceeds the demanded error of the system, an external diode between VAREF and VDD can reduce the overload current; See Figure 19. Application Note 40 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND 6.3 RAREF Calculation based on the Formula in the Data Sheet The calculation of RAREF in the Data Sheets of the C500/C166 Family is based on the assumption that there is no external capacitance between VAREF and VAGND. The electrical model for the calculation is shown in the figure below. Voltage Reference IAREF A/D Converter VERROR Comparator VAREF RAREF_INT RAREF VRF RAIN VMSB VAREF CMSB CLSB CAIN Conversion Control VAGND Central Analog Ground Figure 21 A/D Converter during Conversion Phase without CAREF During charge-redistribution time (successive approximation), all capacitors of the A/D converter are charged with VAREF/2 and compared with the sampled voltage from analog input ANx. The successive approximation is started with the MSB and finished with the LSB. The capacitor of the MSB needs most charge from the voltage reference due to the binary weighting. The available time to charge the MSB to VAREF/2 and to compare the MSB voltage with the sampled voltage is 4*tBC (tBC: Basic Clock frequency can be controlled via register ADCON). Typically half the time can be used to charge MSB to VAREF/2 (value depends on device type and on technology). The other half is necessary for the comparison of the values by the comparator of the A/D converter. The worst case for the maximum allowed resistance between voltage reference VRF and input VAREF of the A/D converter is the conversion of the MSB. The capacitance CMSB is charged with VAREF/2 and the voltage wave form at the comparator input is: Application Note 41 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND –t --------------------------------------------------------------------- (R +R ) ⋅ C MSB V MSB (t) = ( V AREF ⁄ 2 ) ⋅ 1 – e AREF AIN The maximum allowed value for RAREF with (VMSB(tBC) = VAREF/2 - ErrorMSB) is: t BC ⋅ 2 – R AIN R AREF ≤ -------------------------------------------------------V AREF C MSB ⋅ ln ------------------------------Error MSB ⋅ 2 The figure below shows the comparator voltage VMSB(t) during the conversion of the MSB. The conversion of the MSB lasts 4*tBC. V VAREF ErrorMSB 2 VMSB(t) VMSB(2*tBC) t tCHARGE tCOMPARE 4 * tBC tCHARGE : time to charge the MSB capacitance to VAREF/2 within 2*tBC tCOMPARE : time to compare MSB voltage with sampled input voltage ANx within 2*tBC Figure 22 Comparator Voltage during Conversion of MSB The formula in the Data Sheet can be derived from the relation above. The typical value CMSB of a 16-bit microcontroller used for the calculation is CMSB = 16.5 pF (CMSB = CAIN / 2). The assumed maximum ErrorMSB caused by RAREF is LSB/2 = VAREF/(2*210). Application Note 42 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND t BC - – R AIN R AREF ≤ ---------------------12 57 ⋅ 10 This relation rounded with RAREF in kΩ and tBC in ns results in the Data Sheet formula: RAREF ≤ΙtBC / 60 - 0.25 6.3.1 Calculation Example: For a system using an fCPU = 25 MHz and a tBC = 160 ns the allowed maximum value for RAREF is: RAREF =Ι160 ns / 60 - 0.25 RAREF =Ι2.4 kΩ with RAREF in kΩ and tBC in ns The minimum current which has to be supplied by the voltage reference is: IAREF ≥ VAREF / RAREF IAREF ≥Ι5 V / 2400 Ω with VAREF = 5 V IAREF ≥Ι2.1 mA The calculated current is not a continuous one. It is a peak current which flows only at the beginning of MSB conversion and becomes smaller with each converted bit down to the LSB. Note: This value of RAREF assumes that no external capacitance between VAREF and VAGND is used. Application Note 43 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Reference Voltage VAREF and VAGND 6.4 Ratiometric Configuration In a non-ratiometric configuration there is no relation between the voltage of the analog source and the reference voltage at pin VAREF. Both, the accuracy of the reference voltage and the accuracy of the analog source have an influence to the accuracy of the total A/D conversion system, because any changes in the supply voltage of the analog source results in a change at the analog input voltage ANx seen by the A/D converter. Since the voltage reference is independent from the analog source excitation, the ADC conversion result will reflect the changed excitation Figure 23 shows the principle of a ratiometric configuration. The same voltage reference source is used for the analog source excitation and the reference voltage input VAREF. Therefore a given change in the analog source excitation causes the same change at the reference voltage VAREF. The A/D converter conversion result is the ratio of the analog input ANx, to the reference voltage VAREF. Since both, the analog input ANx and the reference voltage VAREF are derived from the same voltage reference source, changes do not cause measurement errors. Hence, the A/D converter conversion result is independent to variations in the analog source excitation or variations in the reference voltage input VAREF. Because of that a stable voltage reference is not necessary to achieve an accurate measurement result. Voltage Reference Analog Source VDD RAREF Ceramic Tantalum VAREF CAREF Central Analog Ground ANx Power Supply 5V VDD GND VSS Microcontroller VAGND Central Digital Ground Figure 23 Ratiometric Configuration Application Note 44 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Overload and Leakage Current 7 Overload and Leakage Current Both, overload and leakage currents are specified in the Data Sheet. Consideration of overload and leakage currents can have an influence on the design of the external components of the analog source. Figure 24 is a simplified electrical model with ESD structure (clamp diodes) and leakage current of an analog input. Analog Source IOZ1 VDD Microcontroller ANx RASRC IOV>0 MUX IOV<0 IOZ1 VLEAK VSS V0 ESD Structure Figure 24 VSS Leakage Source A/D Converter Input with ESD Structure and Leakage Source Note: The ESD structure of the reference voltage VAREF and the reference ground VAGND is the same as shown in the Figure 24. 7.1 Leakage Current The maximum input leakage current of the A/D converter is specified in the Data Sheet in section ’DC Characteristics’. The input leakage current is the sum of all currents which can flow into or out of an input pin caused by parasitic effects of the input structure, see Figure 24. The symbols in the Data Sheets of the C500 and C166 Family used for the input leakage current of the A/D converter are different. For the C166 Family it is IOZ1 and for the C500 Application Note 45 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Overload and Leakage Current Family it is ILI. In this Application Note the symbol IOZ1 is used for the input leakage current. The specified value of the A/D converter input leakage current depends on the device type. Please refer to the Data Sheet for the exact value. The input leakage current has to be taken into account for the calculation of the maximum allowed error of the A/D converter result referred to the analog source. Because the resistance of the analog source RASRC and the input leakage current IOZ1 can cause an additional error via the external ’leakage voltage’ VLEAK. VLEAK = IOZ1 * RASRC The leakage voltage VLEAK can cause an additional unadjusted error AUELEAK. AUELEAK = VLEAK / 1LSB 7.1.1 Calculation Example Assumed system values: AUELEAK = 0.25 LSB Assumed maximum additional unadjusted error caused by resistance of the analog source RASRC VAREF =5V IOZ1 = |± 200 nA| 1 LSB = 4.9 mV (10-bit A/D converter) Specified maximum input leakage current What is the allowed maximum resistance of the analog source RASRC ? RASRC = VLEAK / IOZ1 RASRC = AUELEAK * 1LSB / IOZ1 RASRC = 0.25 LSB * 4.9 mV / 200 nA RASRC = 6125 Ω Note: The specified maximum Input Leakage Current of |+/- 200 nA| can reduce the conversion accuracy when the external resistance has a high value (>10 kOhm). Application Note 46 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Overload and Leakage Current 7.2 Overload Current An overload condition is not a normal operating condition. It occurs if the standard operating conditions are exceeded, i.e. the voltage on an A/D converter input pin VAINx exceeds the specified range (VAINx > VDD + 0.5 V or VAINx < VSS - 0.5 V). The supply voltage must remain within the specified limits. In case of an overload condition on an A/D converter input pin, one of the clamp diodes becomes conductive. If VAINx > VDD + 0.5 V then the clamp diode connected to VDD begins to conduct. If VAINx < VSS - 0.5 V then the clamp diode connected to VSS begins to conduct; See Figure 24. The overload current has to be taken into account for the calculation of external resistors which protect the microcontroller inputs. These external resistors guarantee that, in case of a system error, the specified maximum value of the overload current will not be exceeded. The calculation also has to consider the specified absolute sum of input overload currents on all port pins of the microcontroller and especially the specified absolute sum of the A/D converter input. 7.2.1 Overload Current and Absolute Maximum Ratings The parameters of the Absolute Maximum Ratings are stress ratings only and functional operation of the microcontroller is not guaranteed at these or other conditions above the ’operation conditions’. Stresses above the absolute maximum ratings may cause permanent damage to the microcontroller. Exposing the microcontroller to absolute maximum rating conditions for extended periods may affect device reliability. When the system is switched off or in periods where it is not necessary to guarantee correct operation, the absolute maximum ratings are the fundamental information for the calculation of the input overload current, which may occur in case of a system error. In those cases the specified maximum overload current is IOV = ±10 mA on any pin and the absolute sum of input overload currents on all port pins is 100 mA. For the exact values, please refer to the Data Sheet. 7.2.1.1 Calculation Example Assumed system values: =0V VDD VErr_max = 12 V IOV_max = |±10 mA| System supply voltage is off (worst case) Maximum voltage of the analog signal in case of a fatal system error Specified absolute maximum rating of the overload current What is the minimum value for the external resistor RP to protect an analog input pin for a short time overload condition? Application Note 47 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Overload and Leakage Current RP = (VErr_max - VDD - 0.5 V) / IOV_max RP = (12 V - 0 V - 0.5 V) / 10 mA RP 7.2.2 = 1150 Ω Overload Current and Operating Conditions The Operating Conditions must not be exceeded in order to ensure correct operation of the microcontroller. The specified operating conditions allow a maximum overload current of IOV = ±5 mA on any pin and the absolute sum over input overload currents on all port pins is |50| mA. The specified TUE of the A/D converter is guaranteed only if the absolute sum of input overload currents on all analog input pins does not exceed 10 mA. For the exact values please refer to the Data Sheet. 7.2.2.1 Calculation Example Assumed system values: VDD = 4.5 V VErr_max = 12 V IOV_max = |±5 mA| Minimum system supply voltage during operating conditions (worst case) Maximum voltage of the analog signal in case of a fatal system error Specified maximum of the overload current during operating conditions What is the minimum value of the external resistor RP to protect an analog input of the microcontroller and to ensure correct operation? RP = (VErr_max - VDD - 0.5 V) / IOV_max RP = (12 V - 4.5 V - 0.5 V) / 5 mA RP Application Note = 1400 Ω 48 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families PCB and Design Considerations 8 PCB and Design Considerations This chapter is a brief introduction in mixed signal board design with a list of guidelines for optimum printed circuit board layout for microcontrollers with on-chip A/D converter. 8.1 Component Placing Partition the board with all analog components grounded together in one area and all digital components in the other. Common power supply related components should be centrally located. Mixed signal components, including the microcontroller, should bridge the partitions with only analog pins in the analog area, and only digital pins in the digital area. Rotating the microcontroller can often make this task easier. 8.2 Power Supply Place the analog power and voltage reference regulators over the analog plane. The same holds for the digital power regulators. Analog power traces should be over the analog ground plane. The same holds for the digital power traces. Decoupling capacitors should be close to the microcontroller pins, or positioned for the shortest connection to pins with wide traces to reduce impedance. If both large electrolytic and small ceramic capacitors are recommended, make the small ceramic capacitor closest to the microcontroller pins. 8.3 Ground Planes Have separate analog and digital ground planes on the same layer, separated by a gap, with the digital components over the digital ground plane, and the analog components over the analog ground plane. Analog and digital ground planes should only be connected at one point (most cases). The best place is below the microcontroller. Have vias available in the board to allow alternative points. The analog to digital ground plane connection should be near to the power supply, or near to the power supply connections to the board, or near to the microcontroller. For boards with more than 2 layers, do not overlap analog related and digital related planes. Do not have a plane that crosses the gap between the analog ground plane and the digital ground plane region. Application Note 49 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families PCB and Design Considerations 8.4 Signal Lines Analog signal traces should be over the analog ground plane. Digital power and digital signal traces should be over the digital ground plane. Regions between analog signal traces should be filled with copper, which should be electrically attached to the analog ground plane. Regions between digital signal traces should be filled with copper, which should be electrically attached to the digital ground plane. These regions should not be left floating, which only makes the interference worse. Using ground plane fill has shown to reduce digital to analog coupling by up to 30 dB. 8.5 Clock Generation Locate quartz crystal, ceramic resonator or external oscillator as close as possible to the microcontroller. Keep digital signal traces, especially the clock signal, as far away from analog input and voltage reference pins as possible. Avoid multiple oscillators or asynchronous clocks. Best results are obtained when all circuits are synchronous to the A/D converter sampling clock. Application Note 50 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Used Short Cuts 9 Used Short Cuts ADC ANx AUELeak : Analog Digital Converter : Analog input X : Additional unadjusted error caused by the leakage current CAIN CAIN_max CAREF CEXT CHOLD CLSB CMSB C-Net C9 - C0 C7’ - C0’ chn : A/D converter input capacitance (internal C-net) : Maximum of the A/D converter input capacitance : External capacitance connected to the reference voltage input VAREF : External capacitance connected to the analog input : Hold capacitance of the A/D converter : LSB of the internal C-net : MSB of the internal C-net : Internal A/D converter capacitor network. : C-net for conversion (10-bit resolution). : C-net for calibration. : Analog channel n DNLE : Differential nonlinearity error E ErrorAINx : Variable for allowed Error to calculate CAREF : Maximum deviation between the voltage on ANx and V0 when the sample time is finished ErrorAINx_real : Real deviation between the voltage on ANx and V0 referred to the actual voltage at ANx ErrorAREF : Maximum voltage error at VAREF caused by CAREF ErrorMSB : Missing voltage to charge the MSB capacitance of the internal C-net to VAREF/2 during charge-redistribution phase ESD : Electrostatic discharge fCPU fC : CPU frequency : Cutoff frequency fCYCLE : Cycle frequency (fCYCLE = 1 / tCYCLEn) Application Note 51 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Used Short Cuts INLE IAREF ILi IOV IOV_max IOZ1 : Integral nonlinearity error : Current of the voltage reference : Input leakage current (C500 Family) : Overload current : Specified maximum rating of the overload current or : Specified maximum of the overload current during operating conditions : Input leakage current (C166 Family) LSB LSBr : Least significant bit (general) : Least significant bit referred to r-bit resolution (LSBr = VAREF / 2r) MSB : Most significant bit Q : Charge for a complete charge-redistribution- and a calibration phase r RAIN RASRC RAREF RAREF_INT RP : Resolution of the A/D converter : Internal series resistance of the A/D converter : Internal resistance of the analog source : Resistance between voltage reference and VAREF input : Internal resistance of the voltage reference : External resistor RP to protect an analog input in case of an overload condition tBC tC tCn tCR tCYCLE tCYCLEn tCHARGE tCOMPARE : A/D converter basic clock : Conversion time : Conversion time of analog channel n : Charge redistribution time : Cycle time : Cycle time of analog channel n : Time to charge the MSB capacitance to VAREF/2 within 2*tBC : Time to compare MSB voltage with sampled input voltage ANx within 2*tBC : Sample time : Internal clock, 2 * TCL = 1 / fcpu : Time constants for the different phases of a conversion : Total unadjusted error tS TCL τ1,=τ2,=τ3 TUE Application Note 52 V 1.0, 2001-05 AP2428.01 C500 / C166 Microcontroller Families Used Short Cuts VAREF VAGND VANx VCAIN VC(t) VC(tCYCLE) VDD VERROR VERR_max VLeak VMSB VMSB(t) VMSB(2tBC) VR VRF VSS VS(t) VS(tS) V0 V∆ : Reference voltage input for the A/D converter : Reference ground for the A/D converter : Voltage at the analog input ANx : Voltage at the internal C-net : Charge curve of CEXT for a total cycle : Voltage of CEXT at the end of a total cycle : Supply voltage : Voltage at RAREF : Maximum voltage of an analog signal in case of a fatal system error : Leakage voltage at RASRC : Voltage at the internal MSB of the C-net : Comparator voltage during conversion of MSB : Comparator voltage after 2*tBC : Missing rest voltage at the end of a conversion cycle : Voltage reference : Digital GND : Voltage during sample time : Voltage at the end of sample time : Voltage of the analog source : Voltage jump at the beginning of the sample time Application Note 53 V 1.0, 2001-05 Infineon goes for Business Excellence “Business excellence means intelligent approaches and clearly defined processes, which are both constantly under review and ultimately lead to good operating results. Better operating results and business excellence mean less idleness and wastefulness for all of us, more professional success, more accurate information, a better overview and, thereby, less frustration and more satisfaction.” Dr. Ulrich Schumacher http://www.infineon.com Published by Infineon Technologies AG