Application Note AN 2012-09 V1.0 September 2012 Resonant LLC Converter: Operation and Design 250W 33Vin 400Vout Design Example Sam Abdel-Rahman Infineon Technologies North America (IFNA) Corp. Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Resonant LLC Converter: Operation and Design Sam Abdel-Rahman Published by Infineon Technologies North America 27703 Emperor Blvd, suite 310 Durham, NC 27703 All Rights Reserved. Attention please! THE INFORMATION GIVEN IN THIS APPLICATION NOTE IS GIVEN AS A HINT FOR THE IMPLEMENTATION OF THE INFINEON TECHNOLOGIES COMPONENT ONLY AND SHALL NOT BE REGARDED AS ANY DESCRIPTION OR WARRANTY OF A CERTAIN FUNCTIONALITY, CONDITION OR QUALITY OF THE INFINEON TECHNOLOGIES COMPONENT. THE RECIPIENT OF THIS APPLICATION NOTE MUST VERIFY ANY FUNCTION DESCRIBED HEREIN IN THE REAL APPLICATION. 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Life support devices or systems are intended to be implanted in the human body, or to support and/or maintain and sustain and/or protect human life. If they fail, it is reasonable to assume that the health of the user or other persons may be endangered. AN 2012-09 V1.0 September 2012 Author: Sam Abdel-Rahman We Listen to Your Comments Any information within this document that you feel is wrong, unclear or missing at all? Your feedback will help us to continuously improve the quality of this document. Please send your proposal (including a reference to this document) to: [[email protected]] 2 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Table of contents 1 Introduction..................................................................................................................................................4 2 Overview of LLC Resonant Converter.......................................................................................................4 3 Design Steps ................................................................................................................................................8 4 Bridge and Rectifier Selection .................................................................................................................11 5 Design Example .........................................................................................................................................13 6 Schematics and Bill of Material ...............................................................................................................17 7 References .................................................................................................................................................19 3 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design 1 V1.0 September 2012 Introduction While a resonant LLC converter has several desired features such as high efficiency, low EMI and high power density, the design of a resonant converter is an involved task, and requires more effort for optimization compared to PWM converters. This document aims to simplify this task, and make it easier to optimally design the resonant tank. This document provides an overview of LLC converter operation and design guidelines. Finally, a comprehensive design example is given along with schematics, bill of materials, experimental results and waveforms. 2 Overview of LLC Resonant Converter This section offers an overview of the LLC converter operation and waveforms in the different modes. Figure 2.1 shows a Full-Bridge LLC converter with Full-Bridge rectifier. In a simplistic discussion, the switching bridge generates a square waveform to excite the LLC resonant tank, which will output a resonant sinusoidal current that gets scaled and rectified by the transformer and rectifier circuit, the output capacitor filters the rectified ac current and outputs a DC voltage. Switching bridge S1 Output Capacitor S3 Cr D1 Lr + Vsw - S2 Transformer and Rectifier LLC tank D3 Np Ns Co Lm S4 D2 Ro + Vo - D4 Figure 2.1 Full-Bridge LLC converter with Full-Bridge rectifier 2.1 Converter Voltage Gain Converter gain= switching bridge gain * resonant tank gain * transformer turn ratio (Ns/Np) Where the switching bridge gain is 1 for a Full-Bridge and 0.5 for a Half-Bridge. The resonant tank gain can be derived by analyzing the equivalent resonant circuit shown in Figure 2.2, the resonant tank gain is the magnitude of its transfer function as in Eq. 1. Lr Cr Vin_ac Lm Vo_ac Rac Figure 2.2 Equivalent resonant circuit K (Q , m, Fx ) Vo _ ac ( s ) Vin _ ac ( s ) Fx m 1 2 m F x 2 1 Fx Fx 1 m 1 Q 2 2 2 4 2 2 Eq. 1 2 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Where, Q Lr C r Rac Rac 8 NP Ro 2 NS2 Reflected load resistance Fx fs fr Normalized switching frequency fr 1 2 Lr Cr m Lr Lm Lr Quality factor 2 Resonant frequency Ratio of total primary inductance to resonant inductance One can plot the resonant tank gain K with the normalized switching frequency for different values of Quality factor Q and any single value of m, as shown in Figure 2.3. The selection of the m value will be discussed in a later section of this document, but m=6 was used as an example. It can be seen in Figure 2.3 that low Q curves belong to lighter load operation while higher Q curves represent heavier loads. It’s also seen that all Q curves (load conditions) cross at the resonant frequency point (at Fx=1 or fs=fr) and have a unity gain. Figure 2.3 shows that all gain curves has peaks which define the boundary between the inductive and capacitive impedances of the resonant tank, hence we can define the inductive and capacitive operation regions as shaded in the plot, the objective of defining both regions is because it is desired to maintain an inductive operation across the entire input voltage and load current ranges, and never fall into the capacitive region operation. Such requirement is due to that Zero Voltage Switching (ZVS) is only achieved in the inductive region, in addition to that capacitive operation means that current leads the voltage, so the current in the MOSFET will reverse direction before the MOSFET turns off, then after the MOSFET turns off the reverse current will flow in the MOSFET’s body diode, which will cause a body diode hard commutation once the other MOSFET in the bridge turns on, which in turn will cause reverse recovery losses and noise, and might cause high current spikes and device failure. The capacitive operation can be prevented and will be discussed in a later section of this document. 3 Light load K ( .2 , m , Fx) Capacitive region K ( .5 , m , Fx) ZCS K ( .3 , m , Fx) m=6 Inductive region ZVS 2 K ( .7 , m , Fx) K ( 1 , m , Fx) K ( 5 , m , Fx) Heavy load 1 0 0.1 1 Fx 10 Figure 2.3 2.2 Modes of Operation Since the LLC network gain is frequency modulated, the converter can operate in three modes depending on input voltage and load current conditions, as listed below and shown in Figure 2.4: 5 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 1. At resonant frequency operation, fs=fr. 2. Above resonant frequency operation fs>fr. 3. Below Resonant frequency operation, fs<fr. 3 Below resonance At resonance 2 K ( .2 , m , Fx) Above resonance K ( 10 , m , Fx) 1 0 0.1 1 Fx 10 Figure 2.4 Despite the aforementioned three modes; which will be explained in details later in this section; the converter has only two possible operations within the switching cycle, as described below. And each of the modes pointed out above may contain one or both of these operations. 1) Power delivery operation, which occurs twice in a switching cycle; first, when the resonant tank is excited with a positive voltage, so the current resonates in the positive direction in the first half of the switching cycle, the equivalent circuit of this mode is shown in Figure 2.5, and second occurance is when the resonant tank is excited with negative voltage, so the current resonates in the negative direction in the second half of the switching cycle, the equivalent circuit of this mode is shown in Figure 2.6. During the power delivery operations, the magnetizing inductor voltage is the positive/negative reflected output voltage and the magnetizing current is charging/discharging respectively. The difference between the resonant current and the magnetizing current passes through the transformer and rectifier to the secondary side, and power is delivered to the load. S1 S3 Cr Lr Np Ns D1 S4 S3 Cr + Vo - Lm S2 S1 D3 D2 Lr Np Ns D1 D3 + Vo - Lm S2 D4 Figure 2.5 S4 D2 D4 Figure 2.6 2) Freewheeling operation, which can occurs following the power delivery operation only if the resonant current reaches the transformer magnetizing current, this only happens when fs<fr, causing the transformer secondary current to reach zero and the secondary side rectifier to disconnect, consequently the magnetizing inductor will be free to enter the resonance with the resonant inductor and capacitor, the frequency of this second resonance is smaller than the original resonant frequency fr, especially at high values of m where Lm>>Lr, thus the primary current during the freewheeling operation will only change slightly, and can be approximated to be unchanged for simplicty. The equivalent circuits of the freewheeling operation in both halves of the switching cycle are shown in Figure 2.7 and Figure 2.8. 6 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design S1 S3 Cr Lr Np D1 Ns S1 D3 S3 Cr Lr + Vo - Lm S2 V1.0 September 2012 S4 D2 Np Ns D1 D3 + Vo - Lm S2 D4 S4 D2 Figure 2.7 D4 Figure 2.8 Table 1 explains the converter modes of operation and shows key waveforms Table 1 At Resonant frequency operation fs=fr. Above resonant frequency operation fs>fr. Below resonant frequency operation fs<fr Each half of the switching cycle contains a complete power delivery operation (described above), where the resonant half cycle is completed during the switching half cycle. By end of the switching half cycle, the resonant inductor current ILr reaches the magnetizing current ILm, and the rectifier current reaches zero. The resonant tank has unity gain and best optimized operation and efficiency, therefore, transformer turns ratio is designed such that the converter operates at this point at nominal input and output voltages. Each half of the switching cycle contains a partial power delivery operation (described above), similar to the resonant frequency operation, but it differs in that the resonant half cycle is not completed and interrupted by the start of the other half of the switching cycle, hence primary side MOSFETs have increased turn off losses and secondary rectifier diodes have hard commutation. The converter operates in this mode at higher input voltage, where a step down gain or buck operation is required. Each half of the switching cycle contains a power delivery operation (described above), at the time when resonant half cycle is completed and resonant inductor current ILr reaches the magnetizing current, the freewheeling operation (as described above) starts and carries on to the end of the switching half cycle, hence primary side have increased conduction losses due to the circulating energy. The converter operates in this mode at lower input voltage, where a step up gain or boost operation is required. Ts/2 Ts=1/fs Ts/2 Ts=1/fs S1,S4 S2,S3 S1,S4 S2,S3 Vin S1,S4 S2,S3 Vin Vin Vds_S2,S3 Vds_S2,S3 ILr Tr/2 Ts/2 Ts=1/fs Vds_S2,S3 ILr ILm ILr ILm ID1,D4 ID1,D4 ID1,D4 ID2,D3 ID2,D3 ID2,D3 t t 7 ILm t Application Note AN 2012-09 Resonant LLC Converter: Operation and Design 3 V1.0 September 2012 Design Steps This section is to explain the impact of design parameters on voltage regulation and efficiency performance, and facilitate the design and selection of such parameters. Our ultimate design objective it to achieve the best performance while reaching the gain requirement for all line and load conditions. And for safe operation, we must determine the minimum switching frequency the controller shall limit in order to maintain the operation in the inductive region. The following are detailed explanation of all design steps; additionally Figure 3.1 shows a design flow chart that summarizes the design methodology. START Step 1: Select Qmax value Step 2: Select m value Step 3: Find Fxmin Increase m value Decrease m value Step 4: Find Kmax Yes No Is Kmax < required gain? No Is Kmax = required gain? Yes Step 5: Solve for resonant components values END Figure 3.1 Design flow chart Step 1: Selecting the Qmax Value Quality factor Q Lr C r depends on the load current. Heavy load conditions operate at high Q values, Rac while lighter loads have lower Q values. It is important to set a value for the Qmax associated with the maximum load point. To illustrate the effect of the Q value on voltage regulation, Figure 3.2 shows an example voltage gain plot for different Q values. Let’s assume that the resonant tank gain is required to range from 0.8 to 1.2 for example, we can see that the low Q value curve (Q=0.3) can reach higher boost gain, but it is less sensitive to frequency modulation in the “above resonance fs>fr” region, hence, switching frequency has to increase much in order to reach the minimum voltage gain (K=0.8), causing extra switching losses, while the higher Q 8 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 value curve (Q=1) can reach the minimum gain (K=0.8) with less switching frequency increase, but unable to reach the maximum gain (K=1.2). Therefore, a moderate Q value of around 0.5 seems to satisfy the voltage gain requirement in this specific case. We conclude that adjusting the Q value can help achieving the maximum gain but increases the frequency modulation range, thus, we should not rely on tuning the Qmax value as a design iteration in order to reach the desired maximum voltage gain, but instead we should rely on tuning the m value as will be explained in the next step. Although there isn’t a direct method for selecting the optimum Q value, we should select Qmax moderately as discussed earlier and based on the specific design in hand. 2 K ( .3 m Fx) K ( .5 m Fx) K ( 1 m Fx) 1.5 1 1.2 0.8 0.5 0 0.1 1 Fx 10 Figure 3.2 Step 2: Selecting the m Value As mentioned above, m Lr Lm , m is a static parameter that we have to start the design by optimizing its Lr value, therefore it’s important to understand the impact of the m ratio on the converter operation. To illustrate the effect of the m value, Figure 3.3 shows the same resonant tank gain plots but for different m values, m= 3, 6 and 12. It is obvious that lower values of m can achieve higher boost gain, in addition to the narrower range of the frequency modulation, meaning more flexible control and regulation, which is valuable in applications with wide input voltage range. Nevertheless, low values of m for the same quality factor Q and resonant frequency fr means smaller magnetizing inductance Lm, hence, higher magnetizing peak-peak current ripple, causing increased circulating energy and conduction losses. We have to start by selecting a reasonable initial value for m (6-10), and then optimize it by few iterations to get the maximum m value that can still achieve the maximum gain requirement for all load conditions. Low m value: Higher boost gain Narrower frequency range More flexible regulation x High m value: Higher magnetizing inductance Lower magnetizing circulating current Higher efficiency 0 .001 10 3 3 3 K ( .2 , m , Fx) K ( .2 , m , Fx) m=3 K ( .2 , m , Fx) m=6 K ( .3 , m , Fx) K ( .3 , m , Fx) K ( .3 , m , Fx) K ( .5 , m , Fx) 2 K ( .5 , m , Fx) K ( .7 , m , Fx) K ( 1 , m , Fx) K ( 5 , m , Fx) 2 K ( .5 , m , Fx) K ( .7 , m , Fx) K ( 1 , m , Fx) 1 0 0.1 K ( 5 , m , Fx) 1 Fx 10 m=12 2 K ( .7 , m , Fx) K ( 1 , m , Fx) 1 K ( 5 , m , Fx) 0 0.1 1 Fx Figure 3.3 9 10 1 0 0.1 1 Fx 10 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Step 3: Finding the Minimum Normalized Switching Frequency After selecting a value for Qmax and an initial value for m, we need to find the minimum normalized switching frequency that will guarantee inductive operation for the Qmax (max load) condition, this minimum frequency will also guarantee inductive operation for all other loads. The minimum normalized switching frequency occurs at the peak gain of the Qmax curve, so it can be found by solving Eq. 2 (assumed Qmax=0.4 and m=6 as an example), or can be visually spotted in the gain plot as in Figure 3.4. d K (Q, m, Fxmin ) 0 dFx Q max0.4 , m6 Eq. 2 Solve for Fx min 3 Light load K ( .2 , m , Fx) m=6 K ( .25 , m , Fx) K ( .3 , m , Fx) Max load curve (Qmax=0.4) 2 K ( .4 , m , Fx) K ( .6 , m , Fx) K ( 5 , m , Fx) 1 0 0.1 Fxmin 1 Fx 10 Figure 3.4 Step 4: Voltage Gain Verification This step is to verify that the maximum gain Kmax reached during maximum load by the selected m value is adequate. This can be done by solving Eq. 3, or can be visually spotted in the gain plot as in Figure 3.4. K max K (Qmax , m, Fxmin ) Eq. 3 3 Light load K ( .2 m Fx) m=6 K ( .25 m Fx ) K ( .3 m Fx) 2 K ( .4 m Fx) Kmax K ( .6 m Fx) K ( 5 m Fx) Max load curve (Qmax=0.4) 1 0 0.1 Fxmin 1 Fx Figure 3.5 10 10 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Few iterations are needed in order to reach the optimized design, as shown in the design flow chart in Figure 3.1. If Kmax is not enough, then we have to reduce the m value and repeat steps 3 and 4, in order to gain a higher boost gain. On the other side, If Kmax is higher than what is required; in that case we can increase the m value and repeat steps 3 and 4 in order to gain a better efficiency. Step 5: Calculating Resonant Components Values After few iterations of the design flow and reaching the optimum m value, we can proceed to calculating the resonant tank components values, Eq. 4 and Eq. 5 can be solved to find Lr and Cr, and then Lm can be calculated using Eq. 6. 2 Rac ,min Qmax 2 8 N V 2 P2 o N S Po max Lr C r Rac ,min Eq. 4 fr 1 2 LrCr Eq. 5 m Lr Lm Lr Eq. 6 It must be noted that selection of the resonant frequency fr was not considered in the design steps above, since it has no impact on the maximum gain and operation region of the resonant converter, however it is selected considering the converter power density and power losses. 4 Bridge and Rectifier Selection An important step to achieve the best converter performance is to choose the right bridge and rectifier circuits. LLC converters can be implemented with a full-bridge or a half-bridge circuit on the primary side, as shown in Figure 4.1, S1 S3 C1 S1 S2 S4 C2 S2 Full-bridge switching circuit Half-bridge switching circuit Figure 4.1 Table 2 shows a comparison between the half-bridge and the full-bridge switching circuits. A half-bridge would have twice the current of what a full-bridge would have, the squared rms current in the half-bridge case is four times, the number of switches in a half bridge is half of that in a full-bridge, therefore, the total FETs conduction losses of a half-bridge is twice compared to the full-bridge. Although a half-bridge requires half the primary number of turns for the same voltage gain and magnetic flux swing, thus half the primary winding resistance, the primary copper losses are still twice compared to the fullbridge because the squared rms current in the half-bridge case is four times. So in applications with high primary currents, where conduction losses are dominant, it is suggested to use a full-bridge switching circuit. 11 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Table 2 Primary Bridge - Half-Bridge compared to Full-Bridge Irms Irms2 Number of FETs Total FETs conduction Np Rpri Transformer primary losses ×2 ×4 ÷2 copper loss ×2 ÷2 ÷2 ×2 *Comparison assumes same FET and same transformer core LLC converters can also be implemented with a full-bridge or a full-wave rectifier circuit on the secondary side, as shown in Figure 4.2 D2 Np Ns D1 D3 Co D2 Np + Vo - Ns + Vo - Ns D4 D1 Full-bridge rectifier Full-wave rectifier Figure 4.2 Table 3 shows a comparison between the full-bridge and full-wave rectifiers. A full-wave rectifier requires diodes that are twice the voltage rating compared to a full-bridge rectifier, but it has only two diodes while the full-bridge rectifier has four diodes, since each diode in both rectifier circuits carries the same average current, the full-wave rectifier has half the total diode conduction losses compared to the full-bridge rectifier. A full-wave rectifier has two secondary windings, hence the resistance is doubled for the same winding area, each winding in a full-wave caries an rms current that is √0.5 of the rms current of the full-bridge circuit, therefore the total secondary windings copper losses of the full-wave rectifier is twice compared to the fullbridge rectifier. In applications with high output voltages, the full bridge rectifier is advantageous since we can use diodes with half the voltage rating compared to the full-wave rectifier. While in low output voltages and high currents application, the full-wave is more common, because of lower total conduction losses and lower component count and cost. Table 3 Secondary Rectifier - Full-Wave compared to Full-Bridge Diode Number of voltage diodes Diode Number of Rsec Irms Transformer secondary conduction secondary per winding per winding copper loss losses windings ×2 × √0.5 ×2 rating ×2 ÷2 ÷2 ×2 *Comparison assumes same diode voltage drop and same transformer core 12 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design 5 Design Example 5.1 Application and Specifications V1.0 September 2012 Our design example is applicable to the dc-dc stage of a solar micro inverter, as shown in Figure 5.1, with specifications as listed in Table 4. According to the discussion in section 4, the LLC converter will be implemented with a full-bridge on the primary side and a full-bridge rectifier on the secondary side, same circuit as shown in Figure 2.1. ~30V 400V DC-DC DC Link Converter DC-AC Inverter 240Vac 60Hz Figure 5.1 Table 4 Specifications Output voltage 400V Input voltage 18V – 36V (33V nominal) Output power 250W Output power derates linearly with input voltage Ex: Output power= 125W @ Vin=18V Resonant frequency 5.2 100kHz Design Steps We start the design by calculating the transformer turn ratio and the minimum and maximum voltage gains of the resonant tank, as follows. ܯ = 1 ܰ ܸ_ ܰ = ∙ ܯ → = 0.0825 ܰ௦ ܸ௨௧ ܰ௦ ܸ_ ܯ ௫ = ∙ ܯ → ܯ ௫ = 1.833 ܸ_ ܸ_ ܯ = ∙ ܯ → ܯ = 0.917 ܸ_ ௫ Next, we follow the design steps discussed in section 3, as follows. Step 1: Selecting the Qmax Value Let’s choose ܳ ௫ = 0.4 Step 2: Selecting the m Value Let’s choose ݉ = 6.3 Step 3: Finding the Minimum Normalized Switching Frequency We can solve the gain equation to find the minimum frequency, which occurs at the peak of the Qmax curve, as shown below, or we can look it up from the gain plot as shown in Figure 5.2. 13 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design ݀ ܳ(ܭ ௫ , ݉ , ܨ )ฬ ݀ܨ ொ , ୀ.ଷ ೌೣ ୀ.ସ =0 V1.0 September 2012 → ܨ = 0.489 → ݂௦_ = ܨ ∙ ݂ = 48.9݇ݖܪ Step 4: Voltage Gain Verification Since the power is derated at lower input voltages as listed in the specifications, we have to calculate the maximum Q value at the minimum input voltage case (ܳ [email protected] ), as follows. Note that this power derating specification is related to the solar panel I-V characterestics. (In case of other applications where power rating is the same across the input voltage range, we only have a single Qmax value). ܸ_ ܳ [email protected] = ܳ ௫ ∙ = 0.2 ܸ_ ௫ Then we can calculate the maximum gain reached at the minimum switching frequency for the ܳ [email protected] condition, or we can look it up from the gain plot as shown in Figure 5.2. ܭ ௫ = ܳ(ܭ [email protected] , ݉ , ܨ ) = 1.974 ܭ ௫ = 1.974 > ܯ ௫ = 1.833 → No need for tuning the m value 3 K ( .2 m Fx) 2 [email protected]=0.2 1.974 Qmax=0.4 K ( .3 m Fx) K ( .4 m Fx) 1 0 0.1 0.489 1 Fx 10 Figure 5.2 Step 5: Calculating Resonant Components Values The reflected load resistance at full load is, ܴ_ = 8 ܰ ଶ ܸ௨௧ଶ ∙൬ ൰ ߨଶ ܰ௦ ܲ, ௫ → ܴ_ = 3.534 Ω Next we solve the equations below to obtain the resonant tank components values ඥܮ/ܥ 3.534 Ω 1 ݂ = 100݇= ݖܪ 2ߨඥܮ ∙ ܥ ܮ + ܮ ݉ = 6.3 = ܮ ∴ ܮ = 2.25ߤܮ ܪ = 11.93ߤܥ ܪ = 1.13ߤܨ ܳ ௫ = 0.4 = 14 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design 5.3 V1.0 September 2012 Experimental Waveforms and Efficiency The design example was implemented with the specification shown in Table 5 Table 5 Prototype specifications Resonant frequency ݂ 110 kHz Resonant capacitor ܥ 0.94 µF Minimum switching frequency ݂௦_ 50 kHz Transformer Specifications Turns ratio ܰ : ܰ௦ 1:12 Magnetizing inductor ܮ 12.2 µH Leakage (Resonant) inductor ܮ 2.2 µH Figure 5.3 through Figure 5.6 shows experimental waveforms at different input voltage conditions, Red channel: Primary FET Vgs Yellow channel: Primary FET Vds Green channel: Resonant current ILr Blue channel: Rectifier output current ID1+ID3 Vin= 33V Po=250W Vin= 18V Po=125W Figure 5.3 Figure 5.4 Vin= 36V Po=250W Vin= 36V Light Load Missing Cycle Mode Figure 5.5 Figure 5.6 15 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Table 6 Efficency data Output Power (% of 250W) Input Voltage 20% 40% 60% 80% 100% 36V 97.1% ** 97.1%** 97.1% 97.1% 97.1% 33V 96.0% 97.2% 97.6% 97.6% 97.4% 24V 94.5% 96.8% 97.1% 97.0% 18V 94.0% 96.3% 96.2% ** Missing cycle mode / Burst mode operation 98.0% 97.6% 97.4% 96.9% 97.0% Efficiency % 97.6% 97.4% 97.5% 96.5% 96.0% 96.0% Vin= 33V 95.5% Vin= 24V 95.0% Vin= 36V 94.5% Vin= 18V 94.0% 93.5% 0 50 100 150 200 Output Power (W) Figure 5.7 Efficency curves 16 250 300 Application Note AN 2012-09 Resonant LLC Converter: Operation and Design 6 V1.0 September 2012 Schematics and Bill of Material C1 INPUT W1 Jumper INPUT INPUT+ C2 1 2 C3 10uF/100V 10uF/100V D1 1N5819HW-7-F R1 Q1 C7 D2 1N5819HW-7-F R2 SA CON1 400V 2.2nF/500VAC R3 10k Q3 D3 IDH05G65C5 Q2 SC 0 C8 BSC028N06NS 0 BSC028N06NS C5 D4 2.2u/450V IDH05G65C5 C6 2.2u/450V R4 10k Q4 + C4 NC 10uF/100V 10uF/100V PGND INPUT- PBSS4032PT,215 C12 1 2 C13 SW2 + C25 NC C10 2.2u/450V PBSS4032PT,215 SW1 + C9 NC W2 Jumper + C27 NC C11 2.2u/450V T1 10uF/100V 10uF/100V OUT+ 1 2 + C26 NC CON5 + C28 NC + C29 NC + C30 NC CON2 C14 0.47uF/250V C15 0.47uF/250V D5 IDH05G65C5 + C31 NC D6 IDH05G65C5 OUT1 2 T2 Trans CT 12V 12V D8 1N5819HW-7-F R6 R7 NC Trans Q6 SD 1 BSC028N06NS 0 BSC028N06NS R9 10k Q7 CT2 Q5 0 CT1 D7 1N5819HW-7-F R5 SB CON6 D9 2 R10 10k Q8 V- V+ AC AC 4 R8 1k C16 1u Bridge R12 1k D10 12V 12V PBSS4032PT,215 1 2 400V Q9 MMBT3904 3 R14 R13 402k 12V_SEC C17 NC 1u PBSS4032PT,215 R15 402k CON3 GND 1 2 R18 D11 CON4 CSCT1 CS- 1 V- 2 AC 4 CS+ V+ 3 AC FB CS+ 4 C20 10n CT2 10 IC1 12V_SEC SFH6186-3T C22 10uF/25V 4 +Vin NC NC NC -Vout 8 6 R25 12V_SEC 12V_SEC 12V_SEC 4 1 3 U2 AP331AWG-7 1 3 NC 5 FB R27 10k 4 R29 10k IC4 1 3 2 2 R32 0 C24 NC 12V 12V 12V C32 C33 IC5 100n 1 2 C34 SA 100n SW1 SW1 SA 3 4 VDD 8 SB LO HB LI HS HI 1 7 VSS HO IC6 100n SB 2 6 5 C35 R33 SC 100n R35 10 SC 3 SW2 SW2 4 VDD LO HB VSS HO LI HS HI 8 SD R34 R36 10 5 10 FB 10 FB 1 2 EN 3 4 Load R37 5 9.1k/1% 6 R39 7 5.6k/1% VREF R43 220k 30k R50 1M R49 C40 2.2u C42 1u C44 C45 1n R41 62k 8 R44 30k R4510 NC 9 C41 10u R56 30k VCC HG SS LG LOAD SHG FREQ SLG Delay GND TD SRD Vmc CL IC7 CS ICE2HS01G Vres VINS Vref 20 19 R57 R58 12k NC 12V R59 0 Figure 6.1 Schematics 17 C37 12V C36 100n 100n 18 17 16 15 R40 NC 14 C38 R38 0 13 12 220k R54 NC R42 47nF C39 470p 11 R46 4.7/1W R53 18k 10n CS+ 0 R48 0 C43 INPUT R52 1k R55 100k Timer EnA R51 NC SD 7 6 CS- CS+ CS- R28 IC3 1k TL431 R31 10k 2 Q10 2N7002 SFH6186-3T C46 NC R26 5.6k 12V_SEC R63 Load R61 402k R24 NC 0 C23 10uF/25V Isolated DC/DC NC R30 R60 402k 330n IC2 TL431 1 5 +Vout R47 402k R23 10k 7 2 3 D12 13V NC 3 -Vin 2 C21 100n 3.3k U1 1 0 NC C19 R21 R11 402k R19 402k NC R22 2 Bridge 12V NC C18 400V C47 R64 3.9k R20 1k 1 3 R17 402k R16 R62 10k Application Note AN 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Table 7 Bill of Material Qty Designator Value Part Number 1 6 9 4 2 3 C1 C2, C3, C7, C8, C12, C13 C4, C9, C25, C26, C27, C28, C29, C30, C31 C5, C6, C10, C11 C14, C15 C16, C17, C44 C18, C24, C42, C46, C47, R7, R14, R24, R30, R40, R45, R54, R58, R63, R64 C19 C20, C43 C21, C32, C33, C34, C35, C36, C37 C22, C23 C38 C39 C40 C41 C45 D1, D2, D7, D8 D3, D4, D5, D6 D9, D11 D10 D12 IC1, IC4 IC2, IC3 IC5, IC6 IC7 Q1, Q2, Q5, Q6 Q3, Q4, Q7, Q8 Q9 Q10 R1, R2, R5, R6, R22, R25, R32, R38, R42, R48, R59 R3, R4, R9, R10, R23, R27, R29, R31, R62 R8, R12, R20, R28, R52 R11, R13, R15, R17, R19, R47, R60, R61 R16 R18, R33, R34, R35, R36 R21 R26 R37 R39 R41 R43, R51 R44, R50, R56 R46 R49 R53 R55 R57 T1 T2 U1 U2 2.2nF/500VAC 10uF/100V NC 2.2u/450V 0.47uF/250V 1u VY1222M47Y5UQ63V0 C5750X7S2A106M B32674D4225K B32652A3474J C3216X7R1H105K NC 330n 10n 100n 10uF/25V 47nF 470p 2.2u 10u 1n BAS 3010A IDH05G65C5 Bridge 12V 13V SFH6186-3T TL431 SO8 ICE2HS01G BSC028N06NS PBSS4032PT,215 MMBT3904 2N7002 GRM319R71H334KA01D GRM319R71H103KA01D GRM319R71H104KA01D TMK316B7106KL-TD GRM31M5C1H473JA01L CC1206KRX7R9BB471 C3216Y5V1H225Z/0.85 TMK316B7106KL-TD GRM3195C1H102JA01D BAS 3010A-03W E6327 IDH05G65C5 BAS3007A-RPP SMAZ12-TP 3SMAJ5928B-TP SFH6186-3T TL431CPK LM5100AM/NOPB ICE2HS01G BSC028N06NS PBSS4032PT,215 MMBT3904FSCT-ND 2N7002 L6327 0 10k 1k 402k 3.9k 10 3.3k 5.6k 9.1k/1% 5.6k/1% 62k 220k 30k 4.7/1W 1M 18k 100k 12k Transformer Trans CT Isolated DC/DC AP331AWG-7 ERJ-8GEY0R00V ERJ-8ENF1002V ERJ-8GEYJ102V ERJ-8ENF4023V ERJ-8GEYJ392V ERJ-8GEYJ100V ERJ-8GEYJ332V ERJ-8ENF5601V ERJ-8ENF9101V ERJ-8ENF5601V ERJ-8ENF6202V ERJ-8ENF2203V ERJ-8ENF3002V ERJ-1TRQF4R7U ERJ-8ENF1004V ERJ-8ENF1802V ERJ-8ENF1003V ERJ-8ENF1202V Custom** B82801B504A50 VBT1-S12-S12-SMT AP331AWG-7 15 1 2 7 2 1 1 1 1 1 4 4 2 1 1 2 2 2 1 4 4 1 1 11 9 5 8 1 5 1 1 1 1 1 2 3 1 1 1 1 1 1 1 1 1 **Transformer built by Midcom-Wurth Electronics, E41/17/12-3C90 18 Application Note AN 2012 2012-09 Resonant LLC Converter: Operation and Design V1.0 September 2012 Figure 6.2 7 References [1] Infineon Technologies: ICE2HS01G datasheet, High Performance Resonant Mode Controller, V1.1, August 2011. [2] Infineon Technologies: Design Guide for LLC Converter with ICE2HS01G, ICE2HS01G, V1.0, July 2011. [3] Infineon Technologies: 300W W LLC Evaluation Board with LLC controller ICE2HS01G, V1.1, August 2011. 19

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