Application Note OptiMOS™ Resonant LLC Converter Operation and Design

Application Note AN 2012-09
V1.0 September 2012
Resonant LLC Converter: Operation and Design
250W 33Vin 400Vout Design Example
Sam Abdel-Rahman
Infineon Technologies North America (IFNA) Corp.
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Resonant LLC Converter: Operation and Design
Sam Abdel-Rahman
Published by Infineon Technologies North America
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Durham, NC 27703
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AN 2012-09
V1.0 September 2012
Author: Sam Abdel-Rahman
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2
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Table of contents
1 Introduction..................................................................................................................................................4
2 Overview of LLC Resonant Converter.......................................................................................................4
3 Design Steps ................................................................................................................................................8
4 Bridge and Rectifier Selection .................................................................................................................11
5 Design Example .........................................................................................................................................13
6 Schematics and Bill of Material ...............................................................................................................17
7 References .................................................................................................................................................19
3
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
1
V1.0 September 2012
Introduction
While a resonant LLC converter has several desired features such as high efficiency, low EMI and high
power density, the design of a resonant converter is an involved task, and requires more effort for
optimization compared to PWM converters. This document aims to simplify this task, and make it easier to
optimally design the resonant tank. This document provides an overview of LLC converter operation and
design guidelines. Finally, a comprehensive design example is given along with schematics, bill of materials,
experimental results and waveforms.
2
Overview of LLC Resonant Converter
This section offers an overview of the LLC converter operation and waveforms in the different modes. Figure
2.1 shows a Full-Bridge LLC converter with Full-Bridge rectifier. In a simplistic discussion, the switching
bridge generates a square waveform to excite the LLC resonant tank, which will output a resonant sinusoidal
current that gets scaled and rectified by the transformer and rectifier circuit, the output capacitor filters the
rectified ac current and outputs a DC voltage.
Switching
bridge
S1
Output
Capacitor
S3
Cr
D1
Lr
+
Vsw
-
S2
Transformer
and Rectifier
LLC tank
D3
Np Ns
Co
Lm
S4
D2
Ro
+
Vo
-
D4
Figure 2.1 Full-Bridge LLC converter with Full-Bridge rectifier
2.1
Converter Voltage Gain
Converter gain= switching bridge gain * resonant tank gain * transformer turn ratio (Ns/Np)
Where the switching bridge gain is 1 for a Full-Bridge and 0.5 for a Half-Bridge.
The resonant tank gain can be derived by analyzing the equivalent resonant circuit shown in
Figure 2.2, the resonant tank gain is the magnitude of its transfer function as in Eq. 1.
Lr
Cr
Vin_ac
Lm
Vo_ac
Rac
Figure 2.2 Equivalent resonant circuit
K (Q , m, Fx ) 
Vo _ ac ( s )
Vin _ ac ( s )
Fx m  1
2

m  F
x
2



 1  Fx  Fx  1  m  1  Q
2
2
2
4
2
2
Eq. 1
2
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Where,
Q
Lr C r
Rac
Rac 
8 NP

 Ro
 2 NS2
Reflected load resistance
Fx 
fs
fr
Normalized switching frequency
fr 
1
2 Lr  Cr
m
Lr  Lm
Lr
Quality factor
2
Resonant frequency
Ratio of total primary inductance to resonant inductance
One can plot the resonant tank gain K with the normalized switching frequency for different values of Quality
factor Q and any single value of m, as shown in Figure 2.3. The selection of the m value will be discussed in
a later section of this document, but m=6 was used as an example.
It can be seen in Figure 2.3 that low Q curves belong to lighter load operation while higher Q curves
represent heavier loads. It’s also seen that all Q curves (load conditions) cross at the resonant frequency
point (at Fx=1 or fs=fr) and have a unity gain.
Figure 2.3 shows that all gain curves has peaks which define the boundary between the inductive and
capacitive impedances of the resonant tank, hence we can define the inductive and capacitive operation
regions as shaded in the plot, the objective of defining both regions is because it is desired to maintain an
inductive operation across the entire input voltage and load current ranges, and never fall into the capacitive
region operation. Such requirement is due to that Zero Voltage Switching (ZVS) is only achieved in the
inductive region, in addition to that capacitive operation means that current leads the voltage, so the current
in the MOSFET will reverse direction before the MOSFET turns off, then after the MOSFET turns off the
reverse current will flow in the MOSFET’s body diode, which will cause a body diode hard commutation once
the other MOSFET in the bridge turns on, which in turn will cause reverse recovery losses and noise, and
might cause high current spikes and device failure. The capacitive operation can be prevented and will be
discussed in a later section of this document.
3
Light load
K ( .2 , m , Fx)
Capacitive
region
K ( .5 , m , Fx)
ZCS
K ( .3 , m , Fx)
m=6
Inductive region
ZVS
2
K ( .7 , m , Fx)
K ( 1 , m , Fx)
K ( 5 , m , Fx)
Heavy load
1
0
0.1
1
Fx
10
Figure 2.3
2.2
Modes of Operation
Since the LLC network gain is frequency modulated, the converter can operate in three modes depending on
input voltage and load current conditions, as listed below and shown in Figure 2.4:
5
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
1. At resonant frequency operation, fs=fr.
2. Above resonant frequency operation fs>fr.
3. Below Resonant frequency operation, fs<fr.
3
Below resonance
At resonance
2
K ( .2 , m , Fx)
Above
resonance
K ( 10 , m , Fx)
1
0
0.1
1
Fx
10
Figure 2.4
Despite the aforementioned three modes; which will be explained in details later in this section; the converter
has only two possible operations within the switching cycle, as described below. And each of the modes
pointed out above may contain one or both of these operations.
1) Power delivery operation, which occurs twice in a switching cycle; first, when the resonant tank is
excited with a positive voltage, so the current resonates in the positive direction in the first half of the
switching cycle, the equivalent circuit of this mode is shown in Figure 2.5, and second occurance is
when the resonant tank is excited with negative voltage, so the current resonates in the negative
direction in the second half of the switching cycle, the equivalent circuit of this mode is shown in
Figure 2.6.
During the power delivery operations, the magnetizing inductor voltage is the positive/negative
reflected output voltage and the magnetizing current is charging/discharging respectively.
The difference between the resonant current and the magnetizing current passes through the
transformer and rectifier to the secondary side, and power is delivered to the load.
S1
S3
Cr
Lr
Np
Ns
D1
S4
S3
Cr
+
Vo
-
Lm
S2
S1
D3
D2
Lr
Np
Ns
D1
D3
+
Vo
-
Lm
S2
D4
Figure 2.5
S4
D2
D4
Figure 2.6
2) Freewheeling operation, which can occurs following the power delivery operation only if the resonant
current reaches the transformer magnetizing current, this only happens when fs<fr, causing the
transformer secondary current to reach zero and the secondary side rectifier to disconnect,
consequently the magnetizing inductor will be free to enter the resonance with the resonant inductor
and capacitor, the frequency of this second resonance is smaller than the original resonant
frequency fr, especially at high values of m where Lm>>Lr, thus the primary current during the
freewheeling operation will only change slightly, and can be approximated to be unchanged for
simplicty. The equivalent circuits of the freewheeling operation in both halves of the switching cycle
are shown in Figure 2.7 and Figure 2.8.
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Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
S1
S3
Cr
Lr
Np
D1
Ns
S1
D3
S3
Cr
Lr
+
Vo
-
Lm
S2
V1.0 September 2012
S4
D2
Np
Ns
D1
D3
+
Vo
-
Lm
S2
D4
S4
D2
Figure 2.7
D4
Figure 2.8
Table 1 explains the converter modes of operation and shows key waveforms
Table 1
At Resonant frequency
operation
fs=fr.
Above resonant frequency
operation
fs>fr.
Below resonant frequency
operation
fs<fr
Each half of the switching cycle
contains a complete power
delivery operation (described
above), where the resonant half
cycle is completed during the
switching half cycle.
By end of the switching half cycle,
the resonant inductor current ILr
reaches the magnetizing current
ILm, and the rectifier current
reaches zero.
The resonant tank has unity gain
and best optimized operation and
efficiency, therefore, transformer
turns ratio is designed such that
the converter operates at this
point at nominal input and output
voltages.
Each half of the switching cycle
contains a partial power delivery
operation (described above),
similar to the resonant frequency
operation, but it differs in that the
resonant half cycle is not
completed and interrupted by the
start of the other half of the
switching cycle, hence primary
side MOSFETs have increased
turn off losses and secondary
rectifier diodes have hard
commutation.
The converter operates in this
mode at higher input voltage,
where a step down gain or buck
operation is required.
Each half of the switching cycle
contains a power delivery
operation (described above), at
the time when resonant half cycle
is completed and resonant
inductor current ILr reaches the
magnetizing current, the
freewheeling operation (as
described above) starts and
carries on to the end of the
switching half cycle, hence
primary side have increased
conduction losses due to the
circulating energy.
The converter operates in this
mode at lower input voltage,
where a step up gain or boost
operation is required.
Ts/2
Ts=1/fs
Ts/2
Ts=1/fs
S1,S4
S2,S3
S1,S4
S2,S3
Vin
S1,S4
S2,S3
Vin
Vin
Vds_S2,S3
Vds_S2,S3
ILr
Tr/2
Ts/2
Ts=1/fs
Vds_S2,S3
ILr
ILm
ILr
ILm
ID1,D4
ID1,D4
ID1,D4
ID2,D3
ID2,D3
ID2,D3
t
t
7
ILm
t
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
3
V1.0 September 2012
Design Steps
This section is to explain the impact of design parameters on voltage regulation and efficiency performance,
and facilitate the design and selection of such parameters. Our ultimate design objective it to achieve the
best performance while reaching the gain requirement for all line and load conditions. And for safe operation,
we must determine the minimum switching frequency the controller shall limit in order to maintain the
operation in the inductive region.
The following are detailed explanation of all design steps; additionally Figure 3.1 shows a design flow chart
that summarizes the design methodology.
START
Step 1:
Select Qmax value
Step 2:
Select m value
Step 3:
Find Fxmin
Increase m
value
Decrease m
value
Step 4:
Find Kmax
Yes
No
Is Kmax <
required
gain?
No
Is Kmax =
required gain?
Yes
Step 5:
Solve for resonant
components values
END
Figure 3.1 Design flow chart
Step 1: Selecting the Qmax Value
Quality factor Q 
Lr C r
depends on the load current. Heavy load conditions operate at high Q values,
Rac
while lighter loads have lower Q values. It is important to set a value for the Qmax associated with the
maximum load point.
To illustrate the effect of the Q value on voltage regulation, Figure 3.2 shows an example voltage gain plot
for different Q values. Let’s assume that the resonant tank gain is required to range from 0.8 to 1.2 for
example, we can see that the low Q value curve (Q=0.3) can reach higher boost gain, but it is less sensitive
to frequency modulation in the “above resonance fs>fr” region, hence, switching frequency has to increase
much in order to reach the minimum voltage gain (K=0.8), causing extra switching losses, while the higher Q
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Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
value curve (Q=1) can reach the minimum gain (K=0.8) with less switching frequency increase, but unable
to reach the maximum gain (K=1.2). Therefore, a moderate Q value of around 0.5 seems to satisfy the
voltage gain requirement in this specific case.
We conclude that adjusting the Q value can help achieving the maximum gain but increases the frequency
modulation range, thus, we should not rely on tuning the Qmax value as a design iteration in order to reach
the desired maximum voltage gain, but instead we should rely on tuning the m value as will be explained in
the next step.
Although there isn’t a direct method for selecting the optimum Q value, we should select Qmax moderately
as discussed earlier and based on the specific design in hand.
2
K ( .3 m Fx)
K ( .5 m Fx)
K ( 1 m Fx)
1.5
1
1.2
0.8
0.5
0
0.1
1
Fx
10
Figure 3.2
Step 2: Selecting the m Value
As mentioned above, m 
Lr  Lm
, m is a static parameter that we have to start the design by optimizing its
Lr
value, therefore it’s important to understand the impact of the m ratio on the converter operation. To illustrate
the effect of the m value, Figure 3.3 shows the same resonant tank gain plots but for different m values, m=
3, 6 and 12. It is obvious that lower values of m can achieve higher boost gain, in addition to the narrower
range of the frequency modulation, meaning more flexible control and regulation, which is valuable in
applications with wide input voltage range. Nevertheless, low values of m for the same quality factor Q and
resonant frequency fr means smaller magnetizing inductance Lm, hence, higher magnetizing peak-peak
current ripple, causing increased circulating energy and conduction losses.
We have to start by selecting a reasonable initial value for m (6-10), and then optimize it by few iterations to
get the maximum m value that can still achieve the maximum gain requirement for all load conditions.
Low m value:
 Higher boost gain
 Narrower frequency range
 More flexible regulation
x
High m value:
 Higher magnetizing inductance
 Lower magnetizing circulating current
 Higher efficiency
0 .001 10
3
3
3
K ( .2 , m , Fx)
K ( .2 , m , Fx)
m=3
K ( .2 , m , Fx)
m=6
K ( .3 , m , Fx)
K ( .3 , m , Fx)
K ( .3 , m , Fx)
K ( .5 , m , Fx)
2
K ( .5 , m , Fx)
K ( .7 , m , Fx)
K ( 1 , m , Fx)
K ( 5 , m , Fx)
2
K ( .5 , m , Fx)
K ( .7 , m , Fx)
K ( 1 , m , Fx)
1
0
0.1
K ( 5 , m , Fx)
1
Fx
10
m=12
2
K ( .7 , m , Fx)
K ( 1 , m , Fx)
1
K ( 5 , m , Fx)
0
0.1
1
Fx
Figure 3.3
9
10
1
0
0.1
1
Fx
10
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Step 3: Finding the Minimum Normalized Switching Frequency
After selecting a value for Qmax and an initial value for m, we need to find the minimum normalized
switching frequency that will guarantee inductive operation for the Qmax (max load) condition, this minimum
frequency will also guarantee inductive operation for all other loads.
The minimum normalized switching frequency occurs at the peak gain of the Qmax curve, so it can be found
by solving Eq. 2 (assumed Qmax=0.4 and m=6 as an example), or can be visually spotted in the gain plot as
in Figure 3.4.
d
K (Q, m, Fxmin )
0
dFx
Q max0.4 , m6
Eq. 2
 Solve for Fx
min
3
Light load
K ( .2 , m , Fx)
m=6
K ( .25 , m , Fx)
K ( .3 , m , Fx)
Max load curve
(Qmax=0.4)
2
K ( .4 , m , Fx)
K ( .6 , m , Fx)
K ( 5 , m , Fx)
1
0
0.1
Fxmin
1
Fx
10
Figure 3.4
Step 4: Voltage Gain Verification
This step is to verify that the maximum gain Kmax reached during maximum load by the selected m value is
adequate. This can be done by solving Eq. 3, or can be visually spotted in the gain plot as in Figure 3.4.
K max  K (Qmax , m, Fxmin )
Eq. 3
3
Light load
K ( .2 m Fx)
m=6
K ( .25 m Fx )
K ( .3 m Fx)
2
K ( .4 m Fx)
Kmax
K ( .6 m Fx)
K ( 5 m Fx)
Max load curve
(Qmax=0.4)
1
0
0.1
Fxmin
1
Fx
Figure 3.5
10
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Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Few iterations are needed in order to reach the optimized design, as shown in the design flow chart in Figure
3.1. If Kmax is not enough, then we have to reduce the m value and repeat steps 3 and 4, in order to gain a
higher boost gain. On the other side, If Kmax is higher than what is required; in that case we can increase the
m value and repeat steps 3 and 4 in order to gain a better efficiency.
Step 5: Calculating Resonant Components Values
After few iterations of the design flow and reaching the optimum m value, we can proceed to calculating the
resonant tank components values, Eq. 4 and Eq. 5 can be solved to find Lr and Cr, and then Lm can be
calculated using Eq. 6.
2
Rac ,min
Qmax 
2
8 N
V
 2  P2  o
 N S Po max
Lr C r
Rac ,min
Eq. 4
fr 
1
2 LrCr
Eq. 5
m
Lr  Lm
Lr
Eq. 6
It must be noted that selection of the resonant frequency fr was not considered in the design steps above,
since it has no impact on the maximum gain and operation region of the resonant converter, however it is
selected considering the converter power density and power losses.
4
Bridge and Rectifier Selection
An important step to achieve the best converter performance is to choose the right bridge and rectifier
circuits.
LLC converters can be implemented with a full-bridge or a half-bridge circuit on the primary side, as shown in
Figure 4.1,
S1
S3
C1
S1
S2
S4
C2
S2
Full-bridge switching circuit
Half-bridge switching circuit
Figure 4.1
Table 2 shows a comparison between the half-bridge and the full-bridge switching circuits.
A half-bridge would have twice the current of what a full-bridge would have, the squared rms current in the
half-bridge case is four times, the number of switches in a half bridge is half of that in a full-bridge, therefore,
the total FETs conduction losses of a half-bridge is twice compared to the full-bridge.
Although a half-bridge requires half the primary number of turns for the same voltage gain and magnetic flux
swing, thus half the primary winding resistance, the primary copper losses are still twice compared to the fullbridge because the squared rms current in the half-bridge case is four times.
So in applications with high primary currents, where conduction losses are dominant, it is suggested to use a
full-bridge switching circuit.
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Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Table 2
Primary Bridge - Half-Bridge compared to Full-Bridge
Irms
Irms2
Number of FETs
Total FETs conduction
Np
Rpri
Transformer primary
losses
×2
×4
÷2
copper loss
×2
÷2
÷2
×2
*Comparison assumes same FET and same transformer core
LLC converters can also be implemented with a full-bridge or a full-wave rectifier circuit on the secondary
side, as shown in Figure 4.2
D2
Np
Ns
D1
D3
Co
D2
Np
+
Vo
-
Ns
+
Vo
-
Ns
D4
D1
Full-bridge rectifier
Full-wave rectifier
Figure 4.2
Table 3 shows a comparison between the full-bridge and full-wave rectifiers.
A full-wave rectifier requires diodes that are twice the voltage rating compared to a full-bridge rectifier, but it
has only two diodes while the full-bridge rectifier has four diodes, since each diode in both rectifier circuits
carries the same average current, the full-wave rectifier has half the total diode conduction losses compared
to the full-bridge rectifier.
A full-wave rectifier has two secondary windings, hence the resistance is doubled for the same winding area,
each winding in a full-wave caries an rms current that is √0.5 of the rms current of the full-bridge circuit,
therefore the total secondary windings copper losses of the full-wave rectifier is twice compared to the fullbridge rectifier.
In applications with high output voltages, the full bridge rectifier is advantageous since we can use diodes
with half the voltage rating compared to the full-wave rectifier. While in low output voltages and high currents
application, the full-wave is more common, because of lower total conduction losses and lower component
count and cost.
Table 3
Secondary Rectifier - Full-Wave compared to Full-Bridge
Diode
Number of
voltage
diodes
Diode
Number of
Rsec
Irms
Transformer secondary
conduction
secondary
per winding
per winding
copper loss
losses
windings
×2
× √0.5
×2
rating
×2
÷2
÷2
×2
*Comparison assumes same diode voltage drop and same transformer core
12
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
5
Design Example
5.1
Application and Specifications
V1.0 September 2012
Our design example is applicable to the dc-dc stage of a solar micro inverter, as shown in Figure 5.1, with
specifications as listed in Table 4. According to the discussion in section 4, the LLC converter will be
implemented with a full-bridge on the primary side and a full-bridge rectifier on the secondary side, same
circuit as shown in Figure 2.1.
~30V
400V
DC-DC DC Link
Converter
DC-AC
Inverter
240Vac
60Hz
Figure 5.1
Table 4 Specifications
Output voltage
400V
Input voltage
18V – 36V (33V nominal)
Output power
250W
Output power derates linearly with input voltage
Ex: Output power= 125W @ Vin=18V
Resonant frequency
5.2
100kHz
Design Steps
We start the design by calculating the transformer turn ratio and the minimum and maximum voltage gains of
the resonant tank, as follows.
‫ ܯ‬௡௢௠ = 1
ܰ௣ ܸ௜௡_௡௢௠
ܰ௣
=
∙ ‫ ܯ‬௡௢௠ →
= 0.0825
ܰ௦
ܸ௢௨௧
ܰ௦
ܸ௜௡_௡௢௠
‫ ܯ‬௠ ௔௫ =
∙ ‫ ܯ‬௡௢௠ → ‫ ܯ‬௠ ௔௫ = 1.833
ܸ௜௡_௠ ௜௡
ܸ௜௡_௡௢௠
‫ ܯ‬௠ ௜௡ =
∙ ‫ ܯ‬௡௢௠ → ‫ ܯ‬௠ ௜௡ = 0.917
ܸ௜௡_௠ ௔௫
Next, we follow the design steps discussed in section 3, as follows.
Step 1: Selecting the Qmax Value
Let’s choose ܳ௠ ௔௫ = 0.4
Step 2: Selecting the m Value
Let’s choose ݉ = 6.3
Step 3: Finding the Minimum Normalized Switching Frequency
We can solve the gain equation to find the minimum frequency, which occurs at the peak of the Qmax curve,
as shown below, or we can look it up from the gain plot as shown in Figure 5.2.
13
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
݀ ‫ܳ(ܭ‬௠ ௔௫ , ݉ , ‫ܨ‬௑௠ ௜௡ )ฬ
݀‫ܨ‬௑
ொ
,௠ ୀ଺.ଷ
೘ ೌೣ ୀ଴.ସ
=0
V1.0 September 2012
→ ‫ܨ‬௑௠ ௜௡ = 0.489
→ ݂௦_௠ ௜௡ = ‫ܨ‬௑௠ ௜௡ ∙ ݂௥ = 48.9݇‫ݖܪ‬
Step 4: Voltage Gain Verification
Since the power is derated at lower input voltages as listed in the specifications, we have to calculate the
maximum Q value at the minimum input voltage case (ܳ௠ [email protected] ௜௡ ), as follows. Note that this power derating
specification is related to the solar panel I-V characterestics. (In case of other applications where power
rating is the same across the input voltage range, we only have a single Qmax value).
ܸ௜௡_௠ ௜௡
ܳ௠ [email protected] ௜௡ = ܳ௠ ௔௫ ∙
= 0.2
ܸ௜௡_௠ ௔௫
Then we can calculate the maximum gain reached at the minimum switching frequency for the ܳ௠ [email protected] ௜௡
condition, or we can look it up from the gain plot as shown in Figure 5.2.
‫ܭ‬௠ ௔௫ = ‫ܳ(ܭ‬௠ [email protected] ௜௡ , ݉ , ‫ܨ‬௑௠ ௜௡ ) = 1.974
‫ܭ‬௠ ௔௫ = 1.974 > ‫ ܯ‬௠ ௔௫ = 1.833
→ No need for tuning the m value
3
K ( .2 m Fx) 2
[email protected]=0.2
1.974
Qmax=0.4
K ( .3 m Fx)
K ( .4 m Fx)
1
0
0.1
0.489
1
Fx
10
Figure 5.2
Step 5: Calculating Resonant Components Values
The reflected load resistance at full load is,
ܴ௔௖_௠ ௜௡ =
8 ܰ௣ ଶ ܸ௢௨௧ଶ
∙൬ ൰
ߨଶ ܰ௦ ܲ௢,௠ ௔௫
→ ܴ௔௖_௠ ௜௡ = 3.534 Ω
Next we solve the equations below to obtain the resonant tank components values
ඥ‫ܮ‬௥/‫ܥ‬௥
3.534 Ω
1
݂௥ = 100݇‫= ݖܪ‬
2ߨඥ‫ܮ‬௥ ∙ ‫ܥ‬௥
‫ܮ‬௥ + ‫ܮ‬௠
݉ = 6.3 =
‫ܮ‬௥
∴ ‫ܮ‬௥ = 2.25ߤ‫ܮ ܪ‬௠ = 11.93ߤ‫ܥ ܪ‬௥ = 1.13ߤ‫ܨ‬
ܳ௠ ௔௫ = 0.4 =
14
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
5.3
V1.0 September 2012
Experimental Waveforms and Efficiency
The design example was implemented with the specification shown in Table 5
Table 5 Prototype specifications
Resonant frequency ݂௥
110 kHz
Resonant capacitor ‫ܥ‬௥
0.94 µF
Minimum switching frequency ݂௦_௠ ௜௡
50 kHz
Transformer Specifications
Turns ratio ܰ௣ : ܰ௦
1:12
Magnetizing inductor ‫ܮ‬௠
12.2 µH
Leakage (Resonant) inductor ‫ܮ‬௥
2.2 µH
Figure 5.3 through Figure 5.6 shows experimental waveforms at different input voltage conditions,
Red channel:
Primary FET Vgs
Yellow channel: Primary FET Vds
Green channel:
Resonant current ILr
Blue channel:
Rectifier output current ID1+ID3
Vin= 33V Po=250W
Vin= 18V Po=125W
Figure 5.3
Figure 5.4
Vin= 36V
Po=250W
Vin= 36V Light Load Missing Cycle Mode
Figure 5.5
Figure 5.6
15
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Table 6 Efficency data
Output Power (% of 250W)
Input Voltage
20%
40%
60%
80%
100%
36V
97.1% **
97.1%**
97.1%
97.1%
97.1%
33V
96.0%
97.2%
97.6%
97.6%
97.4%
24V
94.5%
96.8%
97.1%
97.0%
18V
94.0%
96.3%
96.2%
** Missing cycle mode / Burst mode operation
98.0%
97.6%
97.4%
96.9%
97.0%
Efficiency %
97.6%
97.4%
97.5%
96.5%
96.0%
96.0%
Vin= 33V
95.5%
Vin= 24V
95.0%
Vin= 36V
94.5%
Vin= 18V
94.0%
93.5%
0
50
100
150
200
Output Power (W)
Figure 5.7 Efficency curves
16
250
300
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
6
V1.0 September 2012
Schematics and Bill of Material
C1
INPUT
W1
Jumper
INPUT
INPUT+
C2
1
2
C3
10uF/100V 10uF/100V
D1
1N5819HW-7-F
R1
Q1
C7
D2
1N5819HW-7-F
R2
SA
CON1
400V
2.2nF/500VAC
R3
10k
Q3
D3
IDH05G65C5
Q2
SC
0
C8
BSC028N06NS
0
BSC028N06NS
C5
D4
2.2u/450V
IDH05G65C5
C6
2.2u/450V
R4
10k
Q4
+ C4
NC
10uF/100V 10uF/100V
PGND
INPUT-
PBSS4032PT,215
C12
1
2
C13
SW2
+ C25
NC
C10
2.2u/450V
PBSS4032PT,215
SW1
+ C9
NC
W2
Jumper
+ C27
NC
C11
2.2u/450V
T1
10uF/100V 10uF/100V
OUT+
1
2
+ C26
NC
CON5
+ C28
NC
+ C29
NC
+ C30
NC
CON2
C14
0.47uF/250V
C15
0.47uF/250V
D5
IDH05G65C5
+ C31
NC
D6
IDH05G65C5
OUT1
2
T2
Trans CT
12V
12V
D8
1N5819HW-7-F
R6
R7
NC
Trans
Q6
SD
1
BSC028N06NS
0
BSC028N06NS
R9
10k
Q7
CT2
Q5
0
CT1
D7
1N5819HW-7-F
R5
SB
CON6
D9
2
R10
10k
Q8
V-
V+
AC
AC
4
R8
1k
C16
1u
Bridge
R12
1k
D10
12V
12V
PBSS4032PT,215
1
2
400V
Q9
MMBT3904
3
R14
R13
402k
12V_SEC
C17 NC
1u
PBSS4032PT,215
R15
402k
CON3
GND
1
2
R18
D11
CON4
CSCT1
CS-
1
V-
2
AC
4 CS+
V+
3
AC
FB
CS+
4
C20
10n
CT2
10
IC1
12V_SEC
SFH6186-3T
C22
10uF/25V
4
+Vin
NC
NC
NC
-Vout
8
6
R25
12V_SEC
12V_SEC
12V_SEC
4
1
3
U2
AP331AWG-7
1
3
NC
5
FB
R27
10k
4
R29
10k
IC4
1
3
2
2
R32
0
C24
NC
12V
12V
12V
C32
C33
IC5
100n
1
2
C34
SA
100n
SW1
SW1
SA 3
4
VDD
8 SB
LO
HB
LI
HS
HI
1
7
VSS
HO
IC6
100n
SB
2
6
5
C35
R33
SC
100n
R35 10
SC 3
SW2
SW2
4
VDD
LO
HB
VSS
HO
LI
HS
HI
8 SD
R34
R36 10
5
10
FB
10
FB
1
2
EN
3
4
Load
R37
5
9.1k/1%
6
R39
7
5.6k/1%
VREF
R43
220k
30k
R50
1M
R49
C40
2.2u
C42
1u
C44
C45
1n
R41
62k
8
R44
30k
R4510
NC
9
C41
10u
R56
30k
VCC
HG
SS
LG
LOAD
SHG
FREQ
SLG
Delay
GND
TD
SRD
Vmc
CL
IC7
CS
ICE2HS01G
Vres
VINS
Vref
20
19
R57 R58
12k NC
12V
R59
0
Figure 6.1 Schematics
17
C37
12V
C36
100n
100n
18
17
16
15
R40
NC
14
C38
R38
0
13
12
220k
R54
NC
R42
47nF
C39
470p
11
R46
4.7/1W
R53
18k
10n
CS+
0
R48
0
C43
INPUT
R52
1k
R55
100k
Timer
EnA
R51
NC
SD
7
6
CS-
CS+
CS-
R28
IC3
1k
TL431
R31
10k
2
Q10
2N7002
SFH6186-3T
C46
NC
R26
5.6k
12V_SEC
R63
Load
R61
402k
R24
NC
0
C23
10uF/25V
Isolated DC/DC
NC
R30
R60
402k
330n
IC2
TL431
1
5
+Vout
R47
402k
R23
10k
7
2
3
D12
13V
NC
3
-Vin
2
C21
100n
3.3k
U1
1
0
NC C19
R21
R11
402k
R19
402k
NC
R22
2
Bridge
12V
NC
C18
400V
C47
R64
3.9k
R20
1k
1
3
R17
402k
R16
R62
10k
Application Note AN 2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Table 7 Bill of Material
Qty
Designator
Value
Part Number
1
6
9
4
2
3
C1
C2, C3, C7, C8, C12, C13
C4, C9, C25, C26, C27, C28, C29, C30, C31
C5, C6, C10, C11
C14, C15
C16, C17, C44
C18, C24, C42, C46, C47, R7, R14, R24, R30,
R40, R45, R54, R58, R63, R64
C19
C20, C43
C21, C32, C33, C34, C35, C36, C37
C22, C23
C38
C39
C40
C41
C45
D1, D2, D7, D8
D3, D4, D5, D6
D9, D11
D10
D12
IC1, IC4
IC2, IC3
IC5, IC6
IC7
Q1, Q2, Q5, Q6
Q3, Q4, Q7, Q8
Q9
Q10
R1, R2, R5, R6, R22, R25, R32, R38, R42, R48,
R59
R3, R4, R9, R10, R23, R27, R29, R31, R62
R8, R12, R20, R28, R52
R11, R13, R15, R17, R19, R47, R60, R61
R16
R18, R33, R34, R35, R36
R21
R26
R37
R39
R41
R43, R51
R44, R50, R56
R46
R49
R53
R55
R57
T1
T2
U1
U2
2.2nF/500VAC
10uF/100V
NC
2.2u/450V
0.47uF/250V
1u
VY1222M47Y5UQ63V0
C5750X7S2A106M
B32674D4225K
B32652A3474J
C3216X7R1H105K
NC
330n
10n
100n
10uF/25V
47nF
470p
2.2u
10u
1n
BAS 3010A
IDH05G65C5
Bridge
12V
13V
SFH6186-3T
TL431
SO8
ICE2HS01G
BSC028N06NS
PBSS4032PT,215
MMBT3904
2N7002
GRM319R71H334KA01D
GRM319R71H103KA01D
GRM319R71H104KA01D
TMK316B7106KL-TD
GRM31M5C1H473JA01L
CC1206KRX7R9BB471
C3216Y5V1H225Z/0.85
TMK316B7106KL-TD
GRM3195C1H102JA01D
BAS 3010A-03W E6327
IDH05G65C5
BAS3007A-RPP
SMAZ12-TP
3SMAJ5928B-TP
SFH6186-3T
TL431CPK
LM5100AM/NOPB
ICE2HS01G
BSC028N06NS
PBSS4032PT,215
MMBT3904FSCT-ND
2N7002 L6327
0
10k
1k
402k
3.9k
10
3.3k
5.6k
9.1k/1%
5.6k/1%
62k
220k
30k
4.7/1W
1M
18k
100k
12k
Transformer
Trans CT
Isolated DC/DC
AP331AWG-7
ERJ-8GEY0R00V
ERJ-8ENF1002V
ERJ-8GEYJ102V
ERJ-8ENF4023V
ERJ-8GEYJ392V
ERJ-8GEYJ100V
ERJ-8GEYJ332V
ERJ-8ENF5601V
ERJ-8ENF9101V
ERJ-8ENF5601V
ERJ-8ENF6202V
ERJ-8ENF2203V
ERJ-8ENF3002V
ERJ-1TRQF4R7U
ERJ-8ENF1004V
ERJ-8ENF1802V
ERJ-8ENF1003V
ERJ-8ENF1202V
Custom**
B82801B504A50
VBT1-S12-S12-SMT
AP331AWG-7
15
1
2
7
2
1
1
1
1
1
4
4
2
1
1
2
2
2
1
4
4
1
1
11
9
5
8
1
5
1
1
1
1
1
2
3
1
1
1
1
1
1
1
1
1
**Transformer built by Midcom-Wurth Electronics, E41/17/12-3C90
18
Application Note AN 2012
2012-09
Resonant LLC Converter: Operation and Design
V1.0 September 2012
Figure 6.2
7
References
[1] Infineon Technologies: ICE2HS01G datasheet, High Performance Resonant Mode Controller, V1.1,
August 2011.
[2] Infineon Technologies: Design Guide for LLC Converter with ICE2HS01G,
ICE2HS01G, V1.0, July 2011.
[3] Infineon Technologies: 300W
W LLC Evaluation Board with LLC controller ICE2HS01G, V1.1, August
2011.
19
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