Application Note AN-1160 Design of Resonant Half-Bridge converter using IRS2795(1,2) Control IC By Helen Ding Table of Contents 1. Introduction & Device Overview 2. LLC Resonant Half Bridge Converter Operation 3. Transformer and Resonant Circuit Design 4. IRS2795 Passive Components Design 5. IRS2795 Power Loss Calculation 6. MOSFET Selection Guide 7. Operating Waveforms and Efficiency 8. Layout Guidelines and Example 9. Appendix Symbol list References www.irf.com AN-1160 1 1. Introduction and Device Overview The IRS2795(1,2) is a self oscillating half-bridge driver IC for resonant half-bridge DC-DC converter applications for use up to 600V. It has a fixed 50% duty-cycle and very wide operating frequency range. The maximum switching frequency can go up to 500kHz. The frequency can be programmed externally through the RT and CT pins. The IC offers flexibility to program the minimum operating frequency, the maximum operating frequency and the frequency sweep at power up for the softstart function. The dead time is programmed by the CT capacitor. The programmable dead-time allows the user to optimize the system with the minimum body-diode conduction time for higher efficiency under full load, while keeping ZVS switching under no load condition. The IC offers over current protection using the on-state resistance of the low-side MOSFET. The protection threshold is 2V for IRS27951 and it is 3V for IRS27952 IC. The IC can be disabled by externally pulling the voltage at the CT/SD pin below its enable voltage threshold. The IC enters “sleep” mode and only consumes micro-power when disabled. IRS2795(1,2) packing in a 8-pin package, it’s easy to use, and drastically reduces external component count for a high efficiency low cost power supply. Figure 1 is the typical application schematic of IRS2795(1,2): DC BUS VIN R1 R2 Rcc D3 Dbs Cbs U1 1 CDC 2 3 4 Dss Rss RT VCC VB RT HO CT/SD VS COM LO 8 M1 Rg1 D1 7 6 Lr 5 IRS2795 M2 Rg2 Rmax Lm CT Css LOAD COUT D2 Cr RTN Rled Rbias Rs1 Cf1 U2 Cf2 Rf2 U3 TL431 Rs2 Figure 1: IRS2795(1,2) typical application circuit www.irf.com AN-1160 2 The pinout of IRS2795(1,2) is shown below. 1 VCC 2 RT 3 CT/SD 4 COM IRS2795 Lead Assignment Pin# Symbol Description 1 VCC Supply voltage 2 RT Oscillator timing resistor VB 8 3 CT/SD HO 7 4 COM Ground VS 6 5 LO Low-side gate drive 6 VS High-side gate drive return/ HV Current Sense LO 5 7 HO High-side gate drive 8 VB High-side floating supply voltage Oscillator timing capacitor/Shutdown Figure 2: IRS2795(1,2) IC pin assignment 2. LLC Resonant Half-Bridge Converter Operation The increasing popularity of the LLC resonant converter in its half-bridge implementation is due to its high efficiency, low switching noise and ability to achieve high power density. This topology is also the most attractive topology for front-end DC bus conversion. It utilizes the magnetizing inductance of the transformer to construct a complex resonant tank with Buck Boost transfer characteristics in the soft-switching region. The typical power stage schematic for this topology is shown below. M1 D1 HO Vin VS Lr n:1:1 Lm M2 LO COUT LOAD D2 Cr Figure 3: Typical schematic of a DC-DC half-bridge resonant converter Devices M1 and M2 operate at 50% duty cycle and the output voltage is regulated by varying the switching frequency of the converter. The converter has two resonant frequencies – a lower resonant frequency (given by Lm, Lr, Cr and the load) and a fixed higher series resonant frequency Fr1 (given by Lr and Cr only). The two bridge devices M1 and M2 can be soft-switched for the entire www.irf.com AN-1160 3 load range by operating the converter under inductive load mode (ZVS region). It can be either above or below the resonant frequency Fr1. The typical AC transfer characteristics1 for a LLC tank resonant converter are shown in Figure 4. The group of curve indicates the gain under different load conditions. Figure 4: Typical frequency response of a LLC resonant converter The characteristics of a LLC resonant converter can be divided into three regions based on the 3 different modes of operation. The first region is for switching frequency above the resonant frequency Fr1. Fr1 = 1 2π Lr ⋅ C r 1 (the purple shaded area) the switching frequency is higher than resonant frequency Fr1. In region ○ The converter operation is very similar to a series resonant converter. Here Lm never resonates with resonant capacitor Cr; it is clamped by the output voltage and acts as the load of the series resonant tank. This is the inductive load region and the converter is always under ZVS operation regardless of the load condition. In the 2nd region, the switching frequency is higher than the lower resonant frequency but lower than 2 is in the pink shaded area in Figure 4. The lower resonant frequency varies with load, Fr1. Region ○ 2 and region ○ 3 traces the peak of the family load vs. gain curves. In this so the boundary of region ○ complex region, the LLC resonant operation can be divided into two time intervals. In the first time interval, Lr resonates with Cr and Lm is clamped by output voltage. When the current in the resonant 1 For this AC analysis, only the fundamental component of the square-wave voltage input to the resonant network contributes to the power transfer to output. The transformer, rectifier and filter are replaced by an equivalent AC resistance, Rac. www.irf.com AN-1160 4 inductor Lr resonates back to the same level as the magnetizing current, Lr and Cr stop resonating. Lm now participates in the resonant operation and the second time interval begins. During this time interval, dominate resonant components change to Cr and Lm in series with Lr. The ZVS operation in 2 is guarantees by operating the converter to the right side of the load gain curve. For a region ○ 2 or region ○ 3 depends on the switching frequency below resonant Fr1, it could falls in either region ○ load condition. 3 below fr1, the LLC resonant converter operates in capacitive mode; M1 and M2 In the ZCS range ○ are under hard switching and have high switching losses. So ZCS operation should always be avoided. The typical operating waveforms of the 3 modes are demonstrated in Figure 5 to Figure 7. Figure 5: Typical waveform of above resonant ZVS switching Figure 6: Typical waveform of below resonant ZVS switching www.irf.com AN-1160 5 The waveforms indicate that the current in secondary rectifier diodes moves from continuous current mode (CCM) to discontinuous current mode (DCM) when the switching frequency varies from above resonant ZVS to below resonant ZVS due to load increasing. The ripple voltage on the resonant capacitor Cr also increases in the below resonant ZVS mode. Figure 7: Typical waveform of below resonant ZCS switching In ZCS mode, the two switching devices M1 and M2 are turned off under zero current condition. The turn-on of the two switches is hard switching (none ZVS). The turn-on switching loss is high especially under high voltage bus voltage. The resonant capacitor Cr also has high voltage stress. ZCS operation should always be avoided. The typical voltage conversion ratio of a LLC resonant converter is shown in Figure 8. Q - Low (no load) 2nVout Vin Q - High (full load) Frequency modulation range @ low line 2nVout Vinmin Frequency modulation range @ high line 2nVout Vinmax Over load or Short Circuit Fr1 fmin fmax Frequency Figure 8: Typical voltage conversion ratio of a LLC resonant converter www.irf.com AN-1160 6 With a fixed input voltage, the converter varies switching frequency to regulate the output voltage over load range – keeping the same conversion ratio over the family of curves with different Q. Given a fixed load condition, the converter varies switching frequency along that load line to regulate output voltage over input voltage range – the conversion ratio increases when input voltage decreases. To design the LLC resonant half-bridge converter, we use the First Harmonic Approximation (FHA) to get equivalent circuit. All the components are put to primary side to simply the analysis. The load equals to a resistor Rac that is in parallel with transformer primary inductance Lm. M1 D1 n:1:1 Cr Vin Lr M2 Lm COUT LOAD D2 Cr 2⋅ Vin Lr n ⋅ Vout⋅ π 4 π Vf und Rac Rac 2 n ⋅ RLOAD⋅ 8 2 π Lm Figure 9: The FHA equivalent circuit The input voltage of the resonant tank is a square wave with amplitude equals to the input DC voltage Vin. The fundamental component of the square waveform is: 2 ⋅ Vin π sin(ϖ ⋅ t ) The output voltage of the resonant tank is the voltage across Lm. It is very close to a square waveform with amplitude swinging from − n ⋅ Vout to + n ⋅ Vout . So the fundamental component of the output square waveform is: 4 ⋅ n ⋅ Vout sin(ϖ ⋅ t ) π www.irf.com AN-1160 7 The power dissipation on the equivalent AC resistor is equal to the power dissipation of RLOAD resistor, thus it can be written as: 4 ⋅ n ⋅ Vout 2 Vout 2π = R LOAD Rac 2 Rearrange the formula and get the equivalent AC resistor: Rac = 8 ⋅ n2 R LOAD π2 The transfer ratio of the equivalent circuit can be obtained as following: j ⋅ ω ⋅ Lm⋅ Rac j ⋅ ω ⋅ Lm+ Rac M 1 j ⋅ ω⋅ Lr + + j ⋅ ω⋅ Cr j ⋅ ω⋅ Lm⋅ Rac j ⋅ ω⋅ Lm + Rac Re-write the formula, 1 M 1+ Lr 1 − Lm + 2 ω ⋅ Lm⋅ Cr jω⋅ Lr Rac − j ω⋅ Cr⋅ Rac With the following definitions, M can be simplified. Fr1 = k= 1 , 2π Lr ⋅ C r Lm , Lr Rac = x= Fsw , Fr1 8 ⋅ n 2 ⋅ RLOAD π x , Lr ⋅ C r ϖ = 2πFsw = 2π ⋅ x ⋅ Fr1 = 2 , Q= 2πFr1 ⋅ Lr 1 = Rac 2πFr1 ⋅ C r ⋅ Rac 1 M 1+ 1 ⋅ 1 − k + j⋅ Q⋅ x − 2 x 1 1 x Or, 1 M 2 1 + 1 ⋅ 1 − 1 + k 2 x www.irf.com AN-1160 Q⋅ x − 1 2 x 8 Per Figure 9, M is also equals to the output voltage to input voltage ratio: M= n ⋅ Vout ⋅ 2⋅ Vin 4 π = Vout ⋅ 2 ⋅ n Vin π So we have the conversion ratio of output voltage Vout to input voltage Vin: Vout M = Vin 2 ⋅ n 3. Transformer and Resonant Circuit Design This section provides the details of how to calculate the key components of a LLC converter, take a 24V output 240W power supply as an example. The system input data Parameter Unit Vinmax V The maximum DC bus voltage 430 Vinmin V The minimum DC bus voltage 350 Vinnom V The nominal DC bus voltage 390 Vout V The DC output voltage 24 Iout A The output load current 10 Fr1 KHz The resonant frequency 100 Fmax KHz The maximum switching frequency ① 150 The maximum duty-cycle 0.5 Dmax Description Value Tss ms The soft start time 10 Fss KHz The soft start frequency 300 Transformer ETD49 ①Note: Typically set Fmax < 2xFr1 as the parasitic capacitance in the system introduced a 3rd resonant frequency which could cause the output voltage to increase with switching frequency at no load if the maximum switching frequency is higher than the limit. Step 1: Calculate the transformer turns ratio Vin max , 2 ⋅ Vout 430 n= = 8.96 2 ⋅ 24 n= The transformer turns ratio is calculated with the maximum input voltage to make sure the output is always under regulation, including the worst case - high-line voltage and no load condition. www.irf.com AN-1160 9 Usually the transfer ratio of the power stage is higher than the theoretical calculated value. This is because of the parasitic capacitance in the system (the coupling capacitor between transformer windings and the junction capacitors of output diodes) affects the resonance, especially at zero load where the switching frequency is much higher than the resonant frequency. So it’s recommended to choose the n to be slightly higher than the calculated value especially if the controller has no burst mode to keep regulation at high line and zero load condition. n=9 Step 2: Choose k value k is the ratio between the transformer magnetic inductance and the resonant inductance. Smaller k value gives steeper Gain curve, especially at the below resonant ZVS region as shown in Figure 10. The output voltage is more sensitive to frequency variation with smaller k factor. 3 2.5 k=5 2 1.5 k=10 1 0.5 0 Figure 10: k factor A higher k value results in higher magnetic inductance and thus lower magnetizing current in the transformer primary winding – that means lower circulating power losses. However, higher magnetic inductance could also cause non-ZVS switching at high line and zero load condition where the circulating current is too small to fully charge / discharge the VS node during dead-time. The recommend range of k is from 3 to 10. Here k = 5 is chosen. Step 3: Calculate Qmax to stay in ZVS operation at the maximum load under the minimum input voltage The input impedance of the equivalent resonant circuit (Figure 9) is given by: www.irf.com AN-1160 10 Zin j ⋅ ω⋅ Lr + 1 j ⋅ ω⋅ Cr 2 2 Zin k ⋅x ⋅Q Q⋅ Rac 2 2 2 + j ⋅ ω⋅ Lm⋅ Rac j ⋅ ω⋅ Lm + Rac + j x − 1 + k ⋅x ⋅Q 1 x x⋅ k + 2 2 2 1 + k ⋅x ⋅Q To keep the converter working in soft switching mode, the operating point should always in the ZVS region as shown in Figure 4. The ZVS ZCS boundary line is defined by the phase angle of Zin Ф(Zin)= 0 (the boundary condition between capacitive and inductive load), i.e. the imaginary part of Zin is zero. With this condition we can calculate the maximum Q which allows the converter to stay in ZVS. The maximum Q happens at the minimum input voltage and the maximum load. 1 + k⋅ 1 − Qmax 1 k 1 ⋅ 2 Mmax 1 + k⋅ 1 − 2 Mmax − 1 1 k ⋅ 1 2 2⋅ n ⋅ Vout Vinmin 2 2⋅ n ⋅ Vout − 1 Vinmin Where Mmax is the maximum conversion ratio at the minimum input voltage, Q max = 0.456 Step 4: Calculate the minimum switching frequency The minimum switching frequency happens at the maximum load and minimum input voltage with the previous calculated maximum Qmax. As Qmax is defined by Im(Zin)=0, x⋅ k x − 1 + 0 x 2 2 2 1 + k ⋅ x ⋅ Qmax The Fmin can be calculated with: xmin 1 1 + k⋅ 1 − 1 2 Mmax 1 1 + k⋅ 1 − 1 2 2n ⋅ Vout Vinmin x min = 0.607 F min = x min⋅ Fr1 = 60.7 KHz Step 5: Calculate Lr, Cr and Lm As Qmax happens at the maximum load, so the resonant components Lr, Cr and Lm can be calculated per the Qmax value that had obtained in step 3: www.irf.com AN-1160 11 RLOAD = Rac = Lr = Cr = Vout 24V = = 2.4Ω Iout 10 A 8 ⋅ n 2 ⋅ RLOAD π2 = 8 × 9 2 × 2 .4 π2 = 157.57Ω Q max⋅ Rac 0.456 × 157.57 = = 114uH 2 ⋅ π ⋅ Fr1 2 ⋅ π ⋅ 100 K 1 1 = = 22.2nF 2 ⋅ π ⋅ Fr1 ⋅ Q max⋅ Rac 2 ⋅ π ⋅ 100 K × 0.456 × 157.57 Choose the nearest standard capacitor value for Cr, C r = 22nF Recalculate Fr1 to keep the same Qmax with the selected Cr capacitor. Fr1 = 1 = 100.7 Khz 2 ⋅ π ⋅ C r ⋅ Q max⋅ Rac Recalculate Lr with the selected Cr and Fr1. Lr = Q max⋅ Rac = 113uH 2 ⋅ π ⋅ Fr1 The actual Lr value should be lower than the calculated value to stay in ZVS region. Now calculate Lm value based on Lr and the k factor that preset in step 2: Lm = Lr ⋅ k = 113 × 5 = 565uH Please note that Lm is the magnetizing inductance of the transformer. The total primary inductance value Lp is the sum of Lm and Lr. L p = Lm + Lr = 678uH To simplify the power stage, the resonant inductor can be integrated into the power transformer by using slotted bobbin, also called two-section or two-chamber bobbin. By separate the primary winding and the secondary winding in the two chambers, the coupling between primary and secondary is much worse than the single section bobbin. Thus the leakage inductance is high and can be used as resonant inductor. The component count is lower and the copper loss is also smaller. Figure 11 is the picture of a two-section bobbin. Figure 11: 2-section Transformer www.irf.com AN-1160 12 When measure the inductance of a transformer, the primary inductance Lp is measured with all secondary windings opened. And the leakage inductance is measured with all the secondary windings shorted. Step 6: Calculate transformer primary and secondary turns The standard half-bridge equation for the transformer turns number calculation is used here: Np = Vin min⋅ D max 2 ⋅ ∆B ⋅ Ae ⋅ F min With ∆B = 0.2T , Ae = 2.11cm (ETD49), F min = 60 KHz , Vin min = 350V , D max = 0.5 2 Np = 350 × 0.5 ×10 = 35 2 × 0.2 × 2.11× 60 Ns = Np 35 = = 3.89 n 9 The number of turns must be an integer and should be higher than the calculated value, so choose Ns = 4 Then recalculate Np: Np = Ns ⋅ n = 4 × 9 = 36 Step 7: Calculate transformer primary and secondary current Most LLC converters design the minimum switching frequency to be below the resonant frequency Fr1, in order to maintain output voltage regulation at low line and full load. When the switching frequency is lower than the resonant frequency Fr1, the current waveform is shown as in Figure 12. Figure 12: Transformer primary current at full load and minimum input voltage I1 is the current where the resonant current in Lr meets the magnetizing current in Lm. This is also the point where Cr and Lr finish resonance for the first half-period of Fr1. At this point, there is no more energy delivered to the load and the output diodes are off. The Cr starts to resonate with Lr + Lm until the switching MOSFETs change states. I1 can be calculated as: www.irf.com AN-1160 13 I1 = n ⋅ Vout = 0.95 A 2 ⋅ Lm ⋅ 2 ⋅ Fr1 The peak and RMS value of primary current can be estimated as: Iout ⋅ π 2 Ipri ( pk ) = + I1 = 1.99 A 2⋅n 2 IpriRMS = Ipri ( pk ) 2 = 1 .4 A The RMS current is calculated by assuming pure sinusoid current waveform. So the actual primary RMS current is higher than the calculated value. The current in each secondary winding is very close to half-sinusoid, thus the peak and RMS current can be estimated by: Iout ⋅ π = 15.7 A 2 Iout ⋅ π Isrms = = 7.85 A 4 Ispk = The wire gauge of primary and secondary windings should be selected properly according to the calculated RMS current. Step 8: Calculate resonant capacitor voltage The Cr waveform is shown as in Figure 13: Figure 13: Typical resonant tank voltage and current waveforms ILm is the magnetizing current of transformer primary, not including the current which is delivered to the secondary load through an ideal transformer in parallel with Lm. The difference between ILr and ILm is the output current. www.irf.com AN-1160 14 Ideal Transformer M1 D1 Cr Lr ILr Lm M2 ILm COUT Iout/n LOAD D2 Figure 14: Lm and ideal transformer The VCr voltage reaches its peak when Lr current is crossing zero and it is at the mid of input voltage when Lr current reached its peak. The Cr voltage is at the maximum value when VS node is zero and it is at the minimum value when VS node is equals to Vin. So VCrmin and VCrmax can be calculated as: VC r max = n ⋅ Vout + Ipri ( pk ) × Lr Cr VC r min = Vin − n ⋅ Vout − Ipri ( pk ) × Lr Cr The peak to peak voltage ripple of VCr is VCrmax-VCrmin. VC rpk _ pk = 2n ⋅ Vout + 2 ⋅ Ipri ( pk ) × Lr − Vin Cr It can be seen that the maximum peak-to-peak voltage happens at the maximum load and the minimum DC input Vinmin, the switching frequency is at the minimum Fmin. In this example: Vcrpk _ pk = 2 × 9 × 24V + 2 × 1.99 A × 113uH − 350V = 368V 22nF The resonant capacitor Cr can be selected according to the capacitance value, together with its voltage and current rating. Polypropylene film capacitor is preferred to use for lower power loss. Please note the polypropylene film capacitor is rated under DC voltage or 50Hz AC voltage and has voltage derating at high frequency and high ambient temperature. The ability of withstanding high frequency voltage is limited by thermal (power dissipation) and peak current capability. Usually the derating starts at 85~90C ambient and is not a concern. But a capacitor with higher voltage rating www.irf.com AN-1160 15 should be chosen if the ambient temperature is higher than 85C. Below is an example of EPCOS MKP capacitor B32621 (630Vdc/400Vac). Figure 15: Vrms vs. frequency curve of MKP capacitor B32621 @ Ta<=90°C 4. IRS2795 Passive Components Design Step 9: Calculate the minimum dead-time to keep ZVS switching at zero load at the maximum input voltage For resonant half-bridge converter, the switching frequency goes to the maximum under no load at the maximum input voltage. Theoretically when the switching frequency is above the resonant frequency Fr1, the operation is ZVS switching. However, above resonance is only one of the necessary conditions for ZVS. The other condition is the equivalent parasitic capacitor of the halfbridge midpoint (junction capacitor of VS node) to be fully (dis-)charged within the dead-time period. Figure 16 demonstrates if the dead-time is not sufficient, the turn-on of the MOSFET has hardswitching even though the converter is working under the below resonant ZVS mode. www.irf.com AN-1160 16 Figure 16: ZVS and none-ZVS waveform of region 2 operation To keep the converter always working under ZVS condition, it is necessary to calculate the minimum time that required to fully (dis-)charging the VS equivalent capacitor during the two switches interleaving period (dead-time). As the equivalent capacitor is (dis-)charged by the circulating current in the transformer primary winding, so the worst case happens at the maximum input voltage and zero load condition where the transformer current is at minimum. At zero load, there is no current transfer to the secondary side and the current in the tank is just the magnetizing current of transformer. In each half-cycle, it is a linear straight line as shown in Figure 17. VS Primary current Figure 17: Transformer primary current at zero load So the primary current under this condition can be calculated as: I ' pri ( pk ) = n ⋅ Vout 4 F max⋅ ( Lr + Lm) I ' pri ( pk ) = 0.53 A www.irf.com AN-1160 17 The total equivalent junction capacitor CHB of VS node is shown in Figure 18. M1 VB HO Coss_ef f_1 IRS2795 VCC Cr VS Cw ell Crss_eff Lr Lm M2 LO Coss_ef f_2 Cs COM Figure 18: VS Equivalent junction capacitor C HB = 2 ⋅ Coss _ eff + Crss _ eff + CWell + Cs It includes: The effective Coss of the two MOSFETs (both high-side and low-side); The Coss_eff as defined in the MOSFET datasheet is the effective capacitance of MOSFET that gives the same charging time as a fixed capacitor while VDS is rising from 0 to 80% of VDS. So the Coss_eff of a 500V MOSFET is defined under 0 to 400V VDS which fits to this application. The effective Crss of the low-side MOSFET; The Crss of MOSFET is typically defined at VDS=25V. The Crss capacitance value reduces as VDS voltage increasing. So the effective Crss can be chose as ½ or 1/3 of Crss. The stray capacitance Cwell of IRS2795(1,2); The stray capacitance of IRS2795(1,2) is the high-side well capacitance of the 600V driver. The value of the stray capacitor is around 5pF. The snubber capacitor Cs (if any) that is connected to the VS node. For example, the Coss_eff of MOSFET STF13NM50N is 110pF, Crss is 5pF, and there is no snubber capacitor to the VS node, the (dis-)charging time of VS node can be calculated as: Coss _ eff = 110 pF , Crss _ eff = 2.5 pF , CWell = 5 pF , Cs = 0 pF Tch = C HB ⋅ Vin max I ' pri ( pk ) Tch = 185ns The dead-time calculation should also include the gate driver falling time. The MOSFET turn-off timing diagram is shown in Figure 19, which using LO and M2 as an example. In the first time interval t1, gate voltage discharges to a plateau voltage V’m, and both VDS voltage and ID current www.irf.com AN-1160 18 stay unchanged in t1. As long as MOSFET gate voltage reaches the miller plateau V’m, miller cap Cgd is discharged and VDS voltage starts increasing. Due to the nonlinearity of Coss capacitor, VDS voltage increase slowly at the beginning, then the slope becomes steeper at higher VDS voltage. The miller plateau is the flat portion of gate driver curve. It varies with drain current. MOSFET turns off at a relative low current level in LLC application, the miller plateau is very close to the gate turn off threshold Vgs(th). The timing that is interested for the dead time calculation is t1, as the charging time of the VS node (i.e. VDS of M2) starting from t2 is already included in the Tch calculation. In t1, VDS voltage is 0V, and MOSFET gate equals to a constant capacitor load to the IC. So the discharge time t1 can be calculated based on the RC time constant of the gate drive loop. t1 = − RC geq ln V 'm VG Where, R = Rdown _ eff + R g + R gFET C geq = (Qg − Qgd − Qgs ) , Please refer to Figure 21. Vgs − Vm V ' m ≈ Vgs (th ) VG = Vcc , IRS2795(1,2) gate output voltage is clamped to Vcc voltage Rdown_eff: IRS2795(1,2) gate driver effective pull down resistance (6Ω) Rg :is the external MOSFET gate drive resistor RgFET: MOSFET gate input resistance Figure 19: MOSFET turn-off equivalent circuit and timing diagram STF13NM50 gate equivalent capacitor is 2.32nF, MOSFET internal gate resistor is 5Ω, Vgs(th) is 3V. Thus if Vcc=15V, Rg=10Ω, gate discharge time t1 is: t1 = 78.4ns The dead-time should be longer than the sum of Tch and t1. For experience, it is recommended to add 50ns to the calculated value. The minimum dead-time TDT is then given by: TDT = Tch + t1 + 50ns = 313ns For most of the design, it’s not recommended to have a dead-time that is longer than 1us, as longer dead-time leads to higher body-diode power losses at full load. So if the calculated dead-time is too long, go back to step 2 and choose a smaller k value. Once the system parameters are defined, the passive components around the IRS2795(1,2) as shown in Figure 20 can be calculated. www.irf.com AN-1160 19 Figure 20: IRS2795(1,2) two-pin Oscillator CT = TDT ⋅ 10 −3 − 40 ⋅10 −12 313 ⋅10 −12 − 40 ⋅ 10 −12 = = 321 pF 0.85 0.85 CT capacitor should be equal or bigger than the calculated value for ZVS operation. Choose a standard capacitor value for CT. C T = 390 pF Calculate the actual dead-time per the selected CT value: t DT = (0.85CT + 40 pF ) ⋅ 2V = 371.5ns 2mA Calculate RT per the minimum switching frequency Fmin and CT: RT = 1 − 1kΩ 2 ⋅ F min⋅ t DT ⋅ 10 −3 RT resistor should be smaller than the calculated value to keep ZVS operation. Calculate Rmax per the maximum switching frequency Fmax and CT, RT: Re q = 1 − 1kΩ , 2 ⋅ F max⋅ t DT ⋅10 −3 R max = RT ⋅ Re q RT − Re q Calculate Rss with the desired soft-start frequency: Rsseq = 1 − 1kΩ , 2 ⋅ Fss ⋅ t DT ⋅10 − 3 Rss = RT ⋅ Rsseq RT − Rsseq Calculate Css based on the desired soft-start time: Css = Tss 3 ⋅ Rss In sleep mode or fault mode, RT pin is discharged to 0V. A diode Dss is put in parallel with Rss to fast discharge Css when IC is shutdown or in fault mode. This is to make sure the system still has soft start when IRS2795(1,2) restarts quickly. Dss can be any general purpose low voltage (10V) and low current (100mA) diode. The bootstrap capacitor CBS is used to hold VBS supply voltage for the high-side driver. The value of CBS is recommended to be 100nF to 220nF. Bigger CBS capacitor causes higher charging current www.irf.com AN-1160 20 during startup and should be avoid. IRS2795(1,2) doesn’t have integrated bootstrap MOSFET. A 600V/1A fast recovery diode is required for bootstrap. 5. IRS2795 Power Loss Calculation 5.1 Low voltage static loss that caused by quiescent current Pd1 = Vcc × Iqcc where Iqcc is 2.5mA maximum per IRS2795(1,2) datasheet. 5.2 The gate driver power losses The gate driver losses of IRS2795(1,2) are the losses when driving the two external MOSFETs M1 and M2. In ZVS mode, MOSFET VDS voltage is 0V prior to the gate turns on, so the “Miller” charge Qgd should be subtracted from the total gate charge. Further, at ZVS operation, the MOSFET is as a constant capacitor load to the driver. The equivalent capacitor value equals to the Cgs+Cgd at VDS=0V condition, which can be obtained from the gate charge curve in a MOSFET data sheet. It is indeed the slope factor of the gate charge curve where VGS is above the miller plateau voltage Vm, as shown in Figure 21. C geq = (Qg − Qgd − Qgs ) Vgs − Vm Figure 21: MOSFET gate charge curve and equivalent gate capacitance at ZVS mode Typically the Qg, Qgd and Qgs value are specified under 10V VGS voltage, Vm is the flat portion voltage of the gate charge curve. For example, STF13NM50 Qg=30nC, Qgd=15nC, Qgs=5nC, Vm=5.7V, its gate equivalent capacitor in ZVS is 2.32nF. The total gate charge in ZVS mode is proportional to the gate voltage: Qgz = Cgeq ⋅ VG www.irf.com AN-1160 21 IRS2795(1,2) gate output voltage is clamped to Vcc voltage. So the total gate driver losses of both high-side and low-side can be calculated by: Pdr = Pdr1 + Pdr 2 = 2 ⋅ Cgeq ⋅ Vcc 2 ⋅ Fsw The total gate driver losses are dissipated in driver IC IRS2795(1,2) and the external gate driver resistor including the MOSFET internal gate resistor. The power loss in IRS2795(1,2) is proportional to the resistor divider value: Pd 2 = ( Rup _ eff Rdown _ eff Pdr + )× Rup _ eff + Rg + R g FET Rdown _ eff + Rg + R gFET 2 Where, Rg :is the external MOSFET gate drive resistor Rup_eff: IRS2795(1,2) gate driver effective pull up resistance (40Ω) Rdown_eff: IRS2795(1,2) gate driver effective pull down resistance (6Ω) RgFET: MOSFET gate input resistance The gate driver pull-up and pull-down resistance used for power loss calculation are given below: Rup = 40Ω, Rdown = 6Ω . They are bigger than datasheet specification (with is defined under 20mA current) as they are the equivalent pull-up and pull-down resistance under high gate current. 5.3 The CMOS switching losses The switching loss in low voltage logic circuit is proportional to the switching frequency and supply voltage Vcc: Pd 3 = Vcc × Fsw × Qcmos For IRS2795(1,2), Qcmos = 6nC ~ 10nC 5.4 The high voltage switching losses The switching losses in high voltage level-shift circuit: Pd 4 = (Vcc + Vin ) × Fsw × Qp Vin is the input bus voltage. Qp is the charge absorbed by the level shifter. For IRS2795(1,2), Qp is 2nC under 300V to 430V bus voltage. 5.5 An example of power loss calculation The total power loss in IRS2795(1,2) is the sum of Pd1 to Pd4. Pd _ total = Pd1 + Pd 2 + Pd 3 + Pd 4 An example of power loss calculation with Vcc=15V, maximum switching frequency =150KHz, MOSFETs =STF13NM50N, input bus voltage = 400V, external gate resistor = 10ohm: Pd 1 = 37.5mW Pdr = 157 mW , Pd 2 = 79.5mW Pd 3 = 18mW Pd 4 = 124.5mW Pd _ total = 259.5mW www.irf.com AN-1160 22 It can be seen that the high voltage switching loss Pd4 and gate driver loss Pd2 are the main source of total power losses. Pd4 is proportional to switching frequency and HV bus voltage. For 400V DC BUS voltage, IRS2795(1,2) can directly drives big MOSFETs (Cgeq ≤ 4.7nF) up to 250KHz switching frequency. It is necessary to clamp the Vcc supply voltage to 15V or lower to reduce gate driver losses when the frequency goes to 300KHz while driving big MOSFETs. For 300KHz to 500KHz switching frequency and 400V applications, it is recommended to use external driver. IC operation current Icc can be obtained by the total low-voltage power loss and Vcc voltage: Icc = ( Pd1 + Pdr + Pd 3) / Vcc 6. MOSFET Selection Guide The power MOSFET should be selected per the breakdown voltage and RDSON value. In addition, the body diode reverse recovery characteristic also plays important role to the selection. The converter usually has a few switching cycles that is under hard switching at the beginning of startup. This is because the resonant capacitor and output capacitors are fully discharged. In this case, longer reverse recovery time could cause shoot through between the two MOSFETs. Thus a MOSFET with fast reverse recovery diode is preferred. As the resonant half-bridge has ZVS switching, the turn-on loss is negligible. If not switching under very high frequency (≤150Khz), the major power loss in MOSFET comes from the conduction loss. The maximum conduction loss can be calculated as: Pcon = Iqrms 2 × Rdson @ Tj Where Iqrms = Ipri ( pk ) , and [email protected] is the MOSFET on-state resistance at the system 2 maximum allowable junction temperature. The calculation of the turn-off loss of MOSFET is complicated due to none linearity of Coss under different VDS voltage. Thus we use the estimated formula: Poff = C HB × Vin 2 × Fsw 24 The total power loss in each MOSFET equals to Pcon + Poff . IRS2795(1,2) uses the Rdson of low side MOSFET for current sensing and over current protection. The product family provides two choices on different over current protection level: the OCP threshold of IRS27951 is 2V and IRS27952 is 3V. Typically the IRS27951 is good for oversized MOSFET where a lower Rdson for better efficiency and the IRS27952 is good for cost effective MOSFET where the Rdson is bigger. A quick estimation for OCP threshold is to use 2.5 to 3 times of the maximum drain current times the Rdson of MOSFET. At startup, the MOSFET current could be a few times higher than the normal working current. To prevent false triggering of over current protection when using large Rdson MOSFET, it is recommended to extend the soft-start time to tens of milliseconds. www.irf.com AN-1160 23 7. Operating Waveforms and Efficiency of the Reference Design The specification of the reference design: Parameter Description Value Vinmax The maximum DC bus voltage 430V Vinmin The minimum DC bus voltage 350V Vinnom The nominal DC bus voltage 390V Vout 1 The DC output voltage 24V Iout 1 The output load current 6A Vout 2 The DC output voltage 12V Iout 2 The output load current 6A Fr1 The resonant frequency 100KHz Fmax The maximum switching frequency 150KHz Dmax The maximum duty-cycle 0.5 Tss The soft start time 30ms Fss The soft start frequency 300KHz Transformer ETD49 Design analysis result: www.irf.com Resonant tank components Cr=22nF, Lr=125uH, Lm=500uH, k=4 Transformer Np=36, N24V=4, N12V=2, n=9 IRS27951 components CT=390pF, RT=18k, Rmax=14k, Rss=3.9k, Css=3.3uF AN-1160 24 Rstart2 Rstart3 270k 270k 270k Rvcc 56 STPS30100 Dg1 1N4148 Rx1 STF13NM50N 24V Dbs COM CVcc2 DSS 4 CSS Dz CT 1 9 TX D2 20 Cout3 + 1 2 3 4 5 6 Cout4 + + Cout5 Rprl1 560 DNP 22nF/1kV Cout2 + Jumper 100nF 1 JP4 Cout1 W1 1mF/35V LO Cr 1mF/35V VS COM 220nF 1mF/35V CT/SD Rgs1 Cbs 7 5 1 2 3 4 5 6 Header 6 Dg2 Rx2 STF13NM50N IRS2795 4.7 1N4148 LO 7 Q2 2 Rg2 19 18 STPS30100 COM2 17 14 10 610uH COM Rgs2 5 13 12 D3 11 20TQ040 12V 1 18V RT 18k 8 6 1 RMAX + HO 1 CVcc1 CDC 2 1 VB RT 1 RSS JP3 3.9k 510K 1N4148 Rdisch 15k 270uF/450V 10 VCC Lf1 3.3uH/10A VTR 1mF/35V 3 390pF 5A/250V 2 VCC 3.3uF 0.33uF/630V + C4 1uF C3 Header 2 N U1 3 100nF C2 33uF/35V F1 L1 4 DNP 2 1 C1 0.1uF/275V-X2 R1 0.1uF/275V-X2 2 5 t° Header 3 VCC 1 RNTC VS Rg1 DNP 1 B1 265VAC JP1 Q1 1 4.7 1N4148 MURS160 GBU4J-BPMS-ND DNP= Do Not Populate 1 Lf2 3.3uH/10A JP5 1 2 3 4 5 6 Cout8 + Cout9 Rprl2 470 + 100nF 1.5mF/25V D4 Cout7 + 1000uF/25V Cout6 Cs 1.5mF/25V COM 1 2 3 4 5 6 Header 6 2.2nF/250V 1 20TQ040 Rs4 15k Rled2 Rled1 5.6k 2.2k Rbias2 Rs1 33k Rbias1 2.2k DNP Cf1 DNP TLP621 FB COMP 1 Cf2 100nF DNP=Do not Populate Rs2 0 Rf2 47k U3 TL431 7.1 Schematic Rs5 3.9k Rs3 3.74k 1 U2 COM2 25 AN-1160 D1 Rstart1 www.irf.com 1 Vbus The reference board has input rectifier and filter, so it can take either DC or AC input. The DC input range is 350V~430V, the AC input voltage range is 250Vac~300Vac. The dummy loads at 24V and 12V output are for cross-regulation purpose. HO D5 Figure 22 – IRS27951 Reference Design Schematic CT 1 1 RT 7.2 BOM Designator Description Quantity Value/Rating Vendor DIGIKEY Part# B1 Single Phase Bridge Rectifier 1 600V/4A GBU4J-BPMS-ND C1, C2 X2 Safety Capacitor 2 100nF/275VAC DIGIKEY P10524-ND C3 Metal Poly Capacitor 1 0.33uF/630V DIGIKEY P12245-ND EET-HC2W271LA C4 Electrolytic Bulk Capacitor TS-HC 1 270uF/450V DIGIKEY Cbs 1206 General Purpose Ceramic SMD 1 220nF/50V DIGIKEY 490-1776-1-ND Cf2, Cout5, Cout9, CVcc2 1206 General Purpose Ceramic SMD 4 100nF/50V DIGIKEY 490-1775-1-ND CDC Electrolytic Capacitor FM Radial 1 33uF/35V DIGIKEY P13475-ND Cf1 Not Used Cout1, Cout2, Cout3, Cout4 Aluminium Electrolytic Capacitor 105°C 4 1000uF/35V DIGIKEY 565-1581-ND Cout6, Cout7 Aluminium Electrolytic Capacitor 105°C 2 1500uF/25V DIGIKEY 565-1557-ND Cout8 Aluminium Electrolytic Capacitor 105°C 1 1000uF/ 25V DIGIKEY 565-1555-ND Cr Polypropylene Capacitor High Ripple 1 22nF/1kV DIGIKEY 495-3552-ND Cs 250VAC Y1 Safety Ceramic Disc Capacitor 1 2.2nF/250V DIGIKEY 445-2411-ND CSS 1206 General Purpose Ceramic SMD 1 3.3uF/16V DIGIKEY 445-4038-1-ND CVcc1 1206 General Purpose Ceramic SMD 1 1uF/25V DIGIKEY 445-1592-1-ND CT 1206 General Purpose Ceramic SMD ±5% 1 390pF/50V DIGIKEY 478-1487-1-ND D1, D2 TO220AB Power Schottky Rectifier 2 100V/30A DIGIKEY STPS30100CT D3, D4 TO220AC Power Schottky Rectifier 2 40V/20A DIGIKEY 20TQ040PBF-ND D5, Dg1, Dg2, DSS Fast Recovery Diode DO-35 4 75V/0.3A DIGIKEY 1N4148DICT-ND Dbs Fast Recttifier diode SMB 1 600V/1A DIGIKEY MURS160-FDICT-ND Dz Zener Diode SMD 1 18V/0.5W DIGIKEY FLZ18VCCT-ND F1 FUSE IEC FA LBC 5x20 1 250V/5A DIGIKEY F2395-ND JP1 CONN HEADER 3POS 0.156 VERT TIN 1 DIGIKEY WM4621-ND JP3 CONN HEADER 2POS 0.1 VERT TIN 1 DIGIKEY WM4200-ND JP4, JP5 CONN HEADER 6POS 0.156 VERT TIN 2 L1 EMI Common Mode Choke 1 DIGIKEY WM4624-ND 16mH/2.6A DIGIKEY 237-1233-ND Lf1, Lf2 PCV Series Drum Core Inductor 10mm 2 4.7uH/12A COILCRAFT PCV-0-472-10L Q1, Q2 TO-220FP N-Channel Power MOSFET 2 500V/12A DIGIKEY STF13NM50N R1, Rbias2, Rgs1, Rgs2 Not Used Rbias1, Rled1 1206 SMD Film RED 1/4W 1% 2 2.2k DIGIKEY RHM2.20kFCT-ND Rdisch Metal Film Power Resistor 2W 5% 1 510k DIGIKEY BC510KW-2CT-ND Rf2 1206 SMD Film RED 1/4W 1% 1 47k DIGIKEY RHM47.0kFCT-ND Rg1, Rg2 1206 SMD Film RED 1/4W 5% 2 10 DIGIKEY RHM10ERCT-ND Rled2 1206 SMD Film RED 1/4W 1% 1 5.6k DIGIKEY RHM5.60kFCT-ND RMAX 1206 SMD Film RED 1/4W 1% 1 15k DIGIKEY RHM15.0kFCT-ND RNTC Inrush Current Limiter 1 5 DIGIKEY 495-2093-ND Rprl1 Metal Film Power Resistor 2W 5% 1 560 DIGIKEY PPC560W-2CT-ND Rprl2 Metal Film Power Resistor 2W 5% 1 470 DIGIKEY PPC470W-2CT-ND Rs1 1206 SMD Film RED 1/4W 1% 1 33k DIGIKEY RHM33.0KFCT-ND Rs2 1206 SMD Film RED 1/4W 1% 1 0 DIGIKEY P0.0ECT-ND Rs3 1206 SMD Film RED 1/4W 1% 1 3.74k DIGIKEY RHM3.74KFCT-ND Rs4 1206 SMD Film RED 1/4W 1% 1 15k DIGIKEY RHM15.0KFCT-ND Rs5, RSS 1206 SMD Film RED 1/4W 1% 2 3.9k DIGIKEY RHM3.90KFCT-ND Rstart1, Rstart2, Rstart3 1206 SMD Film RED 1/4W 1% 3 270k DIGIKEY RHM270KFCT-ND RT 1206 SMD Film RED 1/4W 1% 1 18k DIGIKEY RHM18.0KFCT-ND Rvcc 1206 SMD Film RED 1/4W 5% 1 56 DIGIKEY RHM56ERCT-ND Rx1, Rx2 1206 SMD Film RED 1/4W 5% 2 4.7 DIGIKEY RHM4.7ERCT-ND TX Resonant Power Transformer 1 ETD49 PRECISION INC 019-4974-00R U1 IRS27951 Control IC 1 IR IRS27951S U2 Photocoupler TRANS-OUT 4-DIP 1 TLP621 DIGIKEY TLP621FT-ND U3 Programmable Voltage Regulator SOT23-3 1 TL431 DIGIKEY 296-17328-1-ND W1 Jumper for Primary Current Sensing Loop 1 www.irf.com AN-1160 AWG22, multi strands 26 7.3 Typical Operating Waveforms Figure 23 – 400Vdc input, 0W load startup Figure 24 - 400Vdc input, 220W load startup Figure 25 - 400Vdc input, 220W load operation www.irf.com AN-1160 27 Figure 26 – 350Vdc input, 220W load operation Figure 27 – 420Vdc input, 220W load operation Figure 28 – 420Vdc input, 0W load operation www.irf.com AN-1160 28 7.4 Short circuit protection Figure 29 –260Vac input, short 12V, IC latched shut down 7.5 Efficiency The average efficiency of the board at 25%, 50%, 75% and 100% load is 92% at 270Vac input: 24Vout 24.176 24.2 24.22 24.24 23.517 24.814 www.irf.com 24V Iout(A) 1.5 3 4.5 6 6 0 12Vout 11.97 11.92 11.9 11.86 12.26 11.63 12V Iout(A) 1.5 3 4.5 6 0 6 AN-1160 270Vac 400Vdc Pout(W) Efficiency Efficiency 54.2 90.8% 91.0% 108.4 92.6% 92.6% 162.5 92.4% 92.9% 216.6 92.2% 92.7% 141.1 92.8% 69.8 89.5% 29 Efficieny vs. Output Power 93.5% Efficiency 93.0% 92.5% 92.0% 400Vdc input 91.5% 270Vac input 91.0% 90.5% 50.0 70.0 90.0 110.0 130.0 150.0 170.0 190.0 210.0 Output Power (W) Figure 30 – Efficiency Plot 8. Layout guidelines and example Ground Plane: In order to minimize noise coupling, the ground plane should not be placed under or near the high voltage floating side. Gate Drive Loops: Current loops behave like antennas and are able to receive and transmit EM noise. In order to reduce the EM coupling and improve the power switch turn on/off performance, the gate drive loops must be reduced as much as possible. For the low-side driver, the return of the drive loop must be directly connected to the COM pin of the IC and separate with signal ground (power ground and signal ground have star connection at COM pin). Supply Capacitor: It is recommended to place a bypass capacitor (CVCC) between the VCC and COM pins. A 1µF ceramic capacitor is suitable for most applications. This component should be placed as close as possible to the pins in order to reduce parasitic elements. CBS Capacitor: The CBS capacitor should be placed as close as possible to the VB and VS pins. Routing and Placement: 1) The 8-pin IC has only one COM pin for both signal return and power return, so it is strongly recommended to route the signal ground and power ground separately with a star connection at the COM pin. www.irf.com AN-1160 30 2) The RT pin provides a current reference for the internal oscillator and needs to be kept as clean as possible to avoid frequency jittering or duty-cycle mismatch between high-side and low-side. The components connected to this pin must keep away from the high frequency switching loop such as the gate driver loop and the VS node. The PCB traces connected to RT pin also need to be kept away from any switching node. 3) Connect CT capacitor directly to COM pin, don’t share the return with any other signal ground. Layout examples VS node Signal components are kept away from switching nodes Supply bypass capacitors are close to IC pins. Star connection at COM pin Figure 31: Single layer board example 9. Appendix Symbols list Lr: primary resonant inductance. It is the primary leakage inductance of transformer when there is no external added resonant inductor. Lm: transformer primary magnetic inductance. It is the measured transformer primary inductance minus the leakage inductance. Cr: primary resonant capacitor and DC blocking capacitor fr1: the resonant frequency between Lr and Cr Rac: Equivalent AC resistance for resonant tank AC analysis RDSon: MOSFET channel ON resistance www.irf.com AN-1160 31 fmax: converter maximum operating switching frequency fmin: converter minimum operating switching frequency Qg: MOSFET total gate charge Qgd: MOSFET gate to drain (Miller) charge Qgs: MOSFET gate to source charge IQCC: IRS2795(1,2) quiescent current Rg: MOSFET gate drive resistance external to IRS2795(1,2) Rup: IRS2795(1,2) gate driver pull up resistance Rdown: IRS2795(1,2) gate driver pull down resistance RgFET: MOSFET gate input resistance PICmax: IRS2795(1,2) maximum power dissipation VCC: Supply voltage on IRS2795(1,2) Vcc pin ICC: IRS2795(1,2) IC supply current References [1] IRS2795(1,2) datasheet www.irf.com AN-1160 32

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