Design of Resonant Half-Bridge converter using IRS2795(1,2) Control IC

Application Note AN-1160
Design of Resonant Half-Bridge converter using
IRS2795(1,2) Control IC
By Helen Ding
Table of Contents
1. Introduction & Device Overview
2. LLC Resonant Half Bridge Converter Operation
3. Transformer and Resonant Circuit Design
4. IRS2795 Passive Components Design
5. IRS2795 Power Loss Calculation
6. MOSFET Selection Guide
7. Operating Waveforms and Efficiency
8. Layout Guidelines and Example
9. Appendix
Symbol list
References
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1. Introduction and Device Overview
The IRS2795(1,2) is a self oscillating half-bridge driver IC for resonant half-bridge DC-DC converter
applications for use up to 600V. It has a fixed 50% duty-cycle and very wide operating frequency
range. The maximum switching frequency can go up to 500kHz. The frequency can be programmed
externally through the RT and CT pins. The IC offers flexibility to program the minimum operating
frequency, the maximum operating frequency and the frequency sweep at power up for the softstart function.
The dead time is programmed by the CT capacitor. The programmable dead-time allows the user to
optimize the system with the minimum body-diode conduction time for higher efficiency under full
load, while keeping ZVS switching under no load condition.
The IC offers over current protection using the on-state resistance of the low-side MOSFET. The
protection threshold is 2V for IRS27951 and it is 3V for IRS27952 IC.
The IC can be disabled by externally pulling the voltage at the CT/SD pin below its enable voltage
threshold. The IC enters “sleep” mode and only consumes micro-power when disabled.
IRS2795(1,2) packing in a 8-pin package, it’s easy to use, and drastically reduces external
component count for a high efficiency low cost power supply.
Figure 1 is the typical application schematic of IRS2795(1,2):
DC BUS
VIN
R1
R2
Rcc
D3
Dbs
Cbs
U1
1
CDC
2
3
4
Dss
Rss
RT
VCC
VB
RT
HO
CT/SD
VS
COM
LO
8
M1
Rg1
D1
7
6
Lr
5
IRS2795
M2
Rg2
Rmax
Lm
CT
Css
LOAD
COUT
D2
Cr
RTN
Rled
Rbias
Rs1
Cf1
U2
Cf2
Rf2
U3
TL431
Rs2
Figure 1: IRS2795(1,2) typical application circuit
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The pinout of IRS2795(1,2) is shown below.
1
VCC
2
RT
3
CT/SD
4
COM
IRS2795
Lead Assignment
Pin#
Symbol
Description
1
VCC
Supply voltage
2
RT
Oscillator timing resistor
VB
8
3
CT/SD
HO
7
4
COM
Ground
VS
6
5
LO
Low-side gate drive
6
VS
High-side gate drive return/ HV Current Sense
LO
5
7
HO
High-side gate drive
8
VB
High-side floating supply voltage
Oscillator timing capacitor/Shutdown
Figure 2: IRS2795(1,2) IC pin assignment
2. LLC Resonant Half-Bridge Converter Operation
The increasing popularity of the LLC resonant converter in its half-bridge implementation is due to its
high efficiency, low switching noise and ability to achieve high power density. This topology is also
the most attractive topology for front-end DC bus conversion. It utilizes the magnetizing inductance
of the transformer to construct a complex resonant tank with Buck Boost transfer characteristics in
the soft-switching region. The typical power stage schematic for this topology is shown below.
M1
D1
HO
Vin
VS
Lr
n:1:1
Lm
M2
LO
COUT
LOAD
D2
Cr
Figure 3: Typical schematic of a DC-DC half-bridge resonant converter
Devices M1 and M2 operate at 50% duty cycle and the output voltage is regulated by varying the
switching frequency of the converter. The converter has two resonant frequencies – a lower
resonant frequency (given by Lm, Lr, Cr and the load) and a fixed higher series resonant frequency
Fr1 (given by Lr and Cr only). The two bridge devices M1 and M2 can be soft-switched for the entire
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load range by operating the converter under inductive load mode (ZVS region). It can be either
above or below the resonant frequency Fr1.
The typical AC transfer characteristics1 for a LLC tank resonant converter are shown in Figure 4.
The group of curve indicates the gain under different load conditions.
Figure 4: Typical frequency response of a LLC resonant converter
The characteristics of a LLC resonant converter can be divided into three regions based on the 3
different modes of operation.
The first region is for switching frequency above the resonant frequency Fr1.
Fr1 =
1
2π Lr ⋅ C r
1 (the purple shaded area) the switching frequency is higher than resonant frequency Fr1.
In region ○
The converter operation is very similar to a series resonant converter. Here Lm never resonates with
resonant capacitor Cr; it is clamped by the output voltage and acts as the load of the series resonant
tank. This is the inductive load region and the converter is always under ZVS operation regardless of
the load condition.
In the 2nd region, the switching frequency is higher than the lower resonant frequency but lower than
2 is in the pink shaded area in Figure 4. The lower resonant frequency varies with load,
Fr1. Region ○
2 and region ○
3 traces the peak of the family load vs. gain curves. In this
so the boundary of region ○
complex region, the LLC resonant operation can be divided into two time intervals. In the first time
interval, Lr resonates with Cr and Lm is clamped by output voltage. When the current in the resonant
1
For this AC analysis, only the fundamental component of the square-wave voltage input to the resonant
network contributes to the power transfer to output. The transformer, rectifier and filter are replaced by an
equivalent AC resistance, Rac.
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inductor Lr resonates back to the same level as the magnetizing current, Lr and Cr stop resonating.
Lm now participates in the resonant operation and the second time interval begins. During this time
interval, dominate resonant components change to Cr and Lm in series with Lr. The ZVS operation in
2 is guarantees by operating the converter to the right side of the load gain curve. For a
region ○
2 or region ○
3 depends on the
switching frequency below resonant Fr1, it could falls in either region ○
load condition.
3 below fr1, the LLC resonant converter operates in capacitive mode; M1 and M2
In the ZCS range ○
are under hard switching and have high switching losses. So ZCS operation should always be
avoided.
The typical operating waveforms of the 3 modes are demonstrated in Figure 5 to Figure 7.
Figure 5: Typical waveform of above resonant ZVS switching
Figure 6: Typical waveform of below resonant ZVS switching
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The waveforms indicate that the current in secondary rectifier diodes moves from continuous current
mode (CCM) to discontinuous current mode (DCM) when the switching frequency varies from above
resonant ZVS to below resonant ZVS due to load increasing. The ripple voltage on the resonant
capacitor Cr also increases in the below resonant ZVS mode.
Figure 7: Typical waveform of below resonant ZCS switching
In ZCS mode, the two switching devices M1 and M2 are turned off under zero current condition. The
turn-on of the two switches is hard switching (none ZVS). The turn-on switching loss is high
especially under high voltage bus voltage. The resonant capacitor Cr also has high voltage stress.
ZCS operation should always be avoided.
The typical voltage conversion ratio of a LLC resonant converter is shown in Figure 8.
Q - Low (no load)
2nVout
Vin
Q - High (full load)
Frequency modulation range
@ low line
2nVout
Vinmin
Frequency modulation range
@ high line
2nVout
Vinmax
Over load or Short
Circuit
Fr1
fmin
fmax
Frequency
Figure 8: Typical voltage conversion ratio of a LLC resonant converter
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With a fixed input voltage, the converter varies switching frequency to regulate the output voltage
over load range – keeping the same conversion ratio over the family of curves with different Q.
Given a fixed load condition, the converter varies switching frequency along that load line to regulate
output voltage over input voltage range – the conversion ratio increases when input voltage
decreases.
To design the LLC resonant half-bridge converter, we use the First Harmonic Approximation (FHA)
to get equivalent circuit. All the components are put to primary side to simply the analysis. The load
equals to a resistor Rac that is in parallel with transformer primary inductance Lm.
M1
D1
n:1:1
Cr
Vin
Lr
M2
Lm
COUT
LOAD
D2
Cr
2⋅
Vin
Lr
n ⋅ Vout⋅
π
4
π
Vf und
Rac
Rac
2
n ⋅ RLOAD⋅
8
2
π
Lm
Figure 9: The FHA equivalent circuit
The input voltage of the resonant tank is a square wave with amplitude equals to the input DC
voltage Vin. The fundamental component of the square waveform is:
2 ⋅ Vin
π
sin(ϖ ⋅ t )
The output voltage of the resonant tank is the voltage across Lm. It is very close to a square
waveform with amplitude swinging from − n ⋅ Vout to + n ⋅ Vout . So the fundamental component of
the output square waveform is:
4 ⋅ n ⋅ Vout
sin(ϖ ⋅ t )
π
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The power dissipation on the equivalent AC resistor is equal to the power dissipation of RLOAD
resistor, thus it can be written as:
 4 ⋅ n ⋅ Vout 


2
Vout
2π 

=
R LOAD
Rac
2
Rearrange the formula and get the equivalent AC resistor:
Rac =
8 ⋅ n2
R LOAD
π2
The transfer ratio of the equivalent circuit can be obtained as following:
j ⋅ ω ⋅ Lm⋅ Rac
j ⋅ ω ⋅ Lm+ Rac
M
1
j ⋅ ω⋅ Lr +
+
j ⋅ ω⋅ Cr
j ⋅ ω⋅ Lm⋅ Rac
j ⋅ ω⋅ Lm + Rac
Re-write the formula,
1
M
1+
Lr
1
−
Lm
+
2
ω ⋅ Lm⋅ Cr
jω⋅ Lr
Rac
−
j
ω⋅ Cr⋅ Rac
With the following definitions, M can be simplified.
Fr1 =
k=
1
,
2π Lr ⋅ C r
Lm
,
Lr
Rac =
x=
Fsw
,
Fr1
8 ⋅ n 2 ⋅ RLOAD
π
x
,
Lr ⋅ C r
ϖ = 2πFsw = 2π ⋅ x ⋅ Fr1 =
2
,
Q=
2πFr1 ⋅ Lr
1
=
Rac
2πFr1 ⋅ C r ⋅ Rac
1
M
1+
1
⋅ 1 −
k 

 + j⋅ Q⋅  x −

2

x 
1
1

x
Or,
1
M
2
1 + 1 ⋅ 1 − 1  +

k 
2 
x 


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Q⋅  x −
 
 
1 
2

x 
8
Per Figure 9, M is also equals to the output voltage to input voltage ratio:
M=
n ⋅ Vout ⋅
2⋅
Vin
4
π = Vout ⋅ 2 ⋅ n
Vin
π
So we have the conversion ratio of output voltage Vout to input voltage Vin:
Vout M
=
Vin 2 ⋅ n
3. Transformer and Resonant Circuit Design
This section provides the details of how to calculate the key components of a LLC converter, take a
24V output 240W power supply as an example.
The system input data
Parameter
Unit
Vinmax
V
The maximum DC bus voltage
430
Vinmin
V
The minimum DC bus voltage
350
Vinnom
V
The nominal DC bus voltage
390
Vout
V
The DC output voltage
24
Iout
A
The output load current
10
Fr1
KHz
The resonant frequency
100
Fmax
KHz
The maximum switching frequency ①
150
The maximum duty-cycle
0.5
Dmax
Description
Value
Tss
ms
The soft start time
10
Fss
KHz
The soft start frequency
300
Transformer
ETD49
①Note: Typically set Fmax < 2xFr1 as the parasitic capacitance in the system introduced a 3rd
resonant frequency which could cause the output voltage to increase with switching frequency at no
load if the maximum switching frequency is higher than the limit.
Step 1: Calculate the transformer turns ratio
Vin max
,
2 ⋅ Vout
430
n=
= 8.96
2 ⋅ 24
n=
The transformer turns ratio is calculated with the maximum input voltage to make sure the output is
always under regulation, including the worst case - high-line voltage and no load condition.
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Usually the transfer ratio of the power stage is higher than the theoretical calculated value. This is
because of the parasitic capacitance in the system (the coupling capacitor between transformer
windings and the junction capacitors of output diodes) affects the resonance, especially at zero load
where the switching frequency is much higher than the resonant frequency. So it’s recommended to
choose the n to be slightly higher than the calculated value especially if the controller has no burst
mode to keep regulation at high line and zero load condition.
n=9
Step 2: Choose k value
k is the ratio between the transformer magnetic inductance and the resonant inductance. Smaller k
value gives steeper Gain curve, especially at the below resonant ZVS region as shown in Figure 10.
The output voltage is more sensitive to frequency variation with smaller k factor.
3
2.5
k=5
2
1.5
k=10
1
0.5
0
Figure 10: k factor
A higher k value results in higher magnetic inductance and thus lower magnetizing current in the
transformer primary winding – that means lower circulating power losses. However, higher magnetic
inductance could also cause non-ZVS switching at high line and zero load condition where the
circulating current is too small to fully charge / discharge the VS node during dead-time.
The recommend range of k is from 3 to 10. Here k = 5 is chosen.
Step 3: Calculate Qmax to stay in ZVS operation at the maximum load under the minimum
input voltage
The input impedance of the equivalent resonant circuit (Figure 9) is given by:
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Zin
j ⋅ ω⋅ Lr +
1
j ⋅ ω⋅ Cr
2 2
Zin
k ⋅x ⋅Q
Q⋅ Rac
2 2
2
+
j ⋅ ω⋅ Lm⋅ Rac
j ⋅ ω⋅ Lm + Rac
+ j x −
1 + k ⋅x ⋅Q


1
x
x⋅ k
+

2 2
2
1 + k ⋅x ⋅Q

To keep the converter working in soft switching mode, the operating point should always in the ZVS
region as shown in Figure 4. The ZVS ZCS boundary line is defined by the phase angle of Zin
Ф(Zin)= 0 (the boundary condition between capacitive and inductive load), i.e. the imaginary part of
Zin is zero. With this condition we can calculate the maximum Q which allows the converter to stay
in ZVS. The maximum Q happens at the minimum input voltage and the maximum load.
1 + k⋅  1 −
Qmax
1
k
1


⋅

2
Mmax
1 + k⋅  1 −

2
Mmax − 1
1
k


⋅

1
2
 2⋅ n ⋅ Vout 


Vinmin 

2
 2⋅ n ⋅ Vout  − 1


Vinmin

Where Mmax is the maximum conversion ratio at the minimum input voltage,
Q max = 0.456
Step 4: Calculate the minimum switching frequency
The minimum switching frequency happens at the maximum load and minimum input voltage with
the previous calculated maximum Qmax. As Qmax is defined by Im(Zin)=0,
x⋅ k
x − 1 +
 0
 x
2 2
2
1 + k ⋅ x ⋅ Qmax 

The Fmin can be calculated with:
xmin
1
1 + k⋅  1 −


1

2
Mmax 
1
1 + k⋅  1 −


1

2
 2n ⋅ Vout 
 Vinmin  

 
x min = 0.607
F min = x min⋅ Fr1 = 60.7 KHz
Step 5: Calculate Lr, Cr and Lm
As Qmax happens at the maximum load, so the resonant components Lr, Cr and Lm can be
calculated per the Qmax value that had obtained in step 3:
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RLOAD =
Rac =
Lr =
Cr =
Vout 24V
=
= 2.4Ω
Iout 10 A
8 ⋅ n 2 ⋅ RLOAD
π2
=
8 × 9 2 × 2 .4
π2
= 157.57Ω
Q max⋅ Rac 0.456 × 157.57
=
= 114uH
2 ⋅ π ⋅ Fr1
2 ⋅ π ⋅ 100 K
1
1
=
= 22.2nF
2 ⋅ π ⋅ Fr1 ⋅ Q max⋅ Rac 2 ⋅ π ⋅ 100 K × 0.456 × 157.57
Choose the nearest standard capacitor value for Cr, C r = 22nF
Recalculate Fr1 to keep the same Qmax with the selected Cr capacitor.
Fr1 =
1
= 100.7 Khz
2 ⋅ π ⋅ C r ⋅ Q max⋅ Rac
Recalculate Lr with the selected Cr and Fr1.
Lr =
Q max⋅ Rac
= 113uH
2 ⋅ π ⋅ Fr1
The actual Lr value should be lower than the calculated value to stay in ZVS region.
Now calculate Lm value based on Lr and the k factor that preset in step 2:
Lm = Lr ⋅ k = 113 × 5 = 565uH
Please note that Lm is the magnetizing inductance of the transformer. The total primary inductance
value Lp is the sum of Lm and Lr.
L p = Lm + Lr = 678uH
To simplify the power stage, the resonant inductor can be integrated into the power transformer by
using slotted bobbin, also called two-section or two-chamber bobbin. By separate the primary
winding and the secondary winding in the two chambers, the coupling between primary and
secondary is much worse than the single section bobbin. Thus the leakage inductance is high and
can be used as resonant inductor. The component count is lower and the copper loss is also
smaller. Figure 11 is the picture of a two-section bobbin.
Figure 11: 2-section Transformer
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When measure the inductance of a transformer, the primary inductance Lp is measured with all
secondary windings opened. And the leakage inductance is measured with all the secondary
windings shorted.
Step 6: Calculate transformer primary and secondary turns
The standard half-bridge equation for the transformer turns number calculation is used here:
Np =
Vin min⋅ D max
2 ⋅ ∆B ⋅ Ae ⋅ F min
With ∆B = 0.2T , Ae = 2.11cm (ETD49), F min = 60 KHz , Vin min = 350V , D max = 0.5
2
Np =
350 × 0.5
×10 = 35
2 × 0.2 × 2.11× 60
Ns =
Np 35
=
= 3.89
n
9
The number of turns must be an integer and should be higher than the calculated value, so choose
Ns = 4
Then recalculate Np:
Np = Ns ⋅ n = 4 × 9 = 36
Step 7: Calculate transformer primary and secondary current
Most LLC converters design the minimum switching frequency to be below the resonant frequency
Fr1, in order to maintain output voltage regulation at low line and full load. When the switching
frequency is lower than the resonant frequency Fr1, the current waveform is shown as in Figure 12.
Figure 12: Transformer primary current at full load and minimum input voltage
I1 is the current where the resonant current in Lr meets the magnetizing current in Lm. This is also
the point where Cr and Lr finish resonance for the first half-period of Fr1. At this point, there is no
more energy delivered to the load and the output diodes are off. The Cr starts to resonate with Lr +
Lm until the switching MOSFETs change states. I1 can be calculated as:
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I1 =
n ⋅ Vout
= 0.95 A
2 ⋅ Lm ⋅ 2 ⋅ Fr1
The peak and RMS value of primary current can be estimated as:
 Iout ⋅ π 
2
Ipri ( pk ) = 
 + I1 = 1.99 A
 2⋅n 
2
IpriRMS =
Ipri ( pk )
2
= 1 .4 A
The RMS current is calculated by assuming pure sinusoid current waveform. So the actual primary
RMS current is higher than the calculated value.
The current in each secondary winding is very close to half-sinusoid, thus the peak and RMS current
can be estimated by:
Iout ⋅ π
= 15.7 A
2
Iout ⋅ π
Isrms =
= 7.85 A
4
Ispk =
The wire gauge of primary and secondary windings should be selected properly according to the
calculated RMS current.
Step 8: Calculate resonant capacitor voltage
The Cr waveform is shown as in Figure 13:
Figure 13: Typical resonant tank voltage and current waveforms
ILm is the magnetizing current of transformer primary, not including the current which is delivered to
the secondary load through an ideal transformer in parallel with Lm. The difference between ILr and
ILm is the output current.
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Ideal Transformer
M1
D1
Cr
Lr
ILr
Lm
M2
ILm
COUT
Iout/n
LOAD
D2
Figure 14: Lm and ideal transformer
The VCr voltage reaches its peak when Lr current is crossing zero and it is at the mid of input voltage
when Lr current reached its peak. The Cr voltage is at the maximum value when VS node is zero and
it is at the minimum value when VS node is equals to Vin. So VCrmin and VCrmax can be calculated
as:
VC r max = n ⋅ Vout + Ipri ( pk ) ×
Lr
Cr
VC r min = Vin − n ⋅ Vout − Ipri ( pk ) ×
Lr
Cr
The peak to peak voltage ripple of VCr is VCrmax-VCrmin.
VC rpk _ pk = 2n ⋅ Vout + 2 ⋅ Ipri ( pk ) ×
Lr
− Vin
Cr
It can be seen that the maximum peak-to-peak voltage happens at the maximum load and the
minimum DC input Vinmin, the switching frequency is at the minimum Fmin.
In this example:
Vcrpk _ pk = 2 × 9 × 24V + 2 × 1.99 A ×
113uH
− 350V = 368V
22nF
The resonant capacitor Cr can be selected according to the capacitance value, together with its
voltage and current rating. Polypropylene film capacitor is preferred to use for lower power loss.
Please note the polypropylene film capacitor is rated under DC voltage or 50Hz AC voltage and has
voltage derating at high frequency and high ambient temperature. The ability of withstanding high
frequency voltage is limited by thermal (power dissipation) and peak current capability. Usually the
derating starts at 85~90C ambient and is not a concern. But a capacitor with higher voltage rating
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should be chosen if the ambient temperature is higher than 85C. Below is an example of EPCOS
MKP capacitor B32621 (630Vdc/400Vac).
Figure 15: Vrms vs. frequency curve of MKP capacitor B32621 @ Ta<=90°C
4. IRS2795 Passive Components Design
Step 9: Calculate the minimum dead-time to keep ZVS switching at zero load at the maximum
input voltage
For resonant half-bridge converter, the switching frequency goes to the maximum under no load at
the maximum input voltage. Theoretically when the switching frequency is above the resonant
frequency Fr1, the operation is ZVS switching. However, above resonance is only one of the
necessary conditions for ZVS. The other condition is the equivalent parasitic capacitor of the halfbridge midpoint (junction capacitor of VS node) to be fully (dis-)charged within the dead-time period.
Figure 16 demonstrates if the dead-time is not sufficient, the turn-on of the MOSFET has hardswitching even though the converter is working under the below resonant ZVS mode.
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Figure 16: ZVS and none-ZVS waveform of region 2 operation
To keep the converter always working under ZVS condition, it is necessary to calculate the minimum
time that required to fully (dis-)charging the VS equivalent capacitor during the two switches
interleaving period (dead-time).
As the equivalent capacitor is (dis-)charged by the circulating current in the transformer primary
winding, so the worst case happens at the maximum input voltage and zero load condition where the
transformer current is at minimum. At zero load, there is no current transfer to the secondary side
and the current in the tank is just the magnetizing current of transformer. In each half-cycle, it is a
linear straight line as shown in Figure 17.
VS
Primary current
Figure 17: Transformer primary current at zero load
So the primary current under this condition can be calculated as:
I ' pri ( pk ) =
n ⋅ Vout
4 F max⋅ ( Lr + Lm)
I ' pri ( pk ) = 0.53 A
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The total equivalent junction capacitor CHB of VS node is shown in Figure 18.
M1
VB
HO
Coss_ef f_1
IRS2795
VCC
Cr
VS
Cw ell
Crss_eff
Lr
Lm
M2
LO
Coss_ef f_2
Cs
COM
Figure 18: VS Equivalent junction capacitor
C HB = 2 ⋅ Coss _ eff + Crss _ eff + CWell + Cs
It includes:
The effective Coss of the two MOSFETs (both high-side and low-side);
The Coss_eff as defined in the MOSFET datasheet is the effective capacitance of MOSFET
that gives the same charging time as a fixed capacitor while VDS is rising from 0 to 80% of
VDS. So the Coss_eff of a 500V MOSFET is defined under 0 to 400V VDS which fits to this
application.
The effective Crss of the low-side MOSFET;
The Crss of MOSFET is typically defined at VDS=25V. The Crss capacitance value reduces
as VDS voltage increasing. So the effective Crss can be chose as ½ or 1/3 of Crss.
The stray capacitance Cwell of IRS2795(1,2);
The stray capacitance of IRS2795(1,2) is the high-side well capacitance of the 600V driver.
The value of the stray capacitor is around 5pF.
The snubber capacitor Cs (if any) that is connected to the VS node.
For example, the Coss_eff of MOSFET STF13NM50N is 110pF, Crss is 5pF, and there is no
snubber capacitor to the VS node, the (dis-)charging time of VS node can be calculated as:
Coss _ eff = 110 pF , Crss _ eff = 2.5 pF , CWell = 5 pF , Cs = 0 pF
Tch =
C HB ⋅ Vin max
I ' pri ( pk )
Tch = 185ns
The dead-time calculation should also include the gate driver falling time. The MOSFET turn-off
timing diagram is shown in Figure 19, which using LO and M2 as an example. In the first time
interval t1, gate voltage discharges to a plateau voltage V’m, and both VDS voltage and ID current
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AN-1160
18
stay unchanged in t1. As long as MOSFET gate voltage reaches the miller plateau V’m, miller cap
Cgd is discharged and VDS voltage starts increasing. Due to the nonlinearity of Coss capacitor, VDS
voltage increase slowly at the beginning, then the slope becomes steeper at higher VDS voltage.
The miller plateau is the flat portion of gate driver curve. It varies with drain current. MOSFET turns
off at a relative low current level in LLC application, the miller plateau is very close to the gate turn
off threshold Vgs(th).
The timing that is interested for the dead time calculation is t1, as the charging time of the VS node
(i.e. VDS of M2) starting from t2 is already included in the Tch calculation. In t1, VDS voltage is 0V,
and MOSFET gate equals to a constant capacitor load to the IC. So the discharge time t1 can be
calculated based on the RC time constant of the gate drive loop.
t1 = − RC geq ln
V 'm
VG
Where, R = Rdown _ eff + R g + R gFET
C geq =
(Qg − Qgd − Qgs )
, Please refer to Figure 21.
Vgs − Vm
V ' m ≈ Vgs (th )
VG = Vcc , IRS2795(1,2) gate output voltage is clamped to Vcc voltage
Rdown_eff: IRS2795(1,2) gate driver effective pull down resistance (6Ω)
Rg :is the external MOSFET gate drive resistor
RgFET: MOSFET gate input resistance
Figure 19: MOSFET turn-off equivalent circuit and timing diagram
STF13NM50 gate equivalent capacitor is 2.32nF, MOSFET internal gate resistor is 5Ω, Vgs(th) is
3V. Thus if Vcc=15V, Rg=10Ω, gate discharge time t1 is:
t1 = 78.4ns
The dead-time should be longer than the sum of Tch and t1. For experience, it is recommended to
add 50ns to the calculated value. The minimum dead-time TDT is then given by:
TDT = Tch + t1 + 50ns = 313ns
For most of the design, it’s not recommended to have a dead-time that is longer than 1us, as longer
dead-time leads to higher body-diode power losses at full load. So if the calculated dead-time is too
long, go back to step 2 and choose a smaller k value.
Once the system parameters are defined, the passive components around the IRS2795(1,2) as
shown in Figure 20 can be calculated.
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19
Figure 20: IRS2795(1,2) two-pin Oscillator
CT =
TDT ⋅ 10 −3 − 40 ⋅10 −12 313 ⋅10 −12 − 40 ⋅ 10 −12
=
= 321 pF
0.85
0.85
CT capacitor should be equal or bigger than the calculated value for ZVS operation. Choose a
standard capacitor value for CT.
C T = 390 pF
Calculate the actual dead-time per the selected CT value:
t DT = (0.85CT + 40 pF ) ⋅
2V
= 371.5ns
2mA
Calculate RT per the minimum switching frequency Fmin and CT:
RT =
1
− 1kΩ
2 ⋅ F min⋅ t DT ⋅ 10 −3
RT resistor should be smaller than the calculated value to keep ZVS operation.
Calculate Rmax per the maximum switching frequency Fmax and CT, RT:
Re q =
1
− 1kΩ ,
2 ⋅ F max⋅ t DT ⋅10 −3
R max =
RT ⋅ Re q
RT − Re q
Calculate Rss with the desired soft-start frequency:
Rsseq =
1
− 1kΩ ,
2 ⋅ Fss ⋅ t DT ⋅10 − 3
Rss =
RT ⋅ Rsseq
RT − Rsseq
Calculate Css based on the desired soft-start time:
Css =
Tss
3 ⋅ Rss
In sleep mode or fault mode, RT pin is discharged to 0V. A diode Dss is put in parallel with Rss to
fast discharge Css when IC is shutdown or in fault mode. This is to make sure the system still has
soft start when IRS2795(1,2) restarts quickly. Dss can be any general purpose low voltage (10V)
and low current (100mA) diode.
The bootstrap capacitor CBS is used to hold VBS supply voltage for the high-side driver. The value of
CBS is recommended to be 100nF to 220nF. Bigger CBS capacitor causes higher charging current
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20
during startup and should be avoid. IRS2795(1,2) doesn’t have integrated bootstrap MOSFET. A
600V/1A fast recovery diode is required for bootstrap.
5. IRS2795 Power Loss Calculation
5.1 Low voltage static loss that caused by quiescent current
Pd1 = Vcc × Iqcc
where Iqcc is 2.5mA maximum per IRS2795(1,2) datasheet.
5.2 The gate driver power losses
The gate driver losses of IRS2795(1,2) are the losses when driving the two external MOSFETs M1
and M2. In ZVS mode, MOSFET VDS voltage is 0V prior to the gate turns on, so the “Miller” charge
Qgd should be subtracted from the total gate charge. Further, at ZVS operation, the MOSFET is as a
constant capacitor load to the driver. The equivalent capacitor value equals to the Cgs+Cgd at
VDS=0V condition, which can be obtained from the gate charge curve in a MOSFET data sheet. It is
indeed the slope factor of the gate charge curve where VGS is above the miller plateau voltage Vm,
as shown in Figure 21.
C geq =
(Qg − Qgd − Qgs )
Vgs − Vm
Figure 21: MOSFET gate charge curve and equivalent gate capacitance at ZVS mode
Typically the Qg, Qgd and Qgs value are specified under 10V VGS voltage, Vm is the flat portion
voltage of the gate charge curve. For example, STF13NM50 Qg=30nC, Qgd=15nC, Qgs=5nC,
Vm=5.7V, its gate equivalent capacitor in ZVS is 2.32nF.
The total gate charge in ZVS mode is proportional to the gate voltage:
Qgz = Cgeq ⋅ VG
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21
IRS2795(1,2) gate output voltage is clamped to Vcc voltage. So the total gate driver losses of both
high-side and low-side can be calculated by:
Pdr = Pdr1 + Pdr 2 = 2 ⋅ Cgeq ⋅ Vcc 2 ⋅ Fsw
The total gate driver losses are dissipated in driver IC IRS2795(1,2) and the external gate driver
resistor including the MOSFET internal gate resistor. The power loss in IRS2795(1,2) is proportional
to the resistor divider value:
Pd 2 = (
Rup _ eff
Rdown _ eff
Pdr
+
)×
Rup _ eff + Rg + R g FET Rdown _ eff + Rg + R gFET
2
Where,
Rg :is the external MOSFET gate drive resistor
Rup_eff: IRS2795(1,2) gate driver effective pull up resistance (40Ω)
Rdown_eff: IRS2795(1,2) gate driver effective pull down resistance (6Ω)
RgFET: MOSFET gate input resistance
The gate driver pull-up and pull-down resistance used for power loss calculation are given below:
Rup = 40Ω, Rdown = 6Ω . They are bigger than datasheet specification (with is defined under
20mA current) as they are the equivalent pull-up and pull-down resistance under high gate current.
5.3 The CMOS switching losses
The switching loss in low voltage logic circuit is proportional to the switching frequency and supply
voltage Vcc:
Pd 3 = Vcc × Fsw × Qcmos
For IRS2795(1,2),
Qcmos = 6nC ~ 10nC
5.4 The high voltage switching losses
The switching losses in high voltage level-shift circuit:
Pd 4 = (Vcc + Vin ) × Fsw × Qp
Vin is the input bus voltage. Qp is the charge absorbed by the level shifter. For IRS2795(1,2), Qp is
2nC under 300V to 430V bus voltage.
5.5 An example of power loss calculation
The total power loss in IRS2795(1,2) is the sum of Pd1 to Pd4.
Pd _ total = Pd1 + Pd 2 + Pd 3 + Pd 4
An example of power loss calculation with Vcc=15V, maximum switching frequency =150KHz,
MOSFETs =STF13NM50N, input bus voltage = 400V, external gate resistor = 10ohm:
Pd 1 = 37.5mW
Pdr = 157 mW , Pd 2 = 79.5mW
Pd 3 = 18mW
Pd 4 = 124.5mW
Pd _ total = 259.5mW
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22
It can be seen that the high voltage switching loss Pd4 and gate driver loss Pd2 are the main source
of total power losses. Pd4 is proportional to switching frequency and HV bus voltage. For 400V DC
BUS voltage, IRS2795(1,2) can directly drives big MOSFETs (Cgeq ≤ 4.7nF) up to 250KHz
switching frequency. It is necessary to clamp the Vcc supply voltage to 15V or lower to reduce gate
driver losses when the frequency goes to 300KHz while driving big MOSFETs. For 300KHz to
500KHz switching frequency and 400V applications, it is recommended to use external driver.
IC operation current Icc can be obtained by the total low-voltage power loss and Vcc voltage:
Icc = ( Pd1 + Pdr + Pd 3) / Vcc
6. MOSFET Selection Guide
The power MOSFET should be selected per the breakdown voltage and RDSON value. In addition, the
body diode reverse recovery characteristic also plays important role to the selection. The converter
usually has a few switching cycles that is under hard switching at the beginning of startup. This is
because the resonant capacitor and output capacitors are fully discharged. In this case, longer
reverse recovery time could cause shoot through between the two MOSFETs. Thus a MOSFET with
fast reverse recovery diode is preferred.
As the resonant half-bridge has ZVS switching, the turn-on loss is negligible. If not switching under
very high frequency (≤150Khz), the major power loss in MOSFET comes from the conduction loss.
The maximum conduction loss can be calculated as:
Pcon = Iqrms 2 × Rdson @ Tj
Where Iqrms =
Ipri ( pk )
, and [email protected] is the MOSFET on-state resistance at the system
2
maximum allowable junction temperature.
The calculation of the turn-off loss of MOSFET is complicated due to none linearity of Coss under
different VDS voltage. Thus we use the estimated formula:
Poff =
C HB × Vin 2 × Fsw
24
The total power loss in each MOSFET equals to Pcon + Poff .
IRS2795(1,2) uses the Rdson of low side MOSFET for current sensing and over current protection.
The product family provides two choices on different over current protection level: the OCP
threshold of IRS27951 is 2V and IRS27952 is 3V. Typically the IRS27951 is good for oversized
MOSFET where a lower Rdson for better efficiency and the IRS27952 is good for cost effective
MOSFET where the Rdson is bigger. A quick estimation for OCP threshold is to use 2.5 to 3 times of
the maximum drain current times the Rdson of MOSFET.
At startup, the MOSFET current could be a few times higher than the normal working current. To
prevent false triggering of over current protection when using large Rdson MOSFET, it is
recommended to extend the soft-start time to tens of milliseconds.
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23
7. Operating Waveforms and Efficiency of the Reference Design
The specification of the reference design:
Parameter
Description
Value
Vinmax
The maximum DC bus voltage
430V
Vinmin
The minimum DC bus voltage
350V
Vinnom
The nominal DC bus voltage
390V
Vout 1
The DC output voltage
24V
Iout 1
The output load current
6A
Vout 2
The DC output voltage
12V
Iout 2
The output load current
6A
Fr1
The resonant frequency
100KHz
Fmax
The maximum switching frequency
150KHz
Dmax
The maximum duty-cycle
0.5
Tss
The soft start time
30ms
Fss
The soft start frequency
300KHz
Transformer
ETD49
Design analysis result:
www.irf.com
Resonant tank components
Cr=22nF, Lr=125uH, Lm=500uH, k=4
Transformer
Np=36, N24V=4, N12V=2, n=9
IRS27951 components
CT=390pF, RT=18k, Rmax=14k, Rss=3.9k,
Css=3.3uF
AN-1160
24
Rstart2
Rstart3
270k
270k
270k
Rvcc
56
STPS30100
Dg1
1N4148
Rx1
STF13NM50N
24V
Dbs
COM
CVcc2
DSS
4
CSS
Dz
CT
1
9 TX
D2
20
Cout3
+
1
2
3
4
5
6
Cout4
+
+
Cout5
Rprl1
560
DNP
22nF/1kV
Cout2
+
Jumper
100nF
1
JP4
Cout1
W1
1mF/35V
LO
Cr
1mF/35V
VS
COM
220nF
1mF/35V
CT/SD
Rgs1
Cbs
7
5
1
2
3
4
5
6
Header 6
Dg2
Rx2
STF13NM50N
IRS2795
4.7
1N4148
LO
7
Q2
2
Rg2
19
18
STPS30100
COM2
17
14
10
610uH
COM
Rgs2
5
13
12
D3
11
20TQ040
12V
1
18V
RT
18k
8
6
1
RMAX
+
HO
1
CVcc1
CDC
2
1
VB
RT
1
RSS
JP3
3.9k
510K
1N4148
Rdisch
15k
270uF/450V
10
VCC
Lf1 3.3uH/10A
VTR
1mF/35V
3
390pF
5A/250V
2
VCC
3.3uF
0.33uF/630V
+ C4
1uF
C3
Header 2
N
U1
3
100nF
C2
33uF/35V
F1
L1
4
DNP
2
1
C1
0.1uF/275V-X2
R1
0.1uF/275V-X2
2
5 t°
Header 3
VCC
1
RNTC
VS
Rg1
DNP
1
B1
265VAC
JP1
Q1
1
4.7
1N4148
MURS160
GBU4J-BPMS-ND
DNP= Do Not Populate
1
Lf2 3.3uH/10A
JP5
1
2
3
4
5
6
Cout8
+
Cout9
Rprl2
470
+
100nF
1.5mF/25V
D4
Cout7
+
1000uF/25V
Cout6
Cs
1.5mF/25V
COM
1
2
3
4
5
6
Header 6
2.2nF/250V
1
20TQ040
Rs4
15k
Rled2
Rled1
5.6k
2.2k
Rbias2
Rs1
33k
Rbias1
2.2k
DNP
Cf1 DNP
TLP621
FB
COMP
1
Cf2
100nF
DNP=Do not Populate
Rs2
0
Rf2
47k
U3
TL431
7.1 Schematic
Rs5
3.9k
Rs3
3.74k
1
U2
COM2
25
AN-1160
D1
Rstart1
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1
Vbus
The reference board has input rectifier and filter, so it can take either DC or AC input. The DC input
range is 350V~430V, the AC input voltage range is 250Vac~300Vac.
The dummy loads at 24V and 12V output are for cross-regulation purpose.
HO
D5
Figure 22 – IRS27951 Reference Design Schematic
CT
1
1
RT
7.2 BOM
Designator
Description
Quantity
Value/Rating
Vendor
DIGIKEY
Part#
B1
Single Phase Bridge Rectifier
1
600V/4A
GBU4J-BPMS-ND
C1, C2
X2 Safety Capacitor
2
100nF/275VAC
DIGIKEY
P10524-ND
C3
Metal Poly Capacitor
1
0.33uF/630V
DIGIKEY
P12245-ND
EET-HC2W271LA
C4
Electrolytic Bulk Capacitor TS-HC
1
270uF/450V
DIGIKEY
Cbs
1206 General Purpose Ceramic SMD
1
220nF/50V
DIGIKEY
490-1776-1-ND
Cf2, Cout5, Cout9, CVcc2
1206 General Purpose Ceramic SMD
4
100nF/50V
DIGIKEY
490-1775-1-ND
CDC
Electrolytic Capacitor FM Radial
1
33uF/35V
DIGIKEY
P13475-ND
Cf1
Not Used
Cout1, Cout2, Cout3, Cout4
Aluminium Electrolytic Capacitor 105°C
4
1000uF/35V
DIGIKEY
565-1581-ND
Cout6, Cout7
Aluminium Electrolytic Capacitor 105°C
2
1500uF/25V
DIGIKEY
565-1557-ND
Cout8
Aluminium Electrolytic Capacitor 105°C
1
1000uF/ 25V
DIGIKEY
565-1555-ND
Cr
Polypropylene Capacitor High Ripple
1
22nF/1kV
DIGIKEY
495-3552-ND
Cs
250VAC Y1 Safety Ceramic Disc Capacitor
1
2.2nF/250V
DIGIKEY
445-2411-ND
CSS
1206 General Purpose Ceramic SMD
1
3.3uF/16V
DIGIKEY
445-4038-1-ND
CVcc1
1206 General Purpose Ceramic SMD
1
1uF/25V
DIGIKEY
445-1592-1-ND
CT
1206 General Purpose Ceramic SMD ±5%
1
390pF/50V
DIGIKEY
478-1487-1-ND
D1, D2
TO220AB Power Schottky Rectifier
2
100V/30A
DIGIKEY
STPS30100CT
D3, D4
TO220AC Power Schottky Rectifier
2
40V/20A
DIGIKEY
20TQ040PBF-ND
D5, Dg1, Dg2, DSS
Fast Recovery Diode DO-35
4
75V/0.3A
DIGIKEY
1N4148DICT-ND
Dbs
Fast Recttifier diode SMB
1
600V/1A
DIGIKEY
MURS160-FDICT-ND
Dz
Zener Diode SMD
1
18V/0.5W
DIGIKEY
FLZ18VCCT-ND
F1
FUSE IEC FA LBC 5x20
1
250V/5A
DIGIKEY
F2395-ND
JP1
CONN HEADER 3POS 0.156 VERT TIN
1
DIGIKEY
WM4621-ND
JP3
CONN HEADER 2POS 0.1 VERT TIN
1
DIGIKEY
WM4200-ND
JP4, JP5
CONN HEADER 6POS 0.156 VERT TIN
2
L1
EMI Common Mode Choke
1
DIGIKEY
WM4624-ND
16mH/2.6A
DIGIKEY
237-1233-ND
Lf1, Lf2
PCV Series Drum Core Inductor 10mm
2
4.7uH/12A
COILCRAFT
PCV-0-472-10L
Q1, Q2
TO-220FP N-Channel Power MOSFET
2
500V/12A
DIGIKEY
STF13NM50N
R1, Rbias2, Rgs1, Rgs2
Not Used
Rbias1, Rled1
1206 SMD Film RED 1/4W 1%
2
2.2k
DIGIKEY
RHM2.20kFCT-ND
Rdisch
Metal Film Power Resistor 2W 5%
1
510k
DIGIKEY
BC510KW-2CT-ND
Rf2
1206 SMD Film RED 1/4W 1%
1
47k
DIGIKEY
RHM47.0kFCT-ND
Rg1, Rg2
1206 SMD Film RED 1/4W 5%
2
10
DIGIKEY
RHM10ERCT-ND
Rled2
1206 SMD Film RED 1/4W 1%
1
5.6k
DIGIKEY
RHM5.60kFCT-ND
RMAX
1206 SMD Film RED 1/4W 1%
1
15k
DIGIKEY
RHM15.0kFCT-ND
RNTC
Inrush Current Limiter
1
5
DIGIKEY
495-2093-ND
Rprl1
Metal Film Power Resistor 2W 5%
1
560
DIGIKEY
PPC560W-2CT-ND
Rprl2
Metal Film Power Resistor 2W 5%
1
470
DIGIKEY
PPC470W-2CT-ND
Rs1
1206 SMD Film RED 1/4W 1%
1
33k
DIGIKEY
RHM33.0KFCT-ND
Rs2
1206 SMD Film RED 1/4W 1%
1
0
DIGIKEY
P0.0ECT-ND
Rs3
1206 SMD Film RED 1/4W 1%
1
3.74k
DIGIKEY
RHM3.74KFCT-ND
Rs4
1206 SMD Film RED 1/4W 1%
1
15k
DIGIKEY
RHM15.0KFCT-ND
Rs5, RSS
1206 SMD Film RED 1/4W 1%
2
3.9k
DIGIKEY
RHM3.90KFCT-ND
Rstart1, Rstart2, Rstart3
1206 SMD Film RED 1/4W 1%
3
270k
DIGIKEY
RHM270KFCT-ND
RT
1206 SMD Film RED 1/4W 1%
1
18k
DIGIKEY
RHM18.0KFCT-ND
Rvcc
1206 SMD Film RED 1/4W 5%
1
56
DIGIKEY
RHM56ERCT-ND
Rx1, Rx2
1206 SMD Film RED 1/4W 5%
2
4.7
DIGIKEY
RHM4.7ERCT-ND
TX
Resonant Power Transformer
1
ETD49
PRECISION INC
019-4974-00R
U1
IRS27951 Control IC
1
IR
IRS27951S
U2
Photocoupler TRANS-OUT 4-DIP
1
TLP621
DIGIKEY
TLP621FT-ND
U3
Programmable Voltage Regulator SOT23-3
1
TL431
DIGIKEY
296-17328-1-ND
W1
Jumper for Primary Current Sensing Loop
1
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AN-1160
AWG22, multi strands
26
7.3 Typical Operating Waveforms
Figure 23 – 400Vdc input, 0W load startup
Figure 24 - 400Vdc input, 220W load startup
Figure 25 - 400Vdc input, 220W load operation
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AN-1160
27
Figure 26 – 350Vdc input, 220W load operation
Figure 27 – 420Vdc input, 220W load operation
Figure 28 – 420Vdc input, 0W load operation
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AN-1160
28
7.4 Short circuit protection
Figure 29 –260Vac input, short 12V, IC latched shut down
7.5 Efficiency
The average efficiency of the board at 25%, 50%, 75% and 100% load is 92% at 270Vac input:
24Vout
24.176
24.2
24.22
24.24
23.517
24.814
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24V Iout(A)
1.5
3
4.5
6
6
0
12Vout
11.97
11.92
11.9
11.86
12.26
11.63
12V Iout(A)
1.5
3
4.5
6
0
6
AN-1160
270Vac
400Vdc
Pout(W) Efficiency Efficiency
54.2
90.8%
91.0%
108.4
92.6%
92.6%
162.5
92.4%
92.9%
216.6
92.2%
92.7%
141.1
92.8%
69.8
89.5%
29
Efficieny vs. Output Power
93.5%
Efficiency
93.0%
92.5%
92.0%
400Vdc input
91.5%
270Vac input
91.0%
90.5%
50.0
70.0
90.0 110.0 130.0 150.0 170.0 190.0 210.0
Output Power (W)
Figure 30 – Efficiency Plot
8. Layout guidelines and example
Ground Plane:
In order to minimize noise coupling, the ground plane should not be placed under or near the high
voltage floating side.
Gate Drive Loops:
Current loops behave like antennas and are able to receive and transmit EM noise. In order to
reduce the EM coupling and improve the power switch turn on/off performance, the gate drive loops
must be reduced as much as possible. For the low-side driver, the return of the drive loop must be
directly connected to the COM pin of the IC and separate with signal ground (power ground and
signal ground have star connection at COM pin).
Supply Capacitor:
It is recommended to place a bypass capacitor (CVCC) between the VCC and COM pins. A 1µF
ceramic capacitor is suitable for most applications. This component should be placed as close as
possible to the pins in order to reduce parasitic elements.
CBS Capacitor:
The CBS capacitor should be placed as close as possible to the VB and VS pins.
Routing and Placement:
1) The 8-pin IC has only one COM pin for both signal return and power return, so it is strongly
recommended to route the signal ground and power ground separately with a star connection at the
COM pin.
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AN-1160
30
2) The RT pin provides a current reference for the internal oscillator and needs to be kept as clean
as possible to avoid frequency jittering or duty-cycle mismatch between high-side and low-side. The
components connected to this pin must keep away from the high frequency switching loop such as
the gate driver loop and the VS node. The PCB traces connected to RT pin also need to be kept
away from any switching node.
3) Connect CT capacitor directly to COM pin, don’t share the return with any other signal ground.
Layout examples
VS node
Signal components are kept
away from switching nodes
Supply bypass capacitors
are close to IC pins.
Star connection at COM pin
Figure 31: Single layer board example
9. Appendix
Symbols list
Lr: primary resonant inductance. It is the primary leakage inductance of transformer
when there is no external added resonant inductor.
Lm: transformer primary magnetic inductance. It is the measured transformer primary
inductance minus the leakage inductance.
Cr: primary resonant capacitor and DC blocking capacitor
fr1: the resonant frequency between Lr and Cr
Rac: Equivalent AC resistance for resonant tank AC analysis
RDSon: MOSFET channel ON resistance
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AN-1160
31
fmax: converter maximum operating switching frequency
fmin: converter minimum operating switching frequency
Qg: MOSFET total gate charge
Qgd: MOSFET gate to drain (Miller) charge
Qgs: MOSFET gate to source charge
IQCC: IRS2795(1,2) quiescent current
Rg: MOSFET gate drive resistance external to IRS2795(1,2)
Rup: IRS2795(1,2) gate driver pull up resistance
Rdown: IRS2795(1,2) gate driver pull down resistance
RgFET: MOSFET gate input resistance
PICmax: IRS2795(1,2) maximum power dissipation
VCC: Supply voltage on IRS2795(1,2) Vcc pin
ICC: IRS2795(1,2) IC supply current
References
[1] IRS2795(1,2) datasheet
www.irf.com
AN-1160
32
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