A3930 and A3931 Datasheet

A3930 and A3931
Automotive 3-Phase BLDC Controller and MOSFET Driver
Features and Benefits
Description
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The A3930 and A3931 are 3-phase brushless DC (BLDC) motor
controllers for use with N-channel external power MOSFETs.
They incorporate much of the circuitry required to design a
cost effective three-phase motor drive system, and have been
specifically designed for automotive applications.
A key automotive requirement is functionality over a wide
input supply range. A unique charge pump regulator provides
adequate (>10 V) gate drive for battery voltages down to 7 V,
and allows the device to operate with a reduced gate drive at
battery voltages down to 5.5 V. Power dissipation in the charge
pump is minimized by switching from a voltage doubling mode
at low supply voltage to a dropout mode at the nominal running
voltage of 14 V.
A bootstrap capacitor is used to provide the above-battery
supply voltage required for N-channel MOSFETs. An internal
charge pump for the high-side drive allows for DC (100% duty
cycle) operation.
Internal fixed-frequency PWM current control circuitry can
be used to regulate the maximum load current. The peak
load current limit is set by the selection of an input reference
voltage and external sensing resistor. The PWM frequency is
set by a user-selected external RC timing network. For added
flexibility, the PWM input can be used to provide speed and
High current 3-phase gate drive for N-channel MOSFETs
Synchronous rectification
Cross-conduction protection
Charge pump and top-off charge pump for 100% PWM
Integrated commutation decoder logic
Operation over 5.5 to 50 V supply voltage range
Extensive diagnostics output
Provides +5 V Hall sensor power
Low-current sleep mode
Package: 48 Lead LQFP with exposed
thermal pad (suffix JP)
Approximate Scale 1:1
Continued on the next page…
Typical Application
3930-DS Rev. 4
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Description (continued)
torque control, allowing the internal current control circuit to set
the maximum current limit.
Efficiency is enhanced by using synchronous rectification. The
power FETs are protected from shoot-through by integrated
crossover control with dead time. The dead time can be set by a
single external resistor.
The A3930 and A3931 only differ in their response to the all-zero
combination on the Hall inputs. In this state, the A3930 indicates
a logic fault, but the A3931 pre-positions the motor in an unstable
starting position suitable for start-up algorithms in microprocessordriven “sensor-less” control systems.
Both devices are supplied in a 48-pin LQFP with exposed thermal
pad. This is a small footprint (81 mm2) power package, that is lead
(Pb) free, with 100% matte tin leadframe plating.
Selection Guide
Part Number
Option
Packing
A3930KJPTR-T
Hall short detection
1500 pieces/reel
A3931KJPTR-T
Pre-positioning
1500 pieces/reel
Terminals
48
Package
LQFP surface mount
Absolute Maximum Ratings
Parameter
Load Supply Voltage
Logic Input/Output Voltage
Output Voltage Range
Symbol
VBB
VRESET
Conditions
VBB pin
Rating
Units
–0.3 to 50
V
RESET pin input
–0.3 to 6
V
Remaining logic pins
–0.3 to 7
V
VGHx
GHA, GHB, and GHC pins
VSx to VSx+ 15
V
VGLx
GLA, GLB, and GLC pins
–5 to 16
V
VCx
CA, CB, and CC pins
VSx+ 15
V
VSx
SA, SB, and SC pins
–5 to 55
V
CSP, CSN, and LSS pins
–4 to 6.5
V
CSO, VDSTH pins
–0.3 to 6.5
V
VDRAIN pin
–0.3 to 55
V
Operating Temperature Range (K)
TA
–40 to 150
°C
Junction Temperature
TJ
150
°C
Transient Junction Temperature
TtJ
175
°C
Storage Temperature Range
TS
–55 to 150
°C
AEC-Q100-002, all pins except CP1
2000
V
AEC-Q100-002, pin CP1
1000
V
AEC-Q100-011, all pins
1050
V
ESD Rating, Human Body Model
ESD Rating, Charged Device Model
Overtemperature event not exceeding 1 s, lifetime duration not exceeding 10 hr; guaranteed by design
characterization
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Functional Block Diagram
VBAT+
CP
CP2
VBB
QV5
V5BD
P
Charge
Pump
Regulator
+5V Ref
V5
CP1
VREG
CREG
CV5
VDRAIN
MODE
Phase A of three phases
COAST
Charge
Pump
V5
BRAKE
CA
CBOOTA
Boostrap
Monitor
RESET
High-Side
Drive
DIR
H1
H2
H3
GHA
&C
RGHA
&B
Control
Logic
H1
SA
H2
&A
H3
VREG
Low-Side
Drive
RDEAD
GLA
RGLA
LSS
PWM
TACHO
R
DIRO
TEST
Blanking
Q
S
ESF
Diagnostics and
Protection
–UVLO
–TSD
–Short to Supply
–Short to Ground
–Shorted Winding
–Low Load
FF1
FF2
OSC
CSP
RSENSE
CSN
Pad
VDSTH
RC
RT
REF
CSOUT
AGND
P
CT
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
3
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
ELECTRICAL CHARACTERISTICS at TJ = –40°C to 150°C, VBB = 7 to 45 V, unless otherwise noted1
Characteristics
Supply and Reference
VBB Functional Operating Range6
Symbol
Min.
Typ.
Max.
Units
5.5
–
50
V
–
–
–
12.10
2 × VBB
–2.7
9
0.4
1.5
11
–
–
13
14
10
5
13.75
mA
μA
mA
V
–
–
V
10
0.7
2.2
–
1.0
2.8
V
V
V
6
10
20
Ω
IDBOOT
ITOCPM
250
–
500
200
750
–
mA
μA
RGSH
250
–
–
kΩ
V5
VBEEXT
I5BD
4.75
–
–
5
–
–
5.25
1
–2
V
V
mA
–
–
3
4.6
1
1.5
–
–
60
40
4
5.6
1.5
2.3
–500
850
–
–
5
6.6
2
3
–
–
ns
ns
Ω
Ω
Ω
Ω
mA
mA
VCx – 0.2
–
–
V
VREG –
0.2
–
–
V
VBB
V5 Quiescent Current
IBBQ
IBBS
IV5Q
VREG Output Voltage
VREG
VBB Quiescent Current
Bootstrap Diode Forward Voltage
Bootstrap Diode Resistance
Bootstrap Diode Current Limit
Top-off Charge Pump Current Limit
High-Side Gate Drive Static Load
Resistance
V5 Output Voltage
VBE of External Transistor QV5
V5BD Base Drive Capability for QV52
Gate Output Drive
Turn-On Rise Time
Turn-Off Fall Time
VfBOOT
rD
tr
tf
Pull-Up On Resistance
RDS(on)UP
Pull-Down On Resistance
RDS(on)DN
Short-Circuit Current – Source2
Short-Circuit Current – Sink
ISC(source)
ISC(sink)
GHx Output Voltage
VGHx
GLx Output Voltage
VGLx
Test Conditions
Function correct, parameters not
guaranteed
RESET = High, outputs = Low
RESET = Low, sleep mode
RESET = High, outputs = Low
VBB ≥ 7.4 V, IREG = 0 to 15 mA
6 V < VBB < 7.4 V
IREG = 0 to 15 mA
5.5 V < VBB < 6 V, IREG < 10 mA
ID = 10 mA
ID = 100 mA
rD(100 mA) = (VfBOOT(150 mA) –
VfBOOT(50 mA)) / 100 mA
CLOAD = 3300 pF, 20% to 80% points
CLOAD = 3300 pF, 80% to 20% points
TJ = 25C, IGHx = –150 mA
TJ = 150C, IGHx = –150 mA
TJ = 25C, IGLx = 150 mA
TJ = 150C, IGLx = 150 mA
TJ = 25C
TJ = 25C
tw < 10 μs
Bootstrap capacitor fully charged
Continued on the next page...
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
ELECTRICAL CHARACTERISTICS (continued) at TJ = –40°C to 150°C, VBB = 7 to 45 V, unless otherwise noted1
Characteristics
Turn-Off Propagation Delay
Dead Time (turn-off to turn-on delay)
Logic Inputs and Outputs
FFx Fault Output (Open Drain)
FFx Fault Output Leakage Current2
TACHO and DIRO Output High Voltage
TACHO and DIRO Output Low Voltage
Input Low Voltage
Input High Voltage (Except RESET)
RESET Input High Voltage
Input Hysteresis
Input Current (Except H1, H2, H3, and
RESET)2
RESET Input Pull-Down Resistor
Hx Input Pull-Up Resistor
Current Sense Differential Amplifier
Input Bias Current2
Input Offset Current2
CSP Input Resistance
CSN Input Resistance
Symbol
tp(off)
tDEAD
VOL
IOH
VOH
VOL
VIL
VIH
VIHR
VIHys
Test Conditions
From Hall input change to unloaded
gate output change
From other control input change to
unloaded gate output change
RDEAD = 5 kΩ
RDEAD = 50 kΩ
RDEAD = 400 kΩ
RDEAD = tied to V5
IOL = 1 mA, fault asserted
VO = 5 V, fault not asserted
IOH = –1 mA
IOL = 1 mA
IIN
RPD
RPU
VIN = 5 V
VIN = 0 V
IIBS
IIOS
RCSP
RCSN
Low Output Voltage Error
Verr
DC Common Mode Gain
Source Resistance
ACMdc
rCSOUT
CSP = CSN = 0 V
CSP = CSN = 0 V
Measured with respect to AGND
Measured with respect to AGND
VID = CSP – CSN, –1.3 V < CSP < 4 V,
–1.3 V < CSN < 4 V
CSP = CSN = 0 V
CSP = CSN = 0 V
CSP = CSN
40 mV < VID < 175 mV, VCM in range
0 < VID < 40 mV,
VCSOUT = (19 × VID) + VOOS + Verr
CSP = CSN = 200 mV
VCSOUT = 2 V, –2 mA < ICSOUT < 0.5 mA
Output Dynamic Range
VCSOUT
–100 μA < ICSOUT < 100 μA
Differential Input Voltage
Output Offset Voltage
Output Offset Voltage Drift
Input Common Mode Range
Differential Input Voltage Gain
Output Current – Sink
Output Current – Source2
VID
VOOS
VOOS(∆t)
VCM
AV
ICSOUT(sink)
ICSOUT(source)
VCSOUT= 2 V ±5%
VCSOUT= 2 V ±5%
Min.
Typ.
Max.
Units
300
500
700
ns
–
150
200
ns
–
835
–
–
180
960
3.3
6
–
1090
–
–
ns
ns
μs
μs
–
–1
V5 – 1 V
–
–
2
2.2
300
–
–
–
–
–
–
–
500
0.4
1
–
0.4
0.8
–
–
–
V
μA
V
V
V
V
V
mV
–1
–
1
μA
–
–
50
100
–
–
kΩ
kΩ
–250
–10
–
–
–200
–
80
4
–150
10
–
–
μA
μA
kΩ
kΩ
0
–
200
mV
100
–
–1.5
18.5
320
100
–
19
550
–
4
19.5
mV
μV/°C
V
V/V
–20
–
20
mV
–
–
–30
80
dB
Ω
0.1
–
–
–
1
–19
–
–
V5
– 0.2
–
–
V
mA
mA
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
ELECTRICAL CHARACTERISTICS (continued) at TJ = –40°C to 150°C, VBB = 7 to 45 V, unless otherwise noted1
Characteristics
Supply Rejection
Small Signal 3 dB Bandwidth Frequency
Symbol
PSRR
f3dB
Settling Time
tSETTLE
AC Common Mode Gain
ACMac
Common Mode Recovery Time
tCMrec
Output Slew Rate
Input Overload Recovery Time
Current Limit
Reference Comparator Input Offset Voltage
Reference Input Clamp Voltage
Comparator Blank Time
REF Input Bias Current
RC Charge Current2
RC HIgh Voltage Threshold
RC Low Voltage Threshold
Protection
VREG Undervoltage Lockout
SR
tIDREC
VIOC
VREFC
tRC
IIBREF
IRC
VRCH
VRCL
VREGUV
Bootstrap Capacitor Undervoltage Lockout
VBOOTUV
Bootstrap Capacitor Undervoltage Lockout
VBOOTUVHys
Hysteresis
V5 Undervoltage Lockout
V5UV
V5 Undervoltage Lockout Hysteresis
V5UVHys
VDSTH Input Voltage Range
VDSTH
VDSTH Input Current2
IDSTH
VDRAIN Input Voltage Range
VDRAIN
Short-to-Ground Threshold Offset3,5
VSTGO
Short-to-Battery Threshold Offset4,5
VSTBO
Low Load Current Detection Voltage
Overtemperature Flag
Overtemperature Flag Hysteresis
VCSOL
TJF
TJFHys
Test Conditions
CSP = CSN = AGND, 0 to 300 kHz
VID=10 mVPP
To within 10%, VCSOUT = 1 VPP square
wave
VICR= 250 mVPP, 0 to 1 MHz
To within 100 mV, VICR= +4.1 to 0 V
step
10% to 90% points, VID= 0 to 175 mV
step
To within 10%, VID=250 mV to 0 V step
External pull-up to 5 V RREF = 200 kΩ
RT= 56 kΩ, CT = 470 pF
VREG rising
VREG falling
VBOOT falling, VCx – VSx
VBOOTUVHys = %VREG
V5 falling
VDSTH > 1 V
VDSTH ≤ 1 V, 7 V ≤ VDRAIN ≤ 30 V
VDSTH > 1 V
VDSTH ≤ 1 V, 7 V ≤ VDRAIN ≤ 30 V
Temperature increasing
Recovery = TJF – TJFHys
Min.
–
–
Typ.
45
1.6
Max.
–
–
Units
dB
MHz
–
400
–
ns
–
–28
–
dB
–
1
–
μs
–
20
–
V/μs
–
500
–
ns
–15
3.75
–
–
–1.15
1.7
0.6
0
4
650
0
–1
1.9
0.7
15
4.2
–
–
–0.85
2.1
0.8
mV
V
ns
μA
mA
V
V
7.5
6.75
59
8
7.25
–
8.5
7.75
69
V
V
%
–
13
–
%
3.4
300
0.3
–1
7
–
–150
–
–150
–
–
–
3.65
400
–
–
VBB
±300
–
±300
–
500
170
15
4.0
500
4
1
45
–
150
–
150
–
–
–
V
mV
V
μA
V
mV
mV
mV
mV
mV
ºC
ºC
1Parameters
are tested at 135C. Values at 150C are guaranteed by design or correlation.
input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin.
3High side on. As V
SX decreases, fault occurs if VBAT – VSX > VSTG .
4Low side on. As V
SX increases, fault occurs if VSX – VLSS > VSTB .
5V
STG threshold is VDTSTH + VSTGO. VSTB threshold is VDTSTH + VSTBO .
6Function is correct but parameters not guaranteed above or below general limits (7 to 45 V).
2For
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
6
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Thermal Characteristics
THERMAL CHARACTERISTICS may require derating at maximum conditions, see Applications Information section
Characteristic
Symbol
Test Conditions*
4-layer PCB, based on JEDEC standard
Package Thermal Resistance
RθJA
Die-to-Exposed Pad Thermal Resistance
RθJP
2-layer PCB, with 3
by thermal vias
in.2
of copper area each side connected
Value Units
23
ºC/W
44
ºC/W
2
ºC/W
*Additional thermal information available on Allegro Web site.
ALLOWABLE PACKAGE POWER DISSIPATION IN WATTS
Power Dissipation versus Ambient Temperature
6.0
5.0
4.0
R
3.0
QJ
A
=
23
°C
/W
2.0
R
QJA
=4
4°C
/W
1.0
0
25
50
75
100
125
AMBIENT TEMPERATURE IN °C
150
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
7
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Functional Description
Basic Operation
The A3930 and A3931 devices provide commutation and current
control for 3-phase brushless DC (BLDC) motors with integrated
Hall-effect (HE) sensors. The motor current is provided by an
external 3-phase N-channel MOSFET bridge which is controlled
by the A3930/A3931, using fixed-frequency pulse width modulation (PWM). The use of PWM with N-channel MOSFETs
provides the most cost-effective solution for a high-efficiency
motor drive.
The A3930/A3931 provides all the necessary circuits to ensure
that the gate-source voltage of both high-side and low-side external MOSFETs are above 10 V, at supply voltages down to 7 V.
For extreme battery voltage drop conditions, functional operation
is guaranteed down to 5.5 V but with a reduced gate drive. The
A3930/A3931 also decodes the commutation sequence from three
HE sensors spaced at 120° in the electrical cycle, and ensure no
cross-conduction (shoot through) in the external bridge. Individual pins provide direction, brake and coast control.
Motor current can be sensed by a low-value sense resistor,
RSENSE, in the ground connection to the bridge, amplified and
compared to a reference value. The A3930/A3931 then limits the
bridge current on a cycle-by-cycle basis. Bridge current can also
be controlled using an external PWM signal with the internal current control either disabled or used to set the absolute maximum
motor current. Specific functions are described more fully in the
following sections.
Power Supplies
Only one power connection is required because all internal
circuits are powered by integrated regulators. The main power
supply should be connected to VBB through a reverse battery
protection circuit.
V5 and V5BD A 5 V supply for external pull-up and bias currents is provided by an integrated 5 V regulator controller and an
external NPN transistor, QV5. The A3930/A3931 provides the
base drive current on the V5BD pin, and the 5 V reference on the
V5 pin. This regulator is also used by the internal logic circuits
and must always be decoupled by at least a 200 nF capacitor,
CV5, between the V5 pin and AGND. For stability, a 100 nF
capacitor, C5BD, also should be connected between V5BD and
AGND. If an external 5 V supply is not required, the V5BD pin
and the V5 pin should be connected together.
CP1, CP2, and VREG The gate drive outputs are powered by
an internal charge pump, which requires a pump capacitor, typically 470 nF, CP, connected between the CP1 and CP2 pins. The
output from the charge pump, 13 V nominal, is used to power
each of the three high- and low-side driver pairs and is also
available on the VREG pin. A sufficiently large storage capacitor, CREG, must be connected to this pin to provide the transient charging current to the low-side drivers. The charge pump
also provides the charging current for the bootstrap capacitors,
CBOOTx.
An additional “top-off” charge pump is provided for each highside drive which allows the high-side drive to maintain the gate
voltage on the external FET indefinitely, ensuring so-called 100%
PWM if required. This is a low-current trickle charge pump
(< 100 μA typical), and is only operated after a high-side driver
has been signaled to turn on. There is a small amount of bias
current (< 20 μA) drawn from the Cx pin to operate the floating
high-side circuit, and the charge pump simply provides enough
drive to ensure that the bootstrap voltage, and hence the gate voltage, will not droop due to this bias current. The charge required
for initial turn-on of the high-side gate is always supplied by
bootstrap capacitor charge cycles.
Hall Effect Sensor Inputs
H1, H2, and H3 Hall-effect sensor inputs are configured for
motors with 120° electrically-spaced HE sensors, but may be
used for 60° electrical spacing with an external inverter. HE sensors usually require an additional pull-up resistor to be connected
between the sensor output and 5 V. This 5 V can be provided by
the integrated 5 V regulator. HE inputs have a hysteresis of typically 500 mV to reduce the effects of switching noise on the HE
connections to the motor. These inputs are also filtered to further
reduce the effects of switching noise. The HE inputs are pulledup to 5 V inside the A3930/A3931 through a high value (100
kΩ typical) resistor in series with a diode. This internal pull-up
makes the HE input appear high if the Hall sensor signal is missing, allowing detection of an HE input logic fault.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
8
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
In order to provide a known start-up position for the motor, an
optional prepositioning function is available in the A3931. When
the Hall inputs are all driven low (H1 = H2 = H3 = 0), the power
FETs in the A phase source current from the supply, and those in
both the B and C phases sink current. This forces the motor to
move to an unstable position midway between two detent points
and allows any start-up algorithm to ensure correct initial direction of rotation. Note that this is only available in the A3931. The
A3930 will indicate a logic fault when all Hall inputs are driven
low. The commutation truth table for these inputs is shown in
table 2. The inputs can also be driven directly from a microcontroller or similar external circuit.
Gate Drive
The A3930/A3931 is designed to drive external N-channel power
MOSFETs. They supply the large transient currents necessary to
quickly charge and discharge the gate capacitance of the external
FETs in order to reduce dissipation in the external FETs during
switching. The charge and discharge rate can be controlled using
external resistors in series with the connections to the gate of the
FETs.
RDEAD Cross-conduction is prevented by the gate drive circuits
which introduce a dead time, tDEAD, between switching one FET
off and the complementary FET on. The dead time is derived
from the value of a resistor, RDEAD, connected between the
RDEAD pin and AGND. If RDEAD is connected to V5, tDEAD
defaults to 6 μs typical.
GLA, GLB, and GLC Low-side gate drive outputs for external
NMOS drivers. External series-gate resistors, RGATE, (as close
as possible to the NMOS gate) can be used to control the slew
rate seen at the power-driver gate, thereby controlling the di/dt
and dv/dt of the Sx outputs. Referring to table 2, GLx = 1 (high)
means that the upper half (PMOS) of the driver is turned on, and
that its drain will source current to the gate of the low-side FET
in the external motor-driving bridge. GLx = 0 (low) means that
the lower half (NMOS) of the driver is turned on, and that its
drain will sink current from the corresponding external FET gate
circuit to the LSS pin.
SA, SB, and SC Directly connected to the motor, these
terminals sense the voltages switched across the load. These
terminals are also connected to the negative side of the bootstrap
capacitors and are the negative supply connections for the
floating high-side drivers. The discharge current from the highside FET gate capacitance flows through these connections,
which should have low-impedance traces to the FET bridge.
GHA, GHB, and GHC High-side gate drive outputs for external NMOS drivers. External series-gate resistors, RGATE, can
be used to control the slew rate seen at the power-driver gate,
thereby controlling the di/dt and dv/dt of the Sx inputs. Referring
to table 2, GHx = 1 (high) means that the upper half (PMOS) of
the driver is turned on, and that its drain will source current to the
gate of the high-side FET in the external motor-driving bridge.
GHx = 0 (low) means that the lower half (NMOS) of the driver
is turned on, and that its drain will sink current from the corresponding external FET gate circuit to the respective Sx pin.
CA, CB, and CC High-side connections for the bootstrap
capacitors and positive supply for high-side gate drivers. The
bootstrap capacitors, CBOOTx, are charged to approximately
VREG when the corresponding Sx terminal is low. When the Sx
output swings high, the voltage on the Cx pin rises with the output to provide the boosted gate voltage needed for the high-side
N-channel power MOSFETs.
VDRAIN High impedance sense input (Kelvin connection) to
the top of the external FET bridge. This input allows accurate
measurement of the voltage at the drain of the high-side FETs and
should be connected directly to the bridge, close to the drain connections of the high-side FETs, with an independent trace.
LSS Low-side return path for discharge of the gate capacitors.
It is connected to the common sources of the low-side external
FETs through an independent low-impedance trace.
Logic Control Inputs
Additional logic-level inputs are provided to enable specific
features described below. These logic inputs all have a nominal
hysteresis of 500 mV to improve noise performance.
RESET Allows minimum current consumption from the VBB
supply. When RESET is low, all internal circuitry is disabled
including the V5 output. When coming out of sleep state, the
protection logic ensures that the gate drive outputs are off until
the charge pump reaches proper operating conditions. The charge
pump stabilizes in approximately 3 ms under nominal conditions.
RESET has an internal pull-down resistor, 50 kΩ typical.
However, to allow the A3930/A3931 to start-up without the
need for an external logic input, the RESET pin can be pulled
to the battery voltage with an external pull-up resistor. Because
RESET also has an internal clamp diode, 6 V typical, to limit the
input current, the value of the external pull-up resistor should be
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9
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
greater than 20 kΩ. The upper limit for the resistor must be low
enough to ensure that the input voltage reaches the input high
threshold, VINR.
COAST An active-low input which turns all FETs off without
disabling the supplies or control logic. This allows the external
FETs and the motor to be protected in case of a short circuit.
MODE Sets the current-decay method. Referring to table 3, when
in slow-decay mode, MODE = 1, only the high-side MOSFET
is switched off during a PWM-off cycle. In the fast-decay mode,
MODE = 0, the device switches both the high-side and low-side
MOSFETs.
Slow decay allows a lower ripple current in the motor at the
PWM frequency, but reduces the dynamic response of the current control. It is suitable for motors which run at a more-or-less
constant speed. Fast decay provides improved current-control
dynamic response, but increases the motor current ripple. It is
suitable for motors used in start-stop and positioning applications.
DIR Determines the direction of motor torque output, as shown in
table 2. For an unloaded, low-inertia motor, this will also usually
be the direction of mechanical rotation. With a motor that has a
high inertial load, the DIR input can be used to apply a controlled
breaking torque, when fast decay is used (MODE = 0).
BRAKE An active-low input that provides a braking function.
When BRAKE = 0 (see table 3), all the low-side FETs are turned
on and the high-side FETs are turned off. This effectively shortcircuits the back EMF in the windings, and brakes the motor.
The braking torque applied depends on the speed. RESET = 0 or
COAST = 0 overrides BRAKE and coasts the motor. Note that
when BRAKE is used to dynamically brake the motor, the windings are shorted with no control over the winding current.
RSENSE, connected between CSP and CSN, the output of the
sense amplifier will be approximately:
VCSOUT ≈ (ILOAD × AV × RSENSE) + VOOS ,
where VOOS is the output offset voltage (the voltage at zero load
current), and AV is the differential voltage gain of the sense
amplifier, 19 typical.
Internal Current Control: REF A fixed reference voltage
can be applied to provide a maximum current limit. A variable
reference voltage will provide a variable torque control. The
output voltage of the current sense differential amplifier, VCSOUT ,
is compared to the reference voltage available on the REF
pin. When the outputs of the MOSFETs are turned on, current
increases in the motor winding until it reaches a trip point value,
ITRIP, given by:
ITRIP = (VREF – VOOS) / (RSENSE × AV) .
At the trip point, the sense comparator resets the source enable
latch, turning off the source driver. At this point, load inductance
causes the current to recirculate until the start of the next PWM
period.
The current path during recirculation is determined by the
configuration of the MODE pin. Torque control can therefore be
implemented by varying the voltage on the REF pin, provided
that the PWM input remains high. If direct control of the torque
or current by PWM input is desired, a voltage can be applied to
the REF pin to set an absolute maximum current limit. The REF
input is internally limited to 4 V, which allows the use of a simple
pull-up resistor to V5, RREF, to set the maximum reference
voltage, avoiding the need for an externally generated reference
voltage. RREF should have a value between 20 kΩ and 200 kΩ.
ESF The state of the enable stop on fault (ESF) pin determines
the action taken when a short is detected. See the Diagnostics
section for details.
Internal PWM Frequency The internal oscillator frequency,
fOSC, is determined by an external resistor, RT, and capacitor, CT,
connected in parallel from the RC pin to AGND. The frequency
is approximately:
TEST Test is for Allegro production use and must be connected
to AGND.
fOSC ≈ 1 / (RTCT + tBLANK + tDEAD) .
Current Regulation
Load current can be regulated by an internal fixed frequency
PWM control circuit or by external input on the PWM pin.
Current Sense Amplifier: CSP, CSN, and CSOUT A differential current sense amplifier with a gain, AV, of 19 typical, is
provided to allow the use of low-value sense resistors or current
shunts as the current sensing elements. Because the output of
this sense amplifier is available at CSOUT, it can be used for
either internal or external current sensing. With the sense resistor,
where fOSC in the range 20 to 50 kHz.
PWM Input Can be used to control the motor torque by an external control circuit signal on the PWM pin. Referring to table 3,
when PWM = 0, the selected drivers are turned off and the load
inductance causes the current to recirculate. The current path during recirculation is determined by the configuration of the MODE
pin. Setting PWM = 1 will turn on selected drivers as determined
by the Hx input logic. Holding PWM=1 allows speed and torque
control solely by the internal current-limit circuit, using the voltage on the REF pin.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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10
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
In some circumstances, it may be desirable to completely disable
the internal PWM control. This can be done by pulling the RC
pin directly to AGND. This will disable the internal PWM oscillator and ensure that the output of the PWM latch is always high.
Blank Time When the source driver is turned on, a current spike
occurs due to the reverse-recovery currents of the clamp diodes
and switching transients related to distributed capacitance in the
load. To prevent this current spike from erroneously resetting
the source enable latch, the current-control comparator output
is blanked for a short period of time, tBLANK, when the source
driver is turned on.
The length of tBLANK is different for internal versus external
PWM. It is set by the value of the timing capacitor, CT, according
to the following formulas:
for internal PWM: tBLANK (μs) = 1260 × CT (μF), and
for external PWM: tBLANK (μs) = 2000 × CT (μF) .
A nominal CT value of 680 pF yields a tBLANK of 1.3 μs for
external PWM, and 860 ns for internal PWM. The user must
ensure that CT is large enough to cover the current spike duration
when using the internal sense amplifier.
Diagnostics
Several diagnostic features integrated into the A3930/A3931
provide speed and direction feedback and indications of fault
conditions.
TACHO and DIRO These outputs provide speed and direction
information based on the HE inputs from the motor. As shown in
figure 1, at each commutation point, the TACHO output changes
state independent of motor direction. The DIRO output is updated
at each commutation point to show the motor direction. When
the motor is rotating in the “forward” or positive direction, DIRO
will be high. When rotation is in the “reverse” or negative direction, DIRO will be low. The actual direction of rotation is determined from the sequence of the three Hall inputs, Hx. Forward
is when the sequence follows table 2 top-to-bottom and reverse
when the sequence follows table 2 bottom-to-top.
DIRO
TACHO
Commutation
Points
"Forward" Motor Rotation
"Reverse" Motor Rotation
Note that there are some circumstances in which the direction
reported on the DIRO output pin and the direction demanded
on the DIR input pin may not be the same. This may happen if
the motor and load have reasonably high inertia. In this case,
changing the state of the DIR pin will cause the torque to reverse,
braking the motor. During this braking, the direction indicated on
the DIRO output will not change.
ESF The state of the enable stop on fault (ESF) pin will determine the action taken when a short is detected. For other fault
conditions, the action is defined by the type of fault. The action
taken follows the states shown in table 1.
When ESF = 1, any short fault condition will disable all the
gate drive outputs and coast the motor. This disabled state will
be latched until the next phase commutation or until COAST or
RESET go low.
When ESF = 0, under most conditions, although the fault flags,
FF1 and FF2, are still activated, the A3930/A3931will not disrupt
normal operation and will therefore not protect the motor or the
drive circuit from damage. It is imperative that the master control
circuit or an external circuit take any necessary action when a
fault occurs, to prevent damage to components.
If desired, the active low COAST input can be used as a crude
disable circuit by connecting the fault flags FF1 and FF2 to the
COAST input and a pull-up resistor to V5.
FF1, FF2, and VDSTH Fault conditions are indicated by the
state of two open drain output fault flags, FF1 and FF2, as shown
in table 1. In addition to internal temperature, voltage, and logic
monitoring, the A3930/A3931 monitors the state of the external
MOSFETs and the motor current to determine if short circuit
faults occur or a low load condition exists. In the event that two
or more faults are detected simultaneously, the state of the fault
flags will be determined by a logical AND of the fault states of
each flag.
• Undervoltage VREG supplies the low-side gate driver and the
bootstrap charge current. It is critical to ensure that the voltages
are sufficiently high before enabling any of the outputs. The
undervoltage circuit is active during power-up, and will pull
both fault flags low and coast the motor (all gate drives low)
until VREG is greater than approximately 8 V. Note that this is
sufficient to turn on the external power FETs at a battery voltage
as low as 5.5 V, but will not normally provide the rated on-resistance of the FET. This could lead to excessive power dissipation
in the external FET.
Figure 1. Direction Indication Outputs
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11
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
In addition to a monitor on VREG, the A3930/A3931 also
monitors both the bootstrap charge voltage, to ensure sufficient
high-side drive, and the 5 V reference voltage at V5, to ensure
correct logical operation. If either of these fall below the lockout voltage level, the fault flags are set.
• Overtemperature This event pulls both fault flags low but
does not disable any circuitry. It is left to the user to turn off
the device to prevent overtemperature damage to the chip and
unpredictable device operation.
• Logic Fault: Hall Invalid The A3930 and the A3931 differ
slightly in how they handle error conditions on the Hall inputs,
Hx. When all Hx are 1s, both devices evaluate this as an illegal
code, and they pull both fault flags, FFx, low and coast the motor. This action can be used, if desired, to disable all FET drives
under bridge or motor fault conditions. The Hall logic fault
condition is not latched, so if the fault occurs while the motor is
running, the external FETs will be reenabled, according to the
commutation truth table (table 2), when the Hx inputs become
valid.
be used to detect if an open load condition is present. If, during
a commutation period, the output from the sense amplifier does
not go above a minimum value, VCSOL, FF1 will go low. No
further action will be taken.
Short Fault Operation Because motor capacitance may cause
the measured voltages to show a fault as the phase switches, the
voltages are not sampled until one tDEAD interval after the external FET is turned on.
If a short circuit fault occurs when ESF = 0, the external FETs
are not disabled by the A3930/A3931. Under some conditions,
some measure of protection will be provided by the internal current limit but in many cases, particularly for a short to ground,
the current limit will provide no protection for the external
FETs. To limit any damage to the external FETs or the motor, the
A3930/A3931 can either be fully disabled by the RESET input
or all FETs can be switched off by pulling the COAST input low.
Alternatively, setting ESF = 1 will allow the A3930/A3931 to disable the outputs as soon as the fault is detected. The fault will be
latched until any of the following conditions occur:
When all Hx are 0s, the A3930 handles this in the same manner
as all 1s, described in the preceding paragraph. The A3931,
however, evaluates this as a prepositioning code, and does not
register it as a fault.
a phase commutation
The Hx inputs have pull-up resistors to ensure that a fault condition will be indicated in the event of an open connection to a
Hall sensor.
This will allow a running motor to coast to the next phase
commutation without the risk of damage to the external power
MOSFETs.
• Short to Ground A short from any of the motor phase connections to ground is detected by monitoring the voltage across
the top FETs in each phase using the appropriate Sx pin and the
voltage at VDRAIN. This drain-source voltage is then compared
to the voltage on the VDSTH pin. If the drain source voltage
exceeds the voltage at the VDSTH pin, FF2 will be pulled low.
Low Load Current Fault Operation No action is taken for
a low load current condition. If the low load occurs due to an
open circuit on a phase connection while the motor is running,
the A3930/A3931 will continue to commutate the motor phases
according to the commutation truth table, table 2.
• Short to Supply A short from any of the motor phase connections to the battery or VBB connection is detected by monitoring the voltage across the bottom FETs in each phase using the
appropriate Sx pin and the LSS pin. This drain-source voltage
is then compared to the voltage on the VDSTH pin. If the drain
source voltage exceeds the voltage at the VDSTH pin, FF2 will
be pulled low.
RESET goes low
COAST goes low
In some cases, this will allow the motor to continue operating at
a much reduced performance. The low load condition is checked
during a commutation period and is only flagged at the next commutation event. The flag is cleared at the end of any subsequentcommutation period where no low load current fault is detected.
• Shorted Motor Winding A short across the motor phase
winding is detected by monitoring the voltage across both the
top and bottom FETs in each phase. This fault will pull FF2 low.
If the motor stalls or is stationary, then the remaining phase connections will usually be insufficient to start rotating the motor. At
start-up or after a reset, the low load condition is flagged until the
first time the motor current exceeds the threshold value, VCSOL.
This allows detection of a possible open phase from startup, even
if the motor is not able to start running.
• Low Load Current The sense amplifier output is monitored
independently to allow detection of a low load current. This can
Note that a low load current condition can also exist if the motor
being driven has no mechanical load.
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12
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Table 1. Fault Action Table
Action*
ESF = 0
ESF = 1
0
0
Undervoltage
Disable
Disable
0
0
Overtemperature
No Action
No Action
0
0
Logic Fault
Disable
Disable
1
0
Short to ground
No Action
Disable
1
0
Short to supply
No Action
Disable
1
0
Shorted motor winding
No Action
Disable
0
1
Low load current
No Action
No Action
1
1
None
No Action
No Action
*Disable indicates that all gate outputs are low and all MOSFETs are
turned off.
FF1
FF2
Fault
Table 2. Commutation Truth Table*
Device
H1
H2
H3
DIR
GLA
GLB
GLC
GHA
GHB
GHC
SA
SB
SC
Both
Both
Both
Both
Both
Both
A3930
A3931
Both
Both
Both
Both
Both
Both
Both
1
1
1
0
0
0
0
0
1
1
1
1
0
0
0
0
0
1
1
1
0
0
0
1
0
0
1
1
1
0
1
0
0
0
1
1
0
0
1
1
0
0
0
1
1
1
1
1
1
1
1
X
X
X
0
0
0
0
0
0
0
0
1
1
0
0
0
0
0
1
0
0
0
0
1
0
0
0
0
1
1
0
1
0
0
1
1
0
0
0
1
1
0
0
0
0
0
1
0
0
0
0
1
1
0
1
0
0
0
0
1
0
1
0
0
0
1
1
0
0
0
1
1
0
0
0
0
0
0
0
0
0
0
1
1
0
0
0
1
1
0
0
0
0
1
1
0
0
0
0
High
Z
Low
Low
Z
High
Z
High
Z
Low
Z
High
High
Z
Low
Z
High
High
Z
Low
Low
Z
Low
Z
Z
Low
Low
Z
High
High
Low
Low
Z
High
High
Z
Z
Low
Z
High
High
Z
Low
Low
Z
*X indicates “don’t care,” Z indicates high impedance state
Table 3. INPUT LOGIC
MODE
PWM
BRAKE
COAST
RESET
Decay
0
0
1
1
1
Fast
Mode of Operation
0
1
1
1
1
Fast
Peak current limit – selected drivers ON
1
0
1
1
1
Slow
PWM chop – current decay with both low-side drivers ON
1
1
1
1
1
Slow
X
X
0
1
1
n/a
PWM chop – current decay with opposite of selected drivers ON
Peak current limit – selected drivers ON
Brake mode - All low-side gates ON
X
X
X
0
1
X
Coast mode - All gates OFF
X
X
X
X
0
X
Sleep mode – All gates OFF, low power state, 5 V OFF
*X indicates “don’t care”
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13
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Applications Information
Power
All supply connections to the A3930/A3931 should have capacitors mounted between the supply pins and the ground pin. These
capacitors will provide the transient currents which occur during
switching and decouple any voltage transients on the pin from the
main supply.
VBB Decouple with at least a 100 nF ceramic capacitor mounted
between the VBB pin and the AGND pin. A larger electrolytic
capacitor, typically 10 μF, in parallel with the ceramic capacitor
is also recommended.
VREG Supplies current for the gate-drive circuit. As the gates
are driven high, they require current from an external capacitor
connected to VREG to support the transients. This capacitor
should be placed as close as possible to the VREG pin with the
ground connection close to the AGND pin. Its value should be at
least 20 times larger than the bootstrap capacitor. The capacitor
should have a very low series resistance (ESR) and inductance
(ESL) to avoid large voltage drops during the initial transient.
The optimum capacitor type is a high quality ceramic such as
X7R. However, when the required capacitance is too large, an
aluminium electrolytic capacitor may be used, with a smaller
ceramic capacitor (≈100 nF) in parallel.
V5 When the 5V regulator is used with an external pass transistor
to provide power to other circuits, a 10 μF decoupling capacitor
should be connected between the V5 pin and AGND as close to
the pins as possible. If an electrolytic capacitor is used, then a
100 nF ceramic capacitor should be added in parallel. To improve
stability, a 100 nF capacitor also should be connected between
the V5BD pin and AGND. If 5V is not required for external
circuits, the external pass transistor may be omitted, but in that
case, V5 must connected directly to V5BD and decoupled with at
least a 220 nF capacitor between V5 and AGND.
AGND The A3930/A3931 has a single ground connection at the
AGND pin. The design ensures that only the operating current
for the controller stage passes through this pin. The charge and
discharge current for the external FETs does not pass though this
pin. The AGND pin is the ground reference for the current trip
threshold, the VDS monitor threshold, and the timing components.
It should therefor be kept as quiet as possible. A suggested ground
connection scheme is described in the layout section below.
Power Dissipation In applications where a high ambient temperature is expected the on-chip power dissipation may become
a critical factor. Careful attention should be paid to ensure the
operating conditions allow the A3930/A3931 to remain in a safe
range of junction temperature.
The power consumed, PTOT , by the A3930/A3931 can be estimated using the following formulas:
PTOT = PBIAS + PCPUMP + PSWITCHING ,
PBIAS = VBB × IBB ,
where IBB is 3 mA, typical, and
PCPUMP = (2 × VBB–VREG) × IAV
where VBB < 15 V, or
PCPUMP = (VBB–VREG) × IAV
where VBB > 15 V, and
IAV = QGATE × N × fPWM ,
PSWITCHING = QGATE × VREG × N × fPWM × Ratio
where N = 2 for slow decay, or N = 4 for fast decay, and
Ratio = 10 / (RGATE + 10)
Bootstrap Capacitors
Bootstrap Capacitor Selection The value for CBOOT must
be correctly selected to ensure proper operation of the device. If
the value is too large, time will be wasted charging the capacitor,
resulting in a limit on the maximum duty cycle and PWM
frequency. If the value is too small, there can be a large voltage
drop at the time when the charge is transferred from CBOOT to the
MOSFET gate.
To keep the voltage drop small, QBOOT  QGATE . A factor of 20 is
a reasonable value. To calculate CBOOT, the following formulas
can be used:
QBOOT = CBOOT × VBOOT ,
= QGATE × 20,
therefore
CBOOT = QGATE × 20 / VBOOT
The voltage drop on the Cx pin as the MOSFET is being turned
on can be approximated by:
ΔV = QGATE / CBOOT
Bootstrap Charging It is good practice to ensure that the highside bootstrap capacitor, CBOOT, is completely charged before a
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14
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
high-side PWM cycle is requested. The minimum time required
to charge the capacitor is approximated by:
tCHARGE(min) ≈ CBOOT × ΔV /250 mA
At power-on, and when the drivers have been disabled for a long
time, the CBOOT may be completely discharged. In these cases,
ΔV can be considered to be the full high-side drive voltage,
12 V. Otherwise, ΔV is the amount of voltage dropped during the
charge transfer, which should be 400 mV or less. The capacitor is
charged whenever the Sx pin is pulled low via a GLx PWM cycle,
and current flows from VREG through the internal bootstrap
diode circuit to CBOOT.
Bootstrap Charge Monitor The A3930 and A3931 provide
automatic bootstrap capacitor charge management. The bootstrap capacitor voltage for each phase, VBOOTx , is continuously
checked to ensure that it is above the bootstrap undervoltage
threshold, VBOOTUV. If VBOOT drops below this threshold, the
A3930 and A3931 will turn on the necessary low-side FET until
the VBOOT exceeds VBOOTUV plus the hysteresis, VBOOTUVHys .
The minimum charge time is typically 7 μs, but may be longer for
very large values of the bootstrap capacitor (CBOOT >1000 nF). If
VBOOT does not exceed VBOOTUV within approximately 200 μs,
an undervoltage fault will be flagged, as shown in table 1.
PWM Control
The A3930 and A3931 have the flexibility to be used in many
different motor control schemes. The internal PWM control can
be used to provide fully integrated, closed-loop current control.
Alternatively, current-mode or voltage-mode control are possible
using external control circuits with either the DIR or the PWM
input pins.
Internal PWM Control The internal PWM current control
function is useful in applications where motor torque control
or simple maximum current limitation is required. However,
for motor speed control applications, it is usually better to use
external PWM control either as a closed- or open-loop system.
External PWM Control When external PWM control is used, it
is possible to completely disable the internal PWM control circuit
by connecting the RC pin to AGND.
With the internal control disabled, however, care should be taken
to avoid excessive current in the power FETs because the A3930/
A3931 will not limit the current. Short-circuit detection will still
be available in case of faults. The output of the sense amplifier is
also available, but provision must be made in the external control
circuits to ignore (blank) the transients at the switching points.
External and Internal Combined PWM Control Where
external PWM control is used but current limitation is still
required, internal PWM current control can be used at the
same time as external PWM control. To do so, usually the
internal PWM frequency is set lower than the external PWM
frequency. This allows the external PWM signal to dominate and
synchronize the internal PWM circuit. It does this by discharging
the timing capacitor, CT, when the PWM pin is low. When
internal and external PWM control are used together, all control
features of the A3930/A3931 are available and active, including:
dead time, current comparator, and comparator blanking.
PWM Frequency Should be set high enough to avoid any
audible noise, but low enough to ensure adequate charging of the
boot capacitor, CBOOT. The external resistor RT and capacitor
CT, connected in parallel from the RC pin to AGND, set the
PWM frequency to approximately:
fOSC ≈ 1 / (RTCT + tBLANK + tDEAD) .
RT should be in the range of 5 to 400 kΩ.
PWM Blank The timing capacitor, CT, also serves as the
means to set the blank time duration. tBLANK. At the end of the
PWM off-cycle, a high-side gate selected by the commutation
logic turns on. At this time, large current transients can occur
during the reverse recovery time of the intrinsic source drain
body diodes of the external power MOSFETs. To prevent false
tripping of the current-sense comparator, the output of the current
comparator is ignored during the blank time.
The length of tBLANK is different for internal versus external
PWM. It is set by the value of the timing capacitor, CT, according
to the following formulas:
for internal PWM: tBLANK (μs) = 1260 × CT (μF), and
for external PWM: tBLANK (μs) = 2000 × CT (μF) .
A nominal CT value of 680 pF will give a blanking time of 1.3 μs
for external PWM and 860 ns for internal PWM. The user must
ensure that CT is large enough to cover the current-spike duration.
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15
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Note that this blank time is only used to mask the internal current comparator. If the current sense amplifier output, CSOUT,
is being used in an external PWM control circuit, then it will
be necessary to externally generate a blank time for that control
loop.
Dead Time The potential for cross-conduction occurs with
synchronous rectification, direction changes, PWM, or after a
bootstrap capacitor charging cycle. To prevent cross-conduction
in any phase of the power FET bridge, it is necessary to have a
dead-time delay, tDEAD, between a high- or low-side turn-off and
the next turn-on event. tDEAD is in the range of between 96 ns and
6.3 μs, and is set by the value of a resistor, RDEAD, between the
RDEAD pin and the GND pin. The maximum dead time of typically 6μs can be set by leaving the RDEAD pin unconnected, or
connected to the V5 pin.
At 25°C, the value of tDEAD (μs) can be approximated by:
tDEAD(nom) ≈ 0.1 + 33 / (5 + IDEAD),
IDEAD = 2000 / RDEAD
where IDEAD is in μA, and RDEAD is between 5 and 400 kΩ. The
greatest accuracy is obtained with values of RDEAD between
10 and 100 kΩ.
The choice of power MOSFET and external series gate resistance
determines the selection of RDEAD. The dead time should be
made long enough to cover the variation of the MOSFET gate
capacitance and the tolerances of the series gate resistance, both
external and internal to the A3930/A3931.
Current
Trip Points
GHx
tDEAD
tDEAD
Synchronous Rectification To reduce power dissipation in
the external MOSFETs, the A3930/A3931 control logic turns
on the appropriate low-side and high-side driver during the load
current recirculation PWM-off cycle. Synchronous rectification
allows current to flow through the FET selected by the MODE
pin setting during the decay time, rather than through the sourcedrain body diode. The body diodes of the recirculating power
FETs conduct only during the dead time that occurs at each PWM
transition. For internal current control using fast decay mode,
reversal of load current is prevented by turning off synchronous
rectification when a zero current level is detected. For external
PWM control using fast decay mode, the load current will not be
limited to zero but will rise to the set current limit in the reverse
direction before disabling synchronous rectification.
Braking. The A3930 and A3931 provide dynamic braking by
forcing all low-side MOSFETs on, and all high-side MOSFETs
off. This effectively short-circuits the back EMF of the motor,
which forces a reverse current in the windings, and creating a
breaking torque.
During braking, the load current can be approximated by:
IBRAKE ≈ VBEMF / RLOAD
Because the load current does not flow through the sense resistor,
RSENSE, during a dynamic brake, care must be taken to ensure
that the power MOSFET maximum ratings are not exceeded.
It is possible to apply a PWM signal to the BRAKE input to
limit the motor braking current. However, because there is
no measurement of this current, the PWM duty cycle must be
determined for each set of conditions. Typically the duty cycle
of such a brake PWM input would start at a value which limits
the current and then drops to 0%, that is, BRAKE goes to low, to
hold the motor stationary.
Setting RESET = 1 and COAST = 0 overrides BRAKE and turns
all motor bridge FETs off, coasting the motor.
GLx
+V
VRCH
RC
VRCL
0
tBLANK
tRC
tOSC
Figure 2. Internal PWM RC Timing
Note: For reasons of
clarity, t DEAD is shown
exaggerated.
Driving a Full-Bridge. The A3930 and A3931 may be used
to drive a full-bridge (for example, a brush DC motor load) by
hard-wiring a single state for the Hall inputs and leaving the
corresponding phase driver outputs floating. For example, with a
configuration of H1 = H2 = 1, and H3 = 0, the outputs CC, GHC,
SC, and GLC would be floated, according to the commutation
truth table, table3, which indicates a state of high-impedence (Z)
for SC with that Hall input configuration. The DIR input
controls the motor rotation, while the PWM and MODE
inputs control the motor current behavior, as described in the
input logic table, table 3.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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16
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Circuit Layout
Because this is a switch-mode application, where rapid current
changes are present, care must be taken during layout of the
application PCB. The following points are provided as guidance
for layout (refer to figure 3). Following all guidelines will not
always be possible. However, each point should be carefully
considered as part of any layout procedure.
Ground connection layout recommendations:
1.
2.
3.
Sensitive connections such as RDEAD and VDSTH, which
have very little ground current, should be referenced to the
Quiet ground, which is connected independently closest to
the AGND pin. The components associated with these sensitive pins should never be connected directly to the Supply
common or to the Power ground; they must be referenced
directly to the AGND pin.
6.
The AGND pin should be connected by an independent low
impedance trace to the Supply common at a single point.
7.
Check the peak voltage excursion of the transients on the
LSS pin with reference to the AGND pin using a closegrounded (tip and barrel) probe. If the voltage at LSS
exceeds the absolute maximum specified in this datasheet,
add additional clamping, capacitance or both between the
LSS pin and the AGND pin.
Other layout recommendations:
1.
Gate charge drive paths and gate discharge return paths may
carry large transient current pulses. Therefore, the traces
from GHx, GLx, Sx, and LSS should be as short as possible
to reduce the inductance of the circuit trace.
2.
Supply decoupling for the supply pins VBB, VREG, and
V5 should be connected to Controller Supply ground, which
is connected independently, close to the AGND pin. The
decoupling capacitors should also be connected as close as
possible to the corresponding supply pin.
Provide an independent connection from LSS to the common
point of the power bridge. It is not recommended to connect
LSS directly to the AGND pin, as this may inject noise into
sensitive functions such as the dead-timer. The LSS connection should not be used for the CSP connection.
3.
The oscillator timing components can be connected to Quiet
ground or Controller Supply ground. They should not be
connected to the Supply common or the Power ground.
The inputs to the sense amplifier, CSP and CSN, should be
independent traces and for best results should be matched in
length and route.
4.
Minimize stray inductance by using short, wide copper runs
at the drain and source terminals of all power FETs. This
includes motor lead connections, the input power bus, and
the common source of the low-side power FETs. This will
minimize voltages induced by fast switching of large load
currents.
5.
Consider the use of small (100 nF) ceramic decoupling
capacitors across the source and drain of the power FETs
4.
The exposed thermal pad on the package should be connected to the AGND pin and may form part of the Controller
Supply ground.
5.
If the layout space is limited, then the Quiet ground and the
Controller Supply ground may be combined, provided that
the ground return of the dead-time resistor, RDEAD, is close
to the AGND pin.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
17
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
to limit fast transient voltage spikes caused by trace inductance.
6.
Ensure that the TEST pin is connected to AGND. This pin is
used for production test only.
The above are only recommendations. Each application is different and may encounter different sensitivities. A driver running
with a few amperes will be less susceptible than one running with
150 A, and each design should be tested at the maximum current,
to ensure any parasitic effects are eliminated.
VBB
VDRAIN
+ Supply
GHC
GHB
VREG
GHA
V5
A3930
A3931
SA
SB
SC
RC
VDSTH
RDEAD
AGND
Quiet Ground
Motor
GLA
GLB
GLC
LSS
RSENSE
Optional components
to limit LSS transients
Power Ground
Supply
Common
Controller Supply Ground
Figure 3. Supply and Ground Connections
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
18
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Gate Drive Outputs
Sense Amplifier
VREG
Cx
18 V
18 V
76k
160μA
22V
18 V
GHx
4 k7
CSN
18 V
19 V
22V
Sx
CSOUT
4 k7
20 V
VREG
160μA
CSP
8.5 V
18 V
18 V
8.5 V
2V
32.4 k7
GLx
20 V
72 k7
4.6 k7
LSS
Supplies
REF
CP1
VDRAIN
CP2
VBB
V5
V5BD
3 k7
REF
18 V
19 V
19 V
19 V
19 V
20 V
20 V
6V
Logic Inputs
COAST
ESF
BRAKE
DIR
PWM
MODE
8V
10 V
Hall Sensor Inputs
3 k7
8V
8V
1 k7
Oscillator RC Pin
3 k7
RESET
3 k7
8.5 V
VDS Monitor Threshold Input
Reset Input
V5
100 k7
H1
H2
H3
8.5 V
6V
8.5 V
6V
Fault Output
V5
100 7
FF1
FF2
40 k7
VDSTH
50 k7
1 k7
RC
8V
8V
8.5 V
8V
RDEAD
Logic Output
100 7
V5
2V
RDEAD
100 7
TACHO
DIRO
8V
8V
8.5 V
8.5 V
8V
8V
Figure 4. Input and Output Structures
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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19
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Pin-out Diagrams
NC 37
LSS 38
ESF 39
25 SC
26 GHC
23 VDSTH
Bootstrapped
High-Side Drives
VREG 40
22 CSP
21 CSN
Current
Sense
Charge
Pump
CP1 42
27 CC
24 VDRAIN
Low
Side
Drives
AGND 41
28 SB
29 GHB
30 CB
31 SA
32 GHA
33 CA
34 GLC
35 GLB
36 GLA
JP Package
20 REF
19 CSOUT
18 RDEAD
CP2 43
17 TEST
DIRO 44
VBB 45
16 RC
Control
Logic
COAST 46
15 MODE
NC 47
14 PWM
Hall
13 NC
H3 12
H2 11
H1 10
DIR 9
BRAKE 8
TACHO 7
FF1 6
FF2 5
V5 4
V5BD 3
NC 1
RESET 2
NC 48
Terminal List Table
Number
Name
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
N.C.
RESET
V5BD
V5
FF2
FF1
TACHO
BRAKE
DIR
H1
H2
H3
N.C.
PWM
MODE
RC
TEST
RDEAD
CSOUT
REF
CSN
CSP
VDSTH
VDRAIN
Description
No connection
Control for sleep mode
5V regulator base drive
5V regulator reference
Fault flag 2
Fault flag 1
Speed output
Brake input
Direction control input
Hall sensor input
Hall sensor input
Hall sensor input
No connection
Control input
Decay control input
PWM oscillator control input
Test pin; tie to AGND
Dead time setting
Current sense output
Current limit setting
Current sense input –
Current sense input +
Fault threshold voltage
High-side drain voltage sense
Number
Name
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
–
SC
GHC
CC
SB
GHB
CB
SA
GHA
CA
GLC
GLB
GLA
N.C.
LSS
ESF
VREG
AGND
CP1
CP2
DIRO
VBB
COAST
N.C.
N.C.
Pad
Description
Motor connection phase C
High-side gate drive phase C
Bootstrap capacitor phase C
Motor connection phase B
High-side gate drive phase B
Bootstrap capacitor phase B
Motor connection phase A
High-side gate drive phase A
Bootstrap capacitor phase A
Low-side gate drive phase C
Low-side gate drive phase B
Low-side gate drive phase A
No connection
Low-side source
Enable stop on fault input
Gate drive supply output
Analog ground
Pump capacitor
Pump capacitor
Direction output
Supply voltage
Coast input
No connection
No connection
Thermal dissipation, tie to AGND
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
20
A3930 and
A3931
Automotive 3-Phase BLDC Controller
and MOSFET Driver
Package JP, 48-pin LQFP with Exposed Thermal Pad
0.30
9.00 ±0.20
0.50
1.70
7.00 ±0.20
7º
4° ±4
0º
+0.05
0.15 –0.06
C
B
9.00 ±0.20 7.00 ±0.20
5.00
5.00±0.04
8.60
0.60 ±0.15
48
(1.00)
A
1
48
2
1 2
0.25
5.00±0.04
SEATING PLANE
GAGE PLANE
5.00
8.60
48X
SEATING
PLANE
0.08 C
0.22 ±0.05
0.50
C
C
For Reference Only
(reference JEDEC MS-026 BBCHD)
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
1.60 MAX
1.40 ±0.05
0.10 ±0.05
PCB Layout Reference View
A Terminal #1 mark area
B Exposed thermal pad (bottom surface)
C Reference land pattern layout (reference IPC7351
QFP50P900X900X160-48M); adjust as necessary to meet
application process requirements and PCB layout
tolerances; when mounting on a multilayer PCB, thermal
vias at the exposed thermal pad land can improve thermal
dissipation (reference EIA/JEDEC Standard JESD51-5)
Copyright ©2006-2013, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
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21
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