A6213 Datasheet

A6213 and A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Features and Benefits
•AEC-Q100 qualified
•Supply voltage 6 to 48 V
•True average output current control
•3 A maximum output over operating temperature range
(1.5 A for A6213-1)
•Cycle-by-cycle current limit
•Integrated MOSFET switch
•Dimming via direct logic input or power supply voltage
•Internal control loop compensation
•Undervoltage lockout (UVLO) and thermal shutdown
protection
•Low power shutdown (1 µA typical)
•Robust protection against:
▫Adjacent pin-to-pin short
▫Pin-to-GND short
▫Component open/short faults
Package 8-pin SOICN with exposed
thermal pad (suffix LJ):
Not to scale
Description
The A6213 is a single IC switching regulator that provides
constant-current output to drive high-power LEDs. It integrates
a high-side N-channel DMOS switch for DC-to-DC step- down
(buck) conversion. A true average current is output using a
cycle-by-cycle, controlled on-time method.
Output current is user-selectable by an external current sense
resistor. Output voltage is automatically adjusted to drive
various numbers of LEDs in a single string. This ensures the
optimal system efficiency.
LED dimming is accomplished by a direct logic input pulsewidth modulation (PWM) signal at the enable pin.
The device is provided in a compact 8-pin narrow SOIC package
(suffix LJ) with exposed pad for enhanced thermal dissipation.
It is lead (Pb) free, with 100% matte-tin leadframe plating.
Applications:
Automotive lighting
•Daytime running lights
•Front and rear fog lights
•Turn/stop lights
•Map light
•Dimmable interior lights
Typical Application Circuit
VIN (6 to 48 V)
GND
C1
1
8
SW
A6213
R1 2
7
TON
BOOT
3
6
GND
EN
PAD
4
VCC 5
CS
VIN
C4
LED+
D1
...
EN
L1
C5
Enable/PWM Dimming
(100 Hz to 2 kHz)
LED–
RSENSE
A62131-DS, Rev. 6
January 21, 2013
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Selection Guide
Operating Ambient
Temperature, TA
Part Number
A6213KLJTR-T
A6213KLJTR-1-T
–40°C to 125°C
–40°C to 125°C
Package
Packing
8-pin SOICN with exposed thermal pad
8-pin SOICN with exposed thermal pad
3000 pieces per 13-in reel
3000 pieces per 13-in reel
Absolute Maximum Ratings
Characteristic
Symbol
Supply Voltage
Bootstrap Drive Voltage
Notes
Rating
Unit
VIN
–0.3 to 50
V
VBOOT
–0.3 to VIN + 8
V
VSW
–1.5 to VIN + 0.3
V
Switching Voltage
Linear Regulator Terminal
VCC
Enable and TON Voltage
VEN , VTON
Current Sense Voltage
VCC to GND
VCS
–0.3 to 7
V
–0.3 to VIN + 0.3
V
–0.3 to 7
V
–40 to 125
°C
TJ(max)
150
°C
Tstg
–55 to 150
°C
Operating Ambient Temperature
TA
Maximum Junction Temperature
Storage Temperature
K temperature range for automotive
Thermal Characteristics*: may require derating at maximum conditions; see application section for optimization
Characteristic
Symbol
Package Thermal Resistance
(Junction to Ambient)
RθJA
Package Thermal Resistance
(Junction to Pad)
RθJP
Test Conditions*
Value
Unit
On 4-layer PCB based on JEDEC standard
35
°C/W
On 2-layer generic test PCB with 0.8 in.2 of copper area each side
62
°C/W
2
°C/W
*Additional thermal information available on the Allegro™ website.
Pinout Diagram
Terminal List Table
Number
8
VIN 1
7
TON 2
SW
BOOT
PAD
EN 3
CS 4
6
5
GND
VCC
Name
Function
1
VIN
Supply voltage input terminals
2
TON
Regulator on-time setting resistor terminal
3
EN
Logic input for Enable and PWM dimming
4
CS
Drive output current sense feedback
5
VCC
Internal linear regulator output
6
GND
Ground terminal
7
BOOT
DMOS gate driver bootstrap terminal
8
SW
Switched output terminals
–
PAD
Exposed pad for enhanced thermal dissipation;
connect to GND
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Functional Block Diagram
CVCC
CBOOT
VCC
VIN
VIN
BOOT
L1
LED String
D1
SW
VREG 5.3 V
VCC
UVLO
Average
On-Time
Current
Generator
TON
On-Time
Timer
Off-Time
Timer
Gate Drive
UVLO
Shutdown
RON
Level Shift
EN
+
IC and Driver
Control Logic
CCOMP
0.2 V
Current Limit
Off-Time
Timer
–
–
Buck Switch
Current Sense
+
VIL = 0.4 V
VIH = 1.8 V
+
+
VCC UVLO
–
–
Thermal
Shutdown
ILIM
CS
PAD
GND
RSENSE
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
ELECTRICAL CHARACTERISTICS: Valid at VIN = 24 V, TA = –40°C to 125°C, typical values at TA = 25°C, unless otherwise noted
Characteristics
Symbol
Input Supply Voltage
VIN Undervoltage Lockout Threshold
VIN Undervoltage Lockout Hysteresis
VUVLO
IIN
IINSD
Buck Switch Current Limit Threshold
ISWLIM
Buck Switch On-Resistance
R
BOOT Undervoltage Lockout
Threshold
BOOT Undervoltage Lockout
Hysteresis
DS(on)
VBOOTUV
Typ.
Max.
Unit
6
–
48
V
–
5.3
–
V
–
150
–
mV
VCS = 0.5 V, EN = high
–
5
–
mA
EN shorted to GND
–
1
10
µA
A6213
3.0
4.0
5.0
A
A6213-1
1.9
2.2
2.5
A
–
0.25
0.4
Ω
1.7
2.9
4.3
V
–
370
–
mV
–
110
150
ns
–
110
150
ns
800
1000
1200
ns
187.5
200
210
mV
–
0.9
–
µA
5.1
5.4
5.7
V
5
20
–
mA
VIN increasing
VBOOT = VIN + 4.3 V, TA = 25°C, ISW = 1 A
VBOOT to VSW increasing
VBOTUVHYS VBOOT to VSW decreasing
Switching Minimum Off-Time
tOFFmin
Switching Minimum On-Time
tONmin
Selected On-Time
Min.
VUVLO_HYS VIN decreasing
VIN Pin Supply Current
VIN Pin Shutdown Current
Test Conditions
VIN
tON
VCS = 0 V
VIN = 24 V, VOUT = 12 V, RON = 137 kΩ
Regulation Comparator and Error Amplifier
Load Current Sense Regulation
Threshold 1
VCSREG
VCS decreasing, SW turns on
Load Current Sense Bias Current
ICSBIAS
VCS = 0.2 V, EN = low
Internal Linear Regulator
VCC Regulated Output
VCC Current
Limit 2
VCC
ICCLIM
0 mA < ICC < 5 mA, VIN > 6 V
VIN = 24 V, VCC = 0 V
Enable Input
Logic High Voltage
VIH
VEN increasing
1.8
–
–
V
Logic Low Voltage
VIL
VEN decreasing
–
–
0.4
V
RENPD
VEN = 5 V
–
100
–
kΩ
tPWML
Measured while EN = low, during dimming
control, and internal references are powered-on
(exceeding tPWML results in shutdown)
10
17
–
ms
EN Pin Pull-down Resistance
Maximum PWM Dimming Off-Time
Thermal Shutdown
Thermal Shutdown Threshold
TSD
–
165
–
°C
Thermal Shutdown Hysteresis
TSDHYS
–
25
–
°C
In test mode, a ramp signal is applied at CS pin to determine the CS pin regulation threshold voltage. In actual application, the average CS pin
voltage is regulated at VCSREG regardless of ripple voltage.
2 The internal linear regulator is not designed to drive an external load
1
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Characteristic Performance
VIN
VIN
VOUT
VOUT
C1,C2
C1,C2
iLED
C3
iLED
C3
VEN
C4
C4
VEN
t
t
Panel 1B. VIN = 24 V
Panel 1A. VIN = 19 V
VIN
VOUT
C1,C2
iLED
C3
C4
VEN
t
Panel 1C. VIN = 30 V
Figure 1: Startup waveforms from off-state at various input voltages; note that the rise time of the LED current depends on input/output
voltages, inductor value, and switching frequency
• Operating conditions: LED voltage = 15 V, LED current = 1.3 A, R1 = 63.4 kΩ (frequency = 1 MHz in steady state),
VIN = 19 V (panel 1A), 24 V (panel 1B) and 30 V (panel 1C)
• Oscilloscope settings: CH1 (Red) = VIN (10 V/div), CH2 (Blue) = VOUT (10 V/div),
CH3 (Green) = iLED (500 mA/div), CH4 (Yellow) = Enable (5 V/div), time scale = 50 µs/div
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
VIN
VOUT
C1,C2
iLED
C3
C4
VEN
t
Panel 2A. Duty cycle = 50% and time scale = 1 ms/div
VIN
VOUT
C1,C2
iLED
C3
C4
VEN
t
Panel 2B. Duty cycle = 2% and time scale = 50 µs/div
Figure 2: PWM operation at various duty cycles; note that there is no startup delay during PWM dimming operation
• Operating conditions: at 200 Hz, VIN = 24 V, VOUT = 15 V, R1 = 63.4 kΩ, duty cycle = 50% (panel 2A) and 2% (panel 2B)
• CH1 (Red) = VIN (10 V/div), CH2 (Blue) = VOUT (10 V/div),
CH3 (Green) = iLED (500 mA/div), CH4 (Yellow) = Enable (5 V/div), time scale = 1 ms/div (panel 2A) and 50 µs/div (panel 2B)
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
6
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
95
95
VIN = 24 V, VOUT = 15 V
VIN = 12 V, VOUT = 5.5 V
85
VIN = 12 V, VOUT = 3.5 V
80
fSW = 500 kHz
90
Efficiency, η (%)
Efficiency, η (%)
90
75
fSW = 1 MHz
85
fSW = 2 MHz
80
75
70
70
0
0.5
1.0
1.5
2.0
2.5
3.0
0
0.5
LED Current, iLED (A)
1.0
1.5
2.0
2.5
3.0
LED Current, iLED (A)
Figure 3: Efficiency versus LED Current at various LED voltages
Operating conditions: fSW = 1 MHz
Figure 4: Efficiency versus LED Current at various switching
frequencies. Operating conditions: VIN = 12 V, VOUT = 5.5 V
LED Current (A)
1
0.1
iLED = 3 A
iLED = 2 A
iLED = 1.4 A
0.01
0.001
0.1
1
10
100
Duty Cycle (%)
Figure 5. Average LED Current versus PWM dimming percentage
Operating conditions: VIN = 12 V, VOUT = 3.5 V, fSW = 1 MHz, fPWM = 200 Hz, L = 10 µH
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
7
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Functional Description
The A6213 is a buck regulator designed for driving a high-current
LED string. It utilizes average current mode control to maintain
constant LED current and consistent brightness. The LED current
level is easily programmable by selection of an external sense
resistor, with a value determined as follows:
iLED = VCSREG / RSENSE
where VCSREG = 0.2 V typical.
Switching Frequency
The A6213 operates in fixed on-time mode during switching. The
on-time (and hence switching frequency) is programmed using
an external resistor connected between the VIN and TON pins, as
2.2
2.0
1.8
fsw (MHz)
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
0
20
40
60
80
100 120 140 160 180 200 220 240 260
RTON (kΩ)
Figure 6: Switching Frequency versus RTON Resistance
• During SW on-time:
iRIPPLE = [(VIN – VOUT) / L] × tON = [(VIN – VOUT) / L] × T × D
where D = tON / T.
• During SW off-time:
iRIPPLE = [(VOUT – VD) / L] × tOFF = [(VOUT – VD) / L] × T × (1 – D)
Therefore (simplified equation for Output Voltage):
VOUT = VIN × D – VD × (1 – D)
If VD << VOUT , then VOUT ≈ VIN × D.
More precisely:
VOUT = (VIN – Iav × RDS(on) ) × D – VD × (1 – D) – RL × Iav
Where RL is the resistance fo the inductor.
CIN
tON = k × (RON + RINT ) × ( VOUT / VIN )
fSW = 1 / [ k × (RON + RINT )]
where k = 0.0139, with fSW in MHz, tON in µs, and RON and RINT
(internal resistance, 5 kΩ) in kΩ (see figure 6).
Enable and Dimming
The IC is activated when a logic high signal is applied to the EN
(enable) pin. The buck converter ramps up the LED current to a
target level set by RSENSE.
When the EN pin is forced from high to low, the buck converter
is turned off, but the IC remains in standby mode for up to 10 ms.
If EN goes high again within this period, the LED current is
turned on immediately. Active dimming of the LED is achieved
by sending a PWM (pulse-width modulation) signal to the EN
pin. The resulting LED brightness is proportional to the duty cycle
( tON / Period ) of the PWM signal. A practical range for PWM dimming frequency is between 100 Hz (Period = 10 ms) and 2 kHz.
At a 200 Hz PWM frequency, the dimming duty cycle can be
varied from 100% down to 1% or lower.
If EN is low for more than 17 ms, the IC enters shutdown mode
to reduce power consumption. The next high signal on EN will
initialize a full startup sequence, which includes a startup delay
of approximately 130 µs. This startup delay is not present during
PWM operation.
The EN pin is high-voltage tolerant and can be directly connected
to a power supply. However, if EN is higher than the VIN voltage
VSW
VIN
0
–VD
t
iL
A6213
VIN
given by the following equation:
i(max)
MOS
SW
D
L
VOUT
iL
RSENSE
iRIPPLE
iav
i(min)
tON
tOFF
t
T
Figure 7: Simplified buck controller equations
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
8
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
at any time, a series resistor (1 kΩ) is required to limit the current
flowing into the EN pin. This series resistor is not necessary if
EN is driven from a logic input.
PWM Dimming Ratio
The brightness of the LED string can be reduced by adjusting the
PWM duty cycle at the EN pin as follows:
Dimming ratio = PWM on-time / PWM period
For example, by selecting a PWM period of 5 ms (200 Hz PWM
frequency) and a PWM on-time of 50 µs, a dimming ratio of 1%
can be achieved.
In an actual application, the minimum dimming ratio is determined by various system parameters, including: VIN , VOUT ,
inductance, LED current, switching frequency, and PWM
frequency. As a general guideline, the minimum PWM on-time
should be kept at 50 µs or longer. A shorter PWM on-time is
acceptable under more favorable operating conditions.
Output Voltage and Duty Cycle
Figure 7 provides simplified equations for approximating output
voltage. Essentially, the output voltage of a buck converter is
approximately given as:
VOUT = VIN × D – VD1 × (1 – D ) ≈ VIN × D, if VD1<< VIN
D = tON / (tON + tOFF )
where D is the duty cycle, and VD1 is the forward drop of the
Schottky diode D1 (typically under 0.5 V).
Minimum and Maximum Output Voltages
For a given input voltage, the maximum output voltage depends
on the switching frequency and minimum tOFF . For example, if
tOFF(min) = 150 ns and fSW = 1 MHz, then the maximum duty
cycle is 85%. So for a 24 V input, the maximum output is 20.3 V.
This means up to 6 LEDs can be operated in series, assuming
Vf = 3.3 V or less for each LED.
The minimum output voltage depends on minimum tON and
switching frequency. For example, if the minimum tON = 150 ns
and fSW = 1 MHz, then the minimum duty cycle is 15%. That
means with VIN = 24 V, the minimum VOUT = 3.2 V (one LED).
To a lesser degree, the output voltage is also affected by other
factors such as LED current, on-resistance of the high-side
switch, DCR of the inductor, and forward drop of the low-side
diode. The more precise equation is shown in figure 7.
As a general rule, switching at lower frequencies allows a wider
range of VOUT , and hence more flexible LED configurations.
This is shown in figure 8.
Figure 8 shows how the minimum and maximum output voltages vary with LED current (assuming RDS(on) = 0.4 Ω, inductor
DCR = 0.1 Ω, and diode Vf = 0.6 V).
If the required output voltage is lower than that permitted by the
minimum tON , the controller will automatically extend the tOFF ,
in order to maintain the correct duty cycle. This means that the
switching frequency will drop lower when necessary, while the
LED current is kept in regulation at all times.
24
9
22
8
20
7
VOUT(max) (V)
16
VOUT ( V )
VOUT ( V )
18
14
12
10
VOUT(max) (V)
6
5
4
3
8
6
2
VOUT(min) (V)
4
VOUT(min) (V)
1
2
0
0
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
fsw (MHz)
Figure 8: Minimum and Maximum Output Voltage versus
Switching Frequency (VIN = 24 V, iLED = 2 A, minimum tON and
tOFF = 150 ns)
0
0.5
1.0
1.5
2.0
2.5
3.0
iLED (A)
Figure 9: Minimum and Maximum Output Voltage versus iLED
current (VIN = 9 V, fSW = 1 MHz, minimum tON and tOFF = 150 ns)
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115 Northeast Cutoff
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A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Thermal Budgeting
The A6213 is capable of supplying a 3 A current through its highside switch (1.5 A for A6213-1). However, depending on the duty
cycle, the conduction loss in the high-side switch may cause the
package to overheat. Therefore care must be taken to ensure the
total power loss of package is within budget. For example, if the
maximum temperature rise allowed is ∆T = 50 K at the device
case surface, then the maximum power dissipation of the IC is
1.4 W. Assuming the maximum RDS(on) = 0.4 Ω and a duty cycle
of 85%, then the maximum LED current is limited to 2 A approximately. At a lower duty cycle, the LED current can be higher.
Fault Handling
The A6213 is designed to handle the following faults:
•Pin-to-ground short
•Pin-to-neighboring pin short
•Pin open
•External component open or short
•Output short to GND
The waveform in Figure 10 illustrates how the A6213 responds
in the case in which the current sense resistor or the CS pin is
shorted to GND. Note that the SW pin overcurrent protection is
tripped at around 4 A, and the part shuts down immediately. The
part then goes through startup retry after approximately 380 µs of
cooldown period.
The A6213-1 has the same protection mechanism, except its
VOUT
C1
A6213-1 tripped SW_ILIM at ~2.4 A
i_LED
Sense Resistor
shorted during
normal operation
C2
overcurrent threshold is 2.2 A. This reduces the risk of inductor
saturation or LED damage during a fault.
As another example, the waveform in Figure 12 shows the fault
case where external Schottky diode D1 is missing or open. As
LED current builds up, a larger-than-normal negative voltage is
developed at the SW node during off-time. This voltage trips the
missing Schottky detection function of the IC. The IC then shuts
down immediately, and waits for a cool-down period before retry.
VEN
Negative voltage
developed at SW pin
during off-time
VSW
VOUT
t
Figure 11: A6313-1 during fault condition where the sense
resistor or CS pin is shorted to GND. Note that its overcurrent
protection threshold is set lower than that of the A6213. Ch1 =
VOUT (5 V/div), Ch2 = i_LED (500 mA/div), t = 200 µs/div.
C1
C1
Cool-down
period
~ 380 µs
C2
VSW
C2
C3
C3
iLED
VOUT
iLED
C4
t
Figure 10: A6213 overcurrent protection tripped in the case of a
fault caused by the sense resistor pin shorted to ground; shows
switch node, VSW (ch1, 10 V/div.), output voltage, VOUT (ch2, 10
V/div.), LED current, iLED (ch3, 1 A/div.), t = 100 µs/div.
t
Figure 12: Startup waveform with a missing Schottky diode;
shows Enable, VEN (ch1, 5 V/div.), swtich node, VSW (ch2, 5 V/
div.), output voltage, VOUT (ch3, 5 V/div.), LED current, iLED (ch4,
500 mA/div.), t = 100 µs/div.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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10
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
1. Determine the saturation current of the inductor. This can be
done by simply adding 20% to the average LED current:
iSAT ≥ iLED × 1.2.
2. Determine the ripple current amplitude (peak-to-peak value). As
a general rule, ripple current should be kept between 10% and
30% of the average LED current:
0.1 < iRIPPLE(pk-pk) / iLED < 0.3.
3. Calculate the inductance based on the following equations:
L = (VIN – VOUT ) × D × T / iRIPPLE , and
D = (VOUT + VD1 ) / ( VIN + VD1 ) ,
where
D is the duty cycle,
T is the period 1/ fSW , and
VD1 is the forward voltage drop of the Schottky diode
D1 (see figure 7).
Inductor Selection Chart
The chart in Figure 13 summarizes the relationship between
LED current, switching frequency, and inductor value. Based on
this chart: Assuming LED current = 2 A and fSW =1 MHz, then
VIN
2.0
1.8
Switching Frequency, f SW (MHz)
Component Selections
The inductor is often the most critical component in a buck converter. Follow the procedure below to derive the correct parameters for the inductor:
1.6
1.4
1.2
L=10 µH
1.0
L=15 µH
0.8
L=22 µH
0.6
L=33 µH
0.4
L=47 µH
0.2
0
0.0
0.5
1.0
1.5
LED Current, ILED
2.0
2.5
3.0
(A)
Figure 13: Inductance selection based on ILED and fSW ;
VIN = 24 V, VOUT = 12 V, ripple current = 30%
the minimum inductance required is L = 10 µH in order to keep
the ripple current at 30% or lower. (Note: VOUT = VIN / 2 is the
worst case for ripple current). If the switching frequency is lower,
then either a larger inductance must be used, or the ripple current
requirement has to be relaxed.
VIN
L1
LED+
Iripple
SW
D1
L1
LED+
Iripple
SW
D1
CS
Vripple
...
...
C1
CS
LED–
RSENSE
Without output capacitor:
Ripple current through LED string is proportional to ripple voltage
at CS pin.
Vripple
LED–
RSENSE
With a small capacitor across LED string:
Ripple current through LED string is reduced, while ripple voltage
at CS pin remains high.
Figure 14. Ripple current and voltage, with and without shunt capacitor
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115 Northeast Cutoff
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11
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Additional Notes on Ripple Current
• For consistent switching frequency, it is recommended to
choose the inductor and switching frequency to ensure the inductor ripple current percentage is at least 10% over normal operating voltage range (ripple current is lowest at lowest VIN).
If ripple current is less than 10%, the switching frequency may
jitter due to insufficient ripple voltage at CS pin. However, the
average LED current is still regulated.
• There is no hard limit on the highest ripple current percentage
allowed. A 60% ripple current is still acceptable, as long as both
the inductor and LEDs can handle the peak current (average current × 1.3 in this case). However, care must be taken to ensure
the valley of the inductor ripple current never drops to zero at the
highest input voltage (which implies a 200% ripple current).
• In general, allowing a higher ripple current percentage enables
lower-inductance inductors to be used, which results in smaller
size and lower cost. The only down-side is the core loss of the
inductor increases with larger ripple currents. But this is typically
a small factor.
Output Filter Capacitor
The A6213 is designed to operate without an output filter capacitor, in order to save cost. Adding a large output capacitor is not
recommended.
In some applications, it may be required to add a small filter
capacitor (up to several µF) across the LED string (between
LED+ and LED-) to reduce output ripple voltage and current. It is
important to note that:
• The effectiveness of this filter capacitor depends on many factors, such as: switching frequency, inductors used, PCB layout,
LED voltage and current, and so forth.
• The addition of this filter capacitor introduces a longer delay
in LED current during PWM dimming operation. Therefore the
maximum PWM dimming ratio is reduced.
• The filter capacitor should NOT be connected between LED+
and GND. Doing so may create instability because the control
loop must detect a certain amount of ripple current at the CS pin
for regulation.
• If lower ripple current is required for the LED string, one solution is to add a small capacitor (such as 2.2 µF) across the LED
string from LED+ to LED– . In this case, the inductor ripple current remains high while the LED ripple current is greatly reduced.
VIN
VIN
VOUT
VOUT
iLED
C1,C2
C3
VEN
C4
C1,C2
iLED
C3
VEN
C4
t
Panel 15A: Operation without using any output
capacitor across the LED string
t
Panel 15B: Operation with a 0.68 µF ceramic
capacitor connected across the LED string
Figure 15: Waveforms showing the effects of adding a small filter capacitor across the LED string
• Operating conditions: at 200 Hz, VIN = 24 V, VOUT = 15 V, fSW = 500 kHz, L = 10 µH, duty cycle = 50%
• CH1 (Red) = VIN (10 V/div), CH2 (Blue) = VOUT (10 V/div),
CH3 (Green) = iLED (500 mA/div), CH4 (Yellow) = Enable (5 V/div), time scale = 1 ms/div
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12
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Application Circuit
The application circuit in Figure 16 shows a design for driving
a 15 V LED string at 1.3 A (set by RSENSE ). The switching frequency is 500 kHz, as set by R1. A 0.68 µF ceramic capacitor is
added across the LED string to reduce the ripple current through
the LEDs (as shown in Figure 15B).
Suggested Components
Symbol
Part Number
Manufacturer
C1
EMZA500ADA470MF80G
United Chemi-Con
C2
UMK316BJ475KL-T
Taiyo Yuden
C3
CGA5L2X5R1H684K160AA
TDK
L1
NR8040T100M
Taiyo Yuden
D1
B250A-13-F
Diodes, Inc.
RSENSE
RL1632R-R150-F
Susumu
VIN = 24 to 48 V
C1
47 µF
50 V
GND
L1
10 µH / 2 A
C2
4.7µF
50V
1
R1
3
4
TON
A6213
EN
SW
BOOT
C4
0.1 µF
7
D1
60 V / 2 A
GND 6
PAD
CS
8
VCC
C3
0.68 µF
50 V
5
C5
0.1 µF
...
140 kΩ
EN
2
VIN
LED+
LED
string
(≈15 V)
LED–
RSENSE
0.15 Ω
Figure 16: Application Circuit Diagram
Additional Application Circuits
The following are some application examples to expand the capability of the A6213:
• Figure 17 shows PWM dimming of LED current by pulsing the
power supply line
VBAT
• Figure 18 shows analog dimming of LED current by an external DC voltage
• Figure 19 shows thermal de-rating of LED current by an NTC
resistor
VIN
LED+
A6213
GND
1
2
3
12 V
100 kΩ
4
VIN
TON
SW
BOOT
EN
GND
CS
VCC
8
7
LED
String
(~6 V)
6
5
VBAT
LED–
0
LED
Current
1A
VBAT pulsed on/off at 200 Hz, with duty cycle
between 1 % and 99%
RSENSE
0.2 Ω
0
Figure 17: PWM Dimming of LED Current by Using Pulsed Power Supply Line
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13
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
VIN = 12 V
C1
47 µF
50 V
L1
47 µH 2 A
C2
4.7 µF
50 V
A6213
GND
1
R1
2
200 kΩ
EN
3
4
VCS = 0.2 V
Analog Dimming
Voltage: 0 to 5.2 V
ADIM
25 kΩ
SW
VIN
BOOT
TON
EN
GND
CS
VCC
8
7
C4
0.1 µF
D1
60 V 2 A
6
LED
String
(~6 V)
C3
open
5
C5
0.1 µF
1 kΩ
iLED:
1.04 A to 0 A
LED+
LED–
VSENSE:
0.22 V to 0 V
iADIM
RSENSE
0.2 Ω
iLED
iLED = (0.2 V – iADIM × 1000)/RSENSE
iADIM = (VADIM – 0.2)/25 k
100%
ADIM
0
0.2 V
5.2 V
Figure 18: Analog Dimming of LED Current with an External DC Voltage
C1
47 µF
50 V
L1
47 µH 2 A
C2
4.7 µF
50 V
GND
A6213
1
R1
2
200 kΩ
3
VCS = 0.2 V
NTC:
220 k @ 25ºC
22 k @ 100ºC 30 kΩ
4
VCC = 5.2 V
iADIM:
0.02 mA @ 25ºC
0.096 mA @ 100ºC
LED+
1 kΩ
VIN
TON
SW
BOOT
EN
GND
CS
VCC
8
7
C4
0.1 µF
0.9 A @ 25ºC
0.52 A @ 100ºC
D1
60 V 2 A
6
LED
String
(~6 V)
C3
open
5
C5
0.1 µF
LED–
VSENSE:
0.18 V @ 25ºC
0.104 V @ 100ºC
RSENSE
0.2 Ω
iLED = (0.2 V – iADIM × 1000)/RSENSE
iADIM = (VCC – 0.2)/(RNTC + 30 k)
Figure 19: Thermal Foldback of LED Current Using NTC Resistor
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14
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Component Placement and PCB Layout Guidelines
PCB layout is critical in designing any switching regulator. A
good layout reduces emitted noise from the switching device,
and ensures better thermal performance and higher efficiency.
The following guidelines help to obtain a high quality PCB
layout. Figure 20 shows an example for components placement.
Figure 21 shows the three critical current loops that should be
minimized and connected by relatively wide traces.
1) When the upper FET (integrated inside the A6213) is on, current flows from the input supply/capacitors, through the upper
FET, into the load via the output inductor, and back to ground as
shown in loop 1. This loop should have relatively wide traces.
Ideally this connection is made on both the top (component) layer
and via the ground plane.
2) When the upper FET is off, free-wheeling current flows from
ground through the asynchronous diode D1, into the load via the
output inductor, and back to ground as shown in loop 2. This loop
should also be minimized and have relatively wide traces. Ideally
this connection is made on both the top (component) layer and
via the ground plane.
3) The highest di/dt occurs at the instant the upper FET turns on
and the asynchronous diode D1 undergoes reverse recovery as
shown in loop 3. The ceramic input capacitors C2 must deliver
this high instantaneous current. C1 (electrolytic capacitor) should
not be too far off C2. Therefore, the loop from the ceramic input
capacitor through the upper FET and asynchronous diode to
ground should be minimized. Ideally this connection is made on
both the top (component) layer and via the ground plane.
4) The voltage on the SW node (pin 8) transitions from 0 V to
VIN very quickly and may cause noise issues. It is best to place
the asynchronous diode and output inductor close to the A6213 to
minimize the size of the SW polygon.
Keep sensitive analog signals (CS, and R1 of switching frequency setting) away from the SW polygon.
6) For accurate current sensing, the LED current sense resistor
RSENSE should be placed close to the IC.
7) Place the boot strap capacitor C4 near the BOOT node (pin 7)
and keep the routing to this capacitor short.
8) When routing the input and output capacitors (C1, C2, and C3
if used), use multiple vias to the ground plane and place the vias
as close as possible to the A6213 pads.
9) To minimize PCB losses and improve system efficiency, the
input (VIN) and output (VOUT) traces should be wide and duplicated on multiple layers, if possible.
Loop 1
Loop 2
Loop 3
L1
SW
VIN
Figure 20: Example layout for the A6213 evaluation board
CIN
D1
COUT
LED
Figure 21: Three different current loops in a buck converter
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15
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
10) Connection to the LED array should be kept short. Excessively long wires can cause ringing or oscillation. When the LED
array is separated from the converter board and an output capacitor is used, the capacitor should be placed on the converter board
to reduce the effect of stray inductance from long wires.
Optimizing Thermal Layout
The features of the printed circuit board, including heat conduction and adjacent thermal sources such as other components,
have a very significant effect on the thermal performance of the
device. To optimize thermal performance, the following should
be taken into account:
Thermal Dissipation
The amount of heat that can pass from the silicon of the A6213
to the surrounding ambient environment depends on the thermal
resistance of the structures connected to the A6213. The thermal
resistance, RθJA , is a measure of the temperature rise created by
power dissipation and is usually measured in degrees Celsius per
watt (°C/W).
•The device exposed thermal pad should be connected to as
much copper area as is available.
The temperature rise, ΔT, is calculated from the power dissipated,
PD , and the thermal resistance, RθJA , as:
ΔT = PD × RθJA
A thermal resistance from silicon to ambient, RθJA , of approximately 35°C/W can be achieved by mounting the A6213 on a
standard FR4 double-sided printed circuit board (PCB) with a
copper area of a few square inches on each side of the board
under the A6213. Additional improvements in the range of 20%
may be achieved by optimizing the PCB design.
•Copper thickness should be as high as possible (for example,
2 oz. or greater for higher power applications).
•The greater the quantity of thermal vias, the better the dissipation. If the expense of vias is a concern, studies have shown
that concentrating the vias directly under the device in a tight
pattern, as shown in Figure 22, has the greatest effect.
•Additional exposed copper area on the opposite side of the
board should be connected by means of the thermal vias. The
copper should cover as much area as possible.
•Other thermal sources should be placed as remote from the
device as possible
•Place as many vias as possible to the ground plane around the
anode of the asynchronous diode.
Signal traces
LJ package
footprint
0.7 mm
0.7 mm
LJ package
exposed
thermal pad
Top-layer
exposed copper
Ø0.3 mm via
Figure 22: Suggested PCB layout for thermal optimization (maximum
available bottom-layer copper recommended)
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115 Northeast Cutoff
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16
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Package LJ, 8-Pin Narrow SOIC
with Exposed Thermal Pad
4.90 ±0.10
0.65
8°
0°
1.75
0.25
0.17
B
2.41 NOM
3.90 ±0.10
6.00 ±0.20
2
1.27
0.40
3.30 NOM
0.51
0.31
1.27 BSC
2
SEATING
PLANE
PCB Layout Reference View
C
For Reference Only; not for tooling use (reference MS-012BA)
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
1.70 MAX
0.15
0.00
C
SEATING PLANE
GAUGE PLANE
Branded Face
0.10 C
1
5.60
3.30
0.25 BSC
8X
2.41
1.04 REF
A
1
1.27
8
8
A
Terminal #1 mark area
B
Exposed thermal pad (bottom surface); dimensions may vary with device
C
Reference land pattern layout (reference IPC7351 SOIC127P600X175-9AM);
all pads a minimum of 0.20 mm from all adjacent pads; adjust as necessary
to meet application process requirements and PCB layout tolerances; when
mounting on a multilayer PCB, thermal vias at the exposed thermal pad land
can improve thermal dissipation (reference EIA/JEDEC Standard JESD51-5)
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115 Northeast Cutoff
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17
A6213 and
A6213-1
Automotive-Grade, Constant-Current
PWM Dimmable Buck Regulator LED Driver
Revision History
Revision
Revision Date
1
October 21, 2013
2
March 27, 2014
3
April 25, 2014
4
December 31, 2014
5
January 29, 2016
6
March 17, 2016
Description of Revision
Updated Tstg.
Revised VCC Spec and Suggested Components Table.
Added new Figure 11.
Added new application circuit diagrams.
Revised VCC Absolute Maximum Rating from 14 to 7 V.
Added Load Current Sense Regulation Threshold footnote (page 4); updated Additional Notes on
Ripple Current (page 12).
Copyright ©2016, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in any devices or systems, including but not limited to life support devices or systems, in which a failure of
Allegro’s product can reasonably be expected to cause bodily harm.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
www.allegromicro.com
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1.508.853.5000; www.allegromicro.com
18
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