A6266 Datasheet

A6266
Automotive, Boost, High Current LED Controller
Features and Benefits
Description
▪ AEC Q100 Grade 0 Automotive Qualified
▪ Constant current LED drive
▪ 5 to 50 V supply
▪ Boost mode
▪ Drives up to 15 LEDs in series
▪ Programmable switching frequency 100 to 700 kHz
▪ Open LED overvoltage indication and protection
▪ Single and multiple LED short indication
▪ LED short to ground and supply protection
▪ PWM dimming control
The A6266 is a boost converter controller, providing a
programmable constant current output for driving high power
LEDs in series. Driving the LEDs in series ensures identical
currents and uniform brightness. For automotive applications,
optimum performance is achieved when driving up to 15 LEDs
at currents up to 1 A.
The A6266 provides a cost-effective solution using an external
logic-level MOSFET and minimum additional external
components. The maximum LED current is set with a single
external sense resistor and can be modified using a current
reference input. Direct PWM control is possible via the Enable
input, which also provides a shutdown mode.
Applications:
▪ Automotive high power LED lighting systems
▪ Fog lights, reversing lights, daytime running lights
▪ Headlights
Integrated diagnostics and two fault outputs give indication of
undervoltage, chip overtemperature, output open circuit, LED
short circuit and LED undercurrent, and can be configured to
provide short to supply and short to ground protection for the
LED connections. A unique feature is the ability to detect one
or more shorted LEDs.
Package: 16-pin TSSOP with exposed
thermal pad (suffix LP)
The device is provided in a 16-pin TSSOP package with exposed
thermal pad (suffix LP). It is lead (Pb) free, with 100% matte
tin leadframe plating.
Not to scale
Typical Application Diagram
VBAT 12 V or 24 V(50 V max)
Power net
Boost Mode
VBAT(min)
Maximum
(V)
Quantity of
LEDs
6
10
7
12
8
13
9
15
Vf of each LED = 3.5 V,
D = 85%
VIN
VREG
Fault
Flags
FF1
FF2
Enable
EN
LN
LP
A6266
SG
IREF
CKOUT
SP
OSC
SN
GND
A6266-DS, Rev. 3
LF
LA
A6266
Automotive, Boost, High Current LED Controller
Selection Guide
Part Number
A6266KLPTR-T
Packing
Package
4000 pieces per 13-in. reel
16-pin TSSOP with exposed thermal pad
Absolute Maximum Ratings With respect to GND at TA = 25°C, unless otherwise specified
Characteristic
Rating
Unit
–0.3 to 50
V
Pins FF1, EN
–0.3 to 50
V
Pins FF2, CKOUT
–0.3 to 6.5
V
Pin OSC
–0.3 to 6.5
V
Pin SG
–0.3 to 6.5
V
–0.3 to 7
V
Load Supply Voltage
Symbol
Notes
VIN
Pins LA, LN
Pin LF
With respect to LA
–6 to 6
V
Pin LP
With respect to LN
–6 to 6
V
Pin SP, SN
–0.3 to 5
V
Pin VREG
–0.3 to 7
V
Pin IREF
Junction Temperature
–0.3 to 7
V
150
°C
TJ(max)
Storage Temperature Range
Tstg
Operating Temperature Range
TA
Range K
–55 to 150
°C
–40 to 150
°C
Thermal Characteristics may require derating at maximum conditions, see application information
Characteristic
Symbol
Package Thermal Resistance
(Junction to Ambient)
RθJA
Package Thermal Resistance
(Junction to Exposed Pad)
RθJP
Test Conditions*
On 4-layer PCB based on JEDEC standard
On 2-layer PCB with 3.8
in.2
of copper area each side
Value
Unit
34
ºC/W
43
ºC/W
2
ºC/W
*Additional thermal information available on the Allegro website
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
A6266
Automotive, Boost, High Current LED Controller
Pin-out Diagram
EN 1
16 LF
FF1 2
15 LA
FF2 3
14 LN
CKOUT 4
IREF 5
PAD
OSC 6
13 LP
12 VIN
11 VREG
SN 7
10 GND
SP 8
9 SG
Terminal List Table
Number
Name
Function
1
EN
Enable chip
2
FF1
Fault flag
3
FF2
4
CKOUT
Fault flag
5
IREF
Current reference
6
OSC
Oscillator input/frequency set
7
SN
Switch current sense –ve input
8
SP
Switch current sense +ve input
9
SG
Switch gate drive
Oscillator output, with phase shift
10
GND
11
VREG
12
VIN
Main supply
13
LP
Load current sense +ve input
14
LN
Load current sense –ve input
15
LA
LED string voltage sense
16
LF
–
PAD
Ground
Internal regulator capacitor
Reference LED voltage sense
Exposed thermal pad
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
3
A6266
Automotive, Boost, High Current LED Controller
Functional Block Diagram
VBAT
Boost
VIN
LN
LF
LP
VREG
LED Open and
Short Detect
VREG
AL
FF2
FF1
EN
Fault
Detect
AE
Shutdown
Control
Logic
LA
AC

IREF
R
S
SG
Q
SP

RSS
AS
Temp
Monitor
IREF
Slope
Gen
Osc
OSC
SN
RSL
GND
CKOUT
ROSC
AL AE AC AS
See Functional Description section.
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
A6266
Automotive, Boost, High Current LED Controller
ELECTRICAL CHARACTERISTICS1 Valid at TJ = –40°C to 150°C, VIN = 6 to 40 V; unless otherwise noted
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
5
–
50
V
Supply and Reference
VIN Functional Operating Range2
VIN Quiescent Current
VREG Output Voltage
IINQ
SG open circuit
–
–
8
mA
IINS
EN = GND
–
4
10
μA
4.75
5
5.25
V
VREG
IREG = 0 to 2 mA
Gate Output Drive
Turn-On Time
tr
CLOAD = 1 nF, 20% to 80%
–
35
–
ns
Turn-Off Time
tf
CLOAD = 1 nF, 80% to 20%
–
35
–
ns
Maximum Duty Cycle
D
tON × fOSC
80
85
–
%
TJ = 25°C, IGHx = –100 mA
–
1.7
–
Ω
TJ = 150°C, IGHx = –100 mA
–
–
3.5
Ω
TJ = 25°C, IGLx = 100 mA
–
1
–
Ω
TJ = 150°C, IGLx = 100 mA
–
–
2
Ω
Pull-Up On Resistance
RDS(on)UP
Pull-Down On Resistance
RDS(on)DN
Output High Voltage
VSGH
ISG = –100 μA
VREG –
0.1
–
VREG
V
Output Low Voltage
VSGL
ISG = 100 μA
–
–
0.1
V
VOL
Logic Inputs and Outputs
Fault Output (Open Drain)
IOL = 1 mA, fault not asserted
–
–
0.4
V
Fault Output FF1 Sink Current1
IOH(snk)
0.4 V < VO < 50 V, fault not asserted
–
1.3
–
mA
Fault Output FF1 Leakage Current1
IOH1(lkg)
VO = 12 V, fault asserted
–1
–
1
μA
Current1
IOH2(lkg)
VO = 5 V, fault asserted
–5
–
5
μA
–
–
0.8
V
Fault Output FF2 Leakage
Input Low Voltage
VIL
Input High Voltage
VIH
2
–
–
V
Input Hysteresis
VIhys
225
330
–
mV
Enable Input Internal Clamp Voltage
VENC
–
8.4
–
V
Enable Input Current Limit Resistor
REN
Between EN and internal clamp
–
200
–
kΩ
Disable Time
tDIS
fOSC = 350 kHz
–
94
–
ms
Oscillator
Oscillator Frequency
Minimum Oscillator
Frequency3
OSC Pin Voltage
fOSC
ROSC = 43 kΩ
–
500
–
kHz
ROSC = 62 kΩ
315
350
385
kHz
ROSC = VREG
–
350
–
kHz
90
–
–
kHz
ROSC = 62 kΩ
1.15
1.2
1.25
V
fMIN
VOSC
CKOUT Output Delay
tDC
–
150
–
ns
OSC Input Low Voltage
VOIL
OSC input rise to CKOUT rise
–
–
0.8
V
OSC Input High Voltage
VOIH
3.5
–
–
V
Continued on the next page…
Allegro MicroSystems, LLC
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
A6266
Automotive, Boost, High Current LED Controller
ELECTRICAL CHARACTERISTICS1 (continued) Valid at TJ = –40°C to 150°C, VIN = 6 to 40 V; unless otherwise noted
Characteristics
Symbol
Test Conditions
Min.
Typ.
Max.
Unit
300
600
–
mV
Oscillator (continued)
OSC Input Hysteresis
VOihys
OSC Watchdog Period
tOSWD
Between successive rising edges
CKOUT Output High Voltage
VCOH
IOH = –1 mA
CKOUT Output Low Voltage
VCOL
7
–
–
μs
VREG – 1
–
VREG
V
IOL = 1 mA
–
–
0.4
V
LP = LN = 0 V
–
–2
–
μA
LED Current Sense
Input Bias Current LN
Input Bias Current LP
ILN
ILP
LP = LN = 0 V
–
–25
–
μA
VIDL
EN = High, VIDL = VLP – VLN
–
100
–
mV
Differential Input Voltage (PWM off)
VIDLO
EN = Low, VIDL = VLP – VLN
–
8
–
mV
Input Common-Mode Range
VCMLL
VLP = VLN
0
–
1
V
Current Error
EISL
[(10 × ILED × RSL) – 1] × 100
–5
–
5
%
Differential Input Voltage (Active)
Switch Current Sense
Input Bias Current
IBIASS
SP = SN = 0 to 2 V
–30
–
–
μA
Maximum Differential Input Voltage3
VIDS
VIDS = VSP – VSN with D = 50%
330
410
490
mV
IINS
VIDS = 400 mV
–
400
–
μA
VSP = VSN
0
–
2
V
Input Source Current
Input Common-Mode Range
VCMS
Diagnostics and Protection
Fault Blank Timer4
VREG Undervoltage Turn-Off
VREG Undervoltage Hysteresis
tFB
VREGUV
Start-up
–
3
–
ms
Decreasing VREG
3.3
3.8
4.0
V
VREGUVhys
100
120
–
mV
LED String Short Voltage
VSCL
430
505
580
mV
Non-Reference LED Short Offset
Voltage
VSCO
160
200
240
mV
Reference LED Short Offset Voltage
VSCOR
430
505
580
mV
VOCL
5
5.5
6
V
LED Open Voltage
LF Bias Current
ILF
LF = 1.7 V
–
8
–
μA
LA Bias Current
ILA
LA = 1.7 V
–
34
–
μA
–
1
–
mV
LED Undercurrent Voltage Difference5
VUCL
Open Fault Time-Out
tOTO
fOSC = 350 kHz
–
94
–
ms
Overtemperature Warning Threshold
TJF
Temperature increasing
–
170
–
ºC
TJhys
Recovery = TJF – TJhys
–
15
–
ºC
Overtemperature Hysteresis
1For
input and output current specifications, negative current is defined as coming out of (sourcing) the specified device pin.
2Function is correct but parameters are not guaranteed below the general limit (6 V).
3Parameters ensured by design.
4Fault Blank timer not enabled for open-LED condition.
5Undercurrent when V
SENSEL < VIDL– VUCL , where VSENSEL is the voltage across the LED current sense resistor RSL.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
6
A6266
Automotive, Boost, High Current LED Controller
Functional Description
The A6266 is a boost converter controller that is designed to
drive series-connected high power LEDs in automotive applications. It provides programmable constant current output at load
voltages and currents limited only by the external components.
For automotive applications optimum performance is achieved
when driving up to 15 LEDs at currents up to 1 A.
indicates detection of a circuit fault. An external pull-up resistor
should be connected to a suitable logic supply. If VREG is not
used, then the logic supply should not be pulled 300 mV above
VREG. Table 1 defines when FF2 is active. If FF2 is pulled low
when an open LED fault is indicated then the output disable will
be overridden.
The A6266 integrates all necessary control elements to provide a cost-effective solution using a single external logic-level
MOSFET and minimum additional external passive components.
OSC Resistor to ground to set the internal oscillator or clock
input from external oscillator. When connected to VREG the
oscillator runs at typically 350 kHz. Higher accuracy in the
frequency is possible by connecting a resistor from this pin to
ground or by driving this pin with an external precision oscillator.
The LED current is set by selecting an appropriate value for the
sense resistor value and using the EN input to provide simple
on-off control or for PWM brightness control using a suitable
externally generated PWM signal. The LED current can be
reduced in a single step by reducing the voltage between the
IREF pin and GND to less than 1 V.
The pin functions and circuit operation are described in detail in
the following sections.
Pin Functions
VIN Supply to the control circuit. A bypass capacitor must be
connected between this pin and GND.
CKOUT Logic output at the oscillator frequency with phase
shift. Used to drive succeeding controllers to interleave switching
instants.
IREF LED current reference modifier. A voltage input that can be
used to reduce the LED current sense voltage. When connected to
VREG, the current sense voltage, VIDL, and the value of the sense
resistor, RSL , define the maximum LED current.
SG Gate drive for external logic-level MOSFET low-side switch
that connects the inductor to ground.
GND Ground reference connection. This pin should be connected
directly to the negative supply.
SP, SN Sense amplifier connections for switch current limit
sense resistor, RSS .
EN Logic input to enable operation. Can be used as direct PWM
input. Chip enters low power sleep mode when low for longer
than the disable time, tDIS.
LP Positive sense amplifier connection for LED current limit
sense resistor, RSL . This pin is also the bias supply for the LED
current sense amplifier.
FF1 Fault Flag output and isolation control. Open drain current
sink output, when high impedance indicates detection of a critical
circuit fault. An external pull-up resistor should be connected to a
suitable logic supply for simple logic fault flag operation or to the
source of the PMOS FET used to isolate the load from the supply.
Table 1 defines when FF1 is active. If FF1 is pulled low when
an output short fault is indicated then the output disable will be
overridden.
LN Negative sense amplifier connection for LED current limit
sense resistor, RSL .
FF2 Fault Flag output. Open drain output, when high impedance
VREG Compensation capacitor for internal 5 V regulator.
LA Anode reference connection to LEDs. Using an external resistor divider with the same ratio as the number of LEDs provides
a measurement of the voltage across all LEDs in the load. This
is compared to the voltage on the LF pin to provide shorted LED
detection. In addition, it is compared against voltage references to
provide open circuit or shorted LED string detection.
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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7
Automotive, Boost, High Current LED Controller
LF Single diode forward voltage reference input. Measures the
forward voltage of the first LED. This value is used as a reference
against the voltage on the LA pin to detect possible shorted LEDs
in the LED string.
Circuit Operation
Converter A constant frequency, current mode control scheme
is used to regulate the current through the LEDs. There are two
control loops within the regulator. The inner loop formed by the
amplifier, AS (see the Functional Block Diagram for AS, AC, AE,
and AL), comparator, AC, and the RS bistable, controls the inductor current as measured through the switch by the switch sense
resistor, RSS .
The outer loop including the amplifier, AL, and the integrating
error amplifier, AE, controls the average LED current by providing a setpoint reference for the inner loop.
The LED current is measured by the LED sense resistor, RSL ,
and compared to the internal reference current to produce an
integrated error signal at the output of AE. This error signal sets
the average amount of energy required from the inductor by the
LEDs. The average inductor energy transferred to the LEDs is
defined by the average inductor current as determined by the
inner control loop.
The inner loop establishes the average inductor current by
controlling the peak switch current on a cycle-by-cycle basis.
Because the relationship between peak current and average current is non-linear, depending on the duty cycle, the reference
level for the peak switch current is modified by a slope generator.
This compensation reduces the peak switch current measurement
by a small amount as the duty cycle increases (refer to figure 1).
The slope compensation also removes the instability inherent in a
fixed frequency current control scheme.
The control loops work together as follows: the switch current,
sensed by the switch current sense resistor, RSS , is compared
to the LED current error signal. As the LED current increases
the output of AE will reduce, reducing the peak switch current
and thus the current delivered to the LEDs. As the LED current
decreases the output of AE increases, increasing the peak switch
current and thus increasing the current delivered to the LEDs.
Under some conditions, especially when the LED current is set to
a low value, the energy required in the inductor may result in the
inductor current dropping to zero for part of each cycle. This is
known as discontinuous mode operation, and results in some low
frequency ripple. The average LED current, however, remains
regulated down to zero. In discontinuous mode, when the inductor current drops to zero, the voltage at the drain of the external
MOSFET rings, due to the resonant LC circuit formed by the
inductor, and the switch and diode capacitance. This ringing is
low frequency and is not harmful.
Switch Current Limit The switch current is measured by the
switch sense resistor, RSS , and the switch sense amplifier, AS
(see the Functional Block Diagram). The input limit of the sense
amplifier, VIDS , and the maximum switch current, ISMAX , define
the maximum value of the sense resistor as:
RSS = VIDS / ISMAX
(1)
This defines the maximum measurable value of the switch (and
inductor) current.
The maximum switch current is modulated by the on-time of the
switch. An internal slope compensation signal is subtracted from
the voltage sense signal to produce a peak sense voltage which
effectively defines the current limit. This signal is applied at a
rate of –50 mV / μs starting with no contribution (t = 0 μs) at the
beginning of each switching cycle. Figure 1 illustrates how the
peak sense voltage (typical values) changes over a period of 3 μs.
For example, the maximum current (typical) through the switch
at t = 1.5 μs (D = 50%) would be 410 mV/RSS , however, if the
switch remained on for a further 1 μs, the maximum current
through the switch would be 360 mV/RSS .
Peak Sense Votlage (mV)
A6266
500
450
400
350
300
250
200
150
100
50
0
0
1
2
3
Period (μs)
Figure 1. Slope compensation for peak switch current control.
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Worcester, Massachusetts 01615-0036 U.S.A.
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8
Automotive, Boost, High Current LED Controller
LED Current Level The LED current is determined by a
combination of the LED sense resistor, RSL , the LED current
threshold voltage, VIDL , and the voltage between the IREF pin
and GND ( VIREF ).
The 100% current level, when the IREF pin is connected to
VREG, is defined as:
ILED(max) = VIDL / RSL
(2)
If VIREF is less than 1 V then the 100% current level is defined as:
ILED(max) = VIREF / (10 × RSL )
(3)
This feature provides direct analog dimming using a voltage from
0 to 1 V. This can be used to provide intensity-matching between
modules or groups of LEDs in critical display or backlighting
applications. It can also be used to provide a soft start, by connecting a capacitor from IREF to GND and a resistor from IREF
to VREG, or one-step dimming by use of a single logic control.
LED Brightness: PWM Dimming LED brightness can
be controlled by changing the current, which affects the light
intensity. However in some applications, for example with amber
LEDs, this will have some effect on the color of the LEDs.
In these cases it is more desirable to control the brightness by
switching the fixed LED current with a pulse width modulated
signal. This allows the LED brightness to be set with little effect
on the LED color and intensity and allows direct digital control
of the LED brightness.
A PWM signal can be applied to the EN input to enable PWM
dimming. The period of this signal should be less than the
minimum disable time, tDIS . During PWM dimming, the A6266
switches the LED current between 100% and typically 8% of
the full current. This ensures that the voltage change across the
LED string is limited to a few volts, depending on the number
of LEDs. This limits the stress on the load capacitor (across
the string of LEDs) due to large changes in voltage. If the load
capacitor is a multilayer ceramic type, then this will reduce any
audible noise due to the piezoelectric effect of the capacitor.
The rate of change of the LED current is also limited, to reduce
any large variations in the input current.
Sleep Mode If EN is held low for longer than the disable time,
tDIS , then the A6266 will shut down and put all sections into a
low-power sleep mode. In this mode the bias current is typically
less than 4 μA.
Note that the disable time is derived from the oscillator period by
a ratio of 32,768, so any variation in the oscillator frequency will
change the disable time.
Oscillator The main oscillator may be configured as a clock
source or it may be driven by an external clock signal. The oscillator is designed to run between 100 and 700 kHz.
When the oscillator is configured as a clock source, the frequency
is controlled by a single external resistor, ROSC (kΩ), between the
OSC pin and the GND pin. The oscillator frequency is approximately:
fOSC = 21700 / ROSC
(kHz)
(4)
Figure 2 shows the resulting fOSC for various values of ROSC.
If the OSC pin is connected to VREG or GND, the oscillator
frequency will be set internally to approximately 350 kHz.
When an external clock source is used to drive the OSC pin, it
can synchronize a number of A6266s operating together. This
ensures that only a single fundamental frequency is detectable
on the supply line, thus simplifying the design of any required
EMC filter. The disadvantage of using a single external clock
source is that all controllers will be switching current from the
supply at the same time. However, this effect may be reduced,
and the EMC performance may be further enhanced, by using
the CKOUT pin of another A6266 as the external clock source.
In this case the switching point of each subsequent A6266 in the
chain will be delayed from that of the previous A6266, and the
current pulses will be spread across the oscillator period.
Oscillator Frequency, fOSC (kHz)
A6266
700
600
500
400
300
200
100
30
50
70
90
110 130 150 170
External Resistor Value, ROSC (kΩ)
190
210
Figure 2. Internal oscillator frequency when set by ROSC
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9
A6266
Automotive, Boost, High Current LED Controller
Diagnostics
The circuit includes several diagnostic and safety functions to
assist in ensuring safe operation of the LEDs, the A6266, and the
external components. When any fault is detected, one or both of
the fault flag outputs, FF1 and FF2, will be inactive (high impedance, open drain) until the fault is removed. The action taken by
the A6266 when a fault occurs is defined in table 1. To be able to
monitor the state of FF1 and FF2, add a suitable external pull-up
resistor.
The A6266 will continue to drive the LEDs under most fault conditions and will only disable the drive to the LEDs when a high
voltage hazard is present or the external components are likely to
be over-stressed. For output short circuits or open LED conditions, the fault status is latched until EN is taken low or a power
cycle occurs. For output short circuits or a shorted LED string,
the fault status is latched until either EN is taken low for a period
greater than the disable time, or a power cycle occurs.
At start-up, a Fault Blank period, tFB , occurs before the fault
detection circuitry becomes active. This period allows steady
state conditions to be established before fault monitoring takes
place.
Note that no fault blanking is applied to open LED faults. This
is generally not an issue because the charging of the output filter
capacitor provides a degree of filtering. In addition, extremely
high voltages are prevented from causing potential device breakdown, for example in the external switching MOSFET.
VREG Undervoltage If the voltage at VREG, VREG , drops
below the specified turnoff voltage, VREGUV , the gate drive
LED Undercurrent Under some circuit conditions, particularly
during a low input voltage condition, it is possible that there
could be insufficient drive to maintain the current to the LEDs
at the required level. If the voltage across the LED current sense
resistor, RSS , falls below the target sense voltage, VIDL , by an
amount that is more than the LED undercurrent voltage difference, VUCL , the A6266 will indicate an LED undercurrent condition by setting FF2 to high impedance. However, the A6266 will
continue to drive the output. When the output again reaches the
required current level, FF2 will go low.
Overtemperature Warning If the chip temperature exceeds
the overtemperature threshold, TJF , fault flag FF2 will be high
impedance. No action will be taken by the A6266 to limit the
chip temperature. An external control circuit must take action
to avoid permanent damage to the A6266 and/or the LEDs. The
temperature will continue to be monitored and the fault flags will
be deactivated when the temperature drops below the recovery
threshold provided by the hysteresis, TJhys .
LED Diagnostics The status of the LEDs in the load can be
determined by monitoring the voltage with respect to ground at
the three pins LP, LF, and LA, namely VLP , VLF , and VLA . These
voltages provide two differential voltage measurements:
• the voltage across a single reference LED:
VLED = VLF – VLP
Table 1. Fault Table
Fault
output, SG, will be driven low and both fault flags, FF1 and
FF2, will be high impedance. VREG must rise above the turn-on
threshold, VREGUV + ΔVREGUV , before the output circuits are
activated. This ensures that the external FET is operating in its
fully enhanced state and avoids permanent damage to the FET,
caused by overheating.
Pin
Action
Latched
FF1
FF2
No Fault
L
L
No Action
–
VREG Undervoltage
Z
Z
Disable*
No
Output Short
Z
L
Disable*
Yes
LED Undercurrent
L
Z
No Action
No
Overtemperature
L
Z
No Action
No
Open LED
L
Z
Disable*
Yes
Shorted LED
L
Z
No Action
No
Shorted LED String
Z
L
Disable*
Yes
* SG low, MOSFET off
L = active pull-down, Z = inactive, open drain
(5)
• the ratio of the voltage across all LEDs in a single string:
VSTR = VLA – VLP
(6)
The voltage, VSTR , is derived from the voltage across all LEDs in
the string, by an external resistor divider with a ratio equal to the
quantity of LEDs in the string. To minimize the effects of the bias
currents introducing an offset voltage, it is recommended that the
resistor between LP and LA should be approximately 560 Ω.
So for example, if eight LEDs were used, the ratio required
would be an eighth, therefore the resistor connected between LA
and the anode end of the LED string would be 3.9 kΩ;
560 / [560 + 3900] = 1/8 .
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A6266
Automotive, Boost, High Current LED Controller
These measurements are used to determine if there is an open
circuit, if one or more LEDs are shorted, if the output is shorted,
or if there is a short across the LED string. Each condition is
described in turn in the following sections.
Open LED–An open circuit is evaluated when:
VSTR > VOCL
(7)
where VOCL is the LED open circuit voltage defined in the Electrical Characteristics table.
Because the output is current-controlled it is possible for an open
circuit on the output to cause extremely high voltages to be present. Therefore, to prevent any hazardous voltages or damage to
the circuits, the gate drive output, SG, is immediately driven low
when an open circuit is detected. After an open circuit fault has
been detected, FF2 will become high impedance, and the open
circuit fault state will remain until the open fault time-out period
toto, expires. When the gate drive output is re-enabled at the end
of the open fault time-out period, the output is again monitored
for an open circuit. If the open circuit is still present, then the
fault will again be flagged and the switch drive disabled. This
cycle will continue, as long as the open circuit condition is present.
Note that the Fault Blank timer is not used when an open LED
fault occurs. This is to avoid potentially damaging voltages
appearing in the power circuitry.
Shorted LED – A short circuit on one or more LEDs is detected
when:
• for the first (reference) LED:
VSTR > VLED + VSCOR
(8)
• for other than the first (reference) LED:
VLED > VSTR + VSCO
becomes a short circuit, then the remaining LEDs will continue
to be lit with the same current through, and voltage across,
each LED.
Note—Accuracy: The output status monitor relies on all the
LEDs in the load having a similar forward voltage drop. Where
possible all the LEDs forming the load for a single controller
should be taken from the same voltage bin. With only two or
three LEDs a wider variation in forward voltage is acceptable,
but the selection of LEDs from the same bin is more critical when
higher numbers of LEDs are used in a single string.
Shorted LED String or Output Short – A short circuit across
the LED string, is detected when:
VSTR < VSCL
(10)
An output short can consist of the LP, LN, or LF terminals of
the LED string being shorted, either to the battery terminal or
to ground. Either a shorted LED string or output short will be
latched and will only be cleared by pulling EN low for a period
greater than the disable time, or by cycling the power.
If either a shorted LED string, or an output short is detected, the
A6266 will stop the switching action by pulling SG low. The
fault flag FF1 will go high impedance and should be pulled up to
the supply with suitable external pull-up resistors to indicate the
fault. Either a shorted LED string or output short will be latched
and will only be cleared by pulling EN low for a period greater
than the disable time, or by cycling the power.
The FF1 output can also be used with pull-up resistors and a
P-channel MOSFET in the supply, to isolate the switching elements and the load from the supply. This MOSFET should be
connected, as shown in figure 3, with the source connected to the
supply and the drain connected to the inductor of the converter.
(9)
where VSTR and VLED are as defined above, VSCO is the nonreference LED short offset voltage, and VSCOR is the reference LED
short offset voltage. VSCO and VSCOR are defined in the Electrical
Characteristics table.
When a short is present, the fault flag FF2 is high impedance, but
the regulator continues to operate and drives the remaining LEDs
with the correct regulated current. FF2 will remain high impedance while the short circuit condition is present.
A short circuit on one or more LEDs will not cause a hazard
because the output is current-controlled. If one LED fails and
VBAT
To VIN
To FF1
Figure 3. Example of a supply isolation MOSFET
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A6266
Automotive, Boost, High Current LED Controller
Two pull-up resistors are used to limit the voltage across the gatesource junction during high input voltages or load dump conditions. If the battery voltage is restricted, one resistor across the
gate-source junction can be used. The FF1 provides a sink current
of up to 1.3 mA.
This circuit can be used to avoid most hazardous conditions and
protect the circuit components from over-stress. Note that under
extreme cases, the circuit cannot protect against certain fault conditions. For example, when any of the following occurs:
• If the cathode end of the LED string is shorted to VBAT, the
LED sense resistor effectively appears between VBAT and
ground. Depending on the current limit of the source supply or
the input fuse rating, the fault current may damage the resistor.
• If the LF node is shorted to VBAT, the reference LED and
the LED sense resistor effectively appear between VBAT and
ground. Depending on the current limit of the source supply or
the input fuse rating, the fault current may damage the resistor
and /or the reference LED.
• If the cathode end of the LED string is shorted to ground. A
fault current determined by the impedance of the shorting link
(now effectively the LED sense resistor) flows through the
power circuit. The fault current will either: be limited by the
maximum switch current sense, VIDS , or if the source supply
cannot maintain this current, be limited by the source supply.
(The source supply will either: fold back, or if the current
exceeds the input fuse rating, the fuse will blow, creating an
open circuit).
To ensure the A6266 inputs (LP and LN) are not damaged during any of the above faults, it is necessary to add a differential
resistor in series with the LN connection (between the sense
resistor and the LN pin). This resistor value should be approximately 150 Ω.
If an output short is detected but it is necessary to keep the output
active, the FF1 output can be pulled low. This will override the
output disable but will not clear the fault.
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A6266
Automotive, Boost, High Current LED Controller
Application Information
Component Selection
External component selection is critical to the successful application of the LED driver. Although the inductor, the switching
MOSFET, and the output capacitor are the most critical elements,
the specification of the rectifying diode and sense resistors should
also be carefully considered.
The starting point for component selection is to define the maximum LED current, the voltage across the LEDs, and the input
operating voltage range. This then allows the average inductor
current under worst case conditions to be calculated. The inductor value is then selected based on the acceptable inductor ripple
current. The amount of ripple current will then determine the
maximum inductor current under worst case conditions. From
this current the switch current sense resistor can be calculated.
LED Current Sense Resistor (RLS) If the voltage at the
IREF pin, VIREF , is greater than 1V, or if IREF is tied to VREG,
then the value of the LED current sense resistor, RLS , can be
calculated from:
RLS = VIDL / ILED(max)
(11)
where VIDL is the differential voltage across the LED current
sense amplifier and ILED(max) is the maximum LED current.
If VIREF is less than 1 V, then the value of the LED current sense
resistor can be calculated from:
RLS = VIREF / (10 × ILED(max) )
(12)
The typical value for VIDL is 100 mV. Examples of various sense
resistor values are given in table 2.
The power loss in the current sense resistor is worse at the lowest
input voltage:
PLOSS = (VLED / VIN(min) ) × RLS × I2LED
(13)
The resistors should be of a low inductance construction. Surface
mount chip resistors are usually the most suitable, however, axial
or radial leaded resistors can be used provided that the lead length
is kept to a minimum.
Table 2. Sense Resistor Values
ILED(max)
(mA)
RLS
(mΩ)
350
286
700
143
1000
100
Inductor Selection Selecting the correct inductance is a
balance between choosing a value that is small enough to help
reduce size and cost, but high enough to ensure that the inductor
current ripple is kept to an acceptable level. A reasonable target
for the ripple current is 20% of the maximum average current.
The maximum average inductor current is approximately:
IL(av)(max) = ILED(max) × VLED / VIN(min)
(14)
The inductor current ripple is approximately:
ILRIP = VIN × (VLED – VIN ) / (fOSC × L × VLED )
(15)
The inductor value is therefore:
L = VIN × (VLED –VIN ) / (fOSC × ILRIP × VLED ) (16)
where:
VLED is the voltage across the LED string,
VIN is the supply voltage,
VIN(min) is the minimum supply voltage,
L is the inductor value, and
fOSC is the oscillator frequency.
With an internal oscillator frequency of 350 kHz, the value of
the inductor for most cases will be between 20 and 50 μH. The
maximum inductor current can then be calculated as:
IL(PK) = IL(av)(max) + (IRIP / 2)
(17)
This defines the minimum peak switch current as set by the
switch current sense resistor.
The current rating for the inductor should be greater, by some
margin, than the peak value above, IL(PK) . When selecting an
inductor from manufacturers datasheets, there are two current
levels usually defined, the smallest value being the figure to work
with:
• Saturation level, where the inductance value typically drops by
10%, or
• Temperature rise, where the part experiences a certain rise in
temperature at full rated current. This parameter can be defined
between a 20°C and 50°C rise in temperature. It is important to
understand how manufacturers define the maximum operating
temperature, because this can often incorporate the self-heating
temperature rise.
In most cases the limiting current is usually the saturation value.
To improve efficiency, the inductor should also have low winding
resistance, typically < 50 mΩ, and the core material will usually
be ferrite, with low losses at the oscillator frequency.
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A6266
Automotive, Boost, High Current LED Controller
Recommended inductor manufacturers/series are:
External Switch MOSFET A logic-level N-channel MOSFET
is used as the switch for the DC-to-DC converter. The voltage at
the drain of the MOSFET is equal to the maximum voltage across
the string of LEDs.
• Coilcraft/ MSS1278T
• TDK/ SLF12575 type H
Diode The diode should have a low forward voltage, to reduce
conduction losses, and a low capacitance, to reduce switching
losses. Schottky diodes can provide both these features if carefully selected. The forward voltage drop is a natural advantage
for Schottky diodes and reduces as the current rating increases.
However, as the current rating increases, the diode capacitance
also increases so the optimum selection is usually the lowest current rating above the required maximum, in this case IL(PK).
Switch Current Sense Resistor (RSS) Neither the absolute
value of the switch current nor the accuracy of the measurement
is important, because the regulator will continuously adjust the
switch current, within a closed loop, to provide sufficient energy
for the output. For maximum accuracy the switch sense resistor
value should be chosen to maximize the differential signal seen
by the sense amplifier. The input limit of the sense amplifier,
VIDS , and the maximum switch current, IS(max), therefore define
the maximum value of the sense resistor as:
RSS = VIDS / IS(max)
(18)
Where IS(max) is the maximum switch current and should be set
above the maximum inductor current, IL(PK) .
This represents the maximum measurable value of the switch
(and inductor) current; however, the peak switch current will
always be less than this, set by the control circuit, depending on
the input voltage and the required load conditions. Because the
switch current control is within a closed loop, it is possible to
reduce the value of the sense resistor to reduce its power dissipation. However this will reduce the accuracy of the regulated LED
current.
The power loss in the switch sense resistor is worse at the lowest
input voltage:
PLOSS = (VLED [VLED – VIN(min)] / VIN(min)2) × RSS × I 2LED (19)
The peak switch current is defined by the maximum inductor current, IL(PK) . However in most cases the MOSFET will be chosen
by selecting low on-resistance, which usually results in a current
rating of several times the required peak current.
In addition to minimizing cost, the choice of MOSFET should
consider both the on-resistance and the total gate charge. The
total gate charge will determine the average current required from
the internal regulator and thus the power dissipation.
Output Capacitor There are several points to consider when
selecting the output capacitor.
Unlike some switch-mode regulators, the value of the output
capacitor in this case is not critical for output stability. The
capacitor value is only limited by the required maximum ripple
voltage.
Due to the switching topology used, the ripple current for this
circuit is high because the output capacitor provides the LED
current when the switch is active. The capacitor is then recharged
each time the inductor passes energy to the output. The ripple
current on the output capacitor will be equal to the peak inductor
current.
Normally this large ripple current, in conjunction with the
requirement for a larger capacitance value for stability, would
dictate the use of large electrolytic capacitors. However in this
case stability is not a consideration, and the capacitor value can
be low, allowing the use of ceramic capacitors.
To minimize self-heating effects and voltage ripple, the equivalent series resistance (ESR), and the equivalent series inductance
(ESL) should be kept as low as possible. This can be achieved by
multilayer ceramic chip (MLCC) capacitors. To reduce performance variation over temperature, low drift types such as X7R
and X5R should be used.
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A6266
Automotive, Boost, High Current LED Controller
The value of the output capacitor will typically be about 10 μF
and it should be rated above the maximum voltage defined by the
series output LEDs.
Reverse Supply Protection Protection for the A6266 is
provided by an external low current diode between the supply
and the VIN pin, as shown in the Functional Block Diagram
section. The isolation MOSFET shown in figure 3 is only able to
provide isolation when the supply polarity is correct. However,
with an additional P-channel MOSFET, it is also possible to
provide reverse battery protection to the switching elements and
the LEDs. The additional FET should be connected, as shown in
figure 4, with the drain to the supply and the source to the source
connection of the original isolation MOSFET.
In the complete circuit, consideration should be given to limiting
the maximum gate-source voltage of the FET. If the supply voltage is likely to exceed 20 V, then either: a Zener clamp must be
added in parallel with the gate-source resistor to prevent damage
to the FET, or a second resistor added as shown in figure 3.
VBAT
To VIN
To FF1
Figure 4. Example of a supply isolation MOSFET
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15
A6266
Automotive, Boost, High Current LED Controller
Package LP 16-Pin TSSOP with Exposed Thermal Pad
0.45
5.00±0.10
16
0.65
16
8º
0º
0.20
0.09
1.70
B
3 NOM
4.40±0.10
3.00
6.40±0.20
6.10
0.60 ±0.15
A
1
1.00 REF
2
3 NOM
0.25 BSC
Branded Face
16X
SEATING
PLANE
0.10 C
0.30
0.19
C
3.00
C
PCB Layout Reference View
For Reference Only; not for tooling use (reference MO-153 ABT)
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
1.20 MAX
0.65 BSC
1 2
SEATING PLANE
GAUGE PLANE
0.15
0.00
A Terminal #1 mark area
B
Exposed thermal pad (bottom surface); dimensions may vary with device
C Reference land pattern layout (reference IPC7351
SOP65P640X110-17M);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as
necessary to meet application process requirements and PCB layout
tolerances; when mounting on a multilayer PCB, thermal vias at the
exposed thermal pad land can improve thermal dissipation (reference
EIA/JEDEC Standard JESD51-5)
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A6266
Automotive, Boost, High Current LED Controller
Revision History
Revision
Revision Date
Rev. 3
April 1, 2013
Description of Revision
Update fMIN
Copyright ©2011-2013, Allegro MicroSystems, LLC
Allegro MicroSystems, LLC reserves the right to make, from time to time, such departures from the detail specifications as may be required to
permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that
the information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, LLC assumes no responsibility for its
use; nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
www.allegromicro.com
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