DN248 - Rail-to-Rail I/O and 2.4V Operation Allow UltraFast Comparators to be Used on Low Voltage Supplies

Rail-to-Rail I/O and 2.4V Operation Allow UltraFast Comparators to be Used on Low Voltage Supplies
Design Note 248
Glen Brisebois
The new LT®1711 to LT1714 family of UltraFast™
comparators have full differential rail-to-rail inputs and
outputs and operate down to 2.4V, allowing unfettered
application on low supply voltages. The LT1711 (single)
and LT1712 (dual) are specified at 4.5ns of propagation
delay and 100MHz toggle frequency. The lower power
LT1713 (single) and LT1714 (dual) are specified at 7ns
of propagation delay and 65MHz toggle frequency. All
of these comparators are fully equipped to support
multiple supply applications and have Latch Enable
(LE) pins and complementary outputs like the popular
LT1016, LT1116, LT1671 and LT1394. They are available in MSOP and SSOP packages, fully specified
over commercial and industrial temperature ranges
on 2.7V, 5V and ±5V supplies.
Simultaneous Full Duplex 75Mbaud Interface
with Only Two Wires
The circuit of Figure 1 shows a simple, fully bidirectional, differential 2-wire interface that gives good
results to 75Mbaud, using the lower power LT1714.
Eye diagrams under conditions of unidirectional and
bidirectional communication are shown in Figures 2
and 3. Although not as pristine as the unidirectional
performance of Figure 2, the performance under simultaneous bidirectional operation is still excellent.
Because the LT1714 input voltage range extends
100mV beyond both supply rails, the circuit works
with a full ±3V (one whole VS up or down) of ground
potential difference.
The circuit works well with the resistor values shown, but
other sets of values can be used. The starting point is the
characteristic impedance, ZO, of the twisted-pair cable.
The input impedance of the resistive network should
match the characteristic impedance and is given by:
RIN = 2 t RO t
R1||(R2 + R3)
RO + 2 t [R1]|(R2 + R3)]
This comes out to 120Ω for the values shown. The
Thevenin equivalent source voltage is given by:
(R2 + R3 – R1)
(R2 + R3 + R1)
RO
t
RO + 2 t [R1]]R2 + R3)]
VTH = VS t
This amounts to an attenuation factor of 0.0978 with
the values shown. (The actual voltage on the lines will
be cut in half again due to the 120Ω ZO.) The reason
this attenuation factor is important is that it is the key
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3V
4
3V
+
14
1/2
LE LT1714
RxD
13 16
3
–
2
2
1
3V
3V
15
7
TxD
+
5
11
1/2
LT1714
8
– LE 10 12
6
9
1
3V
R2A
2.55k
R1A
499Ω
R1C
499Ω
R2C
2.55k
R3A
124Ω
ROA
140Ω
ROB
140Ω
R3C
124Ω
R1D
499Ω
R3D
124Ω
R3B
124Ω
R2B
255k
R1B
499Ω
DIODES: BAV99
w4
3
+
9
R2D
2.55k
DN252 F01
Figure 1. 75Mbaud Full Duplex Interface on Two Wires
01/01/248_conv
–
7
1/2
LT1714
8
LE –
12 6
10
14
1/2
LT1714 LE
3V
5
11
6-FEET
TWISTED PAIR
ZO 120Ω
4
+
TxD
15
16 13
Rx
C4
100pF
R5
7.5k
1MHz
AT-CUT R4
210Ω
VS
R1
1k
R2
1k
DN249 F02
Figure 2. Performance of Figure 1's Circuit When
Operated Unidirectionally. Eye is Wide Open
R6
162Ω
C3
100pF
–
R9
2k
3
+
C2
0.1μF
R8
2k
VS
2
3
1
+
–
7
LT1713
LE
5
6
4
SQUARE
8
VS
2
VS
R7
15.8k
7
6
LT1806
SINE
1
4
DN252 F04
R3
1k
C1
0.1μF
Figure 4. LT1713 Comparator is Configured as a Series
Resonant Xtal Oscillator. LT1806 Op Amp is Configured
in a Q = 5 Bandpass with fC = 1MHz
3V/DIV
DN249 F03
Figure 3. Performance When Operated Simultaneous
Bidirectionally (Full Duplex). Crosstalk Appears as Noise.
Eye is Slightly Shut But Performance is Still Excellent.
to deciding the ratio between the R2-R3 resistor divider
in the receiver path. This divider allows the receiver
to reject the large signal of the local transmitter and
instead sense the attenuated signal of the remote
transmitter. Note that in the above equations, R2 and
R3 are not yet fully determined because they only appear as a sum. This allows the designer to now place
an additional constraint on their values. The R2-R3
divide ratio should be set to equal half the attenuation
factor mentioned above or:
R3/R2 = 1/2 • 0.09761.
Having already designed R2 + R3 to be 2.653k (by
allocating input impedance across RO, R1 and R2 +
R3 to get the requisite 120Ω), R2 and R3 then become
2529Ω and 123.5Ω respectively. The nearest 1% value
for R2 is 2.55k and that for R3 is 124Ω.
1MHz Series Resonant Crystal Oscillator
with Square and Sinusoid Outputs
Figure 4 shows a classic 1MHz series resonant crystal
oscillator. At series resonance, the crystal is a low
impedance and the positive feedback connection is
what brings about oscillation at the series resonant
frequency. The RC feedback around the other path
ensures that the circuit does not find a stable DC operating point and refuse to oscillate. The comparator
output is a 1MHz square wave (top trace of Figure 5)
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1V/DIV
1V/DIV
200ns/DIV
DN249 F05
Figure 5. Oscillator Waveforms with VS = 3V. Top is
Comparator Output. Middle is Xtal Feedback to Pin 2 at
LT1713 (Note the Glitches). Bottom is Buffered, Inverted
and Bandpass Filtered with a Q = 5 by LT1806
with jitter measured at 28psRMS on a 5V supply and
40psRMS on a 3V supply. At Pin 2 of the comparator,
on the other side of the crystal, is a clean sine wave
except for the presence of the small high frequency
glitch (middle trace of Figure 5). This glitch is caused
by the fast edge of the comparator output feeding
back through crystal capacitance. Amplitude stability
of the sine wave is maintained by the fact that the sine
wave is basically a filtered version of the square wave.
Hence, the usual amplitude control loops associated
with sinusoidal oscillators are not necessary.2 The
sine wave is filtered and buffered by the fast, low noise
LT1806 op amp. To remove the glitch, the LT1806
is configured as a bandpass filter with a Q of 5 and
unity-gain center frequency of 1MHz, with its output
shown as the bottom trace of Figure 5. Distortion was
measured at –70dBc and –55dBc on the second and
third harmonics, respectively.
1 Using the design value of R2 + R3 = 2.653k rather than the implementation
value of 2.55k + 124Ω = 2.674k.
2 Amplitude will be a linear function of comparator output swing, which is
supply dependent and therefore adjustable. The important difference here is
that any added amplitude stabilization or control loop will not be faced with the
classical task of avoiding regions of nonoscillation versus clipping.
For applications help,
call (408) 432-1900
dn248f_conv LT/TP 0101 340K • PRINTED IN THE USA
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