LM2595 D

LM2595
1.0 A, Step-Down Switching
Regulator
The LM2595 regulator is monolithic integrated circuit ideally suited
for easy and convenient design of a step−down switching regulator
(buck converter). It is capable of driving a 1.0 A load with excellent
line and load regulation. This device is available in adjustable output
version and it is internally compensated to minimize the number of
external components to simplify the power supply design.
Since LM2595 converter is a switch−mode power supply, its
efficiency is significantly higher in comparison with popular
three−terminal linear regulators, especially with higher input voltages.
The LM2595 operates at a switching frequency of 150 kHz thus
allowing smaller sized filter components than what would be needed
with lower frequency switching regulators. Available in a standard
5−lead TO−220 package with several different lead bend options, and
D2PAK surface mount package.
The other features include a guaranteed $4% tolerance on output
voltage within specified input voltages and output load conditions, and
$15% on the oscillator frequency. External shutdown is included,
featuring 50 mA (typical) standby current. Self protection features
include switch cycle−by−cycle current limit for the output switch, as
well as thermal shutdown for complete protection under fault
conditions.
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5
Heatsink surface connected to Pin 3
TO−220
T SUFFIX
CASE 314D
1
5
Pin
Features
•
•
•
•
•
•
•
•
•
•
Adjustable Output Voltage Range 1.23 V − 37 V
Guaranteed 1.0 A Output Load Current
Wide Input Voltage Range up to 40 V
150 kHz Fixed Frequency Internal Oscillator
TTL Shutdown Capability
Low Power Standby Mode, typ 50 mA
Thermal Shutdown and Current Limit Protection
Internal Loop Compensation
Moisture Sensitivity Level (MSL) Equals 1
Pb−Free Packages are Available
1
5
February, 2009 − Rev. 2
Output
Vin
Ground
Feedback
ON/OFF
D2PAK
D2T SUFFIX
CASE 936A
ORDERING INFORMATION
Simple High−Efficiency Step−Down (Buck) Regulator
Efficient Pre−Regulator for Linear Regulators
On−Card Switching Regulators
Positive to Negative Converter (Buck−Boost)
Negative Step−Up Converters
Power Supply for Battery Chargers
© Semiconductor Components Industries, LLC, 2009
1.
2.
3.
4.
5.
Heatsink surface (shown as terminal 6 in
case outline drawing) is connected to Pin 3
Applications
•
•
•
•
•
•
TO−220
TV SUFFIX
CASE 314B
1
See detailed ordering and shipping information in the package
dimensions section on page 23 of this data sheet.
DEVICE MARKING INFORMATION
See general marking information in the device marking
section on page 23 of this data sheet.
1
Publication Order Number:
LM2595/D
LM2595
12 V
Unregulated
DC Input
R1=1K
Feedback
+Vin
4
LM2595
L1
68 mH
Cff
R2=3.0K
Output
2
5V@1A
Regulated
Output
1
Cin
220 mF/
50 V
3
GND
5
ON/OFF
Cout
220 mF
D1
1N5822
Figure 1. Typical Application
Figure 2. Representative Block Diagram
MAXIMUM RATINGS
Rating
Symbol
Value
Unit
Maximum Supply Voltage
Vin
45
V
ON/OFF Pin Input Voltage
ON/OFF
−0.3 V ≤ V ≤ +Vin
V
Output
−1.0
V
PD
Internally Limited
W
Thermal Resistance, Junction−to−Ambient
RqJA
65
°C/W
Thermal Resistance, Junction−to−Case
RqJC
5.0
°C/W
PD
Internally Limited
W
Thermal Resistance, Junction−to−Ambient
RqJA
70
°C/W
Thermal Resistance, Junction−to−Case
RqJC
5.0
°C/W
Tstg
−65 to +150
°C
Minimum ESD Rating (Human Body Model: C = 100 pF, R = 1.5 kW)
−
2.0
kV
Lead Temperature (Soldering, 10 seconds)
−
260
°C
Maximum Junction Temperature
TJ
150
°C
Output Voltage to Ground (Steady−State)
Power Dissipation
Case 314B and 314D (TO−220, 5−Lead)
Case 936A
(D2PAK)
Storage Temperature Range
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
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LM2595
PIN FUNCTION DESCRIPTION
Pin
Symbol
Description (Refer to Figure 1)
1
Output
This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is typically 1.0 V. It should be
kept in mind that the PCB area connected to this pin should be kept to a minimum in order to minimize coupling to
sensitive circuitry.
2
Vin
This pin is the positive input supply for the LM2595 step−down switching regulator. In order to minimize voltage transients and to supply the switching currents needed by the regulator, a suitable input bypass capacitor must be present
(Cin in Figure 1).
3
GND
4
Feedback
This pin is the direct input of the error amplifier and the resistor network R2, R1 is connected externally to allow programming of the output voltage.
5
ON/OFF
It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total input supply
current to approximately 50 mA. The threshold voltage is typically 1.6 V. Applying a voltage above this value (up to
+Vin) shuts the regulator off. If the voltage applied to this pin is lower than 1.6 V or if this pin is left open, the regulator
will be in the “on” condition.
Circuit ground pin. See the information about the printed circuit board layout.
OPERATING RATINGS (Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee
specific performance limits. For guaranteed specifications and test conditions, see the Electrical Characteristics.)
Rating
Symbol
Value
Unit
Operating Junction Temperature Range
TJ
−40 to +125
°C
Supply Voltage
Vin
4.5 to 40
V
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LM2595
SYSTEM PARAMETERS
ELECTRICAL CHARACTERISTICS Specifications with standard type face are for TJ = 25°C, and those with boldface type apply
over full Operating Temperature Range −40°C to +125°C
Symbol
Characteristics
Min
Typ
Max
Unit
LM2595 (Note 1, Test Circuit Figure 16)
Feedback Voltage (Vin = 12 V, ILoad = 0.2 A, Vout = 5.0 V, )
VFB_nom
Feedback Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A, Vout = 5.0 V)
VFB
1.193
1.18
h
−
Symbol
Min
Efficiency (Vin = 12 V, ILoad = 1.0 A, Vout = 5.0 V)
Characteristics
1.23
Feedback Bias Current (Vout = 5.0 V)
Ib
1.267
1.28
V
81
−
%
Typ
Max
Unit
25
100
200
nA
150
165
180
kHz
1.2
1.3
V
Oscillator Frequency (Note 2)
fosc
Saturation Voltage (Iout = 1.0 A, Notes 3 and 4)
Vsat
1.0
Max Duty Cycle “ON” (Note 4)
DC
95
Current Limit (Peak Current, Notes 2 and 3)
ICL
Output Leakage Current (Notes 5 and 6)
Output = 0 V
Output = −1.0 V
IL
Quiescent Current (Note 5)
Standby Quiescent Current (ON/OFF Pin = 5.0 V (“OFF”))
(Note 6)
135
120
1.2
1.15
V
%
2.1
2.4
2.6
A
0.5
13
2.0
30
IQ
5.0
10
mA
Istby
50
200
250
mA
mA
ON/OFF PIN LOGIC INPUT
1.6
Threshold Voltage
Vout = 0 V (Regulator OFF)
VIH
Vout = Nominal Output Voltage (Regulator ON)
VIL
V
2.2
2.4
V
1.0
0.8
V
ON/OFF Pin Input Current
ON/OFF Pin = 5.0 V (Regulator OFF)
IIH
−
15
30
mA
ON/OFF Pin = 0 V (regulator ON)
IIL
−
0.01
5.0
mA
1. External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.
When the LM2595 is used as shown in the Figure 16 test circuit, system performance will be as shown in system parameters section.
2. The oscillator frequency reduces to approximately 30 kHz in the event of an output short or an overload which causes the regulated output
voltage to drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by
lowering the minimum duty cycle from 5% down to approximately 2%.
3. No diode, inductor or capacitor connected to output (Pin 1) sourcing the current.
4. Feedback (Pin 4) removed from output and connected to 0 V.
5. Feedback (Pin 4) removed from output and connected to +12 V to force the output transistor “off”.
6. Vin = 40 V.
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LM2595
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16)
0.8
0.6
Vout, OUTPUT VOLTAGE CHANGE (%)
Vout , OUTPUT VOLTAGE CHANGE (%)
1.0
Vin = 20 V
ILoad = 200 mA
Normalized at TJ = 25°C
0.4
0.2
0
-0.2
-0.4
-0.6
-0.8
-1.0
−50
−25
0
25
50
75
100
125
0.6
Vout = 5 V
0.4
0.2
0
−0.2
−0.4
−0.6
0
5.0
10
15
20
25
30
Figure 4. Line Regulation
35
40
3.0
SWITCHING CURRENT LIMIT (A)
ILoad = 1 A
0.5
ILoad = 200 mA
0
L = 68 mH
R_ind = 30 mW
−25
0
25
60
75
100
2.0
1.0
10
30
50
70
90
110
TJ, JUNCTION TEMPERATURE (°C)
TJ, JUNCTION TEMPERATURE (°C)
Figure 5. Dropout Voltage
Figure 6. Current Limit
160
Vout = 5 V
Measured at GND Pin
TJ = 25°C
11
10
9
ILoad = 1.0 A
8
7
ILoad = 200 mA
6
5
5
Vin = 12 V
0.0
−50 −30 −10
125
I stby , STANDBY QUIESCENT CURRENT (μA)
INPUT - OUTPUT DIFFERENTIAL (V)
0.8
Figure 3. Normalized Output Voltage
12
I Q, QUIESCENT CURRENT (mA)
1.0
Vin, INPUT VOLTAGE (V)
1.0
4
0
ILoad = 200 mA
TJ = 25°C
1.2
TJ, JUNCTION TEMPERATURE (°C)
1.5
−0.5
−50
1.4
10
15
20
25
30
35
40
140
VON/OFF = 5.0 V
120
100
80
Vin = 40 V
60
40
Vin = 12 V
20
0
−50
−25
0
25
60
75
100
Vin, INPUT VOLTAGE (V)
TJ, JUNCTION TEMPERATURE (°C)
Figure 7. Quiescent Current
Figure 8. Standby Quiescent Current
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130
125
LM2595
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16)
1.0
1.2
NORMALIZED FREQUENCY (%)
Vsat , SATURATION VOLTAGE (V)
1.3
1.1
1.0
0.9
−40°C
0.8
25°C
0.7
125°C
0.6
0.5
0
0.2
0.4
0.6
−3.0
−4.0
−5.0
−6.0
−7.0
0.8
−9.0
−50
1.0
0
25
50
75
100
TJ, JUNCTION TEMPERATURE (°C)
Figure 9. Switch Saturation Voltage
Figure 10. Switching Frequency
100
4.5
80
Ib , FEEDBACK PIN CURRENT (nA)
5.0
4.0
3.5
3.0
2.5
2.0
Vout ' 1.23 V
ILoad = 200 mA
1.5
1.0
0.5
0
-50
−25
SWITCH CURRENT (A)
-25
0
25
50
75
100
40
20
0
-20
-40
-60
-80
-100
-50
125
-25
0
25
12 V, 1 A
85
5 V, 1 A
80
75
3.3 V, 1 A
0
5
10
75
100
Figure 12. Feedback Pin Current
95
70
50
TJ, JUNCTION TEMPERATURE (°C)
TJ, JUNCTION TEMPERATURE (°C)
90
15
20
25
30
VIN, INPUT VOLTAGE (V)
Figure 13. Efficiency
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125
60
Figure 11. Minimum Supply Operating Voltage
EFFICIENCY (%)
V in, INPUT VOLTAGE (V)
−2.0
−8.0
0.4
0.3
0.0
−1.0
35
40
45
125
LM2595
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 16)
A
10 V
100 mV
Output
0
Voltage
Change
- 100 mV
0
1.2 A
B
0.6 A
0
0.5 A
1.2 A
C
D
Load
Current
0.6 A
0.1 A
0
0
2 ms/div
100 ms/div
Figure 14. Switching Waveforms
Figure 15. Load Transient Response
Vout = 5 V
A: Output Pin Voltage, 10 V/div
B: Switch Current, 0.6 A/div
C: Inductor Current, 0.6 A/div, AC−Coupled
D: Output Ripple Voltage, 50 mV/div, AC−Coupled
Horizontal Time Base: 2.0 ms/div
Adjustable Output Voltage Versions
Feedback
Vin
LM2595
2
L1
68 mH
Output
3
8.5 V - 40 V
Unregulated
DC Input
4
GND
5
Vout
5.0 V/1.0 A
1
ON/OFF
CFF
Cin
100 mF
D1
1N5822
Cout
220 mF
R2
Load
R1
V out + V
ǒ
R2 + R1
ǒ1.0 ) R2
Ǔ
R1
ref
Ǔ
V out
1.0
V
ref
Where Vref = 1.23 V, R1
between 1.0 k and 5.0 k
Figure 16. Typical Test Circuit
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LM2595
PCB LAYOUT GUIDELINES
On the other hand, the PCB area connected to the Pin 1
As in any switching regulator, the layout of the printed
(emitter of the internal switch) of the LM2595 should be
circuit board is very important. Rapidly switching currents
kept to a minimum in order to minimize coupling to sensitive
associated with wiring inductance, stray capacitance and
circuitry.
parasitic inductance of the printed circuit board traces can
Another sensitive part of the circuit is the feedback. It is
generate voltage transients which can generate
important to keep the sensitive feedback wiring short. To
electromagnetic interferences (EMI) and affect the desired
assure this, physically locate the programming resistors near
operation. As indicated in the Figure 16, to minimize
to the regulator, when using the adjustable version of the
inductance and ground loops, the length of the leads
LM2595 regulator.
indicated by heavy lines should be kept as short as possible.
For best results, single−point grounding (as indicated) or
ground plane construction should be used.
DESIGN PROCEDURE
Buck Converter Basics
This period ends when the power switch is once again
turned on. Regulation of the converter is accomplished by
varying the duty cycle of the power switch. It is possible to
describe the duty cycle as follows:
The LM2595 is a “Buck” or Step−Down Converter which
is the most elementary forward−mode converter. Its basic
schematic can be seen in Figure 17.
The operation of this regulator topology has two distinct
time periods. The first one occurs when the series switch is
on, the input voltage is connected to the input of the inductor.
The output of the inductor is the output voltage, and the
rectifier (or catch diode) is reverse biased. During this
period, since there is a constant voltage source connected
across the inductor, the inductor current begins to linearly
ramp upwards, as described by the following equation:
I L(on) +
t
d + on , where T is the period of switching.
T
For the buck converter with ideal components, the duty
cycle can also be described as:
V
d + out
V in
Figure 18 shows the buck converter, idealized waveforms
of the catch diode voltage and the inductor current.
ǒV IN * VOUTǓton
Von(SW)
L
Power
Switch
Diode Voltage
During this “on” period, energy is stored within the core
material in the form of magnetic flux. If the inductor is
properly designed, there is sufficient energy stored to carry
the requirements of the load during the “off” period.
L
Vin
Cout
D
Power
Switch
Off
VD(FWD)
Power
Switch
On
Power
Switch
On
RLoad
Time
Figure 17. Basic Buck Converter
Inductor Current
Ipk
The next period is the “off” period of the power switch.
When the power switch turns off, the voltage across the
inductor reverses its polarity and is clamped at one diode
voltage drop below ground by the catch diode. The current
now flows through the catch diode thus maintaining the load
current loop. This removes the stored energy from the
inductor. The inductor current during this time is:
I L(off) +
Power
Switch
Off
ILoad(AV)
Imin
Diode
Power
Switch
Diode
Power
Switch
Time
Figure 18. Buck Converter Idealized Waveforms
ǒV OUT * VDǓtoff
L
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LM2595
PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2595)
Procedure
Example
Given Parameters:
Vout = Regulated Output Voltage
Vin(max) = Maximum DC Input Voltage
ILoad(max) = Maximum Load Current
Given Parameters:
Vout = 5.0 V
Vin(max) = 12 V
ILoad(max) = 1.0 A
1. Programming Output Voltage
To select the right programming resistor R1 and R2 value (see
Figure 1) use the following formula:
1. Programming Output Voltage (selecting R1 and R2)
Select R1 and R2:
ǒ
Ǔ
R2
1.0 )
V out + V
ref
R1
ǒ
V out + 1.23 1.0 )
ǒ
where Vref = 1.23 V
R2 + R1
Resistor R1 can be between 1.0 k and 5.0 kW. (For best
temperature coefficient and stability with time, use 1% metal
film resistors).
ǒV
R2 + R1
V out
ref
* 1.0
V out
V
ref
Ǔ
R2
R1
Select R1 = 1.0 kW
Ǔ ǒ
* 1.0
+
Ǔ
5V
* 1.0
1.23 V
R2 = 3.07 kW, choose a 3.0k metal film resistor.
Ǔ
2. Input Capacitor Selection (Cin)
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
input pin +Vin and ground pin GND This capacitor should be
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
2. Input Capacitor Selection (Cin)
A 220 mF, 50 V aluminium electrolytic capacitor located near
the input and ground pin provides sufficient bypassing.
For additional information see input capacitor section in the
“Application Information” section of this data sheet.
3. Catch Diode Selection (D1)
A. Since the diode maximum peak current exceeds the
regulator maximum load current the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design, the diode should have a
current rating equal to the maximum current limit of the
LM2595 to be able to withstand a continuous output short.
B. The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
3. Catch Diode Selection (D1)
A. For this example, a 1.0 A (for a robust design 3.0 A diode
is recommended) current rating is
adequate.
B. For Vin = 12 V use a 20 V 1N5817 (1N5820) Schottky
diode or any suggested fast recovery diode in the Table 2.
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LM2595
PROCEDURE (ADJUSTABLE OUTPUT VERSION: LM2595) (CONTINUED)
Procedure
Example
4. Inductor Selection (L1)
A. Calculate E x T [V x ms] constant:
4. Inductor Selection (L1)
A. Use the following formula to calculate the inductor Volt x
microsecond [V x ms] constant:
E
ǒ
T+ V
IN
*V
OUT
*V
Ǔ
SAT
V
V
IN
OUT
*V
)V
SAT
D
)V
1000
D
150 kHz
ǒV
+I
)
T + ǒ12 * 5 * 1.0Ǔ
E
T + ǒ6Ǔ
msǓ
B. Match the calculated E x T value with the corresponding
number on the vertical axis of the Inductor Value Selection
Guide shown in Figure 19. This E x T constant is a
measure of the energy handling capability of an inductor and
is dependent upon the type of core, the core area, the
number of turns, and the duty cycle.
C. Next step is to identify the inductance region intersected by
the E x T value and the maximum load current value on the
horizontal axis shown in Figure 19.
D. Select an appropriate inductor from Table 3.
The inductor chosen must be rated for a switching
frequency of 150 kHz and for a current rating of 1.15 x ILoad.
The inductor current rating can also be determined by
calculating the inductor peak current:
I
E
5.5
11.5
5 ) 0.5
1000
12 * 1 ) 0.5
150 kHz
6.7ǒV
ǒV
msǓ
msǓ
B. E x T = 19.2 [V x ms]
C. ILoad(max) = 1.0 A
Inductance Region = L30
D. Proper inductor value = 68 mH
Choose the inductor from Table 3.
ǒVin * VoutǓ ton
p(max) Load(max)
2L
where ton is the “on” time of the power switch and
V
t on + out x 1.0
V
f osc
in
5. Output Capacitor Selection (Cout)
A. Since the LM2595 is a forward−mode switching regulator
with voltage mode control, its open loop has 2−pole−1−zero
frequency characteristic. The loop stability is determined by
the output capacitor (capacitance, ESR) and inductance
values.
5. Output Capacitor Selection (Cout)
A. In this example, it is recommended to use a Nichicon PM
capacitor: 220 mF/25 V
For stable operation use recommended values of the output
capacitors in Table 1.
Low ESR electrolytic capacitors between 180 mF and
1000 mF provide best results.
B. The capacitors voltage rating should be at least 1.5 times
greater than the output voltage, and often much higher
voltage rating is needed to satisfy low ESR requirement
6. Feedforward Capacitor (CFF)
It provides additional loop stability mainly for higher input voltages.
For Cff selection use Table 1. The compensation capacitor between
0.6 nF and 15 nF is wired in parallel with the output voltage setting
resistor R2, The capacitor type can be ceramic, plastic, etc..
6. Feedforward Capacitor (CFF)
In this example, it is recommended to use a feedforward
capacitor 4.7 nF.
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LM2595
LM2595 Series Buck Regulator Design Procedures (continued)
Table 1. RECOMMENDED VALUES OF THE OUTPUT CAPACITOR AND FEEDFORWARD CAPACITOR
(Iload = 1.0 A)
Nichicon Pm Capacitors
Vin (V)
Capacity/ESR/Voltage Range (mF/mW/V)
1000/60/10
1000/60/10
1000/60/10
470/120/10
220/110/25
180/290/25
180/290/25
82/190/35
82/190/35
35
1000/60/10
1000/60/10
1000/60/10
220/110/25
180/140/25
120/200/25
120/200/25
82/190/35
82/190/35
26
1000/60/10
470/120/10
220/110/25
220/110/25
180/140/25
120/200/25
120/200/25
82/190/35
20
1000/60/10
470/120/10
220/110/25
220/110/25
180/140/25
120/200/25
120/200/25
18
1000/60/10
470/120/10
220/110/25
220/110/25
180/140/25
120/200/25
120/200/25
12
470/120/10
470/120/10
220/110/25
220/110/25
180/140/25
10
470/120/10
470/120/10
220/110/25
220/110/25
Vout
2
3
4
6
9
12
15
24
28
Cff (nF)
10
4.7
4.7
4.7
1.5
1.5
1
0.6
0.6
E*T(V*us)
40
MAXIMUM LOAD CURRENT (A)
Figure 19. Inductor Value Selection Guides (For Continuous Mode Operation)
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LM2595
Table 2. DIODE SELECTION
1A Diodes
Surface Mount
VR
Schottky
20V
SK12
Ultra Fast
Recovery
3A Diodes
Through Hole
Schottky
Surface Mount
Ultra Fast
Recovery
Schottky
Ultra Fast
Recovery
1N5817
Through Hole
Schottky
Ultra Fast
Recovery
1N5820
SR102
SK32
SR302
MBR320
30 V
SK13
MBRS130
40 V
SK14
MBRS140
50 V
or
More
All of these
diodes are
rated to at
least 50 V
MURS120
10BF10
1N5818
SR103
11DQ03
All of these
diodes are
rated to at
least 50 V.
MUR120
SK33
All of these
diodes are
rated to at
least 50 V.
MURS320
30WF10
1N5821
MBR330
31DQ03
1N5822
1N5819
SK34
10BQ040
SR104
MBRS340
MBR340
10MQ040
11DQ04
30WQ04
31DQ04
MBRS160
SR105
SK35
SR305
10BQ050
MBR150
MBR360
MBR350
10MQ060
11DQ05
30WQ05
31DQ05
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12
SR304
All of these
diodes are
rated to at
least 50 V.
MUR320
30WF10
LM2595
Table 3. INDUCTOR MANUFACTURERS PART NUMBERS
Renco
Pulse Engineering
Coilcraft
Inductance
(mH)
Current
(A)
Through Hole
Surface Mount
Through Hole
Surface Mount
Through Hole
L4
68
0.32
RL−1284−68−43
RL1500−68
PE−53804
PE−53804−S
−
DO1608−68
L5
47
0.37
RL−1284−47−43
RL1500−47
PE−53805
PE−53805−S
−
DO1608−473
L6
33
0.44
RL−1284−33−43
RL1500−33
PE−53806
PE−53806−S
−
DO1608−333
L9
220
0.32
RL−5470−3
RL1500−220
PE−53809
PE−53809−S
−
DO3308−224
L10
150
0.39
RL−5470−4
RL1500−150
PE−53810
PE−53810−S
−
DO3308−154
Surface Mount
L11
100
0.48
RL−5470−5
RL1500−100
PE−53811
PE−53811−S
−
DO3308−104
L12
68
0.58
RL−5470−6
RL1500−68
PE−53812
PE−53812−S
−
DO3308−683
L13
47
0.70
RL−5470−7
RL1500−47
PE−53813
PE−53813−S
−
DO3308−473
L14
33
0.83
RL−1284−33−43
RL1500−33
PE−53814
PE−53814−S
−
DO3308−333
L15
22
0.99
RL−1284−22−43
RL1500−22
PE−53815
PE−53815−S
−
DO3308−223
L16
15
1.24
RL−1284−15−43
RL1500−15
PE−53816
PE−53816−S
−
DO3308−153
L17
330
0.42
RL−5471−1
RL1500−330
PE−53817
PE−53817−S
−
DO3316−334
L18
220
0.55
RL−5471−2
RL1500−220
PE−53818
PE−53818−S
−
DO3316−224
L19
150
0.66
RL−5471−3
RL1500−150
PE−53819
PE−53819−S
−
DO3316−154
L20
100
0.82
RL−5471−4
RL1500−100
PE−53820
PE−53820−S
−
DO3316−104
L21
68
0.99
RL−5471−5
RL1500−68
PE−53821
PE−53821−S
−
DO3316−683
L22
47
1.17
RL−5471−6
−
PE−53822
PE−53822−S
−
DO3316−473
L23
33
1.40
RL−5471−7
−
PE−53823
PE−53823−S
−
DO3316−333
L24
22
1.70
RL−1283−22−43
−
PE−53824
PE−53824−S
RFB0810−220L
DO3316−223
L26
330
0.80
RL−5471−1
−
PE−53826
PE−53826−S
RFB0810−331L
DO3340P−334ML
L27
220
1.00
RL−5471−2
−
PE−53827
PE−53827−S
RFB0810−221L
DO3340P−224ML
L28
150
1.20
RL−5471−3
−
PE−53828
PE−53828−S
RFB0810−151L
DO3340P−154ML
L29
100
1.47
RL−5471−4
−
PE−53829
PE−53829−S
RFB0810−101L
DO3340P−104ML
L30
68
1.78
RL−5471−5
−
PE−53830
PE−53830−S
RFB0810−680L
DO3340P−683ML
L35
47
2.15
RL−5473−1
−
PE−53935
PE−53935−S
RFB0810−470L
DO3340P−473ML
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LM2595
APPLICATION INFORMATION
EXTERNAL COMPONENTS
regulator loop stability. The ESR of the output capacitor and
the peak−to−peak value of the inductor ripple current are the
main factors contributing to the output ripple voltage value.
Standard aluminium electrolytics could be adequate for
some applications but for quality design, low ESR types are
recommended.
An aluminium electrolytic capacitor’s ESR value is
related to many factors such as the capacitance value, the
voltage rating, the physical size and the type of construction.
In most cases, the higher voltage electrolytic capacitors have
lower ESR value. Often capacitors with much higher
voltage ratings may be needed to provide low ESR values
that, are required for low output ripple voltage.
Input Capacitor (Cin)
The Input Capacitor Should Have a Low ESR
For stable operation of the switch mode converter a low
ESR (Equivalent Series Resistance) aluminium or solid
tantalum bypass capacitor is needed between the input pin
and the ground pin, to prevent large voltage transients from
appearing at the input. It must be located near the regulator
and use short leads. With most electrolytic capacitors, the
capacitance value decreases and the ESR increases with
lower temperatures. For reliable operation in temperatures
below −25°C larger values of the input capacitor may be
needed. Also paralleling a ceramic or solid tantalum
capacitor will increase the regulator stability at cold
temperatures.
Feedfoward Capacitor
(Adjustable Output Voltage Version)
This capacitor adds lead compensation to the feedback
loop and increases the phase margin for better loop stability.
For CFF selection, see the design procedure section.
RMS Current Rating of Cin
The important parameter of the input capacitor is the RMS
current rating. Capacitors that are physically large and have
large surface area will typically have higher RMS current
ratings. For a given capacitor value, a higher voltage
electrolytic capacitor will be physically larger than a lower
voltage capacitor, and thus be able to dissipate more heat to
the surrounding air, and therefore will have a higher RMS
current rating. The consequence of operating an electrolytic
capacitor beyond the RMS current rating is a shortened
operating life. In order to assure maximum capacitor
operating lifetime, the capacitor’s RMS ripple current rating
should be:
The Output Capacitor Requires an ESR Value
That Has an Upper and Lower Limit
As mentioned above, a low ESR value is needed for low
output ripple voltage, typically 1% to 2% of the output
voltage. But if the selected capacitor’s ESR is extremely low
(below 0.05 W), there is a possibility of an unstable feedback
loop, resulting in oscillation at the output. This situation can
occur when a tantalum capacitor, that can have a very low
ESR, is used as the only output capacitor.
At Low Temperatures, Put in Parallel Aluminium
Electrolytic Capacitors with Tantalum Capacitors
Irms > 1.2 x d x ILoad
Electrolytic capacitors are not recommended for
temperatures below −25°C. The ESR rises dramatically at
cold temperatures and typically rises 3 times at −25°C and
as much as 10 times at −40°C. Solid tantalum capacitors
have much better ESR spec at cold temperatures and are
recommended for temperatures below −25°C. They can be
also used in parallel with aluminium electrolytics. The value
of the tantalum capacitor should be about 10% or 20% of the
total capacitance. The output capacitor should have at least
50% higher RMS ripple current rating at 150 kHz than the
peak−to−peak inductor ripple current.
where d is the duty cycle, for a buck regulator
V
t
d + on + out
T
V in
|V out|
t on
and d +
+
for a buck*boost regulator.
T
|V out| ) V
in
Output Capacitor (Cout)
For low output ripple voltage and good stability, low ESR
output capacitors are recommended. An output capacitor
has two main functions: it filters the output and provides
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LM2595
Catch Diode
Locate the Catch Diode Close to the LM2595
The LM2595 is a step−down buck converter; it requires a
fast diode to provide a return path for the inductor current
when the switch turns off. This diode must be located close
to the LM2595 using short leads and short printed circuit
traces to avoid EMI problems.
Use a Schottky or a Soft Switching
Ultra−Fast Recovery Diode
Since the rectifier diodes are very significant sources of
losses within switching power supplies, choosing the
rectifier that best fits into the converter design is an
important process. Schottky diodes provide the best
performance because of their fast switching speed and low
forward voltage drop.
They provide the best efficiency especially in low output
voltage applications (5.0 V and lower). Another choice
could be Fast−Recovery, or Ultra−Fast Recovery diodes. It
has to be noted, that some types of these diodes with an
abrupt turnoff characteristic may cause instability or
EMI troubles.
A fast−recovery diode with soft recovery characteristics
can better fulfill some quality, low noise design requirements.
Table 2 provides a list of suitable diodes for the LM2595
regulator. Standard 50/60 Hz rectifier diodes, such as the
1N4001 series or 1N5400 series are NOT suitable.
VERTRICAL RESOLUTION 0.4 A/DIV
ripple voltage. On the other hand it does require larger
inductor values to keep the inductor current flowing
continuously, especially at low output load currents and/or
high input voltages.
To simplify the inductor selection process, an inductor
selection guide for the LM2595 regulator was added to this
data sheet (Figure 19). This guide assumes that the regulator
is operating in the continuous mode, and selects an inductor
that will allow a peak−to−peak inductor ripple current to be
a certain percentage of the maximum design load current.
This percentage is allowed to change as different design load
currents are selected. For light loads (less than
approximately 300 mA) it may be desirable to operate the
regulator in the discontinuous mode, because the inductor
value and size can be kept relatively low. Consequently, the
percentage of inductor peak−to−peak current increases. This
discontinuous mode of operation is perfectly acceptable for
this type of switching converter. Any buck regulator will be
forced to enter discontinuous mode if the load current is light
enough.
0.4 A
Inductor
Current
Waveform 0 A
Inductor
0.8 A
Power
Switch
Current
Waveform 0 A
The magnetic components are the cornerstone of all
switching power supply designs. The style of the core and
the winding technique used in the magnetic component’s
design has a great influence on the reliability of the overall
power supply.
Using an improper or poorly designed inductor can cause
high voltage spikes generated by the rate of transitions in
current within the switching power supply, and the
possibility of core saturation can arise during an abnormal
operational mode. Voltage spikes can cause the
semiconductors to enter avalanche breakdown and the part
can instantly fail if enough energy is applied. It can also
cause significant RFI (Radio Frequency Interference) and
EMI (Electro−Magnetic Interference) problems.
HORIZONTAL TIME BASE: 2.0 ms/DIV
Figure 20. Continuous Mode Switching Current
Waveforms
Selecting the Right Inductor Style
Some important considerations when selecting a core type
are core material, cost, the output power of the power supply,
the physical volume the inductor must fit within, and the
amount of EMI (Electro−Magnetic Interference) shielding
that the core must provide. The inductor selection guide
covers different styles of inductors, such as pot core, E−core,
toroid and bobbin core, as well as different core materials
such as ferrites and powdered iron from different
manufacturers.
For high quality design regulators the toroid core seems to
be the best choice. Since the magnetic flux is contained
within the core, it generates less EMI, reducing noise
problems in sensitive circuits. The least expensive is the
bobbin core type, which consists of wire wound on a ferrite
rod core. This type of inductor generates more EMI due to
the fact that its core is open, and the magnetic flux is not
contained within the core.
When multiple switching regulators are located on the
same printed circuit board, open core magnetics can cause
Continuous and Discontinuous Mode of Operation
The LM2595 step−down converter can operate in both the
continuous and the discontinuous modes of operation. The
regulator works in the continuous mode when loads are
relatively heavy, the current flows through the inductor
continuously and never falls to zero. Under light load
conditions, the circuit will be forced to the discontinuous
mode when inductor current falls to zero for certain period
of time (see Figure 20 and Figure 21). Each mode has
distinctively different operating characteristics, which can
affect the regulator performance and requirements. In many
cases the preferred mode of operation is the continuous
mode. It offers greater output power, lower peak currents in
the switch, inductor and diode, and can have a lower output
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15
LM2595
interference between two or more of the regulator circuits,
especially at high currents due to mutual coupling. A toroid,
pot core or E−core (closed magnetic structure) should be
used in such applications.
inductor and/or the LM2595. Different inductor types have
different saturation characteristics, and this should be kept
in mind when selecting an inductor.
Exceeding an inductor’s maximum current rating may
cause the inductor to overheat because of the copper wire
losses, or the core may saturate. Core saturation occurs when
the flux density is too high and consequently the cross
sectional area of the core can no longer support additional
lines of magnetic flux.
This causes the permeability of the core to drop, the
inductance value decreases rapidly and the inductor begins
to look mainly resistive. It has only the DC resistance of the
winding. This can cause the switch current to rise very
rapidly and force the LM2595 internal switch into
cycle−by−cycle current limit, thus reducing the DC output
load current. This can also result in overheating of the
VERTICAL RESOLUTION 25 mA/DIV
Do Not Operate an Inductor Beyond its
Maximum Rated Current
0.05 A
Inductor
Current
Waveform
0A
0.05 A
Power
Switch
Current
Waveform
0A
HORIZONTAL TIME BASE: 2.0 ms/DIV
Figure 21. Discontinuous Mode Switching Current
Waveforms
GENERAL RECOMMENDATIONS
Output Voltage Ripple and Transients
Source of the Output Ripple
Minimizing the Output Ripple
In order to minimize the output ripple voltage it is possible
to enlarge the inductance value of the inductor L1 and/or to
use a larger value output capacitor. There is also another way
to smooth the output by means of an additional LC filter (3 mH,
100 mF), that can be added to the output (see Figure 31) to
further reduce the amount of output ripple and transients.
With such a filter it is possible to reduce the output ripple
voltage transients 10 times or more. Figure 22 shows the
difference between filtered and unfiltered output waveforms
of the regulator shown in Figure 31.
The lower waveform is from the normal unfiltered output
of the converter, while the upper waveform shows the output
ripple voltage filtered by an additional LC filter.
Since the LM2595 is a switch mode power supply
regulator, its output voltage, if left unfiltered, will contain a
sawtooth ripple voltage at the switching frequency. The
output ripple voltage value ranges from 0.5% to 3% of the
output voltage. It is caused mainly by the inductor sawtooth
ripple current multiplied by the ESR of the output capacitor.
Short Voltage Spikes and How to Reduce Them
The regulator output voltage may also contain short
voltage spikes at the peaks of the sawtooth waveform (see
Figure 22). These voltage spikes are present because of the
fast switching action of the output switch, and the parasitic
inductance of the output filter capacitor. There are some
other important factors such as wiring inductance, stray
capacitance, as well as the scope probe used to evaluate these
transients, all these contribute to the amplitude of these
spikes. To minimize these voltage spikes, low inductance
capacitors should be used, and their lead lengths must be
kept short. The importance of quality printed circuit board
layout design should also be highlighted.
The Surface Mount Package D2PAK and its
Heatsinking
The other type of package, the surface mount D2PAK, is
designed to be soldered to the copper on the PC board. The
copper and the board are the heatsink for this package and
the other heat producing components, such as the catch
diode and inductor. The PC board copper area that the
package is soldered to should be at least 0.4 in2 (or
100 mm2) and ideally should have 2 or more square inches
(1300 mm2) of 0.0028 inch copper. Additional increasing of
copper area beyond approximately 3.0 in2 (2000 mm2) will
not improve heat dissipation significantly. If further thermal
improvements are needed, double sided or multilayer PC
boards with large copper areas should be considered.
Voltage spikes
caused by
switching action
of the output
switch and the
parasitic
inductance of the
output capacitor
VERTRICAL
RESOLUTION
20 mV/DIV
Filtered
Output
Voltage
Unfiltered
Output
Voltage
Thermal Analysis and Design
The following procedure must be performed to determine
the operating junction temperature. First determine:
1. PD(max) maximum regulator power dissipation in the
application.
2. TA(max) maximum ambient temperature in the
application.
HORIZONTAL TIME BASE: 5.0 ms/DIV
Figure 22. Output Ripple Voltage Waveforms
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LM2595
Packages Not on a Heatsink (Free−Standing)
3. TJ(max)
maximum allowed junction temperature
(125°C for the LM2595). For a conservative
design, the maximum junction temperature
should not exceed 110°C to assure safe
operation. For every additional +10°C
temperature rise that the junction must
withstand, the estimated operating lifetime
of the component is halved.
4. RqJC
package thermal resistance junction−case.
package thermal resistance junction−ambient.
5. RqJA
(Refer to Maximum Ratings on page 2 of this data sheet or
RqJC and RqJA values).
For a free−standing application when no heatsink is used,
the junction temperature can be determined by the following
expression:
TJ = (RqJA) (PD) + TA
Where (RqJA) (PD) represents the junction temperature rise
caused by the dissipated power and TA is the maximum
ambient temperature.
Packages on a Heatsink
If the actual operating junction temperature is greater than
the selected safe operating junction temperature determined
in step 3, than a heatsink is required. The junction
temperature will be calculated as follows:
The following formula is to calculate the approximate
total power dissipated by the LM2595:
TJ = PD (RqJA + RqCS + RqSA) + TA
PD = (Vin x IQ) + d x ILoad x Vsat
Where RqJC is the thermal resistance junction−case,
RqCS is the thermal resistance case−heatsink,
RqSA is the thermal resistance heatsink−ambient.
If the actual operating temperature is greater than the
selected safe operating junction temperature, then a larger
heatsink is required.
where d is the duty cycle and for buck converter
V
t
d + on + O ,
V in
T
IQ
(quiescent current) and Vsat can be found in the
LM2595 data sheet,
Vin is minimum input voltage applied,
VO is the regulator output voltage,
ILoad is the load current.
The dynamic switching losses during turn−on and
turn−off can be neglected if proper type catch diode is used.
The junction temperature can be determined by the
following expression:
Some Aspects That can Influence Thermal Design
It should be noted that the package thermal resistance and
the junction temperature rise numbers are all approximate,
and there are many factors that will affect these numbers,
such as PC board size, shape, thickness, physical position,
location, board temperature, as well as whether the
surrounding air is moving or still.
Other factors are trace width, total printed circuit copper
area, copper thickness, single− or double−sided, multilayer
board, the amount of solder on the board or even color of the
traces.
The size, quantity and spacing of other components on the
board can also influence its effectiveness to dissipate the
heat.
TJ = (RqJA) (PD) + TA
where (RqJA)(PD) represents the junction temperature rise
caused by the dissipated power and TA is the maximum
ambient temperature.
12 to 25 V
Unregulated
DC Input
Cin
100 mF/50 V
R4
Feedback
+Vin
L1
100 mH
LM2595
CFF
ON/OFF
GND
D1
1N5819
R3
Cout
220 mF
Figure 23. Inverting Buck−Boost Develops −12 V
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17
−12 V @ 0.7 A
Regulated
Output
LM2595
ADDITIONAL APPLICATIONS
Using a delayed startup arrangement, the input capacitor
can charge up to a higher voltage before the switch−mode
regulator begins to operate.
The high input current needed for startup is now partially
supplied by the input capacitor Cin.
It has been already mentioned above, that in some
situations, the delayed startup or the undervoltage lockout
features could be very useful. A delayed startup circuit
applied to a buck−boost converter is shown in Figure 28.
Figure 30 in the “Undervoltage Lockout” section describes
an undervoltage lockout feature for the same converter
topology.
Inverting Regulator
An inverting buck−boost regulator using the LM2595 is
shown in Figure 23. This circuit converts a positive input
voltage to a negative output voltage with a common ground
by bootstrapping the regulators ground to the negative
output voltage. By grounding the feedback pin, the regulator
senses the inverted output voltage and regulates it.
In this example the LM2595 is used to generate a −12 V
output. The maximum input voltage in this case cannot
exceed +28 V because the maximum voltage appearing
across the regulator is the absolute sum of the input and
output voltages and this must be limited to a maximum of
40 V.
This circuit configuration is able to deliver approximately
0.25 A to the output when the input voltage is 12 V or higher.
At lighter loads the minimum input voltage required drops
to approximately 4.7 V, because the buck−boost regulator
topology can produce an output voltage that, in its absolute
value, is either greater or less than the input voltage.
Since the switch currents in this buck−boost configuration
are higher than in the standard buck converter topology, the
available output current is lower.
This type of buck−boost inverting regulator can also
require a larger amount of startup input current, even for
light loads. This may overload an input power source with
a current limit less than 1.0 A.
Such an amount of input startup current is needed for at
least 2.0 ms or more. The actual time depends on the output
voltage and size of the output capacitor.
Because of the relatively high startup currents required by
this inverting regulator topology, the use of a delayed startup
or an undervoltage lockout circuit is recommended.
The inverting regulator operates in a different manner
than the buck converter and so a different design procedure
has to be used to select the inductor L1 or the output
capacitor Cout.
The output capacitor values must be larger than what is
normally required for buck converter designs. Low input
voltages or high output currents require a large value output
capacitor (in the range of thousands of mF).
The recommended range of inductor values for the
inverting converter design is between 68 mH and 220 mH. To
select an inductor with an appropriate current rating, the
inductor peak current has to be calculated.
The following formula is used to obtain the peak inductor
current:
I
(V ) |V |)
O ) V in x t on
[ Load in
2L 1
V
in
|V |
O
where t on +
x 1.0 , and f osc + 52 kHz.
V ) |V | f osc
in
O
I
peak
Under normal continuous inductor current operating
conditions, the worst case occurs when Vin is minimal.
R4
Feedback
12 to 40 V
Unregulated
DC Input
Cin
100 mF/50 V
Design Recommendations:
+Vin
C1
0.1 mF
L1
100 mH
LM2595
CFF
ON/OFF
GND
D1
1N5819
R2
47k
R3
Cout
220 mF
−12 V @ 0.25 A
Regulated
Output
Figure 24. Inverting Buck−Boost Develops with Delayed Startup
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LM2595
+V
+Vin
+Vin
Shutdown
Input
5.0 V
0
Cin
R1
100 mF 47 k
0
LM2595
7
Off
On
R2
5.6 k
5
ON/OFF 6
+Vin
GN
D
+Vin
7
LM2595
Cin
100 mF
On
Off
Shutdown
Input
R3
470
Q1
2N3906
R2
47 k
5
ON/OFF 6
-Vout
R1
12 k
MOC8101
NOTE: This picture does not show the complete circuit.
GN
D
-Vout
NOTE: This picture does not show the complete circuit.
Figure 25. Inverting Buck−Boost Regulator Shutdown
Circuit Using an Optocoupler
Figure 26. Inverting Buck−Boost Regulator Shutdown
Circuit Using a PNP Transistor
With the inverting configuration, the use of the ON/OFF
pin requires some level shifting techniques. This is caused
by the fact, that the ground pin of the converter IC is no
longer at ground. Now, the ON/OFF pin threshold voltage
(1.3 V approximately) has to be related to the negative
output voltage level. There are many different possible shut
down methods, two of them are shown in Figures 25 and 26.
Negative Boost Regulator
This example is a variation of the buck−boost topology
and it is called negative boost regulator. This regulator
experiences relatively high switch current, especially at low
input voltages. The internal switch current limiting results in
lower output load current capability.
The circuit in Figure 27 shows the negative boost
configuration. The input voltage in this application ranges
from −5.0 V to −12 V and provides a regulated −12 V output.
If the input voltage is greater than −12 V, the output will rise
above −12 V accordingly, but will not damage the regulator.
R4
Cout
470 mF
Feedback
+Vin
Cin
100 mF/
50 V
−12 V
Unregulated
DC Input
LM2595
ON/OFF
GND
D1
1N5822
R3
−12 V @ 0.25 A
Regulated
Output
L1
100 mH
Figure 27. Negative Boost Regulator
Design Recommendations:
values for the negative boost regulator is the same as for
inverting converter design.
Another important point is that these negative boost
converters cannot provide current limiting load protection in
the event of a short in the output so some other means, such
as a fuse, may be necessary to provide the load protection.
The same design rules as for the previous inverting
buck−boost converter can be applied. The output capacitor
Cout must be chosen larger than would be required for a what
standard buck converter. Low input voltages or high output
currents require a large value output capacitor (in the range
of thousands of mF). The recommended range of inductor
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LM2595
Delayed Startup
There are some applications, like the inverting regulator
already mentioned above, which require a higher amount of
startup current. In such cases, if the input power source is
limited, this delayed startup feature becomes very useful.
To provide a time delay between the time when the input
voltage is applied and the time when the output voltage
comes up, the circuit in Figure 28 can be used. As the input
voltage is applied, the capacitor C1 charges up, and the
voltage across the resistor R2 falls down. When the voltage
on the ON/OFF pin falls below the threshold value 1.3 V, the
regulator starts up. Resistor R1 is included to limit the
maximum voltage applied to the ON/OFF pin. It reduces the
power supply noise sensitivity, and also limits the capacitor
C1 discharge current, but its use is not mandatory.
When a high 50 Hz or 60 Hz (100 Hz or 120 Hz
respectively) ripple voltage exists, a long delay time can
cause some problems by coupling the ripple into the
ON/OFF pin, the regulator could be switched periodically
on and off with the line (or double) frequency.
+Vin
R1
47 k
5
ON/OFF 6
R2
10 k
R1
10 k
ǒ
Vth ≈ 13 V
Figure 29. Undervoltage Lockout Circuit for
Buck Converter
+Vin
LM2595
2
R2
15 k
GN
D
Cin
100 mF 5
R3
47 k
Z1
1N5242B
R2
47 k
Q1
2N3904
ON/OFF 3
GND
Vth ≈ 13 V
R1
15 k
Vout
NOTE: This picture does not show the complete circuit.
Figure 30. Undervoltage Lockout Circuit for
Buck−Boost Converter
Some applications require the regulator to remain off until
the input voltage reaches a certain threshold level. Figure 29
shows an undervoltage lockout circuit applied to a buck
regulator. A version of this circuit for buck−boost converter
is shown in Figure 30. Resistor R3 pulls the ON/OFF pin
high and keeps the regulator off until the input voltage
reaches a predetermined threshold level with respect to the
ground Pin 3, which is determined by the following
expression:
Z1
GND
NOTE: This picture does not show the complete circuit.
Undervoltage Lockout
[V
ON/OFF 3
Q1
2N3904
Figure 28. Delayed Startup Circuitry
th
Cin
100 mF 5
R3
47 k
Z1
1N5242B
NOTE: This picture does not show the complete circuit.
V
LM2595
2
LM2595
7
Cin
100 mF
+Vin
+Vin
+Vin
C1
0.1 mF
+Vin
Adjustable Output, Low−Ripple Power Supply
A 1.0 A output current capability power supply that
features an adjustable output voltage is shown in Figure 31.
This regulator delivers 1.0 A into 1.2 V to 35 V output.
The input voltage ranges from roughly 3.0 V to 40 V. In order
to achieve a 10 or more times reduction of output ripple, an
additional L−C filter is included in this circuit.
Ǔ
(Q1)
) 1.0 ) R2 V
R1 BE
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20
LM2595
40 V Max
Unregulated
DC Input
Feedback
4
+Vin
LM2595
2
Cin
100 mF
Output
3
GND
5
L1
100 mH
L2
3 mH
1
ON/OFF
CFF
2 to 35 V @ 1.0 A
R2
50 k
Cout
220 mF
D1
1N5822
R1
1.21 k
C1
100 mF
Optional Output
Ripple Filter
Figure 31. 2 to 35 V Adjustable 1.0 A Power Supply with Low Output Ripple
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21
Output
Voltage
LM2595
THE LM2595 STEP−DOWN VOLTAGE REGULATOR WITH 5.0 V @ 1.0 A OUTPUT POWER CAPABILITY.
TYPICAL APPLICATION WITH THROUGH−HOLE PC BOARD LAYOUT
4
Unregulated
DC Input
+Vin
+Vin = 10 V to 40 V
Feedback
L1
68 mH
LM2595
2
Output
3
GND
5
Regulated
Output Filtered
1
ON/OFF
C1
100 mF
/50 V
R2
3.0 k
D1
1N5819
ON/OFF
C2
470 mF
/25 V
−
−
−
−
−
−
−
100 mF, 50 V, Aluminium Electrolytic
470 mF, 25 V, Aluminium Electrolytic
1.0 A, 40 V, Schottky Rectifier, 1N5819
100 mH, DO3340P, Coilcraft
1.0 kW, 0.25 W
3.0 kW, 0.25 W
See Table 1
Vout2 = 5.0 V @ 1.0 A
R1
1.0 k
V
C1
C2
D1
L1
R1
R2
Cff
CFF
ǒ
Ǔ
R2
out + V ref ) 1.0 ) R1
Vref = 1.23 V
R1 is between 1.0 k and 5.0 k
Figure 32. Schematic Diagram of the 5.0 V @ 1.0 A Step−Down Converter Using the LM2595−ADJ
NOTE: Not to scale.
NOTE: Not to scale.
Figure 33. Printed Circuit Board Layout With
Component
Figure 34. Printed Circuit Board Layout
Copper Side
References
•
•
•
•
National Semiconductor LM2595 Data Sheet and Application Note
National Semiconductor LM2595 Data Sheet and Application Note
Marty Brown “Practical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990
Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995
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22
LM2595
ORDERING INFORMATION
Package
Shipping†
TO−220
(Pb−Free)
50 Units / Rail
LM2595TVADJG
TO−220 (F)
(Pb−Free)
50 Units / Rail
LM2595DSADJG
D2PAK
(Pb−Free)
50 Units / Rail
LM2595DSADJR4G
D2PAK
(Pb−Free)
800 / Tape & Reel
Device
LM2595TADJG
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
MARKING DIAGRAMS
TO−220
TV SUFFIX
CASE 314B
TO−220
T SUFFIX
CASE 314D
LM
2595T−ADJ
AWLYWWG
LM
2595T−ADJ
AWLYWWG
D2PAK
DS SUFFIX
CASE 936A
LM
2595−ADJ
AWLYWWG
1
1
5
1
A
WL
Y
WW
G
5
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
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23
5
LM2595
PACKAGE DIMENSIONS
TO−220
TV SUFFIX
CASE 314B−05
ISSUE L
C
B
−P−
Q
OPTIONAL
CHAMFER
E
A
U
K
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 0.043 (1.092) MAXIMUM.
L
S
W
F
5X
0.24 (0.610)
D
0.10 (0.254)
M
T P
DIM
A
B
C
D
E
F
G
H
J
K
L
N
Q
S
U
V
W
J
5X
G
V
M
H
T
N
M
−T−
SEATING
PLANE
INCHES
MIN
MAX
0.572
0.613
0.390
0.415
0.170
0.180
0.025
0.038
0.048
0.055
0.850
0.935
0.067 BSC
0.166 BSC
0.015
0.025
0.900
1.100
0.320
0.365
0.320 BSC
0.140
0.153
--0.620
0.468
0.505
--0.735
0.090
0.110
MILLIMETERS
MIN
MAX
14.529 15.570
9.906 10.541
4.318
4.572
0.635
0.965
1.219
1.397
21.590 23.749
1.702 BSC
4.216 BSC
0.381
0.635
22.860 27.940
8.128
9.271
8.128 BSC
3.556
3.886
--- 15.748
11.888 12.827
--- 18.669
2.286
2.794
TO−220
T SUFFIX
CASE 314D−04
ISSUE F
−T−
B
−Q−
B1
DETAIL A-A
A
U
C
L
J
H
G
5 PL
0.356 (0.014)
M
T Q
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 10.92 (0.043) MAXIMUM.
DIM
A
B
B1
C
D
E
G
H
J
K
L
Q
U
1234 5
K
D
E
SEATING
PLANE
M
B
B1
DETAIL A−A
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24
INCHES
MIN
MAX
0.572
0.613
0.390
0.415
0.375
0.415
0.170
0.180
0.025
0.038
0.048
0.055
0.067 BSC
0.087
0.112
0.015
0.025
0.977
1.045
0.320
0.365
0.140
0.153
0.105
0.117
MILLIMETERS
MIN
MAX
14.529 15.570
9.906 10.541
9.525 10.541
4.318
4.572
0.635
0.965
1.219
1.397
1.702 BSC
2.210
2.845
0.381
0.635
24.810 26.543
8.128
9.271
3.556
3.886
2.667
2.972
LM2595
PACKAGE DIMENSIONS
D2PAK
D2T SUFFIX
CASE 936A−02
ISSUE C
−T−
OPTIONAL
CHAMFER
A
E
U
S
K
B
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. TAB CONTOUR OPTIONAL WITHIN DIMENSIONS A
AND K.
4. DIMENSIONS U AND V ESTABLISH A MINIMUM
MOUNTING SURFACE FOR TERMINAL 6.
5. DIMENSIONS A AND B DO NOT INCLUDE MOLD
FLASH OR GATE PROTRUSIONS. MOLD FLASH
AND GATE PROTRUSIONS NOT TO EXCEED 0.025
(0.635) MAXIMUM.
TERMINAL 6
V
H
1 2 3 4 5
M
D
0.010 (0.254)
M
T
L
G
INCHES
MIN
MAX
0.386
0.403
0.356
0.368
0.170
0.180
0.026
0.036
0.045
0.055
0.067 BSC
0.539
0.579
0.050 REF
0.000
0.010
0.088
0.102
0.018
0.026
0.058
0.078
5 _ REF
0.116 REF
0.200 MIN
0.250 MIN
DIM
A
B
C
D
E
G
H
K
L
M
N
P
R
S
U
V
P
N
R
C
SOLDERING FOOTPRINT*
8.38
0.33
MILLIMETERS
MIN
MAX
9.804
10.236
9.042
9.347
4.318
4.572
0.660
0.914
1.143
1.397
1.702 BSC
13.691
14.707
1.270 REF
0.000
0.254
2.235
2.591
0.457
0.660
1.473
1.981
5 _ REF
2.946 REF
5.080 MIN
6.350 MIN
1.702
0.067
10.66
0.42
16.02
0.63
3.05
0.12
SCALE 3:1
1.016
0.04
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
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LM2595/D