DN06040/D Design Note – DN06040/D Universal Input, 20 W, LED Ballast Device Application Input Voltage Output Power Topology I/O Isolation NCP1351 Solid State Lighting 85 – 265 Vac 20 W Flyback Yes Other Specifications Output 1 Maximum Output Voltage Ripple Nominal Current 33 V Not Given 700 mA PFC (Yes/No) No Target Efficiency 80% at nominal load Max Size 125 x 37 x 35 mm Operating Temp Range Cooling Method/Supply Orientation Signal Level Control 0 to +70°C Convection No Other Requirements Circuit Description Key Features The NCP1351 controller provides for a low cost, variable frequency, flyback converter. It incorporates a very low quiescent current allowing for high value resistors to be used as a start-up circuit direct from the HV rail. The design comprises and input filter, bridge rectifier (using low cost 1N4007 diodes), bulk capacitors and line inductor in π-filter arrangement, the power stage, rectifier diode and smoothing capacitors. Feedback is CVCC, constant current drive for the LED’s with a constant voltage in the event of an open circuit output. In order to stay below IEC6100-3-2 Class C, the design has been optimized at <25 W, so assuming 80% efficiency the maximum output power is ~20 W. y Wide input voltage range – 85 Vac to 265 Vac y Small size, and low cost y Good line regulation y High efficiency y Overload and short circuit protection. Number of LED’s in series LED Current 350 mA 700 mA 1A 1.5 A ® 11 ® LUXEON III 10 6 4 LUXEON® Rebel 10 6 4 LUXEON® K2 11 6 4 2 8 5 Note 1 12 8 Note 1 Note 1 12 7 5 Note 1 VZ (D10) 45 V 33 V 22 V 12 V R12 & R13 3R6 1R8 1R2 0R8 LUXEON I Cree XR-E Cree XP-E ® ® OSRAM Platinum Dragon® 12 Note 1 Note 1 Note 1 Note 1 Out of LED specification September 2008, Rev. 2 www.onsemi.com 1 DN06040/D Schematic September 2008, Rev. 2 www.onsemi.com 2 DN06040/D LED Current The light output of an LED is determined by the forward current so the control loop will be constant current, with a simple Zener to limit the maximum output voltage. Typical forward voltages vary by LED supplier, below are the nominal forward voltage characteristics of the LUXEON® K2 at different operating currents. IF The output current is sensed by a series resistance, once the voltage drop across this reaches the baseemitter threshold of the PNP transistor current flows in the opto-coupler diode and thus in the FB pin of the NCP1351. The LED current is thus set by: I LED = VF 350 mA 3.42 V 700 mA 3.60 V 1000 mA 3.72 V 1500 mA 3.85 V 0.6V .................................................. (Eq.1) RSENSE Total sense resistor power dissipation is: PD = I LED × 0.6V ............................................. (Eq.2) So for 700 mA we need a 0.9 Ω sense resistor capable of dissipating 420 mW, two 330 mW surface mount resistors, 1.8 Ω each in parallel, are used. Driving eight LED’s at 700mA thus gives an output power of 20.2 W at 28.8 V. Inductor selection In a flyback converter the inductance required in the transformer primary is dependant on the mode of operation and the output power. Discontinuous operation requires lower inductance but results in higher peak to average current waveforms, and thus higher losses. For low power designs, such as this ballast, the inductance is designed to be just continuous (or just discontinuous) under worst case conditions, that is minimum line and maximum load. The specification for this ballast is as follows: • Universal input – 85 Vac to 265 Vac • 25 W maximum input power – PFC limit • Assuming 80% efficiency – 20 W output power • 700 mA output current • 100 kHz operation at full load This gives us a minimum DC input voltage of 120 V, there will be some sag on the DC bulk capacitors so an allowance will be made for this by using 80 V as the minimum input voltage, including MOSFET drop etc. First we need to calculate the turn’s ratio, this is set by the MOSFET drain rating, line voltage and reflected secondary voltage. Since this is a constant current circuit we are designing, with a varying output voltage, we need the maximum output voltage. ¾ VIN(max) is the maximum rectified input = 375 V. ¾ VIN(min) is the minimum rectified input = 80 V. ¾ VOUT is 35 V (20 W @ 700 mA is 29 V plus a margin for safety). With a 600 V MOSFET and derating of 80%, our maximum allowable drain voltage is: V D (max ) = 600 × 0.8 = 480 V ..........................(Eq.3) Good results are obtained if we set VCLAMP, at ~150% of the reflected secondary: kC = ¾ VCLAMP × N (VOUT + V f ) = 1.5 ................................... (Eq.5) Vf = 0.7 V as we will need a high voltage diode. Re-arranging for N: N = N S 1.5 × (35 + 0.7 ) = NP 105 .............................. (Eq.6) = 0.51 We will use a ratio of 0.5 or 2:1, this will give a good transformer construction. We can now calculate the maximum duty cycle running in CCM: δ MAX = VOUT VOUT (35 + 0.7 ) = + VIN (min ) N (35 + 0.7 ) + 80 × 0.5 = 0.47 .......................................................................... (Eq.7) And thus headroom, VCLAMP for the reflected secondary voltage and leakage spike of: VCLAMP = VD (max ) − VIN (max ) = 480 − 375 = 105 V September 2008, Rev. 2 ..........(Eq.4) www.onsemi.com 3 DN06040/D I AVE = I I1 I1 = ∆IL IVALLEY 25 = 313 mA ................... (Eq.12) 80 I AVE = δ max 0.313 = 662 mA ...................... (Eq.13) 0.47 Demonstrating that ∆IL does equal twice I1 and that the peak primary current is 1.32 A. IAVE We can calculate the RMS current in the MOSFET and sense resistor for dissipation purposes. For a steppedsawtooth waveform of this type the equation is: t δTSW TSW Looking at the waveform of the current flowing in the primary of the inductor (above) if we define a term k equal to; ΔI k = L ...........................................................(Eq.8) I1 (V δ MAX ) Thus: I RMS .........................................(Eq.9) Then we can determine the inductance we require. If k = 2 then we are in boundary conduction mode as the ripple current equals twice the average pulse current, so setting k to 2: = 283 μH ............(Eq.10) Thus we can now find the primary ripple current assuming operation in boundary conduction mode: VIN (min)TON L 2 = We can also determine the current sense resistor, allowing for a drop across the resistor of 0.8 V: RSENSE = VDROP 0 .8 = = 0.61 Ω ................. (Eq.16) I PK 1.32 The total power dissipation is: 2 2 100 × 10 3 × 2.0 × 25 ΔI L = 1 ⎛ 1.32 ⎞ = 0.665 × 0.47 × 1 + ⎜ ⎟ 3 ⎝ 2 × 0.665 ⎠ ......................................................................... (Eq.15) f SW kPIN (80 × 0.47 ) 2 ⎞ ⎟⎟ ........................ (Eq.14) ⎠ = 526 mA 2 IN (min) 1 ⎛ ΔI δ 1 + ⎜⎜ L 3 ⎝ 2I1 I RMS = I 1 And use the equation: L= VIN (min) = The average pulse current, I1, is: IPK L= PIN VIN (min)δ max Lf SW 80 × 0.47 = = 1.32 A 283 ×10 − 6 ×100 × 10 3 ........(Eq.11) PD ( sense ) = I RMS RSENSE = 0.526 2 × 0.61 ≅ 170 mW ........ (Eq.17) Two 1.2 Ω resistors in parallel will be used as sub 1 Ω resistors typically cost more. The threshold voltage for the current sense is set by an offset resistor; this has a bias current of 270 µA in it so we can determine the resistor value: ROFFSET = VSENSE 0.8 = ≅ 3.0 kΩ ...... (Eq.18) 270 × 10 −6 I BIAS The average input current, IAVE, is: September 2008, Rev. 2 www.onsemi.com 4 DN06040/D Rectifier snubber Testing demonstrated the need for snubbing on the rectifier as there was a large amount of ringing present after the rectifier turns off. The snubber consists of a resistor and capacitor in series, and knowing the junction capacitance and ringing frequency we can determine the necessary values: L .......................................................(Eq.19) Cj Rs = Cs = 2π LC j Rs ..............................................(Eq.20) Knowing that: f = 1 2π LC j ................................................(Eq.21) We can determine L, the stray inductance which then allows us to calculate the necessary snubber resistor. ¾ f = 14.5 MHz (measured on oscilloscope) ¾ Cj = 80 pF (datasheet figure for MUR840 at 62 V) L= 1 4C j (πf ) 2 = 1 4 × 80 ×10 −12 ( × π × 14.5 ×10 6 ) 2 = 1.51 μH Rs = 1.51 × 10 −6 = 137 Ω ......................... (Eq.23) 80 × 10 −12 2 × π × 1.51 × 10 −6 × 80 × 10 −12 Cs = = 504 pF 137 .............................................................................. (Eq.24) The nearest standard values are 470 pF and 140 Ω, inserting these into the circuit eliminated the ringing due to the rectifier. Auxiliary winding Normally in a flyback converter the auxiliary winding would be in the form of a flyback winding, i.e. in phase with the output winding, and thus provide a semi-regulated voltage to supply the controller. As this ballast is current controlled and the output voltage can vary over a considerable range depending on the number of LED’s connected, a forward phased winding is used. The auxiliary will therefore vary with line rather than output voltage. Since neither option could supply sufficient volts at low input/output voltage whilst still staying below the maximum VCC figure of 28 V, a voltage regulator is used formed by Q1 and D6. Below ~20 V the regulator does nothing other than act as a small volt drop, however as the voltage rises it clamps the voltage to around 20.7 V, since the current is very low into the VCC pin there is very little loss. ...............................................................................(Eq.22) September 2008, Rev. 2 www.onsemi.com 5 DN06040/D MAGNETICS DESIGN DATA SHEET Project / Customer: ON Semiconductor Part Description: 25 W Transformer Schematic ID: - Core Type: EE25 Core Gap: Gap for 250 µH Inductance: 250 µH Bobbin Type: NIC 10-pin vertical Windings (in order): Winding # / type Turns / Material / Gauge / Insulation Data N1, Primary Start on pin 1 and wind 20 turns, of 0.28 mm triple insulated wire (e.g. Tex-E), in one neat layer across the entire bobbin width. Finish on pin 2. N2, Secondary Start on pins 9&10 and wind 20 turns, of 0.8 mm Grade II ECW, distributed evenly across the entire bobbin width. Finish on pins 6&7. N3, Primary Start on pin 2 and wind 20 turns, of 0.28 mm triple insulated wire (e.g. Tex-E), in one neat layer across the entire bobbin width. Finish on pin 3. N4, Primary (Aux) Start on pin 4 and wind 5 turns, of 0.28 mm triple insulated wire, in one neat layer spread evenly across the entire bobbin width. Finish on pin 5. Sleeving and insulation between primary and secondary as required to meet the requirements of double insulation. Primary leakage inductance (pins 6&7 and 9&10 shorted together) to be < 6 µH NIC part number: NLT282224W3P4020S5P10F Hipot: 3 kV between pins 1, 2, 3, 4 & 5 and pins 6, 7,8, 9 & 10 for 60 seconds. Lead Breakout / Pinout Schematic 1 N1 2 N2 N3 3 5 6 4 7 3 8 2 9 1 10 6, 7 9, 10 5 mm 4 N4 5 September 2008, Rev. 2 15 mm www.onsemi.com 6 DN06040/D Bill of Materials Ref Part Type / Value Comment Footprint Description C1 C2 Manufacturer Part Number 220nF X2 275VAC 18X10mm, 15mm pitch X-class EMI suppression capacitor NIC NPX224M275VX2MF 47uF 400V Ø16mm, 7.5mm pitch General purpose high voltage electrolytic NIC NRE-H470M400V16X31.5F C3 470pF 100V X7R 1206 Ceramic chip capacitor NIC NMC1206X7R471K100F C4 100nF 50V X7R 0603 Ceramic chip capacitor NIC NMC0603X7R104K50F C5 220nF 50V X7R 0805 Ceramic chip capacitor NIC NMC0805X7R224K50F C6 4.7uF 35V Ø5mm, 2mm pitch General purpose low voltage electrolytic NIC NRWA4R7M50V5X11F C7 180pF 50V NP0 0603 Ceramic chip capacitor NIC NMC0603NPO181J50F NMC0603X7R473K50F C8 47nF 50V X7R 0603 Ceramic chip capacitor NIC C9 220nF 50V X7R 0805 Ceramic chip capacitor NIC NMC0805X7R224K50F C10 10nF (0.01 uF) 1kV 1210 Ceramic chip capacitor JOHANSON 102S41W103KV4E C11 1uF 50V Ø5mm, 2mm pitch General purpose low voltage electrolytic NIC NRWA1R0M50V5X11F C12 1nF Y1 Radial, pitch 10mm Ceramic Y-class capacitor Murata DE1E3KX102MN4AL01 C13 470uF 63V Ø12.5mm, 5mm pitch Miniature low impedance electrolytic NIC NRSZ471M63V12.5X25F C14 Not Inserted C15 220nF 100V X7R 1206 Ceramic chip capacitor NIC NMC1206X7R224K100F C16 1uF 50V 1206 Ceramic chip capacitor NIC NMC1206X7R105K50F D1 1N4007 1A, 1000V Axial Axial Lead, Standard Recovery ON Semiconductor 1N4007RLG D2 1N4007 1A, 1000V Axial Axial Lead, Standard Recovery ON Semiconductor 1N4007RLG D3 1N4007 1A, 1000V Axial Axial Lead, Standard Recovery ON Semiconductor 1N4007RLG D4 1N4007 1A, 1000V Axial Axial Lead, Standard Recovery ON Semiconductor 1N4007RLG D5 MMSD4148 200mA, 100V SOD-123 Switching diode ON Semiconductor MMSD4148T1G D6 20V 1.5W SMA Zener Diode ON Semiconductor 1SMA5932BT3G D7 MURA160 1A, 600V SMA Ultrafast rectifier ON Semiconductor MURA160T3G D8 200mA, 100V SOD-123 Switching diode ON Semiconductor MMSD4148T1G 8A, 400V TO-220 Ultrafast Power Rectifier ON Semiconductor MUR840G D10 MMSD4148 MUR840 (MUR860 Alt) 33V 5%, 200mW SOD323 Zener diode ON Semiconductor MM3Z33VT1G IC1 NCP1351B - SOIC8 Variable Off-Time PWM Controller ON Semiconductor NCP1351BDR2G IC2 HCPL-817 Wide pitch HCPL-817-300E Opto-coupler HCPL-817 Agilent HCPL-817-W0AE - WE-LF 662/SH Common Mode Choke Wurth/Midcom 744 662 0027 D9 L1 AC 2-Way 5mm pitch - Screw Terminal Keystone 8718 LED 2-Way 5mm pitch - Screw Terminal Phoenix 1985881 M1 25.9°C/W - - Heatsink Aavid 577102B00000G M2 25.9°C/W - - Heatsink Aavid 577102B00000G Q1 BC847 45V SOT-23 General purpose NPN ON Semiconductor BC847ALT1G Q2 IRFBC40A 600V TO-220 MOSFET IR IRFBC40A Q3 BC857 -45V SOT-23 General purpose PNP ON Semiconductor BC857ALT1G R1 150R 0.33W, 5% 1210 Resistor thick film NRC NIC NRC25J151F R2 2k2 0.1W, 5% 0603 Resistor thick film NRC NIC NRC06J222F R3 3k0 0.1W, 5% 0603 Resistor thick film NRC NIC NRC06J302F R4a 1R2 1W, 5% 2512 Resistor thick film NRC NIC NRC100J1R2F R4b 1R2 1W, 5% 2512 Resistor thick film NRC NIC NRC100J1R2F R5 1M 0.5W, 5% Axial Metal Film Resistor Vishay SFR2500001004J-R500 R6 1M 0.5W, 5% Axial Metal Film Resistor Vishay SFR2500001004J-R500 R7 2k2 0.125W,5% 0805 Resistor thick film NRC NIC NRC10J222BF R8 10R 0.25W,5% 1206 Resistor thick film NRC NIC NRC12J100F R9 6k8 0.1W,5% 0603 Resistor thick film NRC NIC NRC06J682TRF R10 12k 2W,5% Axial Carbon film resistor NIC NCF200J123TRF R11 200R 0.125W,5% 0805 Resistor thick film NRC NIC NRC10J201F R12 1R8 0.33W,1% 1210 Resistor thick film NRC NIC NRC25J1R8F R13 1R8 0.33W,1% 1210 Resistor thick film NRC NIC NRC25J1R8F R14 2K2 0.125W,5% 0805 Resistor thick film NRC NIC NRC10J222BF R15 4k3 0.125W,5% 0805 Resistor thick film NRC NIC NRC10J432F R16 0 ohm Short 0.125W 0805 Resistor Thick Film Chip Vishay CRCW08050000Z0EA Tx1 25W LED TRANSFORMER - NIC 10 pin vertical 25W Flyback transformer NIC NLT282224W3P4020S5P10F September 2008, Rev. 2 www.onsemi.com 7 DN06040/D Component Placement and PCB Layout Top view Bottom view Completed Demo Board, Side View September 2008, Rev. 2 www.onsemi.com 8 DN06040/D Typical Operational Results 5 µs 100 V V IN = 230 V V IN = 120 V AC Drain waveform at 120 Vac and 230 Vac 500 ns 100 V V IN = 265 V AC V PK = 462 V V IN = 230 V AC V PK = 414 V V IN = 120 V AC V PK = 256 V Turn-off in detail at 120 Vac, 230 Vac and 265 Vac September 2008, Rev. 2 www.onsemi.com 9 DN06040/D Typical Evaluation Results Efficiency versus Line and Load @ 700mA Ta = 21˚C / 70˚F 90% 80% 70% Efficiency (%) 60% 50% 40% 30% 115 Vac 230 Vac 20% 10% 0% 0 5 10 15 20 25 30 LED Voltage (Vdc) Current Regulation versus Forward Voltage @ 700mA Ta = 21˚C / 70˚F 0.8 LED Current (A) 0.7 0.6 0.5 0.4 230 Vac 0.3 115 Vac 0.2 0.1 0 0 3.5 7 10.5 14 17.5 21 24.5 28 31.5 35 LED Forward Voltage (Vdc) September 2008, Rev. 2 www.onsemi.com 10 DN06040/D Modifying the Board for Other LED currents The constant current constant voltage secondary control loop is very flexible and is implemented using a PNP (Q3) with a pair of current sense resistors (R12 & R13) to regulate the current and provide control of the optocoupler to the NCP1351. In addition, there is a maximum voltage control loop that is implemented using zener D10. To modify this circuit for alternate current / voltage configurations, these components should be modified. The table on the front page shows several other possible configuration options. Note because this design is ultimately power limited based on the transformer design and FET used, as the current decreases, the maximum voltage capability increases. For example, for 20W output, the maximum voltage at 350 mA could be as high as 57 Vdc. Under UL1310, Class 2 power supplies for use in dry/damp environments are allowed to have a maximum output voltage of 60 Vdc. On the demo board, Q3 is implemented using a BC857 transistor which has a maximum VCEO of -45 Vdc. If a higher operating voltage is required, this transistor can be changed to a BC856 (maximum VCEO of -65 Vdc). The figure below shows the current regulation performance for a nominal 350 mA output current with the component changes as noted. Typical Current Regulation versus Load, Ta = 21˚C / 70˚F R12 & R13 = 3.6 ohms each, D10 = MMSZ5263B (56V), Q3 = BC856 0.40 0.35 LED Current (A) 0.30 0.25 0.20 0.15 0.10 0.05 0.00 0 5 10 15 20 25 30 35 40 45 50 55 60 LED Forward Voltage (Vdc) 1 © 2008 ON Semiconductor. Disclaimer: ON Semiconductor is providing this design note “AS IS” and does not assume any liability arising from its use; nor does ON Semiconductor convey any license to its or any third party’s intellectual property rights. This document is provided only to assist customers in evaluation of the referenced circuit implementation and the recipient assumes all liability and risk associated with its use, including, but not limited to, compliance with all regulatory standards. ON Semiconductor may change any of its products at any time, without notice. Design note created by Anthony Middleton, e-mail: [email protected] September 2008, Rev. 2 www.onsemi.com 11