350 mA Buck Boost LED Driver using Bipolar Junction Transistors

AND8305/D
350 mA Buck Boost LED
Driver using Bipolar
Junction Transistors (BJTs),
High Side Current Sensing
and a NCP3063 Controller
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Prepared by: DENNIS SOLLEY
ON Semiconductor
INTRODUCTION
(typically absorbed by avalanching alternator rectifiers or
by a transient suppressor). Equally wide input variations can
be expected when the driver is powered from a 12 Vac line
transformer and bridge rectifier. Depending on the
illumination required, a particular application may require
a driver to supply 350 mA to 3, 4 or 5 LEDs in series. From
Table 1, the driver must support output variations between
7 volts and 21 volts.
Hence, a constant current converter with both a wide input
(9-19 V) and wide overlapping output (7-21 V) range, is
preferred.
This application note targets a current regulated, non
inverting buck boost converter. In automotive applications
(e.g. emergency vehicles), a high side current sensing
scheme can simplify wiring by returning the LED string to
chassis ground. The basic buck boost topology, consisting of
a buck and boost converter cascaded together, is illustrated
in Figure1.
Unlike traditional lighting, LEDs require driver solutions
that address the challenges of providing a constant current
to a load whose output voltage can vary by ±30% because of
process and temperature effects. This application note and
associated demo board will focus on driving multiple LEDs,
at a regulated 350 mA, from low voltage DC or AC sources
commonly used in lighting applications.
LED Characteristics
Due to the steep V/I curve of the LED and to achieve
optimum performance, it is critical to drive LEDs with a
constant current to achieve the specified brightness and
color. For high brightness power LEDs, the specified current
may be in the range 150 - 1500 mA, 350 mA being a
common value.
By combining LED manufacturer's data, taken from
several product families, it is possible to come up with
minimum and maximum forward voltage drops for a
“generic” LED, operating at a specified current. This
voltage variation is presented in Table 1, and extended to
include 3 to 5 LED combinations.
+Vin
Table 1. Output Voltage Variation for a
“Generic” 350 mA LED
Generic
LED
# String
D1
Current (A)
VMIN (V)
@ TJ(max)5C
(Note 1)
VMAX (V)
@ 255C
1 LED
0.35
2.30
4.23
3 LEDs
0.35
6.90
12.69
4 LEDs
0.35
9.62
16.92
5 LEDs
0.35
11.50
21.15
Q2
Figure 1. Buck Boost Converter
Theory of Operation
To minimize power dissipation in the power circuit, low
ripple current is required. So the converter is run in
continuous current mode (CCM). For this analysis, all
power components are assumed ideal. During the first
switching interval D*TSW, Q1 and Q2 are turned ON by the
controller across the input Vin and allow energy to be stored
in the inductor. The current flow is illustrated in Figure 2.
Driver Definition
A typical automotive input requirement may require
continuous operation between 9 V and 16 V, excursions
between 18 V and 19 V for one hour, a double battery jump
start to 26 V for one minute and finally a load dump to 70 V
December, 2007 - Rev. 1
+Vout
MOSFETs or BJTs can be selected as the primary switches
Q1/Q2. However, in this lower power application (to
7watts) BJTs offer a cost effective solution. (See application
note AND8306/D for higher power applications (to 20
watts) using FETs).
1. TJ(max) based on LED manufacturer's maximum rating
© Semiconductor Components Industries, LLC, 2007
D2
Q1
1
Publication Order Number:
AND8305/D
AND8305/D
+Vin
V in @ D @ T SW + V out @ (1 * D) @ T SW
+Vout
Q1
(eq. 1)
Rearranging Equation 1 the voltage gain of buck boost is
given by:
D * TSW
Q2
D * TSW
V out + V in @
During the second switching interval (1-D)*TSW,
switches Q1 and Q2 are turned off by the controller,
allowing diodes D1 and D2 to conduct and deliver the
energy stored in the inductor to the load. The current flow
during this interval is illustrated in Figure 3.
D2
(eq. 2)
Varying the duty cycle will vary the output. When D is
below 0.5, the converter is in buck mode, when D is above
0.5, the converter is in boost mode and when D equals 0.5,
the voltage gain Vout/Vin is unity.
The ripple current in the inductor is given by expression
Figure 2. Switch Conduction During First
Switching Interval D*TSW
+Vin
D
1*D
DI L1 +
V in @ D @ T SW
L1
(eq. 3)
For a typical design case, where Vin = 12 V and D*TSW
= 0.5*5 ms, a value for L1 of 150 mH (Equation 3) will
maintain ±30% ripple current in a 350 mA application,
thereby ensuring CCM operation.
+Vout
(1-D) * TSW
BJT Refresher
A BJT is a current controlled device. The turn on, turn off,
saturation voltage and storage time of a BJT are all
determined by the magnitudes of turn on IB1 and turn off IB2
base currents. These currents are identified in Figure 5. The
collector current rise time is controlled by the magnitude of
IB1. The ratio Ic/IB1 controls the VCE(sat) of the BJT but there
is a trade off as a large IB1 will be associated with a long
storage time TS. This is the time interval before the BJT
comes out of saturation.
D1
(1-D) * TSW
Figure 3. Diode Conduction During the Second
Switching Interval (1-D)*TSW
(1-D) * TSW
Collector
Current IC
Vout
Turn On
Base Current
Vin
Turn Off
Base Current
Collector
Current IC
=0A
IB2
IB1
TS
VCE(sat)
D * TSW
Figure 4. Voltage Waveform Across the Inductor
Figure 5. Turn On IB1 and Turn Off IB2 Base Currents
For the inductor flux (V*ms) to remain in equilibrium each
switching cycle, the V*ms product across the inductor during
each switch interval must balance (see Figure 4).
A simplified power stage showing how the Q1 and Q2
base drives are derived is illustrated in Figure 6.
Vin
D2
L1
Q1
Vout
IB2
IB1
Cout
D1
Cin
D
IB1
Q2
IB2
Figure 6. Simplified Power Stage showing BJT Base Drives
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AND8305/D
output inductor and D2, but here, power continues to flow
between the input and output. This mode of operation is
preferred since higher conversion efficiency is possible.
Depending on the drive resistors selected, the storage time
of a typical BJT maybe 1 or 2 ms. Hence the converter's duty
cycle D is modified by an additional term DDLY because of
storage effects. Further the storage times for Q1 and Q2 may
be significantly different, impacting converter operation.
+Vin
Q1 OFF FIRST
Inductor Energy Cycles via D1 and Q2
Key Component Selection
NSS40500UW3T2G and NSS40501UW3T2G from
ONSemiconductor's e-PowerEdge family of BJTs were
chosen for cost/performance criteria. They feature ultra low
saturation voltage at a 10:1 drive ratio (Figure 8) and the
WDFN3 package provides excellent thermal performance
(RqJL = 23°C/W).
+Vout
Q2
D1
+Vin
VCE(sat), COLLECTOR EMITTER
SATURATION VOLTAGE (V)
1.8
D2
Q1
Q2 OFF FIRST
Inductor Energy Flows Input to Output
IC/IB = 100
1.6
1.4
1.2
1.0
0.8
0.6
IC/IB = 10
0.4
0.2
Figure 7. Energy Flow Depending on Whether Q1
or Q2 Turns Off First
0
0.001
If Q1 turns off first, energy flows between D1, the output
inductor and Q2, until Q2's storage interval is completed.
This mode of operation (shown in Figure 7) generates loss
but no power flows between the input and output.
Alternatively, if Q2 turns off first, losses still occur in Q1, the
0.01
0.1
The controller used is ON Semiconductor's NCP3063. A
functional block diagram is shown in the Figure 9.
NCP3063
1
TSD
N.C.
Switch Collector
SET Dominant
R
Q
S
7
Comparator
+
2
S
Q
Switch Emitter
SET Dominant
R
+ 0.2 V
Oscillator
6
3
Timing Capacitor
CT
+VCC
5
10
Figure 8. Collector Emitter Saturation Voltage vs.
Collector Current
8
Ipk Sense
1.0
IC, COLLECTOR CURRENT (A)
1.25 V
Reference
Regulator
Comparator
+
-
4
GND
Inverting Input
Figure 9. Block Diagram of NCP3063
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3
AND8305/D
This device consists of a 1.25 V reference, comparator,
oscillator, an active current limit circuit, a driver and a high
current output switch. In its traditional operating mode, the
NCP3063 is a hysteretic, regulator that uses a gated
oscillator to control the output. Voltage feedback from the
output is sensed at pin 5, and gates the oscillator on/off to
regulate the output. The oscillator frequency and off-time of
the output switch are programmed by the value selected for
the timing capacitor; CT. CT is charged and discharged by a
1 to 6 ratio internal current source and sink, generating a
ramp at pin 3. The ramp is controlled by two comparators
whose levels are set 500 mV apart. In normal operation, D
is fixed at 6/7 or 0.86. In this application, the “gated
oscillator” mode is only used to protect the LED string if a
LED fails “open”.
The NCP3063 can also operate as a conventional PWM
controller, by injecting current into the CT pin. The control
current may be developed either from the input source,
providing voltage feedforward or from the output current
sensing circuit. In either case, the slope of the oscillator ramp
changes causing D to be modulated as shown in Figure 10.
VCC
IFF
important to select a device and package that will maintain
the device temperature in the particular application to avoid
thermal runaway. The effect is shown in Figure 11.
100
IR, REVERSE CURRENT (mA)
TJ = 150°C
100°C
1
75°C
0.1
25°C
0.001
0
10
20
30
40
50
VR, REVERSE VOLTAGE (V)
60
When the driver is in boost mode driving multiple LEDs,
maximum power (6 watts) is delivered through diode D2.
Because of the storage delays discussed previously, both D2
and Q1 conduct for an extended duty cycle compared to D1
and Q2. In order to process 6 watts on a 1 in. x 1 in. demo
board, MBRD340 was selected for D2. At lower power
levels, D2 could be replaced with MBRA340 at board
location D4.
The schematic of the power stage is shown in Figure 12.
Note the addition of the speed up diode D3 to ensure Q3
turns off ahead of Q1. As Q3 approaches saturation, the IB1
base current is diverted through D3 holding the transistor out
of saturation. This technique reduces Q3's storage time an
order of magnitude at the expense of incurring additional
VCE(sat) losses
NCP3063
Figure 10. Current Injection into CT Pin Providing
Continuous Duty Cycle Modulation
Schottky Diode Selection
Schottky diodes have reverse leakage current which
increases with reverse voltage and temperature. Hence it is
ISENSP
ISENSN
D2
VIN
TP1
R14
Q1
L1
TP2
LED+
R4
D4
R2
D1
R3
D3
+ C1
Q3
U1
1
R1
CLK
2
3
4
C2
RTN
SWC
NC
SWE
ISENS
CT
VCC
GND CMPINV
8
C3
R5
7
6
R7
5
R8
70
Figure 11. Reverse Leakage Characteristic of
MBRD360
6 ICHARGE
CT
125°C
0.01
ICHARGE
IFB
10
LED-
C4
I_CNTRL
Figure 12. Schematic of Power Stage
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4
AND8305/D
I_SENSP
R11
I_SENSN
R10
Q6B
Q6A
Q5
I_CNTRL
R8
Q7A
VIN
R9
C5
Q7B
U2
R12
R13
Figure 15. Boost Mode from 12 Vin to 16 Vout
It is evident from Figures 14 and 15 that the inductor
waveforms differ from the classic buck boost illustrated in
Figure4. We define TS to be the difference in storage times
of Q1 and Q2. During buck operation (Figure 14) the voltage
across the inductor is clamped at (Vin-Vout) for the duration
TS. During this interval Q2 is off and Q1 is on for the
remainder of its storage time. During this period, power is
delivered to the output via Q1 and D2 as previously
discussed. In boost mode (Figure 15), the inductor voltage
is clamped at (Vout-Vin) for the interval TS. The effect is
shown in Figure 16.
Figure 13. High Side Current Sensing Control Circuit
In Figure 12, the current sense resistor R4 is placed in
series with LED+, to satisfy the high side sensing
requirement. The control circuit is illustrated in Figure 13.
Here the bandgap reference U2, together with dual NPN
transistors Q7A, Q7B and R12, R13 create two equal current
sinks. These currents flow through the PNP matched current
pair of Q6A and Q6B configured as a current mirror. At the
same time current flowing through resistor R10 creates a
voltage reference VSENSE. When the current sense signal
ILED*R4 equals VSENSE, Q6A turns on. The voltage
follower Q5 controls the current flowing into the CT pin of
U1, thereby regulating the LED current at the required
value. Capacitor C5 provides loop compensation. The
voltage reference VSENSE can be made small (150 mV) to
limit dissipation in the current sense resistor R4.
Modifications to VSENSE and the 350 mA set point, can be
made by adding a parallel resistor at location R11 on the
demo board. For less demanding applications, the 1.25 V
bandgap reference U2 can be replaced with dual series
switching diodes (BAV99LT1) having a similar drop.
Buck Mode < 0.5
(Vin - Vout) * TS
(Vout - Vin) * TS
Boost Mode > 0.5
Converter Waveforms
The voltage waveforms at both the input (upper trace) and
output (lower trace) of the inductor L1 were measured while
the difference waveform (middle trace) gives the voltage
across the inductor. Figure 14 shows the converter operating
in buck mode, while Figure 15 illustrates boost operation.
Figure 16. Voltage Across Inductor when Storage
Interval TS is Included
If we define DDLY = TS / TSW, the flux balance expression
given in Equation 1 is modified as follows:
(eq. 4)
V in @ D @ T SW " (V in * V out) @ T S + V out(1 * D * T DLY) @ T SW
The transfer function given in Equation 2 is also modified
and becomes:
V out + V in @
(D ) D DLY)
(1 * D)
(eq. 5)
The term DDLY appearing in Equation 5 expresses
mathematically the fact that components Q1 and D2 have an
extended duty cycle. Put another way, to achieve the same
converter gain as in the classical case (Equation 2), switch
Q2's duty cycle D is reduced.
Figure 14. Buck Mode from 12 Vin to 8 Vout
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AND8305/D
Demo Board
The top side component layout of the NCP3063 buck boost demo board is shown in Figure 17.
Figure 17. Top Side Component Layout
The bottom side component layout is shown in Figure 18. Note that the copper pours mounting the power components Q1,
Q2, D1, D2 and L1 have been maximized within the 1in. x 1in. footprint of the board.
Figure 18. Bottom Side Component Layout
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AND8305/D
Test Data
82
81
A 12 V source is connected between VIN (positive) and
RTN (negative) and the LED string, consisting of 3, 4 or 5,
350mA rated devices are connected across LED+ and
LED-.
Efficiency data, measured over an extended overlapping
input and output voltage range, is shown in Figure 19. As can
be seen from the efficiency curves, the driver efficiency
varies between 75% and 80 % over a wide input and output
range. For Vin equal to 10 V, the efficiency starts to fall as
Vout is increased above 14 V. Under these operating
conditions, the driver performance is limited by the base
drive supplied to BJT Q1.
18 V
EFFICIENCY (%)
80
16 V
79
12 V
14 V
78
77
Vin = 10 V
76
75
74
73
72
8
10
12
14
Vout (V)
16
18
20
Figure 19. 350 mA Buck-Boost LED Driver
Efficiency over Line and Load (3-7 W)
The BOM for the NCP3063 buck_boost demo board is given in Table 2. Generic resistors and capacitors are referenced by
Digi-Key part numbers.
Table 2. BOM for NCP3063 Buck_boost Demo Board
Designator
Quantity
Manufacturer
Manufacturer Part Number
U1
1
ON Semiconductor
NCP3063DR2G
U2
1
ON Semiconductor
TLV431ASN1T1G
Q1
1
ON Semiconductor
NSS40500UW3T2G
Q3
1
ON Semiconductor
NSS40501UW3T2G
Q5
1
ON Semiconductor
2N7002LTIG
Q6
1
ON Semiconductor
NST30010MXV6T1G
Q7
1
ON Semiconductor
MBT3904DW1T1G
D1
1
ON Semiconductor
MBRA340T3G
D2
1
ON Semiconductor
MBRD340T4G
D3
1
ON Semiconductor
BAT54T1G
C1
1
100 mF/25 V
P10413TB-ND
C2
1
3900 pF/50 V
478-1222-2-ND
C3
1
10 mF/25 V
490-3373-2-ND
C4
1
1 mF/25 V
587-1248-2-ND
C5
1
47 nF/35 V
587-1248-2-ND
R1
1
Not required
NA
R2, R5
2
100/0603
541-100HTR-ND
R3
1
200/0805
P200CTR-ND
R4
1
IRC
LRC-LR1206-01-R400-F
R6
1
2.49 k/0603
P2.49KHTR-ND
R7
1
41.2 k/0603
P41.2HTR-ND
R8, R12, R13
3
2.21 k/0603
541-2.21KHTR-ND
R9
1
4.99 k/0603
311-4.99KHTR-ND
R10
1
499 k/0603
P499HTR-ND
R11
1
Not Required
NA
R14
1
IRC
LRC-LR1206-01-R100-F
L1
1
TDK
SLF10145T-151MR79-PF
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AND8305/D
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